JP2003204294A - Transmission diversity delay correction system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide radio equipment capable of simple, highly accurate and highly reliable transmission diversity delay correction. <P>SOLUTION: RF signals amplified in amplifiers 105a and 105b are distributed in distributors 106a and 106b, one is outputted from transmission parts 100a and 100b and the other is inputted to a synthesizer 108 as it is on the side of the distributor 106a and is inputted to the synthesizer 108 after being passed through a delay line 107 on the side of the distributor 106b. The respective RF signals inputted to the synthesizer 108 are synthesized, then inputted to a detector 109 and converted to a detection voltage. The detection voltage is inputted to a delay amount control circuit 110 and compared with base band signals, compared output is outputted to a delay circuit 102a as control signals and control is performed such that the phase difference of the RF signals outputted from the transmission parts 100a and 100b is within a prescribed range. <P>COPYRIGHT: (C)2003,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a diversity type transmitter used in mobile communication, and more particularly, to transmitting the same modulated waves from two transmitters and receiving the modulated waves at a receiving point. The present invention relates to a transmission diversity method for obtaining a gain.

[0002]

2. Description of the Related Art Conventionally, in a transmission diversity system in which the same modulated wave is transmitted from two transmitting sections at the same time and their modulation timings are matched at a receiving point to obtain a diversity gain, modulation is performed at a transmission output end. Timings must match (at least within acceptable limits). Therefore, the transmitter of the diversity system used for such a purpose needs to suppress the delay time difference of each modulated wave data within an allowable range.

The cause of the difference in the delay times of these modulators is that the delay times of the circuit parts vary. Especially, the influence of the variation in the delay amount in the IF filter mounted on the intermediate frequency (IF) before being converted into the RF frequency is large. Therefore, in order to realize a diversity transmitter, it is necessary to control the delay time in each transmitter.

Therefore, conventionally, there has been adopted a method of measuring the delay amount of the transmitter actually used to control the delay amount and correcting the delay at the time of production. However, in this method, it is necessary to measure the delay amount at the time of production, so that the production efficiency is poor. Also, since it is only set during production,
There is a problem that the expected diversity gain may not be obtained if the difference in delay amount changes due to temperature fluctuations or aging changes during operation.

An example of a transmission diversity delay correction method that solves such a problem is Japanese Patent Application No. 2000-177110.
It has been filed as an issue. FIG. 7 is a block diagram showing a transmission diversity delay correction method proposed in the application.

In FIG. 7, the baseband signal generator 1
The baseband signal generated in 01 is transmitted by the two transmitters 1
00a, 100b. The baseband signals input to the respective transmission units are delayed by the delay circuits 102a and 10a.
It is input to the modulators 103a and 103b through 2b.
The modulators 103a and 103b generate a modulated wave from the baseband signal. The modulated waves are frequency converters 104a, 10
4b is converted into an RF signal, and the amplifiers 105a and 105b
Is amplified to the specified power by the distributors 106a and 106b.
It is output from the transmission unit via.

On the other hand, a part of each transmission output distributed by the distributors 106a and 106b is detected by the detectors 109a and 109b and output to the comparator 113. The comparator 113 compares the detection outputs of the distributors 106 a and 106 b and outputs the comparison signal to the delay amount control circuit 110. The delay amount control circuit 110 uses the delay circuit of each transmission unit so that the difference between the delay times of the RF signals a and b output from each transmission unit 100a and 100b falls within an allowable value based on the input comparison signal. The delay amounts of 102a and 102b are automatically controlled.

[0008]

In the transmission diversity delay correction method shown in FIG. 7, the RF signals distributed by the distributors 106a and 106b are input to the detectors 109a and 109b to be converted into detected voltages, and then detected. Voltage comparator 11
3 and performs mutual comparison, inputs the difference between the two as a comparison signal to the delay amount control circuit 110, and controls the delay circuits of the two transmission units in real time so that the modulation timings match at the transmission output end. Therefore, even if the difference in delay amount changes due to temperature fluctuation or aging change during operation, the delay amount is automatically corrected, and the expected diversity gain can be obtained.

However, in the case of this method, it is necessary to provide the detectors 109a and 109b for each transmitting section. Diode detection is an example of the detectors 109a and 109b, but since temperature fluctuations are large, it is necessary to provide a temperature compensation circuit for each. Further, the diodes themselves have large variations, and it is not easy to match the characteristics of the respective detector circuits.

Further, even if the active element other than the diode is used, the active element causes temperature fluctuation or variation, and the possibility of failure increases. Further, as an example of the comparator 113, there is a comparison using an OP amplifier, but since this is also an active element, temperature fluctuations and variations occur, and the possibility of failure increases.

In view of the above problems, an object of the present invention is to improve the production efficiency, obtain the expected diversity gain even when there is a temperature change or a secular change during operation, and it is simple and It is an object of the present invention to provide a wireless device capable of highly accurate and highly reliable transmission diversity delay correction.

[0012]

A radio apparatus equipped with a transmission diversity system of the present invention allows only one system to pass through a delay line, and then synthesizes mutual transmission signals by a synthesizer, and synthesizes the synthesized transmission signals. Is detected by a detection circuit and controlled until the combined transmission signal reaches a predetermined power, so that the transmission diversity delay correction is performed with high accuracy, high reliability and in real time with a simple configuration.

More specifically, the transmission diversity delay correction method of the present invention is a baseband signal generator for outputting a baseband signal, and a first for outputting a baseband signal as an input and converting it into a predetermined signal for output to the outside. And a second transmitter,
The first transmitter receives the baseband signal and the control signal as input, the second transmitter receives the baseband signal as input, the delay circuit for delaying the baseband signal, and the baseband signal via the delay circuit as input. A modulator that generates a modulated wave, a frequency converter that receives the modulated wave and converts it into an RF signal, an amplifier that receives the RF signal as an input and amplifies it to a predetermined power, and an amplified RF signal as an input, and one of the A first and a second transmitter for external output, and a distributor for distributing the other for delay correction, and a delay line for delaying the delay correction RF signal output from the second transmitter as an input. , A combiner for combining the RF signal for delay correction via the delay line and the RF signal for delay correction output from the first transmitter as inputs, and a combined RF signal as an input for conversion into a detection voltage Detector , Characterized in that it is composed of a delay amount control circuit for outputting a control signal by comparing the two inputs the detection voltage and the baseband signal to the delay circuit of the first transmission unit.

[0014]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a block diagram showing an embodiment of the present invention.

In FIG. 1, a baseband signal generator 1
The baseband signal generated in 01 is transmitted by the transmitter 100a.
And 100b and the delay amount control circuit 110 are input. The baseband signal input to the transmitter 100 a and the control signal output from the delay amount control circuit 110 are the delay circuit 102.
Input to a. The baseband signal input to the transmitter 100b is input to the delay circuit 102b.

The baseband signals passed through the delay circuits 102a and 102b, respectively, are modulated by the modulators 103a and 103b.
Entered in. The modulators 103a and 103b generate a modulated wave from the baseband signal. The modulated wave is converted into an RF signal by the frequency converters 104a and 104b. The RF signal is amplified by the amplifiers 105a and 105b to a predetermined power.

R amplified by the amplifiers 105a and 105b
The F signal is distributed by the distributors 106a and 106b, one of which is output from the transmission units 100a and 100b, and the other of which is directly input to the combiner 108 on the distributor 106a side.
On the side of the distributor 106b, it is input to the combiner 108 after passing through the delay line 107. The respective RF signals input to the combiner 108 are combined and then input to the detector 109 and converted into a detection voltage. The detected voltage is input to the delay amount control circuit 110 and compared with the baseband signal, and the comparison output is output to the delay circuit 102a as a control signal.

Next, the present embodiment uses the W-CDMA modulation method (chip rate 3.84 MHz) and has an accuracy of ± 1/1.
An embodiment applied to a transmitter which requires 6 chips and an adjustment range of ± 1/2 chip will be described.

In FIG. 1, a baseband signal generator 1
The baseband signal generated in 01 is transmitted by the transmitter 100.
a, 100b. The baseband signals input to the respective transmission units are delayed by the delay circuits 102a and 102b.
Entered in. Here, the delay circuits 102a and 102b are previously delayed by 1/2 chip. The reason is 1 /
By delaying by 2 chips, the delay circuit 102
This is because, in the phase adjustment of a, not only the phase can be delayed but also the phase can be advanced, so that the delay can be corrected by adjusting only the delay circuit 102a side and the circuit can be simplified.

Next, the baseband signal is transmitted to the modulator 103.
a and 103b are input to generate a modulated wave. This modulated wave is converted into an RF signal by the frequency converters 104a and 104b. This RF signal is transmitted to the amplifiers 105a and 105b.
It is amplified up to the specified power at. Amplifiers 105a, 105
The RF signal amplified by b is distributed to the distributors 106a and 106b.
, And one is output from the transmitting units 100a and 100b, and the other is directly on the side of the distributor 106a.
8 and the distributor 106b side is input to the combiner 108 after passing through the delay line 107.

Here, the delay line 107 has a length of ½ wavelength. Note that the method of obtaining the 1/2 wavelength is, for example, RF
When the frequency is 2150 MHz and the relative permittivity of the substrate that realizes the delay line is 4.8, the following formula is used. 1/2 wavelength = speed of light / ((√ relative permittivity) × RF frequency × 2) = (3 × 10 ^ 8) / ((√4.8) × (2150 × 10 ^ 6) × 2) = 0.0318 [m] Becomes Since the delay line 107 is a passive element, there is an advantage that there are few failures and variations.

FIG. 2 shows a transmitter 100a in the synthesizer 108.
2 is a table showing the relationship between the chip shift (phase shift) of 100b and the output power. Here, the chip shift is caused by the transmission unit 10
Consider 0a as a reference. The relationship between chip, phase, and wavelength is 1 chip = 360 ° = 1 wavelength.

First, consider a case where there is no delay difference between the transmission units 100a and 100b. The RF signal input to the combiner 108 is halved by the delay line 107 on the transmitting unit 100b side.
Since the wavelength is delayed (−180 °), the transmitter 100
The phase difference (θ) between the RF signals on the a side and the transmitter 100b side is 1
It becomes 80 °. Therefore, the output power of the combiner 108 is 1 / √2 × (sin (ωt) + sin (ωt + 180 °)) = 1 / √2 × (2 × sin ((2ωt + θ) / 2) × cos ( -θ / 2)) = 2 / √2 × (sin (ωt + θ / 2) × cos (-90 °)) = 0, and the output power is irrespective of the input power of the combiner 108.
It becomes 0 (∞ in the input / output power ratio).

Next, consider a case where there is a delay difference between the transmission units 100a and 100b. For example, only the transmitter 100a is 180
° If there is a phase advance (8/16 chip shift),
The RF signal input to the combiner 108 is transmitted by the transmitter 100b.
Since the delay line 107 delays the wavelength by ½ wavelength (−180 °), the phase difference (θ) between the transmitting unit 100a side and the transmitting unit 101b side is 360 °. Therefore, the output power of the combiner 108 is 1 / √2 × (sin (ωt) + sin (ωt + 180 °)) = 2 / √2 × (sin (ωt + θ / 2) × cos (360 °)) = 2 / √2 × (sin (ωt + θ / 2), so 20 × log (2 / √2) ≈3.01dB, and the output power is +
It becomes 3.01 dB.

If only the transmitter 100a has a phase delay of 22.5 ° (chip displacement of 1/16), the RF signal input to the combiner 108 is 1 by the delay line 107 on the transmitter 100b side. Since the wavelength is delayed by 1/2 wavelength (−180 °), the phase difference (θ) between the transmitting unit 100a side and the transmitting unit 101b side is 157.5 °. Therefore, the output power of the combiner 108 is 1 / √2 × (sin (ωt) + sin (ωt + 180 °)) = 2 / √2 × (sin (ωt + θ / 2) × cos (-78.75 °) ) ≈ 0.2759 × (sin (ωt + θ / 2), so 20 × log (0.2759) ≈ -11.18 dB, and the output power is-
It becomes 11.18 dB. A Wilkinson type or the like is used as the synthesizer 108, and since it is a passive element, there is an advantage that there are few failures and variations.

FIG. 3 shows an example of the relationship between the input power of the detector 109 and the output detection voltage. Synthesizer 10
The combined RF signal that outputs 8 is input to the detector 109, and the detector 109 converts the input power into a detection voltage. An example of the detector 109 is diode detection. This is a method of performing half-wave rectification with a diode and smoothing with a time constant to convert electric power into a voltage. In the case of the system like the embodiment, the chip misalignment of ± 1 shown in the table of FIG.
Since it only needs to be converged within / 16 chips, the combiner 10
It may be determined whether the output power is 11.18 dB or less with respect to the input power of 8.

Therefore, the absolute accuracy required for the detector 109 is that the input power is close to 1/16 chip or 2/16 ch.
It suffices to be able to determine whether it is close to ip. In order to judge, the combined power at 1.5 / 16 chip deviation (-7.73) -1/16 chip difference (-11.18) = 3.45 dB, so if the accuracy is within ± 3.45 dB, 1 / 1
Since 6 chips are not erroneously detected as 2/16 chips, it is not necessary to use high-precision parts or circuits in order to improve accuracy.

Since the error in the detector 109 is already the combined RF signal on the transmitting unit 100a side and the transmitting unit 100b side, the error for detecting the phase difference between the transmitting units 100a and 100b does not occur here. Next, the detection voltage output from the detector 109 is input to the delay amount control circuit 110. The baseband signal is also input to the delay amount control circuit 110.

The power of the baseband signal and the combiner 10
8 is a relative relationship between the input power of 8 and the relationship between the input power and the output power of the combiner 108 is the transmitter 100a,
Since it depends on the delay amount of 100b, the delay amount control circuit 11
0 is the power of the baseband signal and the combiner 10 in advance.
By remembering the relationship of the input power of No. 8, the delay amount can be known in real time by satisfying the following formula or constantly calculating. (Power of baseband signal [dBm])-(Input power of detector 109 [dB
m] + 11.18dB)> 0 If the delay difference between the transmitters 100a and 100b is 1 / 16c
If there are more than hip, the above formula is not satisfied. Therefore, in this case, a control signal is output to the delay circuit 102a to perform delay correction.

FIG. 4 is a flow chart showing an example of the operation of the delay control circuit 110. Delay amount control circuit 110
Calculates the detection voltage Pn input from the detector 109 and the power Pm of the baseband signal. (Step S1) Next, it is calculated whether Pm- (Pn + 11.18 dB)> 0 is satisfied. (Step S2) If the calculation formula is satisfied, no particular control is performed,
Returning to step S1, Pn and Pm are obtained again.

Therefore, as long as the formula is satisfied, step S
Since 1 and S2 are repeated, the delay amount is not controlled. If the calculation formula is not satisfied, first delay circuit 1
The delay amount of 00a is increased by 1/16 chip. (Step S3) Then, Pn and Pm are obtained again (Step S4),
It is calculated whether Pm- (Pn + 11.18 dB) has increased (step S5).

If it becomes larger, step S2
Return to and judge whether Pm- (Pn + 11.18dB)> 0 is satisfied. If the calculation formula is satisfied, the process returns to step S1, the delay control is temporarily terminated, and steps S1 and S2 are repeated. If the calculation formula is not satisfied, delay control is performed again in step S3. If the calculation formula is not satisfied in step S5, the delay amount of the delay circuit 100a is reduced to 1 /.
Decrease by 16 chips (step S6).

Then, Pn and Pm are obtained again (step S
7), it is determined whether Pm- (Pn + 11.18dB)> 0 is satisfied. (Step S8) If the calculation formula is satisfied, the process returns to step S1. If the calculation formula is not satisfied, delay control is performed again in step S6. As a result, the transmission unit 100 is automatically
The delay time difference between a and 100b is controlled to fall within the convergence allowable range of FIG.

FIG. 5 is a block diagram showing another embodiment of the present invention. In the above embodiment, it is assumed that the powers of the baseband signals of the transmission units 100a and 100b are equal, whereas in the present embodiment, the transmission unit 100 is used.
It is assumed that the powers of the baseband signals of a and 100b are different, and if the powers of the baseband signals of the transmitters 100a and 100b are different, if the RF signal is directly input to the combiner 108, the relationship shown in FIG. Therefore, the variable attenuator ATT112 is inserted in front of the delay line 107 to adjust the power, and the RF input to the combiner 108 is obtained.
The feature is that a means for making the signals the same power is added.

In FIG. 5, the baseband signal generator 1
Of the baseband signals on the transmission unit 100a side that output 01, the first baseband signal is input to the delay circuit 102a, and the subsequent transmission unit 100a processes the same as in FIG. Further, the second baseband signal is input to the delay amount control circuit 110, and thereafter, processed in the same manner as in FIG.
Also, the third baseband signal is the variable ATT control circuit 1
11 is input. Further, one RF signal output from the distributor 106a is directly input to the combiner 108 in the example of FIG. 1, but is input after being passed through the ATT 113.

Next, one of the baseband signals output from the baseband signal generator on the side of the transmitter 100b is input to the delay circuit 102b, and is processed in the transmitter 100b thereafter in the same manner as in FIG. The other is input to the variable ATT control circuit 111. The variable ATT control circuit 111 is
The powers of the baseband signals of the transmitters 100a and 100b are compared, and the difference is output to the variable ATT 112 as a variable ATT control voltage. FIG. 6 shows an example of the relationship between the control amount of the variable ATT 112 and the control voltage.

For example, the variable ATT 112 and ATT 113 are set to 20 dB in advance, and the variable ATT control circuit 1
The power compared at 11 is 3 dB at the transmitter 100a side.
If it is low, the control voltage is lowered so that the control amount becomes 17 dB. If the power compared by the variable ATT control circuit 111 is 3 dB higher on the transmitting unit 100a side, the control voltage is increased so that the control amount becomes 23 dB.
By this control, the transmission unit 10 input to the synthesizer 108
The RF signals of 0a and 100b have the same power, and the delay amount shown in FIG. 2 can be adjusted.

In FIG. 5, the variable ATT 112 is inserted between the distributor 106b and the delay line 107, and the ATT 113 is
Is inserted between the distributor 106a and the combiner 108, but the variable ATT 112 is connected to the distributor 106a and the combiner 108.
, And the ATT 113 may be inserted between the distributor 106b and the delay line 107.

[0039]

According to the present invention, the delay correction target R
Until the F signals are compared with each other, it can be realized by a circuit that does not include an active element. Therefore, it is possible to perform highly accurate phase detection with few failures and without temperature fluctuation or variation, and simple, highly accurate, and highly reliable transmission. Diversity delay correction can be performed in real time.

Further, the means for detecting the phase by comparing the RF signals to be delay-corrected with each other is adopted because the phase is inverted only through one system through the delay line and then the phase is inverted. Only one circuit is required, which enables highly accurate phase detection.

[Brief description of drawings]

FIG. 1 is a configuration block diagram of a transmission diversity delay correction method in an embodiment of the present invention.

FIG. 2 is a diagram showing a relationship between phase shifts of transmitters 100a and 100b and output power of combiner 108.

FIG. 3 is a diagram showing a relationship between an input power of a detector 109 and a detection voltage.

FIG. 4 is a flowchart showing the operation of the delay amount control circuit 110.

FIG. 5 is another configuration block diagram of a transmission diversity delay correction method in the embodiment of the present invention.

FIG. 6 is a diagram showing a relationship between a control amount of a variable ATT 112 and a control voltage.

FIG. 7 is a block diagram showing a configuration of a conventional example.

[Explanation of symbols]

100a, 100b transmitter 101 Baseband signal generator 102a, 102b delay circuit 103a, 103b modulator 104a, 104b frequency converter 105a, 105b amplifier 106a, 106b distributor 107 delay line 108 synthesizer 109, 109a, 109b Detector 110 Delay amount control circuit 111 Variable ATT control circuit 112 Variable attenuator 113 attenuator 114 comparator

Claims (6)

[Claims] 1. A baseband signal generator for outputting a baseband signal, a variable delay circuit capable of inputting the baseband signal, adjusting a delay amount thereof and outputting the same, and a baseband via the variable delay circuit. A first modulator that generates a modulated wave by a signal; a first frequency converter that converts the modulated wave into an RF signal; a first amplifier that amplifies the RF signal to a predetermined power; A first transmitter having a first distributor for distributing the RF signal to an external transmission output signal and a delay correction signal; a delay circuit for inputting the baseband signal and giving a predetermined delay; A second modulator that generates a modulated wave by the baseband signal that has passed through the delay circuit, a second frequency converter that converts the modulated wave into an RF signal, and a second amplifier that amplifies the RF signal to a predetermined power. Amplifier and the amplified RF was
A second transmitter having a second distributor that distributes the signal to an external transmission output signal and a delay correction signal; and a predetermined value for the delay correction signal output from the second distributor. A delay line that gives a delay, a combination that combines the delay correction signal output from the first distributor and the delay correction signal output from the second distributor delayed by the delay line , A detector for detecting the output of the combiner, the baseband signal, and the detection output of the detector, and a delay amount for outputting a control signal for controlling the delay amount of the variable delay circuit. A transmission diversity delay correction method comprising: a control circuit. 2. The first and second transmitters are WC
2. The transmission diversity delay according to claim 1, wherein transmission is performed by a DMA modulation method, and the delay circuit and the variable delay circuit are set in a state of delaying the baseband signal by 1/2 chip in advance. Correction method. 3. An attenuator having a constant attenuation amount inserted between the first distributor and the combiner, and a variable attenuator inserted between the second distributor and the delay line. The baseband signals input to the first and second transmission units are input, and the R output from the first and second transmission units input to the combiner by comparing the input signal levels.
3. The transmission diversity delay correction according to claim 1, further comprising a variable ATT control circuit that outputs a control voltage that controls the amount of attenuation of the variable attenuator so that the F signal has the same power. method. 4. A variable attenuator inserted between the first distributor and the combiner, and an attenuator having a constant attenuation amount inserted between the second distributor and the delay line. The baseband signals input to the first and second transmission units are input, and the R output from the first and second transmission units input to the combiner by comparing the input signal levels.
3. The transmission diversity delay correction according to claim 1, further comprising a variable ATT control circuit that outputs a control voltage that controls the amount of attenuation of the variable attenuator so that the F signal has the same power. method. 5. The delay line is the RF of the second transmitter.
The transmission diversity delay correction method according to any one of claims 1 to 4, wherein the phase of the signal is inverted. 6. The delay amount control circuit controls the delay amount of the variable delay circuit according to whether the difference between the baseband signal level and the output level of the detector is above or below a predetermined level. The transmission diversity delay correction method according to any one of claims 1 to 5.