JP2002142230A - Photosensor signal processing for projection-type video display - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a projection-type display having a focusing measuring device that forms an output signal for measurement by amplifying a sensor current signal and an interference voltage signal, allowing the output signal to have a sensor signal component and an interference signal component, and allows the amplitude of the sensor signal component to be much larger than that of the interference signal component. SOLUTION: A processor for photosensor signals in the projection-type display device includes photosensors (S1-S8) for generating a sensor signal (Iill) having a current component for indicating a projection raster is being applied. The sensor signal includes a crosstalk voltage component (Vinf) related to scanning. In response to the sensor signal (Iill), a differential amplifier (U280A) generates an output signal (Vs). The sensor current component is converted to an amplified sensor voltage component, and the crosstalk voltage component is differentially amplified. The amplitude of the sensor voltage component is larger than that of the crosstalk voltage component in the output signal.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a video projection type (pro
jection) In the field of displays, in particular,
rojected image) and unwanted interference signals (unwanted
It relates to processing of a photo generated signal generated when an interfering signal is present.

[0002]

2. Description of the Related Art A projection type video display has a geometric raster distortion due to a physical arrangement of a cathode ray tube. Such raster distortions are further exacerbated by the use of a cathode ray tube with a unique optical projection path magnification having a curved concave illuminant surface. The projected image is
It consists of three scanning rasters that need to be placed on the display screen on average.
In order to accurately overwrite these three projected images, it is necessary to adjust the multiplexed waveform to compensate for geometric distortion and to facilitate the superposition of the three projected images. Is labor intensive,
Also, it is not possible to hand over to the user without using highly complex test equipment. An automated convergence system that simplifies alignment during manufacture and facilitates adjustments at the user's site
Sets raster edges around the display screen location to determine raster size and convergence.
edge) A measuring device can be used. However, such self-focusing systems may encounter problems when operating in the presence of interfering signals, eg, frequencies associated with scanning, having high frequency energy. Usually, the malfunction is when the interference signal is a light-generated calibration marker signal (pho
to generated calibration maker signal) which is processed as M and detected. Such erroneous photosensor signals can cause the autofocus system to fail. Generally, such unnecessary interference signals are projected marker images (proje
a high frequency voltage source that is capacitively coupled to low signal level circuitry related to the amplification and detection of the sensor signal generated by the cted maker image). Capacitive coupling signal pickup
Often originates from adjacent circuits located near the photosensor signal amplifier. Thus, the amplifier amplifies both the required sensor signal and the unwanted interference signal with a high gain, so that the amplitude of the interference signal is equal to or greater than the amplitude of the required sensor signal.
The problem of capacitively coupled crosstalk is exacerbated by the use of integrated circuits that include multiple individual operational amplifiers. These multiple amplifier sections are often used to amplify a focused signal having a very high frequency component. Thus, by using such an operational amplifier section for the high gain amplification required for the photosensor signal, for example, through the stray capacitance associated with the circuits and conductors connected to the photosensor signal amplifier, the Unwanted signals are easily combined. Therefore, at the time of automatic positioning, the amplitude of the unnecessary high-frequency component signal becomes extremely large, making it difficult to detect a marker generation sensor signal with high reliability. As a result of unreliable marker edge detection, incorrect autofocusing will be completed and a focusing error will occur. To ensure reliable marker detection,
The photosensor signal-to-interference or noise ratio must be improved. For example, by minimizing the interference signal and selectively amplifying the photosensor signal, the sensor signal-to-interference ratio can be improved.

[0003]

Accordingly, it is an object of the present invention to provide a photosensor amplifier for a projection video display.

[0004]

SUMMARY OF THE INVENTION Projection displays with a raster edge sensor are subject to interference from signals associated with scanning, including high frequency energy that impairs the sensor signal to interference ratio. A processor for the photosensor signal is included in the projection display device. The processor for a photosensor signal includes a photosensor that generates a sensor signal having a current component indicative of projected raster illumination. The sensor signal includes a crosstalk voltage component associated with the scan. A differential amplifier produces an output signal in response to the sensor signal. The sensor current component is converted to an amplified sensor voltage component, and the crosstalk voltage component is differentially amplified. The amplitude of the sensor voltage component is larger than the amplitude of the crosstalk voltage component in the output signal.

[0005] The present invention according to claim 1 is a processor for a photo sensor signal in a projection display device, wherein the processor generates a current component indicating a projected raster illumination. A photosensor for generating a sensor signal including a crosstalk voltage component related to scanning, and a differential amplifier for generating an output signal in response to the sensor signal,
The sensor current component is converted into an amplified sensor voltage component, the crosstalk voltage component is differentially amplified, and the amplitude of the sensor voltage component is larger than the amplitude of the crosstalk voltage component in the output signal. It is characterized by.

According to a second aspect of the present invention, there is provided a projection-type display device having a convergence measuring device exposed to signal crosstalk, wherein the convergence measuring device is disposed adjacent to an edge of a projection screen. A photosensor for generating a sensor current signal in response to incident illumination, an interfering voltage signal source, and an amplifier having first and second inputs arranged differentially, the sensor comprising: To amplify a current signal, only the first input is coupled to the photosensor, the first and second inputs are coupled to the interference voltage signal as a common mode signal input, and the amplifier comprises: Amplifying a sensor current signal and said interference voltage signal to form an output signal for measurement, said output signal having a sensor signal component and an interference signal component, wherein said sensor signal component Wherein the amplitude is much larger than the amplitude of the interference signal component.

[0007]

Embodiments of the present invention will be described below in detail with reference to the drawings.

FIG. 1 is a front view of a video projection display device. The projection display includes a plurality of cathode ray tubes having a raster scan image projected on a screen 700. A cabinet supports and surrounds screen 700, providing an image display area 800 that is slightly smaller than the screen. Screen 700 is shown in dashed lines to show an edge area, indicated by area OS, hidden in cabinet C.
The edge region can be illuminated with a raster scanned image when operating in an overscan mode. The photo sensor has an image display area 800
In the concealed edge area outside of the screen 700, but adjacent to the periphery of the screen 700, but projecting a raster scan image, the surface not suspended on the screen, i. It is also possible to generate an image display on a surface that is concealed. This image display method is known as a front projection display. For front projection devices,
The photosensors are located as described above, but are located adjacent to the perimeter of the screen and not obscured.
The automatic focus correction system (automatic conve
The operation of the rgence correction system) is equally applicable to front display projection or back display projection.

FIG. 1 shows eight sensors located at the four corners of the screen edge and at the center of each edge. With the sensors located at these locations, an electronically generated test pattern, such as a peak video value block M, is measured and the image width, height, and certain geometric errors (eg, rotational errors) are measured. , Bow error, trapezoidal error, pincushion error, etc.), whereby the displayed images are aligned so that they are superimposed on average over the entire screen area. Measurements are performed on each of the three projected color images in both the horizontal and vertical directions, so that at least 48 measurements are obtained.

The operation of the measurement system and the positioning system will be described with reference to FIG. FIG. 2 is a block diagram showing a portion of a raster-scan video projection display. Referring to FIG. 2, a raster scan monochromatic image is formed by three cathode ray tubes of R, G and B, and the formed raster scan monochromatic image is guided through a separate lens system and focused, and a single image is formed on a screen 700. A display image 800 is formed. Each CRT shown in FIG. 2 has four coil sets that provide horizontal deflection, vertical deflection, horizontal focusing and vertical focusing. The horizontal deflection coil set is driven by a horizontal deflection amplifier 600, and the vertical deflection coil set is driven by a vertical deflection amplifier 650. Both the horizontal deflection amplifier and the vertical deflection amplifier are controlled in amplitude and waveform via a data bus 951, and are driven using a signal source selected for display and a synchronized deflection waveform signal. Exemplary green channel horizontal focusing coil 6
15 and the vertical focusing coil 665 are converged by the amplifiers 610 and 660, respectively.
orrection waveform) signal. The convergence correction waveform signals GHC and GVC can be considered to represent DC and AC convergence signals, such as static and dynamic convergence signals, but for example, all measurement positions By correcting the address by the same value or offset and moving the entire raster to achieve a distinct static focusing or centering effect, the properties of these functions can be enhanced. Similarly, by correcting the position address of a particular measurement position, a dynamic focusing effect can be provided. The correction waveform signals GHC and GVC for the green channel are generated by exemplary digital / analog converters 311 and 312 that convert digital values read from memory 550.

An input display signal selector selects two signal sources IP1 and IP2, eg, a broadcast video signal and a display signal generated by an SVGA computer, via a bus 951. The video display signals RGB are derived from a display video selector and are electronically generated message information such as user control information, display setup and alignment signals, and a controller 301 connected via buses 302 and 951; Messages generated in response to commands from 900 and 950 can be combined by on-screen display generator 500. During auto-sensitivity calibration or focus alignment, the controller 900 commands the controller 301 via the data bus 302 to include an exemplary green channel including an exemplary black level signal having a rectangular block M having a predetermined video amplitude value. It commands the video generator 310 to generate a calibration video test signal AV. Controllers 900 and 301 may also determine the horizontal and vertical timing for positioning block M within the scan display raster, or by moving the scan raster or a portion of the scan raster including marker block M. Block M is positioned and illuminates exemplary sensor S1. Green channel test signal AV
Is output from IC 300 and is combined at amplifier 510 with the green channel output signal from on-screen display generator 500. Therefore, the output signal of amplifier 510 is
A display source video and / or OSD generation signal and / or IC 30 coupled to an exemplary green cathode ray tube GCRT
0 generated calibration video test signal
eo test signal) AV.

The controller 301 further executes programs stored in the program memory 305 and containing various algorithms. To facilitate initial setup adjustments, controller 301 outputs digital word D on data bus 303 coupled to controllable current source 250. Digital word D is current source 250
Represents the sensor-specific current supplied to the sensor detector 275.

As previously described, a setup block M has been generated and coupled to the exemplary green CRT to facilitate adjustment and registration of the three color images. A test pattern block M proximate to sensor S1 is shown in FIG. 1 and each sensor is identified by a timed marker block in the video signal projected on the overscan raster, as described above. Or
The irradiation can be performed by positioning the scanning raster so that the marker block M irradiates the sensor S1. By using a particular display signal input, such as a computer display format signal, substantially all of the scanned area can be used for signal display, thus greatly eliminating overscan raster operation. During operation with a computer display format signal, raster overscan is limited to a nominal few percent, eg, 1%.
Therefore, under such substantially zero overscan conditions, the exemplary sensor S1 can be irradiated by adjusting the raster position of the block M. Obviously, by combining the timing of the video signal with the raster position adjustment, ie temporary raster enlargement, the individual sensors can be easily illuminated.

Each sensor generates a stream of electrons and allows conduction that is substantially linearly proportional to the illuminance incident on the sensor, but may be individual for various reasons, such as differences in the brightness of the light emitters of the individual CRTs. The illuminance of these sensors is greatly different, and a lens is provided for that. Therefore, there is an optical path difference between the three monochrome images. Each CRT
The luminance of the light-emitting body deteriorates due to aging, and dust accumulates in the optical projection path over time, and the illuminance of the sensor decreases. Further, the illuminance of the sensor also varies due to the variability of the sensor current source due to sensitivity variations between individual sensors and their inherent spectral sensitivities.

Referring to FIG. 2, video generator 310
Generates an exemplary green video block M having an initial non-peak video value and positioned on a substantially black or black level background, under instructions from the control logic 301. Simultaneously generated and superimposed on the screen to produce a white image block on a substantially black background,
Similar video blocks with non-peak video values can be generated in each color channel. Therefore, the exemplary green block M
Is generated and coupled to the green CRT via the amplifier 510. The video generator 310 is controlled by the microcontroller 301 and controls the green blocks M in horizontal and vertical screen positions so that a particular sensor, for example sensor S1, is illuminated by green light from block M
Has been generated. When the sensor is illuminated, the photogenerated current (pho
to generated current) flows. As described below, the photogenerated current is processed by the amplifier U280 to generate a pulse Isen, as shown in FIG.

As already explained, significantly different photogenerating sensor currents are advantageously compensated, calibrated and measured by the control loop 100 shown in FIG. A sensor processor is shown in a circuit block 200, and the details thereof are shown in FIG. Briefly, a reference current Iref is generated by a digitally controlled current source, and when the sensor is not illuminated, the reference current Iref is used as a current Isw as the sensor detector 27.
5 is supplied. The current Isw biases the detector 275 so that the output state is “low”. The output state "low" has been selected to represent the unlit state of the sensor. For example, when the sensors S1 to S8 are illuminated, the photogenerated charge is processed and a negative-going pulse Isen is formed at the output of the amplifier 280. The negative pulse Isen shunts the constant current reference Iref to reduce the switch current Isw and turn off the sensor detector 275. When the detector 275 is pulsed off, its output goes to the "high" state, which is the nominal power supply voltage potential, selected to indicate that the sensor is lit. The output of sensor detector 275 is a positive rising pulse signal 202, which is coupled to the input of digital focusing IC 300. The rising edge of the pulse signal 202 is sampled, stopping the horizontal and vertical rate counters. This provides a count for determining the location of the illuminated sensor in the measurement matrix.

The sensor current is advantageously measured by the controllable increased reference current Iref until the sensor detector 275 switches and indicates a loss of sensor illumination. The reference current value at which the detector 275 reaches the sensor irradiation loss indicates the irradiation level incident on the sensor. This current can thus be processed and stored as a sensor and color specific threshold. The stored reference current value varies from sensor to sensor and from color to color, but the switching of the detector is の of the measured Isen switching value.
Are equalized so that switching occurs at the irradiation value of.

The details of the sensor processing block 200 shown in FIG. 2 are shown in FIG. 3B, and include a digital control current source 250, a sensor detector 275, and a photosensor amplifier 280. Digital control current source 250 generates a controlled current Iref having a magnitude determined by digital control word D. Data word D is generated by controller 301 and includes eight parallel data signals D0-D7, each representing the lowest to highest order. The individual data bits are connected in series by resistors R1, R3, R5, R7, R10, R13, R16.
And a corresponding PNP transistor Q via R19
1, Q2, Q3, Q4, Q5, Q6, Q7 and Q8, respectively. The emitter of each transistor is coupled to a positive power supply + V, and each collector is coupled via various resistors to the emitter of PNP current source transistor Q9. Thus, the current from transistor Q9 as a current source is controlled by emitter resistor R22 and a digitally selected resistor network coupled in parallel. Collector resistances of current switching transistors R2, R4, R6, R8 and R9, R11 and R1
2, R14 and R15, R17 and R18, R20 and R21
Are selected to have increasing resistance in a binary sequence. For example, the parallel coupling resistances R20 and R21 are about 400Ω, and the resistance couplings R17 and R18 are about 800Ω.
Ω. Therefore, the digital words D0-D7 are 200Ω when all transistors are turned on.
The resistance value can be selected between 100 kΩ by the resistor R22 when all the transistors are turned off. Digital words D0-D7 are zero and 3.
It has a voltage value of 3V and the voltage of the data bit is 0V
At this time, the resistor can be selected. When the voltage of the data bit is 3.3 V, the resistor cannot be selected. Therefore, resistance R22 and the base potential of transistor Q9 determine the magnitude of reference current Iref generated at the collector of transistor Q9.

The digitally determined current Iref is
Coupled to the base of transistor Q10 via resistor R26, turning on transistor Q10. The emitter of transistor Q10 is grounded and the collector is N
It is connected to the emitter of the PN transistor Q11 to form a cascaded amplifier. The base of transistor Q11 is biased by a voltage divider formed by resistors R24 and R23. The resistor R24 is connected to a positive power supply, and R23 is grounded. When the base / emitter junction of transistor Q11 is non-conductive, the junction of resistors R23 and R24 biases the bases of transistors Q9 and Q11 to about 1.65V. Transistor Q
11 collectors are digital focusing integrated circuit IC300,
For example, an output signal 202 representing the illuminated state of sensor S1, ie, the illuminated or unilluminated state, for coupling to an input of a type STV2050 or a microprocessor.
Has been generated.

The sensor detector 275 shown in FIG. 3B operates as follows. The reference current Iref is equal to the switch current I.
Although coupled to the base of transistor Q10 as sw, for example, when sensors S1-S8 are illuminated by marker block M, they are shunted through resistors R27, R28 and capacitors C4, C3 to form sensor current Isen. The transistor Q1 is switched by the switch current Isw.
0 turns on and saturates, forcing the collector of transistor 10 to a nominal ground potential Vcesat of about 50 mV. Therefore, the emitter of the transistor Q11 is
Nominally grounded through the saturated collector / emitter junction of transistor Q10, transistor Q11 is turned on and the collector of transistor Q11 is at a nominal 100 mV or (Q3Vcesat + Q4Vcesat) potential. The collector of the transistor Q11 is the output signal 20.
2, when the output signal 202 is nominally 0V, indicating that the sensor is not illuminated, and when the output signal 202 is at the nominal power supply voltage, it indicates that the sensor is illuminated.

When the transistor Q10 saturates, the emitter / base potential of the transistor Q11 changes to the resistance voltage divider R
23 and R24 reduce the nominal 1.65V to a voltage of about 0.7V formed by the base / emitter junction voltage of transistor Q11 and the saturation voltage of transistor Q10. Since the bases of current source transistor Q9 and cascade transistor Q11 are coupled, the base bias of current source transistor Q9 is also nominally 0.7V
To decrease. Due to the change in the base potential of the transistor Q9, the constant current Iref increases about three times.

The operation of the photosensor amplifier block 280 will be described later. For example, when the sensor S1 is illuminated by the projection marker block, the amplifier block 28
As a result of the advantageous amplitude and frequency response processing by zero, a negative-going current pulse Isen is formed.
Since the reference current Iref is constant, it is irradiated (lit).
The sensor pulse current Isen is shunted from the base current (Isw) of the transistor Q10 to turn off the transistor Q10. When the transistor Q10 turns off,
Transistor Q11 turns off and transistor Q1
The collector voltage of 1 rises to the power supply voltage to produce an output signal 202 with a nominal 3.3V amplitude, indicating that the sensor is illuminated. As previously described, when transistors Q10 and Q11 are turned off, the base bias of current source transistor Q9 causes the resistance divider (R2
3 and R24), and as a result, the magnitude of the constant current Iref is reduced by about 66%.
Thus, by reducing the reference current Iref, the detection of the illuminated sensor is advantageously maintained by terminating the detection and establishing a smaller switching threshold for representing the off or unlit state of the sensor. , That is, latched.

The operation of the photosensor amplifier block 280 is as follows. As already explained, the photosensors S1 to S8 are arranged around the periphery of the display screen 700 and can be connected in a parallel arrangement to a single amplifier, for example U280, or individually to the corresponding amplifier. However, the choice of connecting the sensors in parallel or individually is not critical with respect to the degradation of the signal-to-noise ratio of the photosensor signal.

The ambient illumination of the display screen and photosensor is due to sunlight and the product of incandescent or fluorescent lights. Typically, ambient lighting produces a slowly changing low frequency waveform signal that is representative of intermittent sunlight and / or artificial lighting that is poured onto the projection screen and sensors. Due to the presence of such ambient light, the resulting photosensor signal contains a low frequency component in addition to the variable amplitude DC component. The presence of artificial lighting creates a broadband noise spectrum associated with power line frequencies spanning the megahertz frequency range. On the other hand, the associated low frequency fluctuation
It appears that sunlight components that can cause loss or degradation of the required sensor signal generated by the projection measurement marker M can be easily removed. FIG. 4 (A)
Simulates a sensor signal that is exposed during measurement of the projection marker M to sunlight having shadows and unnecessary illumination by artificial illumination. The waveform selected to simulate shaded or intermittent sunlight is a triangular wave with a peak-to-peak amplitude of 3 mA and a frequency of about 2 Hz. A higher frequency noise component indicated by cross hatching is superimposed on the triangular wave. FIG. 4 shows necessary sensor signals corresponding to the projection marker M generated by the CRT.
It is shown in (B). The period of the signal derived from the simulated marker is selected to be 4 ms in order to facilitate the measurement of four markers for each display area. The sensor signal derived from the simulated marker is
The peak amplitude is 50 μA, the rise time is about 50 μs, and the delay time is nominally 1 ms. Thus, it can be seen that the required signal amplitude and the unwanted signal amplitude are rather reversed and the ratio of unwanted signal amplitude to required signal amplitude is about 60: 1.

The sensor signal input to the amplifier U280 includes not only necessary and unnecessary signal components but also other unrelated signals. The amplitude of the unwanted signal component greatly obscures the intermittent blinking of the projection measurement block M. As already explained, the slowly changing low frequency signal is due to the ambiguity of various ambient light sources, such as changes in cloud layers, shrubs or tree movements, or changes in human shadows. Typically, broadband noise is emitted from artificial light sources or sunlight.

Therefore, while recognizing that the ratio of the required signal amplitude to the unnecessary signal amplitude is about 60: 1, the signal of the photo sensor is coupled to the amplifier block 280, and unnecessary signal components are obtained by signal processing. Substantially removed. Block 280 includes eight photo sensors S1 to S8.
Are shown, the respective emitters connected in parallel are coupled via a low-pass filter and the operational amplifier U2
80, for example, at the low impedance node formed at the input terminal of type TL082. FIG.
(B) includes an interfering voltage source
The stray capacitance or the parasitic capacitance Cs connected in series with Vinf is shown. The source of the interference signal is shown at the emitter junction of the sensor, but the parasitic capacitance and the coupled interference signal are spread across the sensor interconnect. The low-pass filter is formed by a ferrite inductor FB1 connected in series and a grounded capacitor C1. Stray capacitance Cs and capacitor C
Depending on the ratio of the value of one, the operation of the amplifier U280 may be simulated or cause damage to components.
For example, coupling voltages or induced voltages Vinf caused by radio frequency interference, scanning frequency signals, or high voltage kinescope arc components are significantly attenuated.

When any arbitrary photosensor is illuminated, a photogenerated current, eg, I
ill flows from ground to the low pass filter via the collector / emitter junction of the illuminated photosensor transistor. The sensor signal current that has passed through the low-pass filter is applied to the inverting input terminal of the operational amplifier U280, and is converted to a low impedance voltage at the output terminal. A feedback resistor R29 is connected from the output terminal of the amplifier to the inverting input terminal, and generates an output voltage proportional to the photosensor input current. The non-inverting input of the amplifier is a voltage source, for example, -1.
It is connected to -0.6V generated by a potential divider formed by resistors R30 and R31 connected between a 2V power supply and 0V or ground potential. The gain of the amplifier U280 for the sensor current is large and is determined by the feedback resistor R29 and the capacitor C2 connected in parallel with the resistor R29. This gain of the amplifier causes the voltage at the inverting input to be the voltage at the non-inverting input, for example -0.6.
Forced to be very equal to V. Therefore, the voltage of the inverting input terminal is applied to the photo sensors S1 to S8,
A constant voltage is applied between both ends of the collector / emitter junction. A low impedance voltage version of the sensor signal is formed at the output of amplifier U280.
The low impedance voltage version is DC-coupled, and the more negatively illuminated sensors and the greater the sensor current, the greater the negative amplitude. Since amplifier U280 is supplied with a negative power supply voltage having a relatively large amplitude, a large negative signal voltage due to a large light generation current generated by a high level of ambient light is allowed with respect to amplifier headroom, that is, fluctuation of an output signal. are doing. The resistance value of the feedback resistor R29 is determined so that a current pulse of, for example, 50 mA drawn from the marker can be resolved by the subsequent detector 275, and a current of, for example, 3 mA related to ambient light is linearly amplified. This avoids losses associated with amplifier overload, feedback loop control and required signal components. Feedback resistor R29 is connected in parallel with capacitor C2 and provides a frequency selective feedback that limits the amplifier high frequency response of amplifier U280 to a cutoff frequency of about 58 KHz. This high frequency feedback advantageously reduces the bandwidth of the amplifier, thereby minimizing unwanted noise and extraneous signals picked up in the sensor signal. The output of amplifier U280 is shown in FIG. 4 (C), where the required marker signal pulses are indicated by small sawtooths.

The output of the amplifier U280 is connected to the capacitor C3.
Is AC-coupled to a grounded load resistor R28. Capacitor C3 and resistor R28 form the first section of the high pass filter. The junction between the capacitor C3 and the resistor R28 is connected to the capacitor C4. Capacitor C4 is connected in series with resistor R27 to form a second high-pass filter section. The first filter section removes the DC component of the ambient light signal and, as a result of the low cut-off frequency of about 60 Hz, significantly reduces the slowly changing signal components associated with the fluctuating shadow illumination of the display screen. However, positive or negative pulses, for example due to the required marker blinking, are coupled to the second filter stage. The negative-going light-generating voltage peak bounces off the exit pupil of the lens when entering the field of view of each sensor location, due to a marker block M that can be considered to blink as a result of periodic scanning of a narrow area of the illuminant. Things. Such measurement markers blink at a nominal 60 Hz rate period, but have rapid rise and fall times that are much shorter than the 60 Hz rate period. The time constant of the first high-pass filter stage is selected to eliminate or significantly reduce the effect of the current charging and discharging capacitor C3 as a result of the slowly changing ambient light level, thereby detecting Overload of the container 275 is avoided. In summary, feedback amplifier U280 and the output high-pass filter arrangement provide a band-pass filter characteristic with a low frequency cutoff of about 60 Hz and a high frequency limit of about 60 KHz.

An amplitude frequency response plot is shown in FIG. Curve A shows the feedback amplifier U2 measured at the second filter section between capacitor C4 and resistor R27 when a sensor current pulse of 50 μA is applied to the inverting input.
80 represents the photosensor signal response. A curve A shown in FIG. 6 indicates a feedback amplifier U2 when an interference signal having an amplitude of 1 V is coupled to an inverting input terminal via a 10 pF capacitor.
80 responses.

The output of the capacitor C3 is amplified and
The signal that has passed the bandpass filter forms a negative-going voltage pulse across resistor R28. These voltage pulses are AC-coupled via a capacitor C4 and are converted into current pulses by a resistor R27. FIG. 4 (D)
Shows these necessary voltage pulses at the junction of the capacitor C4 and the resistor R27. Capacitor C
4 and resistor R7 are connected in series and form a second section of the high-pass filter stage. Capacitor C4
Blocks the DC current Iref and is charged to the base potential of the detector transistor Q10. Both positive and negative impulses present in the filtered sensor signal are coupled to the base of transistor Q10. The positive impulse is applied to the transistor Q10 via the resistor R26.
Clamped by the base / emitter junction of
A negative-going current pulse drawn from the marker illumination of the sensor shunts the current from the constant current Iref, turning off transistor Q10. As described above, when the transistor Q10 is turned off, the collector of the transistor Q11 becomes a value of logic "1", and FIG.
An output signal 202 having a voltage value of 3.3 V, which represents the marker illumination of the sensor, is formed as shown in (E). Thus, an amplifier having bandpass frequency characteristics according to the present embodiment substantially eliminates unwanted ambient light components from the photosensor signal, thereby automatically setting up while the screen is being illuminated by the ambient light. be able to.

The circuit shown in FIG. 3C includes a sensor signal amplifier U280A with an exemplary crosstalk mechanism Css.
The high frequency interference signal Vhf coupled to is shown. When this crosstalk component is amplified, the signal-to-noise ratio of the sensor is reduced, and the convergence detection in the subsequent circuit becomes spurious. According to the arrangement of this embodiment, this crosstalk signal is substantially removed by coupling to the differential input of amplifier U280A and utilizing the common mode rejection of the operational amplifier. In FIG. 3C, FIG.
The same component names as in (B) are used, and the new components and values are indicated by three-digit numbers. The common mode input connection is realized by, for example, a 20Ω resistor R320 connected between the differential inputs of the amplifier U280A. Bias voltage divider resistors R300 and R310
Is increased by a factor of 2 with respect to the value shown in FIG. The operation of the device according to the present embodiment is as follows.

FIG. 3C shows, via an exemplary stray capacitance Css, for example, an IC package, for example an amplifier U28.
Shown is a high frequency interference crosstalk signal Vhf coupled between adjacent terminals of another amplifier section (not shown) of type TL082 including OA. Alternatively, between adjacent circuit board conductors, or between photosensor amplifiers U
Crosstalk can occur between circuits coupled to the inverting input of 280A. Resistor R320 is advantageously arranged to couple a majority of the interfering signal to the non-inverting input of amplifier U280A to form a common mode input signal. When substantially the same signal is applied to both inputs, the crosstalk component Vx caused by the signal Vhf
Is generated, the amplitude of which is greatly reduced.
However, the feedback surrounding amplifier U280A attempts to maintain the two inputs at the same potential, since resistor R320 forms part of the attenuator at the non-inverting input.
The input signal will definitely be different. Due to this difference,
The negative feedback signal at the inverting input is partially coupled to the non-inverting input to form a positive feedback that provides a signal peaking effect.

Amplifier U for crosstalk signal Vhf
The signal gain of 280A is 1-2 for an interference signal in the 30 KHz range and is divided by a capacitive voltage divider formed by capacitors Css and C1 having the exemplary values shown in FIG. This gain value is much less than the open loop gain level of the amplification provided by the capacitive voltage divider formed by the capacitors Cs and C1 dividing the interference signal Vinf in the circuit arrangement of FIG. The value of the coupling resistor R320 is determined by the amplifier U280.
A is selected based on the input voltage offset specification of A. The offset voltage of amplifier U280A is divided by resistors R300 and R31 divided by common mode coupling resistor R320.
Zero parallel resistance [(R300 // R310) / R320]
Therefore, the signal is amplified by the ratio of the attenuator formed at the non-inverting input terminal. For example, for the resistance values shown in FIG. 3C, the ratio is about 70: 1, so for an exemplary input offset voltage of +/− 5 mV, the operational amplifier will increase the exemplary offset signal by a factor of 70. Amplify, about +/- 350 m at the inverting input of amplifier U280A
V. Photo sensors S1 to S
It is important to maintain the bias voltage between 0.5 and 3 V between 0.5V and 3V. This bias voltage is formed at the inverting input as a result of the feedback action of the operational amplifier trying to maintain the two inputs at the same potential. Thus, the voltage at the non-inverting input is tracked by the inverting input. The fluctuation of the amplifier output voltage due to the offset is larger than the fluctuation of the non-inverting input terminal as a result of the attenuation described above. However, since the capacitor C3 of the low-pass filter blocks the DC component at the output terminal of the amplifier, this amplification is performed. The offset voltage is not important. A nominal -0.8 V photosensor bias is formed by the potential divider resistors R300 and R310. This bias value provides enough headroom and a photosensor transistor bias of 500
It is selected to be maintained at a value greater than mV.

Negative feedback is provided by a parallel coupled resistor R29 and capacitor C2 coupled from the output of amplifier U280A to the inverting input. This feedback forces the voltage across the common mode resistor R320 to be substantially zero, thus reducing the voltage amplitude of the interference signal Vinf as well. By feedback, practically 0V
Is generated across the common mode resistor R320, the sensor current Iill flowing through the resistor R320 is substantially blocked, effectively flowing through the feedback resistor R29 and the sensor signal voltage at the output of the amplifier U280A. Vs
Is generated.

An amplitude frequency response plot is shown in FIG. Curve B represents the photosensor signal response of feedback amplifier U280A measured at the second filter section between capacitor C4 and resistor R27 when a sensor current pulse of 50 μA is input. Curve B shown in FIG.
The response of the feedback amplifier U280A when the interference signal having the amplitude of 1 V is coupled to the inverting input terminal via the capacitance of 10 pF is shown.

Examining curves A in FIGS. 5 and 6, respectively, the processor of circuit 280 converts the sensor signal to about 2:
It can be seen that the interference ratio is 1 or 6 dB. Therefore, the circuit 280 shown in FIG. 3B certainly suppresses the sensor response to ambient light and the pickup of the sensor harness significantly.
The minimum sensor signal-to-interference ratio compromises the ability to provide reliable projection marker detection, as shown by curve A in FIG. The processing arrangement according to the present invention of circuit 280A utilizes a common mode input to prevent interfering signal pickup, and furthermore, coupling resistor R320 provides a plot of the required and unwanted signals shown in curves B of FIGS. As shown, it advantageously provides feedback that peaks the amplitude frequency response. Comparing curve B, the high frequency response of the bandpass processor has been significantly reduced from about 60 KHz to about 8 KHz, and the interfering signals associated with the scan are located outside the passband of the processing array of circuit 280A. I understand. Resistor R320 not only allows for common mode input coupling, but also provides positive feedback from the output of the amplifier to the non-inverting input via resistor R29. This positive feedback results in a resonance occurring at about 7 KHz, a peaking effect in the bandpass frequency range, which increases the required signal by about 2.5 times compared to circuit 280. Curve B in FIG. 5 shows that a 50 μA sensor input signal is advantageously converted to an output signal with an amplitude of about 53 mV. For the interfering signal, the resulting output voltage amplitude is reduced to about 1/3 or 3 mV when compared to the performance of circuit 280. The reduction in processor bandwidth and the introduction of passband peaking has advantageously improved the required to unwanted signal ratio. FIG.
Comparing curve B with FIG. 6, it can be seen that circuit 280A provides a sensor signal to interference ratio of about 16: 1, or 24 dB.

The current / current combined with common mode rejection of the interference voltage signal for the sensor current signal according to this embodiment
The combination of voltage conversion and peaking of the resulting bandpass response ensures that the optimized sensor signal to interference ratio is coupled to the detector 275. The sensor signal coupled to the detector is not substantially different from the signal described for the arrangement shown in FIG. 3 (B), but the effects of high frequency crosstalk interference are greatly reduced and excellent ambient illumination Maintain removal.

[Brief description of the drawings]

FIG. 1 is a front view of a projection type video display to which the present invention is applied.

FIG. 2 is a block diagram illustrating a video image projection display device according to an embodiment of the present invention.

FIG. 3 is a circuit diagram showing the present embodiment in more detail, wherein FIG. 3B shows a digital control current source, a sensor signal detector and a sensor signal processor, and FIG. (C) is a diagram showing another sensor signal processor.

4A and 4B are diagrams showing simulation results according to the present embodiment, wherein FIG. 4A is a diagram showing a simulation of sensor signal processing when there is interference due to ambient light, and FIG. Another diagram showing a simulation of sensor signal processing when there is interference due to ambient light. FIG. 11C is another diagram showing a simulation of sensor signal processing when interference due to ambient light exists. (D) is another diagram showing a simulation of sensor signal processing in the presence of interference by ambient light, and (E) of this figure is another diagram showing a simulation of sensor signal processing in the presence of interference by ambient light. It is.

FIG. 5 shows processors 280 and 2 according to the present embodiment.
FIG. 9 is a diagram showing a simulation of amplitude versus frequency response at 80 A for an input current of 50 μA.

FIG. 6 shows processors 280 and 2 according to the present embodiment.
FIG. 10 is a diagram illustrating a simulation of amplitude versus frequency response when the amplitude of an input interference signal is 1 V for 80A.

[Explanation of symbols]

AV Green channel calibration video test signal Cs Stray capacitance (parasitic capacitance) Css Crosstalk mechanism (stray capacitance) FB1 Ferrite inductor GCRT Green cathode ray tube Iill Light generation current Isen Negative falling pulse Iref Reference current Isw Switch current M Marker block S1 S8 Photosensor Vinf Interference voltage source (interference signal) Vhf High-frequency interference signal (crosstalk signal) Vo Amplifier output signal Vs Sensor signal voltage Vx Crosstalk component 100 Control loop 200 Circuit block (sensor processing block) 202 Positive rising pulse 250 Controllable current source (Digital control current source) 275 Sensor detector 280, U280, U280A, 510, 610, 66
0 Amplifier 300 IC 301 Controller (control logic) 302, 303, 951 Data bus 308 Program memory 310 Video generator 311, 312 Digital / analog converter 500 On-screen display generator 550 Memory 600 Horizontal deflection amplifier 615 Horizontal focusing coil 650 Vertical Deflection amplifier 665 Vertical focusing coil 700 Screen 800 Image display area 900 Focusing (CONV.) Microcontroller 950 Chassis microprocessor (μP)

──────────────────────────────────────────────────続 き Continued on the front page (51) Int.Cl. ⁷ Identification symbol FI Theme coat ゛ (Reference) H04N 9/31 H04N 9/31 A 17/04 17/04 D (71) Applicant 300000708 46, Quai A, Le Gallo F-92648 Boulogne Cedex France F term (reference) 5C060 BC05 BD02 CB00 CH19 HA18 HB19 JA00 5C061 BB01 EE03 EE13 5C082 AA03 AA34 BA27 BA34 BA41 BD01 CB01 MM10

Claims (2)

[Claims] 1. A processor for a photo sensor signal in a projection display device, the processor having a current component indicating that a projection raster has been illuminated, and a crosstalk voltage component associated with scanning. A photosensor for generating a sensor signal including the sensor signal, and a differential amplifier for generating an output signal in response to the sensor signal, wherein the sensor current component is converted into an amplified sensor voltage component, and the crosstalk voltage Wherein the amplitude of the sensor voltage component is greater than the amplitude of the crosstalk voltage component in the output signal.
Processor for photosensor signals. 2. A projection display device having a focusing measurement device exposed to signal crosstalk, said focusing measurement device being positioned adjacent an edge of a projection screen and responsive to incident illumination. A photosensor for generating a sensor current signal, an interference voltage signal source, and an amplifier having first and second inputs arranged in a differential manner. Is coupled to the photosensor, the first and second inputs are coupled to the interference voltage signal as a common mode signal input, and the amplifier amplifies the sensor current signal and the interference voltage signal Forming an output signal for measurement, the output signal having a sensor signal component and an interference signal component, wherein the amplitude of the sensor signal component is greater than the amplitude of the interference signal component. Wherein the large or, projection display apparatus provided with a focusing measurement device.