JP2000295054A - Output power control circuit and transmission circuit - Google Patents

Abstract

(57) Abstract: Provided is an output power control circuit capable of suppressing output fluctuations due to temperature fluctuations and temporal changes and controlling output power in response to traffic fluctuations. SOLUTION: A first branch element for branching an input of a controlled part to be output power controlled, a second branch element for branching an output of the controlled part to be output power controlled, A first square detection circuit that squares the output of the first branch element, a second square detection circuit that squares the output of the second branch element, and a logarithmic output of the first square detection circuit. A first logarithmic amplifier that amplifies, a second logarithmic amplifier that logarithmically amplifies the output of the second square detection circuit, a difference generation circuit that takes a difference between the outputs of the first and second logarithmic amplifiers, A control voltage generation circuit for generating a difference between the output of the difference generation circuit and the reference voltage;

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an output power control circuit, and more particularly to an output power control circuit capable of suppressing output fluctuations due to temperature fluctuations and aging, and controlling output power in response to traffic fluctuations. The present invention relates to an output power control circuit.

[0002] Since wireless communication is a method of radiating radio waves to free space and performing communication between two points or between multiple points, it is a component that inhibits communication of its own channel except for signals of its own channel. At the same time, the signal of its own channel is also a component that hinders communication with signals of other channels.

[0003] In the case of the frequency division multiplex communication system, a filter is used to separate a frequency assigned to its own channel from a frequency assigned to another channel. Assuming that the characteristics of the filter used are stable, the level of the signal component of the other channel passing through the filter that extracts the frequency assigned to its own channel is the carrier frequency of the other channel. Accuracy and output power.

Therefore, in the case of frequency division multiplex communication, stabilization of the carrier frequency and stabilization of the output power are essential requirements for maintaining high communication quality.

[0005] By the way, although relatively recent in the history of wireless communication, the frequency component of a transmission signal is spread over a wide frequency band by a pseudo-random code called a spread code to reduce the power per unit frequency band to a very small level. Code division multiplexing (Code Division) that performs multiplex communication using a single carrier by spreading a plurality of speech signals with a spreading code orthogonal to each other (meaning that the correlation is low). Multiple Access; C for short
It is called DMA. ) Method communication has been put to practical use.

In the communication of the code division multiple access system, the frequency component of a transmission signal is spread over a wide frequency band by a spreading code which is originally a pseudo-random code and transmitted, and despread at the receiving side by the same spreading code as the transmitting side. It is excellent in confidentiality for restoring a transmission signal by using the code division multiple access method in the near future.

[0007] Among the received signals in this code division multiple access system, signals of a plurality of other channels that hinder communication with signals of the own channel are included in the total level more than signals of the own channel. Have been. In order to extract the transmission signal of its own channel from among them, the reception signal is despread by the same spreading code as the spreading code used on the transmission side.

At this time, since the number of bits of the spreading code allocated to each channel is finite, the correlation between each spreading code is not completely zero, and the correlation between the spreading codes of other channels is microscopic. In some cases, a correlation can be obtained between the received signal and its own spreading code. Therefore, a signal obtained by despreading includes an unnecessary component caused by a signal of another channel in addition to a transmission signal of its own channel.

This unnecessary component is an accumulation of signals obtained by despreading a signal spread by independent spreading codes of a plurality of channels with its own spreading code, and thus can be approximately treated as noise.

Here, this noise level depends on the spectrum intensity of another channel. For example, even if one of the plurality of channels is transmitting at an extremely high level, the signal-to-noise ratio is reduced for the other channels.

That is, there is a close relationship between the output power of each channel and the channel capacity in the code division multiple access system. Therefore, also in the code division multiple access system, it is important to stabilize the output power.

In order to completely stabilize the output power, not only the initial setting of the output power must be performed accurately, but also
It is important to suppress both fluctuations in output power due to temperature fluctuations and fluctuations in output power due to aging.

Further, in a code division multiple access communication system, when the number of communicating channels is small and the total transmission power of the communication system is smaller than the allowable total transmission power, the transmission power per channel is kept constant. , That is, transmission at a level proportional to the number of communication channels is also necessary.

Therefore, when viewed throughout wireless communication, there is a strong realization of an output power control circuit capable of suppressing output fluctuations due to temperature fluctuations and aging, and controlling output power in response to traffic fluctuations. Has been requested.

[0015]

2. Description of the Related Art FIG. 15 shows a conventional output power control circuit.

In FIG. 15, reference numeral 101 denotes an input amplifier, 1
02 is an output amplifier, 103 is a variable gain amplifier, and 104 is a control voltage generation circuit.

The input amplifier 101, the variable gain amplifier 103, and the output amplifier 102 form a part of a main signal section (hereinafter, referred to as a controlled section) that is controlled in output power. The gain amplifier 103 and the control voltage generation circuit 104 constitute an output power control unit.

In the configuration shown in FIG. 15, an element having a temperature / voltage characteristic such as a PIN diode or a thermistor, such as a PIN diode or a thermistor, is used for the control voltage generation circuit 104, and a voltage which can offset the temperature / gain characteristic of the controlled part is provided. The variable gain amplifier 103 is generated and supplied to the variable gain amplifier 103, and the gain of the variable gain amplifier 103 is controlled to suppress the fluctuation of the gain due to the temperature fluctuation.

FIG. 18 is a schematic configuration of a conventional transmission circuit.

In FIG. 18, reference numeral 105 denotes a baseband modulation circuit for converting an input baseband signal into a signal of a form suitable for transmission (in the figure, it is denoted as a BB modulation circuit, but is the same circuit). ), 106 is a radio frequency modulation circuit for converting the output of the baseband modulation circuit 105 to a frequency suitable for radio transmission by usually performing two-stage frequency conversion, and 107 is the output of the radio frequency modulation circuit 106 A power amplification circuit 108 amplifies the output of the power amplification circuit to a sufficient level. An antenna 108 radiates the output of the power amplification circuit to space. The output power control circuit often constitutes a part of the radio frequency modulation circuit 106.

[0021]

FIG. 16 is a diagram for explaining a problem (part 1) of the conventional technique.

In FIG. 16, the horizontal axis is temperature and the vertical axis is gain.

The gain / temperature characteristic denoted by reference character a is a gain / temperature characteristic required for the output power control circuit, which is the inverse characteristic of the temperature / gain characteristic of the controlled part, and denoted by reference character b. The gain-temperature characteristic shown in FIG.
FIG. 16 is a gain-temperature characteristic of the variable gain amplifier 103 of FIG.

The problem to be described here is that the temperature / gain characteristics generally required for the controlled part are nonlinear characteristics,
15 is that it is difficult to make the gain / temperature characteristics of the variable gain amplifier 103 of FIG. 15 controlled by the voltage supplied by the control voltage generation circuit 104 of FIG.

That is, the first problem is that, depending on the configuration of FIG. 15, the temperature compensation can be accurately performed over a specific temperature range, but the temperature compensation is not accurate over a wide temperature range. That's what it means. This is shown in FIG. 16 which is an example of temperature compensation, in which the accuracy of temperature compensation is high near normal temperature and low in the low and high temperature sides.

FIG. 17 shows a problem of the prior art (No. 2).
FIG.

In FIG. 17, the horizontal axis represents temperature, and the vertical axis represents gain.

The gain / temperature characteristics denoted by reference numerals a-1 to a-3 indicate the gain / temperature characteristics required of the output power control circuit when there is an individual difference in the temperature characteristics of the controlled part. The gain / temperature characteristics denoted by the reference character b are the gain / temperature characteristics of the controlled part in FIG. 15 controlled by the voltage supplied by the control voltage generation circuit 104 in FIG.

The problem to be described here is that there is generally an individual difference in the temperature / gain characteristics of the controlled part.
When the gain of the variable gain amplifier 103 is controlled by the voltage supplied by the control voltage generation unit 104, even if the temperature compensation can be relatively accurately compensated for a specific controlled unit, the temperature compensation is performed for all the controlled units. It is difficult to compensate accurately.

That is, the second problem is that it is difficult to absorb all the individual differences of the controlled parts depending on the configuration of FIG.

A third problem is that the configuration shown in FIG. 15 attempts to compensate for the temperature / gain characteristics of the controlled part by using an element having temperature characteristics in the voltage / current characteristics. If there is an individual difference in the temperature characteristics of the element to be manufactured, it is difficult to absorb the difference and compensate the temperature.

Although only the temperature compensation is described above, it is extremely difficult to compensate for a change with time in the configuration shown in FIG.

Further, the fourth problem is that, particularly in the case of the code division multiple access system, when the output power is controlled in accordance with the traffic congestion state (in accordance with the increase or decrease in the number of simultaneously communicating channels), the number of communication channels is reduced. Although the range in which the output power may be proportional to the input power is good, it is difficult to suppress the transmission power in a range where the traffic is congested and the total transmission power of the communication system exceeds the total transmission power allowed for the communication system. That is the point.

In view of the above problems, the present invention suppresses output power fluctuations due to temperature fluctuations and temporal changes, including individual differences between controlled parts and output power control parts, and responds to traffic fluctuations. It is an object to provide an output power control circuit that can control output power.

As described above, since the output power control circuit constitutes a part of the radio frequency modulation circuit 106 shown in FIG. 18, the problem of the output power control circuit immediately becomes the problem of the transmission circuit.

Accordingly, an object of the present invention is to provide a transmission circuit which can solve the conventional problems by applying an improved output power control circuit.

[0037]

The first principle of the present invention is as follows.
A voltage obtained by logarithmic amplification of an output obtained by square detection of a signal branched at an output terminal of a controlled unit which is an output power control target, and a signal obtained by square detection of a signal branched at an input terminal of the controlled unit. A technique of generating a voltage difference between a voltage obtained by logarithmically amplifying an output and supplying a difference between the difference voltage and a voltage taken by the difference voltage in a reference state as a voltage for controlling the gain of the controlled part. is there. Note that the reference state refers to a state where the temperature is the reference temperature and does not change with time.

According to the first principle of the present invention, the voltage obtained by logarithmically amplifying the output obtained by square-detecting the signal branched at the output terminal of the controlled unit is dBm (the absolute value of the signal power in which 1 mW is 0 dBm). It is a linear function of the input power of the controlled part expressed in the unit. Also, the voltage obtained by logarithmically amplifying the output obtained by squaring detection of the signal branched at the input terminal of the controlled unit is a linear function of the input power expressed in dBm. It becomes a linear function of the same slope as the slope of the linear function relating the voltage obtained by logarithmic amplification of the output obtained by square detection of the branched signal and the input power expressed in dBm.

Therefore, the voltage of the difference is a voltage irrelevant to the input power.

Therefore, if the difference between the voltage of the difference and the voltage obtained by the voltage of the difference in the reference state is supplied as a voltage for controlling the gain of the controlled unit and negative feedback is applied, the voltage of the controlled unit is controlled. It is possible to suppress the temperature dependence of the gain and the change with time.

A second principle of the present invention is that a voltage obtained by logarithmically amplifying a signal branched at an output terminal of a controlled unit whose output power is to be controlled and a signal branched at an input terminal of the controlled unit. Generate a voltage difference between the voltage obtained by logarithmically amplifying and the voltage of the difference, and the voltage of the difference between the voltage of the difference voltage in the reference state as a voltage for controlling the gain of the controlled part. Technology to supply.

According to the second principle of the present invention, the voltage obtained by logarithmically amplifying the signal branched at the output terminal of the controlled unit is also a linear function of the input power of the controlled unit expressed in dBm. . Also, the voltage obtained by logarithmically amplifying the signal branched at the input terminal of the controlled part is also a linear function of the input power expressed in dBm. It becomes a linear function of the same slope as the slope of the linear function that relates the voltage obtained by logarithmically amplifying the square-detected output and the input power expressed in dBm.

For this reason, the difference voltage is a voltage irrelevant to the input power.

Therefore, if the difference between the voltage of the difference and the voltage taken by the voltage of the difference in the reference state is supplied as a voltage for controlling the gain of the controlled unit and negative feedback is applied, the voltage of the controlled unit is reduced. It is possible to suppress the temperature dependence of the gain and the change with time.

A modification (Part 1) of the second principle of the present invention is as follows.
In the second principle of the present invention, a circuit for compensating for the input / output slope of the logarithmic amplifier is provided on the output side of at least one logarithmic amplifier, and a branch is made at the output terminal of the controlled part, which is the output power control target. The difference between the voltage obtained by logarithmically amplifying the signal and the voltage obtained by logarithmically amplifying the signal branched at the input terminal of the controlled part with respect to the input power expressed in dBm is adjusted. Generate a voltage of
This technique supplies a difference between the voltage of the difference and the voltage taken by the voltage of the difference at a reference temperature as a voltage for controlling the gain of the controlled part.

According to the modification (Part 1) of the second principle of the present invention, even if there is an individual difference between the two logarithmic amplifiers and the slope of the above-mentioned linear function is different, this is compensated for and compensated. The gain of the control unit can be controlled.

It is needless to say that the above technique can be applied to the first principle of the present invention.

A modification (part 2) of the second principle of the present invention is as follows.
In the second principle of the present invention, a circuit for compensating for the input / output inclination of the logarithmic amplifier is provided on the output side of both logarithmic amplifiers, and the output power is controlled at the output terminal of the controlled part to be controlled. The slope of the voltage obtained by logarithmically amplifying the signal and the voltage obtained by logarithmically amplifying the signal branched at the input terminal of the controlled part with respect to the input power expressed in dBm are matched, and the offset voltage of the logarithmic amplifier is adjusted. And compensating for the difference, and generating a difference voltage between the two, and supplying the difference between the difference voltage and the voltage taken by the difference voltage in a reference state as a voltage for controlling the gain of the controlled part. It is.

According to a modification (Part 2) of the second principle of the present invention, the slope and the offset of the linear function may be different due to the individual difference between the two logarithmic amplifiers. Even if there are temperature characteristics, the gain of the controlled part can be controlled after compensating for them.

It is needless to say that the above technique can be applied to the first principle of the present invention.

A modification (3) of the second principle of the present invention is as follows.
In addition to the first and second principles, receiving a voltage instructing to adjust the output power according to external conditions while detecting a change in input power, and changing the voltage taken by the difference voltage in the reference state This is a technique for performing gain control.

According to the modification (Part 3) of the second principle of the present invention, it is possible to keep the output power constant even when the input power itself of the controlled part fluctuates. The output power can be controlled even when the output power must be made variable.

In particular, in the code division multiple access system, since it is necessary to control the output power according to the congestion state of the traffic, it is important that the output power can be varied according to external conditions.

It is needless to say that the above technique can be applied to the first principle of the present invention.

[0055]

FIG. 1 shows the first principle of the present invention.

In FIG. 1, 1-1 is a first coupler, 1
-2 is a second coupler, which outputs a combined power from terminal C when signal power is applied to terminals A and B, and applies power to terminals B and C when signal power is applied to terminal A. Is a device for branching and outputting. 2-1 is a first amplifier, 2-2 is a second amplifier, 3 is a variable gain amplifier, and the first coupler 1-1 and the second coupler 1--2.
2, the first amplifier 2-1 and the second amplifier 2-2,
The variable gain amplifier 3 forms a controlled part.

Reference numeral 4-1 denotes a first square detection circuit, and reference numeral 4-2 denotes a second square detection circuit, each of which outputs an output voltage proportional to the square of the input voltage.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier, each of which is a circuit for generating an output voltage proportional to the logarithm of the input voltage.

7 is a difference generating circuit for taking the difference between the two inputs,
Reference numeral 8 denotes a control voltage generation circuit that generates a voltage for controlling the gain of the variable gain amplifier.

The first coupler 1-1, the variable gain amplifier 3, the second coupler 1-2, the first square detection circuit 4-1 and the second square detection circuit 4-2 , The first logarithmic amplifier 5-1, the second logarithmic amplifier 5-2, the difference generation circuit 7, and the control voltage generation circuit 8 constitute an output power control unit.

In the configuration of FIG. 1, the first coupler 1-
Numeral 1 divides the input power of the controlled part at a predetermined ratio and supplies it to the first square detection circuit 4-1.
Divides the output power of the controlled unit at a predetermined ratio and supplies the branched output power to the second square detection circuit 4-2. Therefore, the output voltage of the first square detection circuit 4-1 has a value proportional to the input power of the controlled unit, and the output voltage of the second square detection circuit 4-2 is proportional to the output power of the controlled unit. Value.

In other words, the output voltage of the first square detection circuit 4-1 is represented by p _IN (p which generally means power). This means that the output voltage is proportional to the input power of the square detection circuit. Note that this is just a voltage, and it is just a voltage.) If the voltage at the input terminal in the configuration of FIG. 1 is v _IN and the proportional coefficient having the 1 / voltage dimension is a _I , p _IN = A _I · (v _IN ) ² (1) holds, and the output voltage of the second square detection circuit 4-2 is defined as p _OUT (also note that this means a voltage for the above-mentioned reason. 1), and let the output voltage of the configuration of FIG. 1 be v _OUT, and let a _{O be} a proportional coefficient having a 1 / voltage dimension, the relationship of p _OUT = a _O · (v _OUT ) ² (2) To establish.

Therefore, assuming that the output voltage of the logarithmic amplifier 5-1 is V _IN and the output voltage of the logarithmic amplifier 5-2 is V _OUT , V _IN = log (a _I (v _IN ) ² ) (3) ) V _OUT = log (a _O (v _OUT ) ² ) (4)

By transforming equation (3) by the rules of logarithmic operation, V _IN = log (a _I ) + log ((v _IN ) ² ) (5) is obtained.

Here, the second term of the equation (5) is an amount proportional to the input power of the configuration of FIG. 1 expressed in dBm. Assuming that this is P _IN , V _IN = log (a _I ) + P _IN (6) That is, the output voltage V _IN of the logarithmic amplifier 5-1 is d
It is a linear function of the input power P _IN expressed in Bm.

Now, _assuming that v _out when the temperature t is t ₀ is v _to and a function representing the temperature characteristic of the gain is f (t), the relation v _OUT = v _to (1 + f (t)) (7) Holds. However, f (t ₀ ) = 0.

If the voltage gain of the controlled part at t = t ₀ is g ₀ , then v _t0 = g ₀ v _IN (8), so that v _OUT = g ₀ v _IN (1 + f (t )) (9)

By rearranging equation (9) into equation (4), That is, the output voltage V _OUT of the logarithmic amplifier 5-2 is also a linear function of the input power P _{IN in} dBm, and its slope is equal to the slope of the output voltage V _IN of the logarithmic amplifier 5-1 with respect to P _IN . Become inclined.

Taking the difference between equations (10) and (6), Get.

The first term of the equation (11) is gain information of the controlled portion having the configuration shown in FIG. 1, the second term is a constant, and only the third term is a temperature-dependent term.

This (V _OUT −V _IN ) is generated by the difference generation circuit 7 of FIG. This is supplied to the inverting input terminal of the control voltage generating circuit 8 in FIG. 1, and the above (V _OUT −V _IN ) is applied to the non-inverting input terminal of the control voltage generating circuit 8 by t = t _0.
Voltage at the time of Supplies, and supplies the output of the control voltage generating circuit 8 to the gain control terminal of the variable gain amplifier 3 of FIG. 1, (V _OUT -
If the gain of the variable gain amplifier 3 is set to a predetermined value when V _IN ) = V _REF , the third term caused by the temperature / gain characteristic of the controlled part in equation (11) is always zero.
In other words, negative feedback is applied so that f (t) = 0, and the output voltage v _OUT of the configuration of FIG. 1 does not fluctuate due to temperature fluctuation.

It is clear that the influence of the change over time can be suppressed by the configuration shown in FIG. 1 if the term indicating the change over time in equation (9) is considered at the same time. Therefore, only the influence of the temperature change is described above. . The same applies to the following.

FIG. 2 is a diagram for explaining the first principle of the present invention.

In FIG. 2, the output voltage V _IN of the logarithmic amplifier 5-1 in FIG. 1 is a linear function of the input power P _IN expressed in dBm, and the output voltage V _OUT of the logarithmic amplifier 5-2 is also dBm.
It becomes a linear function of the input power P _IN expressed in units, and its slope is the output voltage V _IN of the logarithmic amplifier 5-1 and P _IN
It shows that the slope becomes the same as that of the case of _IN , and V _REF which is the value of (V _OUT −V _IN ) with respect to the reference temperature is displayed.

FIG. 3 shows the second principle of the present invention.

In FIG. 3, 1-1 is the first coupler, 1
-2 is a second coupler. 2-1 is the first amplifier, 2
-2 is a second amplifier, 3 is a variable gain amplifier, the first coupler 1-1 and the second coupler 1-2, the first amplifier 2-1 and the second amplifier 2-2 The variable gain amplifier 3 forms a controlled unit.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

7 is a difference generation circuit for taking the difference between the two inputs,
Reference numeral 8 denotes a control voltage generation circuit that generates a voltage for controlling the gain of the variable gain amplifier.

Then, the first coupler 1-1, the variable gain amplifier 3, the second coupler 1-2, the first logarithmic amplifier 5-1 and the second logarithmic amplifier 5-2, An output power control unit is configured by the difference generation circuit 7 and the control voltage generation circuit 8.

In the configuration of FIG. 3, the first coupler 1-
1 branches the input power of the controlled unit at a predetermined ratio and supplies it to the first logarithmic amplifier 5-1, and the second coupler 1-2 branches the output power of the controlled unit at a predetermined ratio. Then, the signal is supplied to the second logarithmic amplifier 5-2. In this case as well, for the following reason, the output voltage of the first logarithmic amplifier 5-1 and the output voltage of the second logarithmic amplifier 5-2 are the primary voltages of the input power of the controlled unit in dBm. Function.

That is, the input voltage of the first logarithmic amplifier 5-1 is set to v _IN , the output voltage of the first logarithmic amplifier 5-1 is set to V _IN , and the input voltage of the second logarithmic amplifier 5-2 is set to V _IN . V _out ,
Assuming that the output voltage of the second logarithmic amplifier is V _OUT , V _IN = log (v _IN ) (13) V _OUT = log (v _out ) (14)

Here, by substituting the relationship of equation (9) into equation (14), Get.

By the way, since log (v _IN ) ² = P _IN as described immediately before the equation (6), by substituting this into the equations (13) and (15), Get.

That is, as shown in FIG. 3, the voltage obtained by logarithmic amplification of the voltage branched by the first coupler 1-1 and the voltage obtained by logarithmic amplification of the voltage branched by the second coupler 1-2 are obtained. The resulting voltage is also the input power P _IN expressed in dBm
Is a linear function of Therefore, the configuration of FIG.
The same operation as the configuration described above can be realized.

Further, the configuration of FIG. 3 has an advantage that a square detection circuit is not required.

FIG. 4 shows a modification (1) of the second principle of the present invention.

In FIG. 4, 1-1 is the first coupler, 1
-2 is a second coupler, 2-1 is a first amplifier, 2-2 is a second amplifier, 3 is a variable gain amplifier, the first coupler 1-1 and the second coupler 1-2. , The first amplifier 2-
The first and second amplifiers 2-2 and the variable gain amplifier 3 form a controlled part which is a part of the main signal part.

5-1 is a first logarithmic amplifier, 5-2 is a second logarithmic amplifier, 6-1 is a slope correction circuit, 7 is a difference generation circuit, and 8 is a control voltage generation circuit.

Then, the first coupler 1-1, the variable gain amplifier 3, the second coupler 1-2, the first logarithmic amplifier 5-1, the second logarithmic amplifier 5-2, An output power control unit is configured by the inclination correction circuit 6-1, the difference generation circuit 7, and the control voltage generation circuit 8.

The feature of the configuration shown in FIG.
1 is provided.

If there is no individual difference between the first logarithmic amplifier 4-1 and the second logarithmic amplifier 4-2, the relationship between these output voltages and the input power in dBm is expressed by the following equation (16). ), The slope will be the same linear function.

However, if there is an individual difference between the two logarithmic amplifiers, there will be a difference in the slope between the output voltage of the two logarithmic amplifiers and the input power in dBm. This difference in slope means that the offset between the output voltages of the two logarithmic amplifiers varies as the input power level varies.

Therefore, by correcting the slope between the output voltages of the two logarithmic amplifiers and obtaining the difference between the output voltages of the two logarithmic amplifiers and supplying the difference to the control voltage generator, the variation of the equivalent offset can be reduced. Thus, the gain of the controlled unit can be made independent of the input power.

This situation is shown in FIG. 5, which is a diagram for explaining a modification (part 1) of the second principle of the present invention.

In FIG. 4, the inclination correction circuit 6-
Although the example in which 1 is arranged on the output side of the logarithmic amplifier 5-2 is shown, the slope correction circuit 6-1 may be arranged on the output side of the logarithmic amplifier 5-1.

It should be noted that a configuration similar to the configuration in FIG. 4 can be applied based on the configuration in FIG.

FIG. 6 shows a second modification of the second principle of the present invention.

In FIG. 6, 1-1 and 1-2 are couplers, 2-1 and 2-2 are amplifiers, 3 is a variable gain amplifier, and the first coupler 1-1 and the second coupler 1-1-2. 2,
The first amplifier 2-1 and the second amplifier 2-2, and the variable gain amplifier 3 form a controlled part that is a part of the main signal part.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

Reference numeral 6-1 denotes an inclination correction circuit, and reference numeral 6-2 denotes a second inclination correction circuit.

7 is a difference generation circuit, and 8 is a control voltage generation circuit.

Then, the first coupler 1-1, the variable gain amplifier 3, the second coupler 1-2, the first logarithmic amplifier 5-1, the second logarithmic amplifier 5-2, An output power control unit is configured by the inclination correction circuit 6-1, the second inclination correction circuit 6-2, the difference generation circuit 7, and the control voltage generation circuit 8.

The feature of the configuration of FIG. 6 is that the inclination correction circuit 6-
1 and the second tilt correction circuit 6-2 are provided on both the output side of the first logarithmic amplifier 5-1 and the output side of the second logarithmic amplifier 5-2.

As a result, the first logarithmic amplifier 5-1
Even if there is an individual difference in the slope of the output of the second logarithmic amplifier 5-2, it is possible to compensate for this difference, as in the configuration of FIG.

The configuration of FIG. 6 has another advantage that even if the two inclination correction circuits themselves have temperature characteristics, they can be compensated for.

That is, even if the inclination correction circuit 6-1 and the second inclination correction circuit 6-2 have temperature characteristics, and the inclination and the offset change with the temperature change, the inclination correction circuit Since it is easy to reduce the difference between the temperature characteristics of 6-1 and the second slope correction circuit 6-2, finally, the slopes of the logarithmic amplifier 5-1 and the logarithmic amplifier 5-2 can be accurately determined. This is because it can be corrected.

Note that a configuration similar to the configuration in FIG. 6 can be applied based on the configuration in FIG.

FIG. 7 is a modification (3) of the second principle of the present invention, which is based on the configuration of FIG.

In FIG. 7, 1-1 is the first coupler, 1
-2 is a second coupler.

2-1 is a first amplifier, 2-2 is a second amplifier, 3 is a variable gain amplifier, the first coupler 1-1 and the second coupler 1-2, the first amplifier 1-2, and the first coupler 1-2. The amplifier 2-1, the second amplifier 2-2, and the variable gain amplifier 3 form a controlled part that is a part of the main signal part.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

Reference numeral 6-1 denotes an inclination correction circuit, and reference numeral 6-2 denotes a second inclination correction circuit.

7 is a difference generation circuit, and 8 is a control voltage generation circuit.

Reference numeral 9 detects a change in input power and receives a voltage for instructing an increase or decrease in output power due to an external condition, and controls a reference voltage supplied to the control voltage generation circuit 8 to control the controlled unit. Is a gain control circuit for controlling the gain of the control signal.

Then, the first coupler 1-1, the variable gain amplifier 3, the second coupler 1-2, the first logarithmic amplifier 5-1, the second logarithmic amplifier 5-2, An output power control unit is configured by the slope correction circuit 6-1, the second slope correction circuit 6-2, the difference generation circuit 7, the control voltage generation circuit 8, and the gain control circuit 9.

A feature of the configuration shown in FIG. 7 is that the gain control circuit 9 is provided.

The gain control circuit 9 detects a change in the input power and receives a voltage instructing an increase or a decrease in the output power due to an external condition, and varies the reference voltage supplied to the control voltage generation circuit 8. Since the reference voltage is a voltage that determines the gain of the controlled unit, by providing the gain control circuit 9, when the input power fluctuates, the gain is reversed to keep the output power constant, It is possible to increase or decrease the output power of the controlled unit depending on external conditions.

In particular, if it becomes possible to increase or decrease the output power of the controlled unit according to external conditions, it becomes possible to control the output power according to the congestion state of the traffic. , The output power can be reduced to increase the channel capacity.

It goes without saying that the configuration of FIG. 7 can be configured based on the configuration of FIG.

FIG. 8 shows a first embodiment of the present invention, which corresponds to the first principle of the present invention shown in FIG.

In FIG. 8, 1-1 is the first coupler, 1
-2 is a second coupler.

2-1 is a first amplifier, 2-2 is a second amplifier, and 3 is a variable gain amplifier.

4-1 is a first square detection circuit, and 4-2 is a second square detection circuit.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

Reference numeral 71 denotes a voltage follower constituted by an operational amplifier, and reference numeral 72 denotes a 1/2 voltage dividing circuit. The difference follower 7 of FIG. Constitute.

Further, 81 is an analog comparator, 8
2 is a low-pass filter (shown as LPF in the figure)
Thus, the analog comparator 81 and the low-pass filter 82 constitute the control voltage generation circuit 8 in FIG.

In FIGS. 1, 3, 4, 6, and 7, the components of the controlled unit and the components of the output power control unit are described in each case. The description of the components of the control unit and the components of the output power control unit will be omitted.

The output V _IN of the logarithmic amplifier 5-1 is applied to the inverting input terminal of the voltage follower 71, and the output V _OUT of the logarithmic amplifier 5-2 is passed through the 1/2 voltage divider 72. It is applied to the non-inverting input terminal of the voltage follower. This is because the voltage follower 71 has a non-inverting gain of 2 and the voltage follower 71 has an inverting gain of −1, so that the difference voltage (V _OUT −V _{OUT) at} the output terminal of the voltage follower 71 is obtained. _IN ) can be obtained.

The output of the voltage follower 71 is supplied to the inverting input terminal of the analog comparator 81, and the difference voltage (V _OUT −V _IN ) is supplied to the non-inverting input terminal of the analog comparator 81 in the reference state. By applying the reference voltage V _REF , a voltage for controlling the gain of the variable gain amplifier 3 at the output of the analog comparator 81 can be obtained.

When the output of the analog comparator 81 is 0 volts, the gain of the controlled unit becomes a predetermined gain, and when the output of the analog comparator 81 is a positive voltage, the gain of the controlled unit is reduced. If the gain characteristic of the variable gain amplifier is set so as to increase the gain of the controlled part when the output of the analog comparator 81 is a negative voltage, the controlled Can be kept constant. Since it is easy to control the gain of the variable gain amplifier 3 as described above, a portion for controlling the gain of the variable gain amplifier 3 is not particularly illustrated and described.

The reason why the low-pass filter 82 is arranged on the output side of the analog comparator 81 is to block an AC component which may be superimposed on the output of the analog comparator 81 by induction. This is to stabilize the gain control.

As described above, the output power can be stabilized by suppressing the gain fluctuation of the controlled unit due to the temperature fluctuation and the aging change.

The configuration shown in FIG. 8 has a very great advantage in circuit test adjustment.

That is, a continuous sine wave is normally input when the configuration of FIG. 8 is tested and adjusted, and a modulated signal is input when the configuration of FIG. 8 is actually used in a communication system.

Although the response of the square detection circuit to the continuous sine wave does not match the response to the modulated signal, the response to the continuous sine wave is modulated by the square detection at the input and output sides of the controlled unit. It is possible to cancel the difference in response to the signal. Therefore,
It can be used in actual communication systems without readjustment after test adjustments made with continuous sine waves.

The detailed description of the square-law detection circuit and the logarithmic amplifier is omitted here, because the square-law detection circuit is a common sense circuit in the communication system. The reason is that it is not necessary to explain it again because it has been commercialized.

In FIG. 8, the most common coupler has been described as an example for branching the input power or the output power.
All meet the object of the present invention. Thus, what has been described as a coupler may generally be referred to as a branching element. This is the same in the following embodiments of the invention.

FIG. 9 shows a second embodiment of the present invention, which corresponds to the second principle of the present invention shown in FIG.

In FIG. 9, 1-1 is the first coupler, 1
-2 is a second coupler. 2-1 is the first amplifier, 2
-2 is a second amplifier, and 3 is a variable gain amplifier.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

Reference numeral 81 denotes an analog comparator, and 82
Is a low-pass filter (shown as LPF in the figure)
Thus, the analog comparator 81 and the low-pass filter 82 constitute the control voltage generation circuit 8 in FIG.

The configuration of FIG. 9 differs from the configuration of FIG. 8 only in that the first square detection circuit 4-1 and the second square detection circuit 4-2 in the configuration of FIG. 8 are eliminated. 8
Since the configuration is completely the same, the description of the operation is omitted here.

However, the configuration shown in FIG. 9 does not require a square-law detection circuit, and thus has the advantage of simplifying the circuit configuration.

Although the response of the logarithmic amplifier to the continuous sine wave and the response to the modulated signal do not match, the response to the continuous sine wave is modulated by logarithmic amplification on the input side and the output side of the controlled part. It is possible to cancel the difference in response to the received signal. Therefore,
It can be used in an actual communication system without re-adjustment after the test adjustment performed by the continuous sine wave, as in the configuration of FIG.

The embodiments of the present invention described hereinafter are all based on the configuration shown in FIG. 9. Therefore, even if the test adjustment performed by the continuous sine wave is not changed, it can be performed in an actual communication system. Figure 9 can be used
The configuration is the same as that described above.

FIG. 10 shows a third embodiment of the present invention.
This corresponds to a modification (part 1) of the second principle of the present invention in FIG.

In FIG. 10, 1-1 is the first coupler,
1-2 is a second coupler.

Reference numeral 2-1 denotes a first amplifier, 2-2 denotes a second amplifier, and 3 denotes a variable gain amplifier.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

Reference numeral 61-1 denotes an operational amplifier, reference numerals 62-1 and 63-1 denote resistors, reference numeral 71 denotes a voltage follower constituted by an operational amplifier, and reference numeral 73 denotes a voltage divider circuit. 4 is constituted by the resistor 62-1, the resistor 63-1 and the voltage dividing circuit 73, and the difference follower 7 of FIG. 1 is constituted by the voltage follower 71 and the voltage dividing circuit 73. Constitute.

Further, 81 is an analog comparator, 8
2 is a low-pass filter (shown as LPF in the figure)
Thus, the analog comparator 81 and the low-pass filter 82 constitute the control voltage generation circuit 8 in FIG.

The operational amplifier 61-1 includes the resistor 62-1.
If and negative feedback is applied by the resistor 63-1, the non-inverting feedback gain respectively a resistance value of the resistor 62-1 and the resistor 63-1 and R ₁ and R ₂ (R ₁ + R ₂ ) /
Since the _{R 1 = 1 + R 2 /} R 1, for example, the resistor 62-
By making the resistance value R ₁ of ₁ variable, the inclination of the second logarithmic amplifier 5-2 can be corrected in one or more regions.

However, when the inclination of the second logarithmic amplifier 5-2 is larger than the inclination of the first logarithmic amplifier 5-1, it is necessary to correct the inclination to be smaller. Therefore, the voltage dividing ratio of the voltage dividing circuit 73 can be corrected.

As a result, the first logarithmic amplifier 5-1
Even if the inclination of any of the second logarithmic amplifier 5-2 and the second logarithmic amplifier 5-2 is large, the inclination can be corrected.

If necessary, by supplying an adjustable voltage to the terminal of the resistor 62-1 on the side opposite to the operational amplifier 61-1, the second logarithmic amplifier 5-2 can be connected to the second logarithmic amplifier 5-2. The offset of the operational amplifier 61-1 can be compensated in the reference state.

Since the other operations are the same as those of the configuration shown in FIG. 8, further description is omitted.

It goes without saying that the configuration shown in FIG. 10 can be configured based on the first principle of the present invention shown in FIG.

FIG. 11 shows a fourth embodiment of the present invention.
This corresponds to a modification (part 2) of the second principle of the present invention in FIG.

In FIG. 11, 1-1 is a first coupler,
1-2 is a second coupler.

Reference numeral 2-1 denotes a first amplifier, 2-2 denotes a second amplifier, and 3 denotes a variable gain amplifier.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

Reference numeral 61-1 denotes an operational amplifier, 62-1 and 63-1 denote resistors, 71 denotes a voltage follower constituted by the operational amplifier, and 72 denotes a 1/2 voltage divider circuit. 1, the resistor 62-1 and the resistor 63-1
6, the operational amplifier 61-2, the resistor 62-2, and the resistor 63-2 constitute the second inclination correction circuit 6-2 in FIG. The voltage follower 71 and the 1/2 voltage dividing circuit 72 constitute the difference generating circuit 7 in FIG.

Further, 81 is an analog comparator, 8
2 is a low-pass filter (shown as LPF in the figure)
Thus, the analog comparator 81 and the low-pass filter 82 constitute the control voltage generation circuit 8 in FIG.

In the configuration of FIG. 11, the operational amplifier 6
1-1, the resistor 62-1 and the slope correction circuit 6-1 including the resistor 63-1;
Since the second inclination correction circuit 6-2 constituted by the resistor 62-2 and the resistor 63-2 is provided, for example, the resistors 62-1 and 62-2 may be variable resistors. Thus, the inclination of the logarithmic amplifier 5-1 and the logarithmic amplifier 5-2 can be arbitrarily corrected.

Further, since two slope correction circuits are provided and the difference between the outputs of the two slope correction circuits is obtained, even if the gain and offset of the two slope correction circuits themselves have temperature characteristics, the two slope correction circuits have the same characteristics. Since it is easy to make the temperature characteristics of the gain and the offset of the two inclination correction circuits substantially equal, the temperature characteristics of the inclination correction circuit itself can be suppressed.

Other operations are the same as those of the configuration shown in FIGS. 8 and 10, and therefore, further description is omitted.

It goes without saying that the configuration shown in FIG. 11 can be configured based on the first principle of the present invention shown in FIG.

FIG. 12 shows a fifth embodiment of the present invention.
This corresponds to a modification (No. 3) of the second principle of the present invention in FIG.

In FIG. 12, 1-1 is the first coupler,
1-2 is a second coupler.

Reference numeral 2-1 denotes a first amplifier, 2-2 denotes a second amplifier, and 3 denotes a variable gain amplifier.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

61-1 is an operational amplifier, 62-1 and 63
-1 is a resistor, 71 is a voltage follower composed of an operational amplifier, 72 is a 1/2 voltage divider circuit, and the operational amplifier 61-1, the resistor 62-1 and the resistor 63-1 in FIG. The operational amplifier 61-2, the resistor 62-2, and the resistor 63-2 constitute a second inclination correction circuit 6-2 in FIG. The difference generating circuit 7 of FIG.

Reference numeral 81 denotes an analog comparator, and 82
Is a low-pass filter (shown as LPF in the figure)
Thus, the analog comparator 81 and the low-pass filter 82 constitute the control voltage generation circuit 8 in FIG.

Further, reference numerals 91 and 92 are voltage followers composed of operational amplifiers, 93 and 94 are resistors, 95
And 96 are 1/2 voltage dividing circuits. The voltage followers 91 and 92, the resistors 93 and 94, and the 1/2 voltage dividing circuits 95 and 96 constitute the gain control circuit 9 in FIG.

The output V _IN 'of the operational amplifier 61-2 is supplied to the inverting input terminal of the voltage follower 91.
The output V _IN ' _{(0) of the} operational amplifier 61-2 in the reference state is applied to the 1/2 voltage dividing circuit 95.

Non-inverting gain of the voltage follower 91
Is 2 and the inversion gain is -1.
At the output of the voltage follower 91, V_IN'And V
_IN’ ₍₀₎Can be obtained.

The voltage follower 92 and the resistor 9 connected to the inverting input terminal of the voltage follower 92
3 and 94 constitute an adding circuit for adding the output of the voltage follower 91 and the external control voltage.

On the other hand, the reference voltage V _REF is supplied to the non-inverting input terminal of the voltage follower 92 via the 1/2 voltage dividing circuit 96.

Therefore, at the output terminal of the voltage follower 92, it is possible to obtain a voltage obtained by subtracting the voltage obtained by adding the external control voltage and the difference voltage (V _IN '-V _IN ' ₍₀₎ ) from the reference voltage V _REF. it can.

The output of the voltage follower 92 is supplied to the non-inverting input terminal of the analog comparator 81,
The difference voltage (V _OUT '─V _IN ') is supplied to the inverting input terminal of the analog comparator 81.

If the difference voltage (V _OUT ─V _IN ) is set to be equal to the reference voltage V _REF when the gain of the controlled part is a predetermined value, then the analog comparator The reference voltage V _REF is fed back to the inverting input terminal 81.

However, in the configuration shown in FIG.
Since a voltage different from the reference voltage V _REF is supplied to the non-inverting input terminal of the comparator 81 by the external control voltage and V _IN ′ ₍₀₎ , the variable gain amplifier 3 Must be changed.

Therefore, in the configuration of FIG. 12, when the input power to the controlled section fluctuates or an external control voltage is applied, the gain of the variable gain amplifier 3 changes. That is,
In the configuration of FIG. 12, it is possible to change the output power when the input power to the controlled unit fluctuates.
The output power can be made variable by an external control voltage.

As a result, the output power can be kept constant even if the input power to the controlled unit changes, or an external control voltage can be supplied by detecting the congestion state of the traffic to thereby control the traffic. The output power can be controlled according to the congestion state.

When the configuration shown in FIG. 12 is applied to a code division multiple access communication system, the following control may be performed.

First, when the number of communication channels is small and the total power of the communication system is lower than the total power allowed for the communication system, the voltage V _IN ′ _{(0 )} Is set to 0 volts and the external control voltage is also set to 0 volts, the reference voltage V _REF is supplied to the non-inverting input terminal of the analog comparator 81.
The gain of the controlled section having the configuration 2 can be kept constant.

When the number of communication channels increases and the total power of the communication system becomes higher than the total power allowed for the communication system, the voltage V _IN supplied to the 圧 voltage dividing circuit 95 is increased. If _{(0) is set} to 0 volt and a voltage other than 0 is supplied to the external control voltage, a voltage different from the reference voltage V _REF is supplied to the non-inverting input terminal of the analog comparator 81. It is possible to suppress the gain of the controlled part having the configuration.

Thus, when the number of communication channels of the communication system of the code division multiple access system is small, the signal-to-noise ratio of each channel is maintained at a required value. It is possible to secure a large channel capacity of the system.

The external control voltage may be determined by a base station of a communication system using a code division multiple access system or a control station that controls the entire system.

When the configuration shown in FIG. 12 is applied to a frequency division multiplex communication system,
By supplying the voltage V _IN ' ₍₀₎ corresponding to the input power in the reference state to 5 and setting the external control voltage to 0 volt, the output power can be controlled to be constant even if the input power fluctuates. .

Here, the configuration of FIG. 12 is obtained by adding a gain control circuit to the configuration of FIG. 11. However, it is noted that the gain control circuit can be added to the configurations of FIG. 8 and FIG. Needless to say.

It is needless to say that a configuration similar to that of FIG. 12 can be configured based on the first principle of the present invention shown in FIG.

FIG. 13 shows a sixth embodiment of the present invention.
It is based on the configuration of FIG.

In FIG. 13, 1-1 is a first coupler,
1-2 is a second coupler.

2-1 is a first amplifier, 2-2 is a second amplifier, and 3 is a variable gain amplifier.

5-1 is a first logarithmic amplifier, and 5-2 is a second logarithmic amplifier.

61-1 is an operational amplifier, 62-1 and 63
-1 is a resistor, 71 is a voltage follower composed of an operational amplifier, 72 is a 1/2 voltage divider circuit, and the operational amplifier 61-1, the resistor 62-1 and the resistor 63-1 in FIG. The operational amplifier 61-2, the resistor 62-2, and the resistor 63-2 constitute a second inclination correction circuit 6-2 in FIG. The difference generating circuit 7 of FIG.

Reference numeral 81 denotes an analog comparator, and 82
Is a low-pass filter (shown as LPF in the figure)
Thus, the analog comparator 81 and the low-pass filter 82 constitute the control voltage generation circuit 8 in FIG.

Further, reference numerals 91 and 92 are voltage followers composed of operational amplifiers, 93 and 94 are resistors, 95
And 96 are 1/2 voltage dividing circuits. The voltage followers 91 and 92, the resistors 93 and 94, and the 1/2 voltage dividing circuits 95 and 96 constitute the gain control circuit 9 in FIG.

Finally, 10 is a local oscillator, 11 is an amplifier for amplifying the local oscillation output to a predetermined level, 12 is a band-pass filter for removing unnecessary waves contained in the output of the local oscillator, and 13 is a local oscillation output. Is modulated by the output of the second amplifier 2-2, and 14 is a band-pass filter that extracts a required sideband from the output of the mixer 13.

The configuration of FIG. 13 is different from the configuration of FIG. 12 in that the local oscillator 10, the amplifier 11, and the band-pass
2. The difference is that a frequency conversion function comprising the mixer 13 and the band-pass filter 14 is added.

However, the principle of the present invention relates to the control of the output power of the controlled part, and has basically no relation to the frequency. In addition, as described above, logarithmic amplification is performed at the input and output of the controlled unit, or logarithmic amplification is performed by square detection, so that the configuration is completely unaffected by the frequency. Therefore, the control of the output power of the controlled part in the configuration of FIG. 13 can be performed in exactly the same manner as in the configuration of FIG.

The technique of controlling the gain by adding the frequency conversion function is based on the configuration shown in FIG. 12, in addition to the configuration shown in FIG. It goes without saying that control is possible.

Further, it is needless to say that a configuration similar to that of FIG. 13 can be configured based on the first principle of the present invention shown in FIG.

FIG. 14 shows a seventh embodiment of the present invention.
This figure shows a device that controls output power among a plurality of channels. It should be noted that detailed circuit diagrams are shown in FIG. 8 and subsequent figures, and the configuration of each part of the diagram showing the principle of the present invention is clarified. In FIG. 14, congestion of the diagram is avoided. For this reason, the circuit configuration is shown using the components used in the drawings illustrating the principle of the present invention.

In FIG. 14, 1-1 is a first coupler,
1-2 is a coupler, 2-1 is a first amplifier, 2-2 is a second amplifier, 3 is a variable gain amplifier, 5-1 is a first logarithmic amplifier, 5-2 is a second logarithmic amplifier, Reference numeral 6-1 denotes an inclination correction circuit, and reference numeral 7 denotes a difference generation circuit. The above components constitute an output power control circuit of the first channel.

Further, 1-1a is a third coupler, 1-2a is a fourth coupler, 2-1a is a third amplifier, 2-2a is a fourth amplifier, 3a is a second variable gain amplifier, 5-1a is a third logarithmic amplifier, 5-2a is a fourth logarithmic amplifier, 6-
1a is a third inclination correction circuit, 7a is a second difference generation circuit, 8a is a second control voltage generation circuit, and the above components constitute an output power control circuit of the second channel.

In the configuration of FIG. 14, the output power of the first output power control circuit is defined by the voltage V _AGC supplied to the variable gain amplifier 3. Therefore, the difference generation circuit 7
Outputs a voltage corresponding to the specified output power.

Since the output voltage of the difference generation circuit 7 is supplied to the control voltage generation circuit 8a as a reference voltage, the second output power control circuit 8a uses the voltage supplied from the difference generation circuit 7 as a reference voltage. The output power of the second output power control circuit is controlled.

That is, by controlling the voltage V _AGC , the output power of the first output power control circuit can be controlled, and at the same time, the output power of the second output power control circuit can be controlled.

The generation of V _AGC is specifically performed by a control voltage generation circuit, and the reference voltage applied to the control voltage generation circuit can be made constant, or the gain control circuit can be used to control the reference voltage. It is also possible.

A feature of the configuration of FIG. 14 is that when the outputs of the two output power control circuits are controlled using the gain control circuit, the gain control circuit is provided only in the first output power control circuit, In the output power control circuit, the output of the difference generation circuit of the first output power control circuit can be used as the reference voltage.

The technique shown in FIG. 14 is not limited to output power control between two output power control circuits, but can be generally applied to output power control between three or more output power control circuits. When the output power of a plurality of channels is controlled by the configuration of FIG. 14, the circuit scale can be reduced.

Although FIG. 14 shows a configuration based on the configuration of FIG. 10, in addition to the configuration based on FIG. 10, the configuration based on FIG. 8, FIG. 11, FIG. It goes without saying that it is possible to do so.

It is needless to say that a configuration similar to that of FIG. 14 can be configured based on the first principle of the present invention shown in FIG.

As described above, according to the present invention, all the conventional problems of the output power control circuit can be solved, and further, an output power control circuit which can be adapted to a wide range of applications can be realized.

Since the output power control circuit normally constitutes a part of the radio frequency modulation circuit which is a main part of the transmission circuit, realization of the output power control circuit as described above solves all the conventional problems relating to the transmission circuit. can do.

[0218]

As described above in detail, according to the present invention, it is possible to realize an output power control circuit capable of maintaining a constant output power while absorbing a characteristic fluctuation due to a temperature fluctuation and a characteristic fluctuation due to a temporal change.

Further, by making the reference voltage for the output power control variable by the external control voltage, it is possible to realize an output power control circuit capable of controlling the output power according to, for example, a traffic congestion state.

Further, the input power fluctuated by making the reference voltage for output power control variable by the difference between the voltage corresponding to the input power in the reference state and the voltage corresponding to the input power in the operating state including the fluctuation. An output power control circuit that keeps the output power constant sometimes can be realized.

Further, it is possible to realize an output power control circuit capable of controlling the output power of a plurality of channels by using the output power control result of a specific channel in common.

As described above, according to the present invention, it is possible to provide an output power control circuit applicable to a wide range of applications.

Further, by applying the output power control circuit according to the present invention to a transmission circuit, the conventional problems concerning the transmission circuit can be solved.

[Brief description of the drawings]

FIG. 1 shows a first principle of the present invention.

FIG. 2 is a diagram illustrating a first principle of the present invention.

FIG. 3 shows a second principle of the present invention.

FIG. 4 is a modification (1) of the second principle of the present invention.

FIG. 5 is a diagram illustrating a modification (part 1) of the second principle of the present invention.

FIG. 6 is a modification (2) of the second principle of the present invention.

FIG. 7 is a modification (3) of the second principle of the present invention.

FIG. 8 shows a first embodiment of the present invention.

FIG. 9 shows a second embodiment of the present invention.

FIG. 10 shows a third embodiment of the present invention.

FIG. 11 shows a fourth embodiment of the present invention.

FIG. 12 shows a fifth embodiment of the present invention.

FIG. 13 shows a sixth embodiment of the present invention.

FIG. 14 shows a seventh embodiment of the present invention.

FIG. 15 shows a conventional output power control circuit.

FIG. 16 shows a problem of the conventional technique (part 1).

FIG. 17 is a problem of the conventional technique (No. 2).

FIG. 18 is a schematic configuration of a transmission circuit.

[Explanation of symbols]

 1-1 First coupler 1-2 Second coupler 1-1a Third coupler 1-2a Fourth coupler 2-1 First amplifier 2-2 Second amplifier 2-1a Third amplifier 2 -2a Fourth amplifier 3 Variable gain amplifier 3a Second variable gain amplifier 4-1 First square detection circuit 4-2 Second square detection circuit 4-1a Third square detection circuit 4-2a Fourth Square detection circuit 5-1 First logarithmic amplifier 5-2 Second logarithmic amplifier 5-1a Third logarithmic amplifier 5-2a Fourth logarithmic amplifier 6-1 Tilt correction circuit 6-2 Second tilt correction circuit 6-1a Third slope correction circuit 7 Difference generation circuit 7a Second difference generation circuit 8 Control voltage generation circuit 8a Second control voltage generation circuit 9 Gain control circuit 10 Local oscillator 11 Amplifier 12 Bandpass filter 13 Mixer 14 Bandpass Filter 61-1 Operational Amplifier 62-1 Resistance 63-1 Resistance 61-2 Operational Amplifier 62-2 Resistance 63-2 Resistance 71 Voltage Follower 72 1/2 Voltage Divider 73 Voltage Divider 81 Analog Comparator 82 Low-Pass Filter 91 Voltage follower 92 Voltage follower 93 Resistance 94 Resistance 95 1/2 voltage divider 96 1/2 voltage divider

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Yoji Nakada 1-3-3 Kita-Nanajo-Nishi, Kita-ku, Sapporo-shi, Hokkaido Within Fujitsu Hokkaido Digital Technology Co., Ltd. Nishi 4-chome 3 Fujitsu Hokkaido Digital Technology Co., Ltd. (72) Inventor Hajime Iwatsuki Kita-ku, Kita-ku, Sapporo-shi, Hokkaido 4-7-1, Fujitsu Hokkaido Digital Technology Co., Ltd. F-term (reference) 5J100 JA01 KA05 LA00 LA09 QA01 SA01 5K060 BB08 DD04 FF06 HH05 HH06 HH10 JJ06 JJ16 KK01 KK04 LL01 LL24 PP05

Claims (6)

[Claims] A first branch element for branching an input of a controlled part to be output power controlled; a second branch element for branching an output of the controlled part to be output power controlled; A first square detection circuit that squares the output of the first branch element, a second square detection circuit that squares the output of the second branch element, and an output of the first square detection circuit. A first logarithmic amplifier for logarithmic amplification, a second logarithmic amplifier for logarithmic amplification of the output of the second square detection circuit, and a difference generation circuit for taking the difference between the outputs of the first and second logarithmic amplifiers, And a control voltage generation circuit for generating a difference between an output of the difference generation circuit and a reference voltage and supplying the difference as a gain control voltage to a variable gain amplifier provided in the controlled section. circuit. 2. A first branch element for branching an input of a controlled part to be output power controlled, a second branch element for branching an output of the controlled part to be output power controlled, A first logarithmic amplifier that logarithmically amplifies the output of the first branch element, a second logarithmic amplifier that logarithmically amplifies the output of the second branch element, and an output of the first and second logarithmic amplifiers. A difference generation circuit that takes a difference; and a control voltage generation circuit that generates a difference between an output of the difference generation circuit and a reference voltage and supplies the difference as a gain control voltage to a variable gain amplifier provided in the controlled unit. An output power control circuit, characterized in that: 3. The output power control circuit according to claim 1, wherein the first logarithmic amplifier and the difference generation circuit are arranged between the first logarithmic amplifier and the difference generation circuit. An output power control circuit, wherein a slope correction circuit for correcting a slope of the input / output of the logarithmic amplifier is arranged between the output power control circuit and the logarithmic amplifier. 4. The output power control circuit according to claim 1, further comprising a gain control circuit for controlling the reference voltage. 5. An output power control circuit comprising a plurality of output power control circuits according to claim 1, wherein an output of the difference generation circuit in a specific output power control circuit is output by another output power control circuit. An output power control circuit, wherein the reference voltage is a reference voltage supplied to the control voltage generation circuit in the output power control circuit. 6. A transmission circuit comprising the output power control circuit according to claim 1. Description: