HK1212115B - Circuits and methods related to radio-frequency receivers having carrier aggregation - Google Patents

Description

Circuits and methods related to radio frequency receivers with carrier aggregation

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority to U.S. Provisional Application No. 61/978,808, filed April 11, 2014, entitled “Circuits and Methods Related to Radio Frequency Receivers with Carrier Aggregation,” the entire contents of which are hereby incorporated by reference.

Technical Field

The present invention relates to carrier aggregation in radio frequency applications.

Background Art

In some radio frequency (RF) applications, cellular carrier aggregation (CA) can involve two or more RF signals being processed via a common path. For example, carrier aggregation can involve using a single path for multiple frequency bands with sufficiently separated frequency ranges. In this configuration, simultaneous operation of more than one frequency band can be achieved.

Summary of the Invention

According to various implementations, the present disclosure relates to a carrier aggregation (CA) circuit comprising a first filter configured to enable operation in a first frequency band and a second filter configured to enable operation in a second frequency band. The CA circuit further comprises a first path implemented between the first filter and a common node, the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open impedance for the second frequency band. The CA circuit further comprises a second path implemented between the second filter and the common node, the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open impedance for the first frequency band.

In some embodiments, the first filter and the second filter may be part of a duplexer. The duplexer may include an input port configured to receive a radio frequency (RF) signal from an antenna. The common node may be configured to couple to an input of a low noise amplifier (LNA). The LNA may be configured to amplify the received RF signal in frequency bands corresponding to the first and second frequency bands. The first frequency band may include, for example, a B3 band having a frequency range of 1.805 to 1.880 GHz, and the second frequency band may include, for example, a B1 band having a frequency range of 2.110 to 2.170 GHz. The second frequency band may further include a B4 band having a frequency range of 2.110 to 2.155 GHz.

In some embodiments, the first frequency band may include a B2 frequency band having a frequency range of 1.930 to 1.990 GHz, and the second frequency band may include a B2 frequency band having a frequency range of 2.110 to 2.155 GHz. The first frequency band may further include a B25 frequency band having a frequency range of 1.930 to 1.995 GHz. In some embodiments, the first frequency band may include a B2 frequency band having a frequency range of 1.930 to 1.990 GHz, and the second frequency band may include a B4 frequency band having a frequency range of 2.110 to 2.155 GHz.

In some embodiments, the first path may include a first phase-shifting circuit, and the second path may include a second phase-shifting circuit. In some embodiments, the first phase-shifting circuit may include, for example, two series capacitors and an inductive shunt path coupled to a node between the two capacitors and ground. The second phase-shifting circuit may include two series capacitors and an inductive shunt path coupled to a node between the two capacitors and ground. In some embodiments, at least some of the capacitors and inductors of the first and second phase-shifting circuits may be implemented as lumped elements. In some embodiments, at least some of the capacitors and inductors of the first and second phase-shifting circuits may be implemented as distributed elements.

In some embodiments, the first phase-shifting circuit may include two series inductors and a capacitive shunt path coupled to a node between the two inductors and ground. The second phase-shifting circuit may include two series inductors and a capacitive shunt path coupled to a node between the two inductors and ground. In some embodiments, at least some of the capacitors and inductors of the first and second phase-shifting circuits may be implemented as lumped elements. In some embodiments, at least some of the capacitors and inductors of the first and second phase-shifting circuits may be implemented as distributed elements.

In some embodiments, each of the first and second paths can include a switch to allow the CA circuit to operate in either CA mode or non-CA mode. The switch for the first path can be located at the output of the first phase-shifting circuit, and the switch for the second path can be located at the output of the second phase-shifting circuit. Both the switches in the first and second paths can be closed for CA mode. One of the two switches can be closed, while the other can be closed for non-CA mode.

In some embodiments, the present disclosure relates to a radio frequency (RF) module having a package substrate configured to accommodate multiple components and a carrier aggregation (CA) circuit implemented on the package substrate. The CA circuit includes a first filter configured to allow operation in a first frequency band, and a second filter configured to allow operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open impedance for the second frequency band. The CA circuit also includes a second path implemented between the second filter and the common node, the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open impedance for the first frequency band.

In some embodiments, each of the first filter and the second filter may include a surface acoustic wave (SAW) filter. The first SAW filter and the second SAW filter may be implemented as a duplexer. In some embodiments, the RF module may further include a low noise amplifier (LNA) implemented on the package substrate. The LNA may be coupled to a common node to receive a combined signal from the first path and the second path. In some embodiments, the RF module may be a front-end module. In some embodiments, the RF module may be a diversity reception (DRx) module.

In some embodiments, the first path may include a first phase-shifting circuit, and the second path may include a second phase-shifting circuit. Each of the first phase-shifting circuit and the second phase-shifting circuit may include a capacitor and an inductor. At least some of the capacitor and inductor components may be implemented as passive devices mounted on or in a package substrate.

In some teachings, the present disclosure relates to a method for manufacturing a radio frequency (RF) module. The method includes providing or forming a package substrate configured to accommodate multiple components and implementing a carrier aggregation (CA) circuit on the package substrate. The CA circuit includes a first filter configured to enable operation in a first frequency band and a second filter configured to enable operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, the first path configured to provide a substantially matched impedance for the first frequency band and a substantially open impedance for the second frequency band. The CA circuit further includes a second path implemented between the second filter and the common node, the second path configured to provide a substantially matched impedance for the second frequency band and a substantially open impedance for the first frequency band.

According to some implementations, the present invention relates to a radio frequency (RF) device having a receiver configured to process RF signals, and an RF module in communication with the receiver. The RF module includes a carrier aggregation (CA) circuit. The CA circuit includes a first filter configured to allow operation in a first frequency band, and a second filter configured to allow operation in a second frequency band. The CA circuit further includes a first path implemented between the first filter and a common node, the first path being configured to provide a substantially matched impedance for the first frequency band and a substantially open impedance for the second frequency band. The CA circuit also includes a second path implemented between the second filter and the common node, the second path being configured to provide a substantially matched impedance for the second frequency band and a substantially open impedance for the first frequency band. The RF device further includes an antenna in communication with the RF module, the antenna being configured to receive the RF signal.

In some embodiments, the RF device may be a wireless device. In some embodiments, the wireless device may be a cellular phone. In some embodiments, the antenna may include a diversity antenna, and the RF module may include a diversity receive (DRx) module. The wireless device may further include an antenna switch module (ASM) configured to route the RF signal from the diversity antenna to the receiver. In some embodiments, the DRx module may be implemented between the diversity antenna and the ASM.

For the purpose of summarizing the present disclosure, certain aspects, advantages and novel features of the present invention are described herein. It should be understood that not necessarily all such advantages can be achieved according to any specific embodiment of the present invention. Therefore, the present invention can be implemented or carried out in a manner that realizes or optimizes an advantage or a group of advantages as taught herein without necessarily realizing other advantages that may be taught or suggested herein.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG1 illustrates a carrier aggregation configuration including a carrier aggregation (CA) circuit configured to accommodate multiple inputs and produce one output.

FIG2 shows that a CA aggregation configuration may include more than two radio frequency (RF) signals.

FIG3 shows a more detailed example in which a CA circuit having one or more features described herein may be implemented with a low noise amplifier (LNA) in a receiver.

FIG4 shows a converged configuration where the RF signal is split at a common input node (RF_IN), and each of the two split RF signals is processed by a bandpass filter and amplified by an LNA.

Figure 5 shows an example of an aggregate configuration where additional filtering of the LNA output can be implemented to better isolate the band distribution between different frequency bands.

FIG. 6 shows a CA configuration, which may be a more specific example of the configuration of FIG. 3 .

FIG. 7 shows the CA configuration of FIG. 6 operating in CA mode.

FIG8 illustrates an example CA configuration in which the first and second signal paths of FIG7 may be configured to provide selected impedances to facilitate CA operation.

FIG9 illustrates two isolated receive (Rx) paths associated with example frequency bands B3 and B1/4.

FIG. 10 shows an example of a Smith graph of complex impedance values of the circuit of FIG. 9 .

11 illustrates two isolated receive (Rx) paths associated with example frequency bands B3 and B1/4, where each path includes a first phase shifting circuit, a filter, and a second phase shifting circuit between its antenna node and output node.

FIG. 12 shows an example of a Smith graph of complex impedance values of the circuit of FIG. 11 .

FIG. 13 shows the two example receive (Rx) paths of FIG. 11 connected at endpoints to produce a common antenna node and a common output node.

FIG. 14 shows an example of a Smith graph of complex impedance values of the circuit of FIG. 13 .

FIG. 15 shows various spectral response curves associated with the examples of FIGs. 9 , 11 , and 13 .

FIG. 16A shows an example of the phase shifting circuit of FIGs. 13-15 .

FIG16B shows further examples of the phase shifting circuits of FIG13-15.

FIG. 17 illustrates a process that may be implemented to manufacture a device having one or more features described herein.

FIG. 18 illustrates an RF module having one or more features described herein.

FIG19 shows an example of an RF structure including one or more features described herein.

FIG. 20 illustrates an example of a wireless device having one or more advantageous features described herein.

FIG21 illustrates that one or more features of the present disclosure may be implemented in a diversity reception module.

FIG. 22 illustrates an example wireless device having the diversity reception module of FIG. 21 .

DETAILED DESCRIPTION

The headings provided herein, if any, are for convenience only and do not necessarily affect the scope or meaning of the claimed invention.

Cellular carrier aggregation (CA) can be supported by allowing two or more radio frequency (RF) signals to be processed through a common path. For example, carrier aggregation can include using a single path for multiple frequency bands with sufficiently separated frequency ranges. In this configuration, simultaneous operation of more than one frequency band is possible.

In the receiver context, carrier aggregation can enable parallel processing of RF signals in multiple frequency bands to provide, for example, high data rate capabilities. In such carrier aggregation systems, it is desirable to maintain a low noise figure (NF) for each RF signal. When the two aggregated frequency bands are close in frequency, it is also desirable to maintain sufficient separation between the two bands.

FIG1 illustrates a carrier aggregation (CA) configuration 100, which includes a CA circuit 110 configured to receive multiple inputs and generate one output. The multiple inputs may include a first RF signal and a second RF signal. The first RF signal may be provided to the CA circuit 110 from a common input node 102 (RF_IN) via a first path 104a including a first filter 106a. Similarly, the second RF signal may be provided to the CA circuit 110 from the common input node 102 (RF_IN) via a second path 104b including a second filter 106b. As described herein, the CA circuit 110 may be configured such that the output at the common output node 114 is a recombined RF signal including two separate frequency bands associated with the first and second RF signals. Furthermore, as described herein, the CA circuit 110 may be configured to produce desired performance characteristics, such as low loss, a low noise figure, and/or high isolation between the two signal paths 104a, 104b.

The various examples herein (including the example of FIG. 1 ) are described in the context of aggregating two frequency bands. However, it should be understood that one or more features of the present disclosure can be implemented in the aggregation of more than two frequency bands. For example, FIG. 2 illustrates a CA aggregation configuration 100 in which three RF signals are separated at a common input node 102 (RF_IN), processed by their respective filters 106 a, 106 b, 106 c, and recombined by CA circuitry 110 to produce a recombined RF signal at a common output node 114 (RF_OUT).

The aggregation configuration 100 of Figures 1 and 2 can be implemented in a variety of RF applications. Figure 3 illustrates a more specific example, in which a CA circuit 110 having one or more features described herein can be implemented in a receiver using a low-noise amplifier (LNA). CA circuit 110 can be configured to receive multiple inputs and generate a single output. The multiple inputs can include a first RF signal and a second RF signal. The first RF signal can be provided to CA circuit 110 from a common input node 102 (RF_IN) via a first path including a first bandpass filter 122. Similarly, the second RF signal can be provided to CA circuit 110 from the common input node 102 (RF_IN) via a second path including a second bandpass filter 124. As described herein, CA circuit 110 can be configured such that the output at common output node 114 is a recombined RF signal that includes two separate frequency bands associated with the first and second RF signals. Furthermore, as described herein, CA circuit 110 can be configured to produce desired performance characteristics, such as low loss, a low noise figure, and/or high isolation between the two input signal paths.

3, the recombined RF signal is shown as being provided to LNA 120 for amplification, thereby producing a low noise amplified output signal at output node 114. LNA 120 may be configured to operate with a sufficiently wide bandwidth to effectively amplify the first and second frequency bands of the recombined RF signal.

In some embodiments, the bandpass filters 122, 124 can be implemented in a variety of ways, including, for example, surface acoustic wave (SAW) filters. It will be understood that other types of filters can also be utilized.

As described herein, the aggregation configuration 100 of FIG. 3 can provide several advantageous features over other aggregation configurations. For example, FIG. 4 illustrates an aggregation configuration 10 in which the RF signal is split at a common input node (RF_IN); each of the two separate RF signals is processed by a bandpass filter (12 or 16) and amplified by an LNA (14 or 18). The separately processed and amplified RF signals (bands "A" and "B") are shown to be combined by a combiner 20 to produce an output RF signal at a common output node (RF_OUT).

In the example of Figure 4, the combined output RF signal includes amplified noise components from each of the two LNAs. Therefore, the noise figure may be degraded by, for example, approximately 3 dB.

Typically, the lack of proper isolation between RF paths (e.g., the paths associated with bands "A" and "B" in FIG. 4 ) and their respective frequency bands contributes to the noise figure of the combined RF signal. FIG. 5 illustrates an example of an aggregate configuration 30 in which additional filtering of the LNA outputs may be implemented to better isolate the frequency band distribution between the "A" and "B" bands. Similar to the example of FIG. 4 , the example aggregate configuration 30 includes the RF signal being separated at a common input node (RF_IN); each of the two separated RF signals is processed by a bandpass filter (32 or 42) and amplified by an LNA (34 or 44). The separately processed and amplified RF signals (bands "A" and "B") are shown as being further filtered by respective filters 36, 46 before being combined to produce a combined RF signal at a common output node (RF_OUT). As a result of this filtering, the total noise output at the common output node (RF_OUT) in band "A" typically includes only components from LNA 34, while the total noise output at the common output node (RF_OUT) in band "B" typically includes only components from LNA 44. While this configuration avoids the 3 dB noise degradation of the previously mentioned example, it typically incurs the excessive cost associated with two LNAs and two post-LNA filters.

Typically, filters constructed from higher-Q resonators provide better band isolation, especially for frequency bands that are relatively close to each other. For example, cellular bands B1 and B3 have ranges of 2.110 to 2.170 GHz and 1.805 to 1.880 GHz, respectively, for receive operation. For such a pair of relatively close frequency bands, good band isolation is generally not possible with low-Q resonators. Therefore, high-Q resonators are often required or desired. However, the use of such additional high-Q resonators downstream of the two LNAs (e.g., 34 and 44 in FIG. 5 ) may be undesirable due to, for example, the additional cost and required space.

Figure 6 shows a CA configuration 100, which may be a more detailed example of the configuration of Figure 3. The CA configuration 100 of Figure 6 may provide a number of desirable features, including those that address some or all of the issues associated with the examples of Figures 4 and 5.

In FIG6 , an example CA configuration 100 includes RF signals being split at a common input node 102 (RF_IN). A first split RF signal is shown as being filtered by a bandpass filter 122, and a second split RF signal is shown as being filtered by a bandpass filter 124. The first and second filtered RF signals are shown as being provided to a CA circuit 110 configured to produce a combined signal at a common node 126.

CA circuit 110 is shown as including a phase circuit, generally indicated at 150, and a switch circuit, generally indicated at 140. Examples of functionality that phase circuit 150 and switch circuit 140 may provide are described in greater detail herein.

The first filtered RF signal from bandpass filter 122 is shown passing through a first phase shifting circuit 152. Similarly, the second filtered RF signal from bandpass filter 124 is shown passing through a second phase shifting circuit 154. Examples of such phase shifting circuits are described in more detail herein.

The first and second RF signals from their respective phase-shifting circuits (152, 154) are shown combined at a common node 126. In some embodiments, a switch S1 can be implemented between the first phase-shifting circuit 152 and the common node 126, and a switch S2 can be implemented between the second phase-shifting circuit 154 and the common node 126. Such switches can allow the CA circuit 110 to operate in either a non-CA mode or a CA mode. For example, in FIG6 , the first switch S1 is shown closed and the second switch S2 is shown open, such that the CA circuit 110 processes the first RF signal in the corresponding frequency band in the non-CA mode. To process a second RF signal in another frequency band in the non-CA mode, the first switch S1 can be opened and the second switch S2 can be closed. In another example, as shown in FIG7 , both the first and second switches can be closed, such that the CA circuit 110 processes the first and second RF signals in their respective frequency bands in the CA mode.

6 and 7 , common node 126 is shown coupled to the input of LNA 120 to allow the processed RF signal (the combined RF signal in CA mode or the single-band RF signal in non-CA mode) to be processed by LNA 120. LNA 120 is shown producing an amplified RF signal as an output (RF_OUT) at node 114.

6 and 7 , the switch circuit 140 may allow the CA circuit 110 to operate in either the non-CA mode or the CA mode. In embodiments where the CA circuit 110 is configured to operate only in the CA mode, the switch circuit 140 may be omitted.

FIG8 illustrates an example CA configuration 100, in which the first and second signal paths can be configured to provide selected impedances to facilitate CA operation. For illustrative purposes, such signal paths may be referred to as "A" and "B" bands, which may include any combination of bands suitable for carrier aggregation. As in the example of FIG7 , switches S1 and S2 may both be in their closed states to facilitate CA operation.

As shown in the example of FIG8 , first and second phase offset circuits 152 and 154 can be used to couple first (A) and second (B) bandpass filters 122 and 124 to LNA 120 and provide desired impedance values for the signals that are combined and routed to LNA 120. Examples of such impedance value adjustment by first and second phase offset circuits 152 and 154 are described in more detail herein.

The impedance of first filter 122 can be tuned to provide a desired impedance for the A-band signal. Thus, the impedance Z _A of the A-band signal at the output of A-band filter 122 is approximately at a matching value of Z _o (e.g., 50 ohms). In the B-band, the impedance Z _B of the B-band signal at the output of A-band filter 122 does not match Z _o . Because the B-band is within the stopband of the A-band filter, the reflection coefficient |Γ _B | of this mismatch is approximately unity. However, the phase of this reflection generally depends on the filter design. Therefore, the impedance Z _B of the B-band signal at the output of A-band filter 122 can be any significantly different mismatch value, much larger or much smaller than Z _o , resulting in a situation where |Γ _B | is approximately 1.

Ideally, the A-band filter 122 should present an open circuit to the B-band signal. However, the A-band filter 122 may not provide this ideal open-circuit impedance for the B-band signal. Therefore, the impedance _ZB of the B-band signal at the output of the A-band filter 122 can be expressed in the complex form _ZB = _RB + _jXB , where the real part (resistance _RB ) and the imaginary part (reactance _XB ) cause the impedance _ZB to be significantly away from the open circuit state (where one or both of _XB and _RB are approximately infinite). As shown in FIG8, the first phase shift circuit 152 can be configured to substantially maintain _Z0 for _ZA and adjust _ZB from _RB + _jXB to (or close to) the open circuit state.

Similarly, the impedance of second filter 124 can be tuned to provide a desired impedance for the B-band signal. Thus, the impedance _ZB of the B-band signal at the output of B-band filter 124 is approximately at a matching value of _Z0 (e.g., 50 ohms). In the A-band, the impedance ZA of the A-band signal at the output of B-band filter 124 does not _{match Z0} . Because the A-band is in the stopband of the B-band filter, the reflection coefficient | _ΓA | of this mismatch is approximately unity. However, the phase of this reflection depends on the filter design. Therefore, the impedance _ZA of the A-band signal at the output of B-band filter 122 can be any significantly different mismatch value, much larger or much smaller than _Z0 , resulting in the situation where | _ΓA | is ≈ 1.

Ideally, B-band filter 124 should appear as an open circuit to A-band signals. However, B-band filter 124 may not provide this ideal open-circuit impedance to A-band signals. Therefore, the impedance Z _A of the A-band signals at the output of B-band filter 124 can be expressed in the complex form Z _A = _RA + jX _A , where the real part (resistance _RA ) and the imaginary part (reactance X _A ) cause the impedance Z _A to be significantly away from an open circuit state (where one or both of X _A and _RA are approximately infinite). As shown in FIG8 , second phase shift circuit 154 can be configured to substantially maintain Z _o for Z _B and adjust Z _A from _RA + jX _A to (or close to) an open circuit state.

As shown in FIG8 , CA configuration 100 with first and second signal paths configured in the aforementioned manner allows the combined A-band and B-band signals from their respective paths to be impedance-matched to the LNA, while substantially blocking out-of-band frequency components. Consequently, noise figure performance can be improved without requiring the use of an additional high-Q filter, such as a SAW filter.

Figures 9-15 illustrate two frequency band paths and examples of how the phase shifting circuit can produce the desired impedance along these paths as described with reference to Figure 8. In Figure 9, two isolated receive (Rx) paths associated with example frequency bands B3 (1,805-1,880 MHz) and B1/4 (2,110-2,170 MHz) are shown. The B3 band path can be considered an example of the general A-band path of Figure 8, and the B1/4 band path can be considered an example of the general B-band path.

In Figure 9, the isolated B3-band path includes a B3 filter "A" between the antenna node (ANT B3) and the output node (RX B3). However, the B3-band path does not include a phase-shifting circuit between the B3 filter and the output node (RX B3). Similarly, the isolated B1/4-band path includes a B1/4 filter "B" between the antenna node (ANT B1) and the output node (RX B1), but does not include a phase-shifting circuit.

10 shows example Smith plots of complex impedance values at the RX B1 node (upper left plot), the BX B3 node (upper right plot), the ANT B1 node (lower left plot), and the ANT B3 node (lower right plot) for the circuit of FIG. 9 over a frequency sweep range between 1.792 GHz and 2.195 GHz. More specifically, the impedance values of points m28 and m29 correspond to the lower limit (1.805 GHz) and the upper limit (1.880 GHz) of the B3 Rx frequency band at the RX B1 node, respectively; the impedance values of points m6 and m14 correspond to the lower limit (2.110 GHz) and the upper limit (2.170 GHz) of the B1 Rx frequency band at the RX B3 node, respectively; the impedance values of points m5 and m7 correspond to the lower limit (1.805 GHz) and the upper limit (1.880 GHz) of the B3 Rx frequency band at the ANT B1 node, respectively; and the impedance values of points m3 and m4 correspond to the lower limit (2.110 GHz) and the upper limit (2.170 GHz) of the B1 Rx frequency band at the ANT B3 node, respectively.

In FIG. 10 , each of the impedance ranges (m28 to m29, m6 to m14, m5 to m7, m3 to m4) can be seen to move significantly from the open circuit impedance position on the Smith plot while remaining close to the outer boundaries of the Smith plot.

FIG11 illustrates two isolated receive (Rx) paths associated with example frequency bands B3 Rx (1,805-1,880 MHz) and B1/4 Rx (2,110-2,170 MHz), each of which includes a first phase-shifting circuit, a filter, and a second phase-shifting circuit between its antenna node and output node. More specifically, the B3 path includes a first phase-shifting circuit 200, a B3 filter, and a second phase-shifting circuit 202 between the antenna node (ANT B3) and the output node (RX B3). Similarly, the B1/4 path includes a first phase-shifting circuit 204, a B1/4 filter, and a second phase-shifting circuit 206 between the antenna node (ANT B1) and the output node (RX B1).

12 shows example Smith plots of complex impedance values for the circuit of FIG. 11 at the RX B1 node (upper left plot), the BX B3 node (upper right plot), the ANTB1 node (lower left plot), and the ANT B3 node (lower right plot). Similar to the example of FIG. 10 , the impedance values of points m28 and m29 correspond to the lower limit (1.805 GHz) and the upper limit (1.880 GHz) of the B3 Rx frequency band at the RX B1 node, respectively; the impedance values of points m6 and m14 correspond to the lower limit (2.110 GHz) and the upper limit (2.170 GHz) of the B1 Rx frequency band at the RX B3 node, respectively; the impedance values of points m5 and m7 correspond to the lower limit (1.805 GHz) and the upper limit (1.880 GHz) of the B3 Rx frequency band at the ANT B1 node, respectively; and the impedance values of points m3 and m4 correspond to the lower limit (2.110 GHz) and the upper limit (2.170 GHz) of the B1 Rx frequency band at the ANT B3 node, respectively.

In FIG12 , it can be seen that each of the four graphs is rotated by an amount such that each of the impedance ranges (m28 to m29, m6 to m14, m5 to m7, and m3 to m4) straddles the Im(Z)=0 line and has a large Re(Z) value such that it is at or near an open-circuit impedance position on the Smith plot. In some embodiments, the amount of delay applied by each delay element in FIG11 can be selected to produce the amount of rotation shown in the corresponding Smith plot.

As described with reference to Figures 11 and 12 , it is desirable that the antenna side of the filter be adjusted to present a high impedance to the relative frequency band signal. In some embodiments, the antenna side of a duplexer (e.g., a B3-B1/4 duplexer) can be configured to present a desired high impedance to the relative frequency band (e.g., a B3-band signal or a B1/4-band signal) of the frequency band filter (e.g., a B1/4-band filter or a B3-band filter) entering the duplexer.

FIG13 shows two example receive (Rx) paths of FIG11 connected at their ends to create a common antenna node (ANT B1/B3) and a common output node (RX B1/B3). In some embodiments, the coupled paths can be configured such that each band path provides a substantially open impedance to out-of-band signals (e.g., a B1/4 band signal in the B3 band path, and a B3 band signal in the B1/4 band path), as shown in FIG11 and FIG12 , and when coupled together as shown in FIG13 , provide matched impedances for both band signals at both the common antenna node (ANT B1/B3) and the common output node (RX B1/B3).

FIG14 shows an example Smith chart of complex impedance values at the common output node RX B1/B3 (upper left panel) and the common antenna node (ANT B1/B3) (lower left panel) for the circuit of FIG13 . For the RX B1/B3 node, the impedance values of points m53 and m54 correspond to the B3 frequency bands of 1.805 GHz and 1.880 GHz, respectively, and the impedance values of points m55 and m56 correspond to the B1 frequency bands of 2.110 GHz and 2.170 GHz, respectively. All points m54, m55, m56, and m56 are clustered around the center of the Smith chart, indicating that the impedance of the RX B1/B3 node is well matched to 50 ohms at substantially all frequencies in both bands B1 and B3. This occurs because each path in its own frequency band is generally not interfered with by the other path; thus the combined circuit exhibits matching in frequency band B1 that is substantially determined by the B1 path alone, and matching in frequency band B3 that is substantially determined by the B3 path alone, even though the paths are physically tied together.

Similarly, for the ANT B1/B3 node, the impedance values of points m46 and m47 correspond to frequencies of 1.805 GHz and 1.880 GHz, respectively; the impedance values of points m48 and m49 correspond to B1 band frequencies of 2.110 GHz and 2.170 GHz, respectively. All points m46, m47, m48, and m49 are clustered around the center of the Smith chart, indicating that the impedance of the ANT B1/B3 node is well matched to 50 ohms at substantially all frequencies in both bands B1 and B3. This occurs because each path in its own frequency band is generally not interfered with by the other path; therefore, the combined circuit exhibits matching in band B1 that is substantially determined by the B1 path alone, and matching in band B3 that is substantially determined by the B3 path alone, even though the paths are physically tied together.

FIG14 further illustrates the distribution of the reflection coefficient S11 (S(4,4) in FIG14 ) at the ANT B1/B3 node (lower right panel), and the distribution of the reflection coefficient S22 (S(5,5) in FIG14 ) at the RX B1/B3 node (upper right panel) for the circuit of FIG13 . In the distributions of S11 (S(4,4)) and S22 (S(5,5)), matching is evident in each of the two RX bands (B3 RX and B1 RX).

Figure 15 shows the spectral response of the B3 receive path and the independent spectral response of the B1 receive path for the circuit of Figure 9 in the upper left panel. The same response applies essentially unchanged to the circuit of Figure 11, as the delay added in Figure 11 generally affects only the phase and not the amplitude of each path. The peak gain in the B3 RX band (e.g., the B3 passband) is indicated as 230, and the peak gain in the B1 RX band (e.g., the B1 passband) is indicated as 232. Note that each path exhibits greater than 30 dB of attenuation in the relative bands.

In Figure 15, the upper right panel shows a single spectral response for the circuit of Figure 13. Note that this single response exhibits two passbands, the B3 RX passband indicated at 234, and the B1 RX passband indicated at 236.

In FIG15 , the lower left panel shows the overlap of B3 RX passband 230 presented by the independent B3 receive paths of FIG9/FIG11 and B3 RX passband 234 presented by the combined circuit of FIG13 . The lower right panel shows the overlap of B1 RX passband 232 presented by the independent B1 receive paths of FIG9/FIG11 and B1 RX passband 236 presented by the combined circuit of FIG13 . In both examples in the lower right and lower left panels, it can be seen that each passband of the combined circuit of FIG13 is substantially similar to the passband of the corresponding independent receive paths from which it is formed, both in bandwidth and characteristic waveform. Furthermore, the gain distribution for a given frequency band before the addition of the corresponding phase-shifting circuit and the combined paths is only slightly higher than the distribution after the addition of the phase-shifting circuit and the combination of the paths. Thus, it can be seen that phase-shifting circuits incorporating one or more of the features described herein can be configured to provide desired functionality with little or no loss.

FIG16A shows a more detailed example of a phase shifting circuit, such as circuits 212 and 216 of FIG13-15 . In FIG16A , CA configuration 100 can be an example of CA configuration 100 of FIG8 . An input RF signal (RF_IN) from an antenna can be received at input node 102 . A duplexer 260 is shown coupled to input node 102 to receive the input RF signal. The received signal can be processed by filters 122 and 124 configured to provide bandpass functionality for Band A and Band B. Examples of these frequency bands are described in more detail herein. In some embodiments, duplexer 260 can be configured to provide impedance matching for the input RF signal, as described herein with reference to FIG11-15 .

The outputs of the bandpass filters 122, 124 are shown as being routed to a first path including a first phase-shifting circuit 152 and a second path including a second phase-shifting circuit 154. The first path is shown as further including a switch S1 between the first phase-shifting circuit 152 and a common output node. The second path is shown as further including a switch S2 between the second phase-shifting circuit 154 and the common output node.

A common output node receiving processed signals from the aforementioned first and second paths is shown coupled to the input of LNA 120. LNA 120 is shown producing an amplified output signal at node 114 (RF_OUT).

The first phase shifting circuit 152 is shown as including capacitors C5 and C6 arranged in series between its input (the output from the Band A filter 122) and the switch S 1. An inductor L5 is shown coupling the node between C5 and C6 to ground.

The second phase shifting circuit 154 is shown as including capacitors C2 and C3 arranged in series between its input (the output from the Band B filter 124) and switch S2. An inductor L4 is shown coupling the node between C2 and C3 to ground.

FIG16B shows a more detailed example of a phase shifting circuit implemented in a lowpass phase shifter configuration, such as circuits 212 and 216 of FIG13-15 . In FIG16B , CA configuration 100 can be an example of CA configuration 100 of FIG8 . An input RF signal (RF_IN) from an antenna can be received at input node 102 . A duplexer 260 is shown coupled to input node 102 to receive the input RF signal. The received signal can be processed by filters 122 and 124 configured to provide bandpass functionality for Band A and Band B. Examples of these frequency bands are described in more detail herein. In some embodiments, duplexer 260 can be configured to provide impedance matching for the input RF signal, as described herein with reference to FIG11-15 .

The outputs of the bandpass filters 122, 124 are shown as being routed to a first path including a first phase-shifting circuit 152 and a second path including a second phase-shifting circuit 154. The first path is shown as further including a switch S1 between the first phase-shifting circuit 152 and a common output node. The second path is shown as further including a switch S2 between the second phase-shifting circuit 154 and the common output node.

A common output node receiving processed signals from the aforementioned first and second paths is shown coupled to the input of LNA 120. LNA 120 is shown producing an amplified output signal (RF_OUT) at node 114.

The first phase shifting circuit 152 is shown to include inductors L5' and L6' arranged in series between its input (the output from the Band A filter 122) and the switch S 1. A capacitor C5' is shown coupling the node between L5' and L6' to ground.

The second phase shifting circuit 154 is shown to include inductors L2' and L3' arranged in series between its input (the output from the Band B filter 124) and switch S2. Capacitor C4' is shown coupling the node between L2' and L3' to ground.

In some embodiments, for example frequency bands B3 RX and B1/4 RX, various functions as described herein with reference to, for example, FIGs. 8 and 13 may be achieved with capacitance and inductance values for the configuration example of FIG. 16A , as listed in Table 1 .

Capacitors/Inductors approximation C2 1.71pF C3 1.71pF C5 5.38pF C6 6.38pF L4 4nH L5 7.668nH

Table 1

It will be appreciated that for other frequency band pairs, the capacitance and inductance values may be selected accordingly.It will also be appreciated that the same or similar functions may be accomplished with appropriate values for the example circuit elements of FIG. 16B.

In some embodiments, some or all of the capacitors and/or inductors may be implemented as part of a signal path or other conductive feature, as a lumped element, or any combination thereof.

FIG17 illustrates a process 280 that may be implemented to manufacture a device having one or more features described herein. At block 282, circuitry having at least duplexer functionality may be assembled or provided on a substrate. In various examples, carrier aggregation (CA) is described in the context of a duplexer; however, it will be understood that CA may also be implemented for more than two frequency bands (e.g., using a multiplexer). In some embodiments, the duplexer may be implemented as a device; and such a device may be assembled on a substrate.

At block 284, a first phase-shifting circuit may be formed or provided between a first output of the duplexer circuit and an input of the first switch. At block 286, a second phase-shifting circuit may be formed or provided between a second output of the duplexer circuit and an input of the second switch. At block 288, the output of the first switch and the output of the second switch may be coupled to a common node. In some embodiments, coupling the first and second phase-shifting circuits to a common node via their respective switches may facilitate operation of the device in either CA mode or non-CA mode.

The common node may be coupled to the input of a low noise amplifier (LNA) at block 290. In some embodiments, this aggregation of two signal paths into one LNA may allow the LNA to operate in either a CA mode or a non-CA mode, as determined by the state of the switch.

In some embodiments, the device depicted in FIG. 17 may be a module configured for RF applications. FIG. 18 shows a block diagram of an RF module 300 (e.g., a front-end module) having a packaging substrate 302, such as a laminate substrate. This module may include one or more LNAs; in some embodiments, the LNAs may be implemented on a semiconductor die 304. The LNAs implemented on the die may be configured to receive RF signals via a signal path as described herein. The LNAs may also benefit from one or more advantageous features related to improved carrier aggregation (CA) functionality as described herein.

The module 300 may further include a plurality of switches implemented on one or more semiconductor dies 306. The switches may be configured to provide various switching functions as described herein, including providing and/or facilitating isolation, enabling/disabling a CA mode of operation, and band selection in a non-CA mode.

Module 300 may further include one or more duplexers and/or multiple filters (collectively denoted as 310) configured to process RF signals. The duplexers/filters may be implemented as a surface mount device (SMD), as part of an integrated circuit (IC), or a combination thereof. The duplexers/filters may include or be based on, for example, SAW filters and may be configured as high-Q devices.

In Figure 18, a plurality of phase shifting circuits are collectively indicated as 308. The phase shifting circuits may include one or more features as described herein to provide improved isolation between paths associated with different frequency bands operating in CA mode, among others.

Figure 19 shows an example 400 of an RF architecture including one or more features as described herein. In some embodiments, the architecture can be implemented on a module 300 such as the example described with reference to Figure 18. It will be understood that the architecture 400 of Figure 19 is not necessarily limited to a module.

The example architecture 400 of FIG19 may include multiple signal paths configured for receiving and/or transmitting RF signals. The architecture 400 may also include antenna switch circuitry 404 coupled to the antenna port 402. The antenna switch circuitry may be configured to route RF signals within the cellular frequency range to multiple paths associated with different cellular frequency bands. In the example shown, the antenna switch circuitry 404 includes a single-pole double-throw (SP2T) switch, wherein the pole is coupled to the antenna port 402.

In the context of the example RX paths, the first path is configured for the B2/B25/4 frequency bands and the second path is configured for the B3/B1/4 frequency bands. RF signals associated with the frequency bands are shown as being processed by their respective filters 406.

Signals in the B2/B25/4 frequency bands (e.g., 1.930 to 1.995 GHz and 2.110 to 2.155 GHz) of the first path can be carrier aggregated as described herein and amplified by the LNAs in LNA bank 410. As described herein, carrier aggregation in the B2/B25/4 frequency bands can include multiple phase shifting circuits implemented between the B2/B25/4 duplexer and the LNAs. Furthermore, as described herein, the paths between the phase shifting circuits and the LNAs can include respective switches to enable operation in both CA and non-CA modes.

Signals in the B3/B1/4 frequency bands (e.g., 1.805 to 1.880 GHz and 2.110 to 2.170 GHz) of the second path can be carrier aggregated as described herein and amplified by the LNAs in LNA bank 410. The LNAs can be configured to provide bandwidth coverage of, for example, 1.805 to 2.170 GHz. As described herein, the carrier aggregation can include multiple phase shifting circuits implemented between the B3/B1/4 duplexer and the LNAs. Furthermore, as described herein, the paths between the phase shifting circuits and the LNAs can include respective switches to enable operation in both CA and non-CA modes.

The amplified signal from the LNA may be routed to a band select switch 412. The band select switch 412 is shown coupled to a node 416 to allow further processing of the amplified RF signal from the selected LNA.

In some implementations, an architecture, device, and/or circuit having one or more features described herein may be included in an RF device, such as a wireless device. The architecture, device, and/or circuit may be implemented directly in the wireless device, in the form of one or more modules as described herein, or in a combination thereof. In some embodiments, the wireless device may include, for example, a cellular phone, a smartphone, a handheld wireless device with or without telephone functionality, a wireless tablet, a wireless router, a wireless access point, a wireless base station, and the like. Although described in the context of a wireless device, it will be understood that one or more features of the present disclosure may also be implemented in other RF systems, such as a base station.

FIG20 schematically illustrates an example wireless device 500 having one or more advantageous features described herein. In some embodiments, the advantageous features may be implemented in a front end (FE) module 300 and/or in an architecture 400 as described herein. One or more of the features may also be implemented in a main antenna switch module (ASM) 514. In some embodiments, the FEM/architecture may include more or fewer components than those indicated by the dashed boxes.

The PAs in the PA module 512 can receive their respective RF signals from the transceiver 510, which can be configured and operated to generate RF signals to be amplified and transmitted, as well as to process received signals. The transceiver 510 is shown as interacting with the baseband subsystem 508, which is configured to provide conversion between data and/or voice signals suitable for the user and RF signals suitable for the transceiver 510. The transceiver 510 is also shown as being connected to the power management component 506, which is configured to manage power for the operation of the wireless device 500. This power management can also control the operation of the baseband subsystem 508 and other components of the wireless device 500.

The baseband subsystem 508 is shown connected to the user interface 502 to facilitate various inputs and outputs of voice and/or data provided to and received from the user. The baseband subsystem 508 may also be connected to the memory 504, which is configured to store data and/or instructions to facilitate the operation of the wireless device and/or provide information storage for the user.

In the example wireless device 500, the front-end module 300/architecture 400 may include one or more carrier aggregation-capable signal paths configured to provide one or more functions as described herein. The signal paths may communicate with the antenna switch module (ASM) 404 via their respective duplexers. In some embodiments, at least some of the signals received via the diversity antenna 530 may be routed from the ASM 404 to one or more low noise amplifiers (LNAs) 518 in the manner described herein. Amplified signals from the LNAs 518 are shown routed to the transceiver 510.

A variety of other wireless device configurations can utilize one or more features described herein. For example, a wireless device need not be a multi-band device. In another example, a wireless device can include additional antennas such as a diversity antenna, and additional connectivity features such as Wi-Fi, Bluetooth, and GPS.

Examples related to diversity reception (DRx) implementation:

Using one or more primary antennas and one or more diversity antennas in a wireless device can improve the quality of signal reception. For example, the diversity antennas can provide additional sampling of RF signals near the wireless device. In addition, the wireless device's transceiver can be configured to process signals received by the primary and diversity antennas to obtain a received signal with higher energy and/or improved fidelity when compared to a configuration using only the primary antennas.

To reduce correlation between signals received by the main and diversity antennas and/or to enhance antenna isolation, the main and diversity antennas can be separated by a relatively large physical distance within the wireless device. For example, the diversity antenna can be located near the top of the wireless device while the main antenna can be located near the bottom of the wireless device, or vice versa.

The wireless device can use the primary antenna to transmit or receive signals by routing corresponding signals from or to the transceiver through the antenna switch module. To meet or exceed design specifications, the transceiver, antenna switch module, and/or primary antenna can be located in relatively close physical proximity to each other in the wireless device. Configuring the wireless device in this manner can provide relatively low signal loss, low noise, and/or high isolation.

In the aforementioned example, the physical proximity of the primary antenna to the antenna switch module may result in the diversity antenna being positioned relatively far from the antenna switch module. In this configuration, the relatively long signal path between the diversity antenna and the antenna switch module may result in significant losses and/or additional losses associated with signals received by the diversity antenna. Therefore, processing signals received by the diversity antenna (including implementing one or more features described herein) in close proximity to the diversity antenna may be advantageous.

21 shows that in some embodiments, one or more features of the present disclosure can be implemented in a diversity receive (DRx) module 300. This module can include a packaging substrate 302 (e.g., a laminate substrate) configured to house a plurality of components and to provide or facilitate electrical connections associated with these components.

In the example of FIG21 , the DRx module 300 can be configured to receive an RF signal at an input 320 from a diversity antenna (not shown in FIG21 ) and route the RF signal to a low noise amplifier (LNA) 332. It will be appreciated that this routing of the RF signal can include carrier aggregation (CA) and/or non-CA configurations. It will also be appreciated that while one LNA (e.g., a wideband LNA) is shown, multiple LNAs may be present in the DRx module 300. Depending on the type of LNA and operating mode (e.g., CA or non-CA), the output 334 of the LNA 332 can include one or more frequency components associated with one or more frequency bands.

In some embodiments, some or all of the aforementioned routing of RF signals between input 320 and LNA 332 can be implemented by components of one or more switches 322 between input 320 and duplexer and/or filter components (collectively designated 324), and components of one or more switches 330 between duplexer/filter components 324 and LNA 332. In some embodiments, switch components 322, 330 can be implemented, for example, on one or more silicon-on-insulator (SOI) die. In some embodiments, some or all of the aforementioned routing of RF signals between input 320 and LNA 332 can be implemented without requiring some or all of the switches associated with switch components 322, 330.

In the example of FIG21 , the duplexer/filter assembly 324 is depicted as including two example duplexers 326 and two independent filters 328. It will be appreciated that the DRx module 300 can have a greater or lesser number of duplexers, and a greater or lesser number of independent filters. The duplexer/filter can be implemented, for example, as a surface mount device (SMD), as part of an integrated circuit (IC), or as a combination thereof. The duplexer/filter can include or be based on, for example, a SAW filter, and can be configured as a high-Q device.

In some embodiments, the DRx module 300 can include a control component, such as a MIPI RFFE interface 340, configured to provide and/or facilitate control functions associated with some or all of the switch components 322, 330 and the LNA 332. The control interface can be configured to operate on one or more I/O signals 342.

22 illustrates that in some embodiments, a DRx module 300 having one or more features as described herein (e.g., the DRx module 300 of FIG. 21 ) may be included in an RF device such as a wireless device 500. In the wireless device, components such as a user interface 502, memory 504, power management 506, a baseband subsystem 508, a transceiver 510, a power amplifier (PA) 512, an antenna switch module (ASM) 514, and an antenna 520 may be substantially similar to the example of FIG. 20 .

In some embodiments, the DRx module 300 can be implemented between one or more diversity antennas and the ASM 514. This configuration can allow RF signals received via the diversity antenna 530 to be processed (including, in some embodiments, amplified by an LNA) with little or no loss and/or with little or no added noise to the RF signals from the diversity antenna 530. The processed signals from the DRx module 300 can then be routed to the ASM via one or more signal paths 532, which can be relatively lossy.

22 , the RF signal from the DRx module 300 can be routed through one or more receive (Rx) paths to the transceiver 510 via the ASM 514. Some or all of the Rx paths can include their own LNAs. In some embodiments, the RF signal from the DRx module 300 may or may not be further amplified using such LNAs.

One or more features of the present disclosure may be implemented with respect to various cellular frequency bands as described herein. Examples of these frequency bands are listed in Table 2. It will be understood that at least some of the frequency bands may be divided into sub-bands. It will also be understood that one or more features of the present disclosure may be implemented with respect to frequency ranges that do not have designations such as the examples in Table 2.

frequency band model Tx frequency range (MHz) Rx frequency range (MHz) B1 FDD 1,920-1,980 2,110-2,170 B2 FDD 1,850-1,910 1,930-1,990 B3 FDD 1,710-1,785 1,805-1,880 B4 FDD 1,710-1,755 2,110-2,155

B5 FDD 824-849 869-894 B6 FDD 830-840 875-885 B7 FDD 2,500-2,570 2,620-2,690 B8 FDD 880-915 925-960 B9 FDD 1,749.9-1,784.9 1,844.9-1,879.9 B10 FDD 1,710-1,770 2,110-2,170 B11 FDD 1,427.9-1,447.9 1,475.9-1,495.9 B12 FDD 699-716 729-746 B13 FDD 777-787 746-756 B14 FDD 788-798 758-768 B15 FDD 1,900-1,920 2,600-2,620 B16 FDD 2,010-2,025 2,585-2,600 B17 FDD 704-716 734-746 B18 FDD 815-830 860-875 B19 FDD 830-845 875-890 B20 FDD 832-862 791-821 B21 FDD 1,447.9-1,462.9 1,495.9-1,510.9 B22 FDD 3,410-3,490 3,510-3,590 B23 FDD 2,000-2,020 2,180-2,200 B24 FDD 1,626.5-1,660.5 1,525-1,559 B25 FDD 1,850-1,915 1,930-1,995 B26 FDD 814-849 859-894 B27 FDD 807-824 852-869 B28 FDD 703-748 758-803 B29 FDD N/A 716-728 B30 FDD 2,305-2,315 2,350-2,360 B31 FDD 452.5-457.5 462.5-467.5 B33 TDD 1,900-1,920 1,900-1,920 B34 TDD 2,010-2,025 2,010-2,025 B35 TDD 1,850-1,910 1,850-1,910

B36 TDD 1,930-1,990 1,930-1,990 B37 TDD 1,910-1,930 1,910-1,930 B38 TDD 2,570-2,620 2,570-2,620 B39 TDD 1,880-1,920 1,880-1,920 B40 TDD 2,300-2,400 2,300-2,400 B41 TDD 2,496-2,690 2,496-2,690 B42 TDD 3,400-3,600 3,400-3,600 B43 TDD 3,600-3,800 3,600-3,800 B44 TDD 703-803 703-803

Table 2

For purposes of this description, it will be understood that "multiplexer," "multiplexing," and the like may include "duplexer," "duplex," and the like.

Unless the context clearly requires otherwise, throughout the specification and claims, the words "comprise", "comprising", etc. are to be interpreted as inclusive, and not in an exclusive or exhaustive sense; that is, in the sense of "including but not limited to". The word "coupled" as generally used herein refers to two or more elements that can be connected directly or through one or more intermediate elements. In addition, the words "herein", "above", "below" and words of similar meaning, when used in this application, shall refer to this application as a whole and not to any specific part of this application. Where the context permits, words used in the singular or plural in the above specific embodiments may also include the plural and singular, respectively. The word "or" when referring to a list of two or more items covers all of the following interpretations of the word: any item in the list, all items in the list, and any combination of items in the list.

The above detailed description of the embodiments of the present invention is not intended to be exhaustive or to limit the invention to the precise form disclosed above. As will be appreciated by those skilled in the art, although specific embodiments and examples of the present invention are described above for illustrative purposes, various equivalent modifications are possible within the scope of the present invention. For example, although processes or blocks are presented in a given order, alternative embodiments may perform routines with steps in a different order, or employ systems with blocks in a different order, and may delete, move, add, subdivide, combine and/or modify some processes or blocks. Each of these processes or blocks may be implemented in a variety of different ways. In addition, although processes or blocks are sometimes shown as being executed serially, alternatively, these processes or blocks may be executed in parallel, or may be executed at different times.

The teachings of the invention provided herein can be applied to other systems, not necessarily the systems described above.The elements and acts of the various embodiments described above can be combined to provide further embodiments.

Although some embodiments of the present invention have been described, these embodiments are presented by way of example only and are not intended to limit the scope of the present disclosure. Indeed, the novel methods and systems described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions, and changes in the form of the methods and systems described herein may be made without departing from the spirit of the present disclosure. The accompanying claims and their equivalents are intended to cover such forms or modifications as would fall within the scope and spirit of the present disclosure.

Claims (15)

1. A carrier aggregation circuit, comprising: The input node is configured to be coupled to the diversity receiving antenna and the common node is configured to be coupled to the input of the low-noise amplifier; A first receiving path between an input node and a common node, the first receiving path including a first filter that provides matched impedance for a first frequency band and mismatched impedance for a second frequency band; A second receiving path between the input node and the common node, the second receiving path including a second filter configured to provide matched impedance for a second frequency band and mismatched impedance for a first frequency band, the first filter and the second filter being part of a duplexer including an input port configured to receive signals from the antenna through the input node; A first phase offset circuit implemented along a first receiving path is used to maintain a matched impedance in a first frequency band of the first receiving path, and to adjust the mismatched impedance in a second frequency band of the first receiving path to an open-circuit impedance in the second frequency band; and A second phase offset circuit implemented along the second receiving path is used to maintain the matched impedance of the second frequency band of the second receiving path, and to adjust the mismatched impedance of the first frequency band of the second receiving path to the open impedance of the first frequency band. 2. The carrier aggregation circuit of claim 1, wherein the low-noise amplifier is configured to amplify the received radio frequency signal in a frequency band corresponding to the first frequency band and the second frequency band. 3. The carrier aggregation circuit of claim 1, wherein the first phase offset circuit comprises two series capacitors and an inductor shunt path coupled to a node between the two capacitors and ground. 4. The carrier aggregation circuit of claim 3, wherein the second phase offset circuit comprises two series capacitors and an inductor shunt path coupled to a node between the two capacitors and ground. 5. The carrier aggregation circuit of claim 1, wherein the first phase offset circuit comprises two series inductors and a capacitive shunt path coupled to a node between the two inductors and ground. 6. The carrier aggregation circuit of claim 5, wherein the second phase offset circuit comprises two series inductors and a capacitive shunt path coupled to a node between the two inductors and ground. 7. The carrier aggregation circuit of claim 1, wherein each of the first receiving path and the second receiving path further comprises a switch to allow the carrier aggregation circuit to operate in carrier aggregation mode or non-carrier aggregation mode. 8. The carrier aggregation circuit of claim 7, wherein the switch for the first receiving path is between the first phase offset circuit and the common node, and the switch for the second receiving path is between the second phase offset circuit and the common node. 9. A radio frequency module, comprising: A packaging substrate configured to accommodate multiple components; and A carrier aggregation circuit implemented on the package substrate includes: an input node configured to be coupled to a diversity receiving antenna and a common node configured to be coupled to an input of a low-noise amplifier; a first receiving path between the input node and the common node, the first receiving path including a first filter providing matched impedance for a first frequency band and mismatched impedance for a second frequency band; and a second receiving path between the input node and the common node, the second receiving path including a second filter configured to provide matched impedance for the second frequency band and mismatched impedance for the first frequency band. The first filter and the second filter are part of a duplexer, which includes an input port configured to receive a signal from an antenna via an input node; a first phase offset circuit implemented along a first receiving path for maintaining a matched impedance in a first frequency band of the first receiving path and adjusting a mismatched impedance in a second frequency band of the first receiving path to an open impedance in the second frequency band; and a second phase offset circuit implemented along a second receiving path for maintaining a matched impedance in a second frequency band of the second receiving path and adjusting a mismatched impedance in a first frequency band of the second receiving path to an open impedance in the first frequency band. 10. The radio frequency module of claim 9, wherein each of the first filter and the second filter comprises a surface acoustic wave filter. 11. The radio frequency module of claim 9, further comprising a low-noise amplifier implemented on the package substrate, the low-noise amplifier having an input coupled to the common node to receive a combined signal from the carrier aggregation circuit. 12. The radio frequency module according to claim 9, wherein the radio frequency module is a diversity receiving module. 13. The radio frequency module of claim 9, wherein each of the first phase offset circuit and the second phase offset circuit comprises a capacitor and an inductor. 14. A wireless device, comprising: The receiver is configured to process radio frequency signals; A carrier aggregation circuit communicating with the receiver, the carrier aggregation circuit comprising: an input node configured to be coupled to a diversity receiving antenna and a common node configured to be coupled to an input of a low-noise amplifier; a first receiving path between the input node and the common node, the first receiving path including a first filter providing matched impedance for a first frequency band and mismatched impedance for a second frequency band; and a second receiving path between the input node and the common node, the second receiving path including a second filter configured to provide matched impedance for the second frequency band and mismatched impedance for the first frequency band. The first filter and the second filter are part of a duplexer, the duplexer including an input port configured to receive a signal from an antenna via an input node; a first phase shift circuit implemented along a first receiving path for maintaining a matched impedance in a first frequency band of the first receiving path and adjusting a mismatched impedance in a second frequency band of the first receiving path to an open-circuit impedance in the second frequency band; and a second phase shift circuit implemented along a second receiving path for maintaining a matched impedance in a second frequency band of the second receiving path and adjusting a mismatched impedance in a first frequency band of the second receiving path to an open-circuit impedance in the first frequency band; and A diversity receiving antenna that communicates with the carrier aggregation circuit, the diversity receiving antenna being configured to receive the radio frequency signal. 15. The wireless device of claim 14, further comprising an antenna switch module configured to route the radio frequency signal from the diversity receiving antenna to the receiver, such that the carrier aggregation circuitry is implemented between the diversity receiving antenna and the antenna switch module.