HK1118649A - Pilot transmission and channel estimation for a communication system utilizing frequency division multiplexing - Google Patents

Description

Pilot transmission and channel estimation for communication systems employing frequency division multiplexing

Background

Priority requirements under 35 U.S.C. § 119

This patent application claims priority from provisional application No.60/659,526 entitled "Estimation for Pilot Design and Channel interleaved frequency Division Multiple Access Communication" filed on 7.3.2005 and assigned to the assignee of the present invention and hereby expressly incorporated by reference herein.

I. Field of the invention

The present invention relates generally to communications, and more particularly to pilot transmission and channel estimation for communication systems.

II. background

Orthogonal Frequency Division Multiplexing (OFDM) is a multi-carrier modulation technique that divides the overall system bandwidth into multiple (K) orthogonal subbands. These subbands are also referred to as tones, subcarriers, and frequency bins. In OFDM, each subband is associated with a respective subcarrier that may be modulated with data.

OFDM has certain desirable characteristics such as high spectral efficiency and robustness to multipath effects. However, a major drawback of OFDM is the high peak-to-average power ratio (PAPR), which means that the ratio of the peak power to the average power of the OFDM waveform can be high. The high PAPR of the OGDM waveform results from the possible in-phase (or coherent) addition of all subcarriers when modulated independently with data. In fact, it can be shown that for OFDM, the peak power can be up to K times the average power.

The high PAPR of the OFDM waveform is undesirable and may degrade performance. For example, large peaks in the OFDM waveform may cause the power amplifier to operate in a highly non-linear region or may clip, which in turn will cause intermodulation distortion and other artifacts that may degrade signal quality. The signal quality degradation will adversely affect the performance of channel estimation, data detection, etc.

There is thus a need in the art for techniques that can mitigate the deleterious effects of high PAPR in multi-carrier modulation.

Summary of the invention

Pilot transmission techniques and channel estimation techniques capable of avoiding high PAPR are described herein. Pilots may be generated based on a polyphase sequence and using single carrier frequency division multiple access (SC-FDMA). A polyphase sequence is a sequence with good temporal characteristics (e.g. a constant temporal envelope) and good spectral characteristics (e.g. a flat spectrum). SC-FDMA includes (1) Interleaved FDMA (IFDMA) where data and/or pilot is transmitted on subbands evenly spaced across the K total subbands and (2) Localized FDMA (LFDMA) where data and/or pilot is typically transmitted on adjacent subbands among the K total subbands.

In one embodiment of pilot transfer using IFDMA, a first sequence of pilot symbols is formed based on a polyphase sequence and the sequence is replicated multiple times to obtain a second sequence of pilot symbols. A phase ramp may be applied to the second pilot symbol sequence to obtain a third output symbol sequence. A cyclic prefix is added to the third sequence of output symbols to form an IFDMA symbol, which is transmitted in the time domain via a communication channel. The pilot symbols may be multiplexed with the data symbols using Time Division Multiplexing (TDM), Code Division Multiplexing (CDM), and/or some other multiplexing scheme.

In one embodiment of pilot transfer using LFDMA, a first sequence of pilot symbols is formed based on a polyphase sequence and transformed to the frequency domain to obtain a second sequence of frequency-domain symbols. A third sequence of frequency-domain symbols is formed by mapping a second sequence of frequency-domain symbols onto a group of subbands used for pilot transmission and zero symbols onto the remaining subbands. The third sequence of symbols is transformed to the time domain to obtain a fourth sequence of output symbols. A cyclic prefix is added to the fourth sequence of output symbols to form an LFDMA symbol, which is transmitted in the time domain via the communication channel.

In one embodiment of channel estimation, at least one SC-FDMA symbol is received via a communication channel and processed (e.g., demultiplexed for TDM pilot or de-channelized for CDM pilot) to obtain received pilot symbols. The SC-FDMA symbols may be IFDMA symbols or LFDMA symbols. The channel estimate is derived based on the received pilot symbols and using a Minimum Mean Square Error (MMSE) technique, a Least Squares (LS) technique, or some other channel estimation technique. Filtering, thresholding, truncation, and/or tap selection may be performed to obtain an improved channel estimate. Channel estimation may also be improved by performing iterative channel estimation or data-aided channel estimation.

Various aspects and embodiments of the invention are described in further detail below.

Brief description of the drawings

The features and nature of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout.

Fig. 1 illustrates an interleaved subband structure for a communication system.

Fig. 2 illustrates generating IFDMA symbols for a set of N subbands.

Fig. 3 shows a narrowband subband structure.

Fig. 4 illustrates generation of LFDMA symbols for a group of N subbands.

Fig. 5A and 5B illustrate two TDM pilot schemes in which pilot and data are multiplexed across multiple symbol periods and multiple sample periods, respectively.

Fig. 5C and 5D illustrate two CDM pilot schemes in which pilot and data are combined across multiple symbol periods and sampling periods, respectively.

Fig. 6 shows a broadband pilot time-division multiplexed with data.

Fig. 7A illustrates a process for generating pilot IFDMA symbols.

Fig. 7B shows a process for generating pilot LFDMA symbols.

Fig. 8 shows a process of performing channel estimation.

Fig. 9 shows a block diagram of a transmitter and a receiver.

Fig. 10A and 10B illustrate Transmit (TX) data and pilot processors for a TDM pilot scheme and a CDM pilot scheme, respectively.

Fig. 11A and 11B show IFDMA and LFDMA modulators, respectively.

Fig. 12A and 12B illustrate IFDMA demodulators for TDM and CDM pilots, respectively.

Fig. 13A and 13B show LFDMA demodulators for TDM and CDM pilots, respectively.

Detailed description of the invention

The term "exemplary" is used herein to mean "serving as an example, instance, or illustration. Any embodiment or design described herein as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments or designs.

The pilot transmission and channel estimation described herein may be used in various communication systems that employ multi-carrier modulation or perform frequency division multiplexing. For example, the techniques may be used for Frequency Division Multiple Access (FDMA) systems, Orthogonal Frequency Division Multiple Access (OFDMA) systems, SC-FDMA systems, IFDMA systems, LFDMA systems, OFDM-based systems, and the like. These techniques may also be used for the forward link (or downlink) and the reverse link (or uplink).

Fig. 1 illustrates an exemplary subband structure 100 that may be used in a communication system. The system has a total bandwidth of BW MHz divided into K orthogonal subbands, which are assigned indices of l to K. The spacing between adjacent subbands is BW/K MHz. In a spectrally shaped system, some subbands on both ends of the system bandwidth are not used for data/pilot transmission, but rather serve as guard subbands that allow the system to meet spectral mask requirements. Alternatively, the K subbands may be defined over a usable portion of the system bandwidth. For simplicity, the following description assumes that all K total subbands may be used for data/pilot transmission.

For subband structure 100, the K total subbands are arranged into S disjoint groups of subbands, which are also referred to as interlaces. The S groups are disjoint or non-overlapping, since each of the K subbands belongs to only one group. Each group contains N subbands that are evenly distributed across the K total subbands, such that successive subbands in the group are spaced apart by S subbands, where K is S · N. Thus, group u contains subbands u, S + u, 2S + u, …, (N-1). S + u, where u is the group index and u ∈ {1, …, S }. The index u is also a subband offset indicating the first subband in the group. The N subbands in each group are interleaved with the N subbands in each of the other S-1 groups.

Fig. 1 shows a specific subband structure. In general, a subband structure may include any number of groups of subbands, and each group may include any number of subbands. Each group may include the same or a different number of subbands. For example, some groups may include N subbands, while other groups may include 2N, 4N, or some other number of subbands. The subbands in each group are uniformly distributed (i.e., spaced evenly) across the K total subbands to achieve the benefits described below. For simplicity, the following description assumes the use of subband structure 100 in FIG. 1.

The S subband groups may be considered as S channels available for data and pilot transmission. For example, each user may be assigned one subband group, and data and pilot for each user may be transmitted on the assigned subband group. The S users may simultaneously transmit data/pilot to the base station on the S subband groups via the reverse link. The base station may also transmit data/pilot to the S users simultaneously on the S subband groups via the forward link. For each link, up to N modulation symbols may be transmitted on N subbands in each group in each symbol period (in time or frequency) without causing interference to other groups of subbands. A modulation symbol is a complex value corresponding to a point in a signal constellation (e.g., M-PSK, M-QAM, and the like).

For OFDM, the modulation symbols are transmitted in the frequency domain. For each subband group, N modulation symbols may be transmitted on the N subbands in each symbol period. In the following description, a symbol period is a duration of one OFDM symbol, one IFDMA symbol, or one LFDMA symbol. One modulation symbol is mapped to each of the N subbands used for transmission, and a zero symbol (i.e., zero signal value) is mapped to each of the K-N unused subbands. The K modulation and zero symbols are transformed from the frequency domain to the time domain by performing a K-point Inverse Fast Fourier Transform (IFFT) on the K modulation and zero symbols to obtain K time domain samples. These time domain samples may have a high PAPR.

Fig. 2 illustrates generating IFDMA symbols for a set of N subbands. The original sequence of N modulation symbols to be transmitted in one symbol period on N subbands in group u is denoted as { d₁，d₂，d₃，…，d_N} (block 210). This original sequence of N modulation symbols is replicated S times to obtain a spread sequence of K modulation symbols (block 212). The N modulation symbols are transmitted in the time domain and occupy a total of N subbands in the frequency domain. The S copies of the original sequence result in N occupied subbands that are S-spaced apart, and adjacent occupied subbands are separated by S-1 zero-power subbands. The spreading sequence has a comb-like spectrum occupying subband group 1 in fig. 1.

The spread sequence is multiplied by a phase ramp to obtain a sequence of frequency-translated output symbols (block 214). Each output symbol in the frequency-translated sequence may be generated as follows:

x_n＝d_n·e^{-j2π(n-1)(u-1)/K}n is 1, …, K, formula (1)

Wherein d is_nIs the nth modulation symbol in the spreading sequence, and x_nIs the nth output symbol in the frequency translated sequence. Phase ramp e^{-j2π·(n-1)·(u-1)/K}With a phase slope of 2 pi (u-1)/K, determined by the first subband in group u. The terms "n-1" and "u-1" in the exponent of the phase ramp are because the indices n and u start from '1' and not from '0'. Multiplying with the phase ramp in the time domain shifts the comb spectrum of the spreading sequence to the high end of the frequency such that the frequency shifted sequence occupies the subband group u in the frequency domain.

The last C output symbols in the frequency translated sequence are copied to the beginning of the frequency translated sequence to form an IFDMA symbol containing K + C output symbols (block 216). The output symbols of these C copies are often referred to as cyclic prefixes or guard intervals, and C is the cyclic prefix length. The cyclic prefix is used to combat inter-symbol interference (ISI) caused by frequency selective fading, i.e., a frequency response that varies across the system bandwidth. The K + C output symbols in the IFDMA symbol are transmitted in K + C sample periods, one output symbol in each sample period. The symbol period of the IFDMA is the duration of one IFDMA symbol and is equal to K + C sample periods. The sampling period is also often referred to as the chip period.

Since the IFDMA symbols are periodic in the time domain (except for the phase ramp), the IFDMA symbols occupy a set of N equally spaced subbands starting from subband u. Similar to OFDMA, users with different subband offsets occupy different subband groups and are orthogonal to each other.

Fig. 3 illustrates an exemplary narrowband subband structure 300 that may be used in a communication system. For subband structure 300, the K total subbands are arranged into S non-overlapping groups. Each group contains N subbands that are contiguous to each other. In general, N > 1, S > 1, and K ═ S · N, where N and S of the narrowband subband structure 300 may be the same as or different from N and S of the interleaved subband structure in fig. 1. The group v contains the sub-bands (v-1). N +1, (v-1). N +2, …, v.N, where v is the group index and v.e {1, …, S }. In general, a subband structure may include any number of groups, each group may include any number of subbands, and each group may include the same or a different number of subbands.

Fig. 4 illustrates generation of LFDMA symbols for a group of N subbands. The original sequence of N modulation symbols to be transmitted over the subband group in one symbol period is denoted as { d₁，d₂，d₃，…，d_N} (block 410). The original sequence of N modulation symbols is converted to the frequency domain with an N-point Fast Fourier Transform (FFT) to obtain a sequence of N frequency-domain symbols (block 412). The N frequency-domain symbols are mapped onto N subbands for transmission and K-N zero symbols are mapped onto the remaining K-N subbands to generate a sequence of K symbols (block 414). The N subbands used for transmission have indices of K +1 through K + N, where 1 ≦ K ≦ K-N. The sequence of K symbols is then converted to the time domain using a K-point IFFT to obtain a sequence of K time-domain output symbols (block 416). The last C output symbols of the sequence are copied to the beginning of the sequence to form an LFDMA symbol containing K + C output symbols (block 418).

The LFDMA symbol is generated such that it occupies a group of N adjacent subbands starting from subband k + 1. Similar to OFDMA, users may be assigned different non-overlapping subband groups, thereby being orthogonal to each other. Each user may be assigned a different subband group in different symbol periods to achieve frequency diversity. The subband groups for each user may be selected based on, for example, a frequency hopping pattern.

Like OFDMA, SC-FDMA has certain desirable characteristics, such as high spectral efficiency and robustness against multipath effects. Furthermore, SC-FDMA does not have a high PAPR because modulation symbols are transmitted in the time domain. The PAPR of an SC-FDMA waveform is determined by the signal points in the signal constellation (e.g., M-PSK, M-QAM, and the like) selected for use. However, due to the non-flat communication channel, the time domain modulation symbols in SC-FDMA are susceptible to inter-symbol interference. Equalization may be performed on the received modulation symbols to mitigate the deleterious effects of inter-symbol interference. Equalization requires a reasonably accurate channel estimate for the communication channel, which can be obtained using the techniques described herein.

The transmitter may transmit pilots to facilitate channel estimation by the receiver. The pilot is the transmission of symbols that are known a priori by both the transmitter and the receiver. As used herein, a data symbol is a modulation symbol corresponding to data, and a pilot symbol is a modulation symbol corresponding to pilot. The data symbols and modulation symbols may be derived from the same or different signal constellations. As will be explained below, the pilots may be transmitted in various ways.

Fig. 5A shows a TDM pilot scheme 500 where pilot and data are multiplexed across multiple symbol periods. For example, can be at D₁Data is transmitted in one symbol period, and then P can be followed₁Pilot is sent in one symbol period, and then may be sent in the next D₁Data is sent in one symbol period, and so on. In general, D₁Not less than 1 and P₁Not less than 1. For the example shown in FIG. 5A, D₁> 1 and P₁1. A sequence of N data symbols may be transmitted on a subband group/burst in each symbol period for data transmission. A sequence of N pilot symbols may be transmitted on a subband group/burst in each symbol period used for pilot transmission. For each symbol period, a sequence of N data or pilot symbols may be converted to an IFDMA symbol or an LFDMA symbol, respectively, as described above with respect to fig. 2 and 4. The SC-FDMA symbols may be IFDMA symbols or LFDMA symbols. The SC-FDMA symbols containing only pilots, which may be pilot IFDMA symbols or pilot LFDMA symbols, are referred to as pilot SC-FDMA symbols. The SC-FDMA symbols containing only data are referred to as data SC-FDMA symbols, which may be data IFDMA symbols or data LFDMA symbols.

Fig. 5B shows a TDM pilot scheme 510 where pilot and data are multiplexed across multiple sample periods. For this embodiment, the data and pilot are multiplexed within the same SC-FDMA symbol. For example, can be at D₂Transmitting data symbols in one sampling period, then P₂Transmitting pilot symbols in one sample period, and thenAt the following D₂Data symbols are sent in one sample period, and so on. In general, D₂Not less than 1 and P₂Not less than 1. For the example shown in FIG. 5B, D₂1 and P₂1. A sequence of N data and pilot symbols may be transmitted on one subband group/burst in each symbol period and may be converted to an SC-FDMA symbol as described with respect to fig. 2 and 4.

The TDM pilot scheme may also multiplex pilot and data across both symbol periods and sampling periods. For example, data and pilot symbols may be transmitted in some symbol periods, only data symbols may be transmitted in other symbol periods, and only pilot symbols may be transmitted in certain symbol periods.

Fig. 5C shows CDM pilot scheme 530 where pilot and data are combined across multiple symbol periods. For this embodiment, a sequence of N data symbols is combined with a first M-chip orthogonal sequence { w }_dMultiplying to obtain M sequences of scaled data symbols, where M > 1. Each scaled data symbol sequence is formed by multiplying the original data symbol sequence by the orthogonal sequence w_dOne chip of. Similarly, a sequence of N pilot symbols is combined with a second M-chip orthogonal sequence { w }_pMultiply to obtain M scaled pilot symbol sequences. Each scaled data symbol sequence is then added to the corresponding scaled pilot symbol sequence to obtain a combined symbol sequence. The M combined symbol sequences are obtained by adding the M scaled data symbol sequences to the M scaled pilot symbol sequences. Each combined symbol sequence is converted into an SC-FDMA symbol.

These orthogonal sequences may be Walsh sequences, OVSF sequences, and the like. For the example shown in fig. 5C, M is 2, and the first orthogonal sequence is { w ═ 2 }_d{ +1+1}, and the second orthogonal sequence is { w { (m) }_p{ +1-1 }. The N data symbols are multiplied by +1 over the symbol period t and by +1 over the symbol period t + 1. The N pilot symbols are multiplied by +1 over the symbol period t and by-1 over the symbol period t + 1. For each symbol period, the N scaled data symbols are compared with the symbol periodThe N scaled pilot symbols are added to obtain N combined symbols corresponding to the symbol period.

Fig. 5D shows CDM pilot scheme 540 where pilot and data are combined across multiple sample periods. For this embodiment, a sequence of N/M data symbols is combined with the M-chip orthogonal sequence { w }_dMultiply to obtain a sequence of N scaled data symbols. Specifically, the first data symbol d in the original sequence is decoded₁(t) multiplication by orthogonal sequence { w_dGet the first M scaled data symbols, and get the next data symbol d₂(t) multiplied by the orthogonal sequence w_dGet the next M scaled data symbols, and so on, and will be the last data symbol d in the original sequence_N/M(t) multiplication by orthogonal sequence { w_dTo obtain the last M scaled data symbols. Similarly, a sequence of N/M pilot symbols is aligned with the M-chip orthogonal sequence { w }_pMultiply to obtain a sequence of N scaled data symbols. The sequence of N scaled data symbols is added to the sequence of N scaled pilot symbols to obtain a sequence of N combined symbols, which is converted to an SC-FDMA symbol.

For the example shown in fig. 5D, M is 2, the orthogonal sequence for the data is { w ═ 2_d{ +1+1}, and the orthogonal sequence for the pilot is { w { (w) }_p{ +1-1 }. A sequence of N/2 data symbols is multiplied by the orthogonal sequence { +1+1} to obtain a sequence of N scaled data symbols. Similarly, a sequence of N/2 pilot symbols is multiplied by the orthogonal sequence { +1-1} to obtain a sequence of N scaled pilot symbols. For each symbol period, the N scaled data symbols are added to the N scaled pilot symbols to obtain N combined symbols corresponding to the symbol period.

As shown in fig. 5C and 5D, a CDM pilot may be transmitted in each symbol period. The CDM pilot may also be sent only in certain symbol periods. The pilot scheme may also employ a combination of TDM and CDM. For example, a CDM pilot may be sent on some symbol periods and a TDM pilot may be sent on other symbol periods. Frequency Division Multiplexed (FDM) pilots may also be sent on a designated set of subbands, e.g., downlink.

For the embodiments shown in fig. 5A through 5D, the TDM or CDM pilot is sent on the N subbands used for data transmission. In general, the subbands used for pilot transmission (or simply, pilot subbands) may be the same as or different from the subbands used for data transmission (or simply, data subbands). Fewer or more subbands may be used to transmit pilot than are used to transmit data. The data and pilot subbands may be static for the entire transmission. Alternatively, the data and pilot subbands may hop across frequency in different time slots to achieve frequency diversity. For example, a physical channel may be associated with a Frequency Hopping (FH) pattern that indicates the particular subband group or groups used for the physical channel in each timeslot. A slot may span one or more symbol periods.

Fig. 6 shows a wideband pilot scheme 600 that would be more suitable for the reverse link. For this embodiment, each user transmits a wideband pilot that is transmitted on all or most of the K total subbands-e.g., all subbands available for transmission. The broadband pilot may be generated in the time domain (e.g., with a pseudo-random number (PN) sequence) or in the frequency domain (e.g., using OFDM). The broadband pilot for each user may be time division multiplexed with the data transfer from that user, and the pilot may be generated using LFDMA (as shown in fig. 6) or IFDMA (not shown in fig. 6). The wideband pilots from all users may be transmitted in the same symbol period, which may avoid interference of data with the pilots used for channel estimation. The broadband pilot from each user may be code division multiplexed (e.g., pseudo-random) with respect to the broadband pilots from other users. This can be achieved by assigning a different PN sequence to each user. The wideband pilot for each user has a low peak-to-average power ratio (PAPR) and spans the entire system bandwidth, which enables the receiver to derive a wideband channel estimate for that user. For the embodiment shown in fig. 6, each data subband hops across frequency at different time slots. For each slot, a channel estimate may be derived for each data subband based on the wideband pilot.

Fig. 5A through 6 illustrate exemplary pilot and data transmission schemes. Pilot and data may also be transmitted in other manners using any combination of TDM, CDM, and/or some other multiplexing scheme.

The TDM and CDM pilots may be generated in various manners. In one embodiment, the pilot symbols used to generate TDM and CDM are modulation symbols from a known signal constellation such as QPSK. The TDM pilot scheme shown in fig. 5A and the CDM pilot scheme shown in fig. 5C may use a sequence of N modulation symbols. The TDM pilot scheme shown in fig. 5B and the CDM pilot scheme in fig. 5D may use a sequence of N/M modulation symbols. The sequence of N modulation symbols and the sequence of N/M modulation symbols may each be selected to have (1) a spectrum that is as flat as possible, and (2) a temporal envelope that varies as little as possible. The flat spectrum ensures that all subbands used for pilot transmission have sufficient power to allow the receiver to correctly estimate the channel gain for these subbands. The constant envelope prevents distortion caused by circuit blocks such as power amplifiers.

In another embodiment, the pilot symbols used to generate the TDM and CDM pilots are formed based on polyphase sequences with good time and spectral characteristics. For example, the pilot symbols may be generated as follows:

n-1, …, N, formula (2)

Wherein the phase *_nCan be derived based on any of the following:

*_npi (n-1) · n, formula (3)

*_n＝π·(n-1)²In the formula (4)

*_n＝π·[(n-1)·(n-N-1)]In the formula (5)

In formula (6), Q and N are relatively prime. Equation (3) corresponds to the Golomb sequence, equation (4) corresponds to the P3 sequence, equation (5) corresponds to the P4 sequence, and equation (6) corresponds to the Chu sequence. The P3, P4, and Chu sequences may be of any length.

Pilot symbols may also be generated as follows:

1, …, T and m 1, …, T, formula (7)

Wherein the phase *_l，mCan be derived based on any of the following:

*_l，m2 pi (l-1) (m-1)/T, formula (8)

*_l，m＝-(π/T)·(T-2l+1)·[(l-1)·T+(m-1)]Of the formula (9)

Equation (8) corresponds to Frank sequences, equation (9) corresponds to P1 sequences, and equation (10) corresponds to Px sequences. Frank, P1, and Px sequences are constrained in length to satisfy N ═ T²Wherein T is a positive integer.

The sequence of pilot symbols generated based on any of the polyphase sequences described above has both a flat frequency spectrum and a constant time-domain envelope. Other polyphase sequences with good spectral characteristics (e.g., flat or known spectrum) and good temporal characteristics (e.g., constant or known temporal envelope) may also be used. The TDM or CDM pilot generated with this sequence of pilot symbols will thus have (1) a low PAPR, which avoids distortion caused by circuit elements such as power amplifiers, and (2) a flat spectrum, which enables the receiver to accurately estimate the channel gain for all subbands used for pilot transmission.

Fig. 7A illustrates a process 700 for generating pilot IFDMA symbols. The first pilot symbol sequence is formed based on a polyphase sequence, which may be any of the polyphase sequences described above or some other polyphase sequence (block 710). The first sequence of pilot symbols is replicated multiple times to obtain a second sequence of pilot symbols (block 712). A phase ramp is applied to the second pilot symbol sequence to obtain a third output symbol sequence (block 714). The phase ramp may be applied digitally to the pilot symbols or implemented by a reliable up-conversion process. A cyclic prefix is added to the third sequence of output symbols to obtain a fourth sequence of output symbols, which is a pilot IFDMA symbol (block 716). The pilot IFDMA symbol is transmitted in the time domain via a communication channel (block 718). Although not shown in fig. 7A for simplicity, pilot symbols may be multiplexed with data symbols using TDM and/or CDM, e.g., as described above with respect to fig. 5A through 5D.

Fig. 7B is a process 750 for generating pilot LFDMA symbols. The first pilot symbol sequence is formed based on a polyphase sequence, which may be any of the polyphase sequences described above or some other polyphase sequence (block 760). The first sequence of N pilot symbols is transformed to the frequency domain with an N-point FFT to obtain a second sequence of N frequency-domain symbols (block 762). The N frequency-domain symbols are then mapped to N subbands for pilot transmission and zero symbols are mapped to the remaining K-N subbands to obtain a third sequence of K symbols (block 764). The third sequence of K symbols is transformed to the time domain with a K-point IFFT to obtain a fourth sequence of K time domain output symbols (block 766). A cyclic prefix is added to the fourth sequence of output symbols to obtain a fifth sequence of K + C output symbols, which is a pilot LFDMA symbol (block 768). The pilot LFDMA symbol is transmitted in the time domain via a communication channel (block 770). Although not shown in fig. 7B for simplicity, pilot symbols may be multiplexed with data symbols using TDM and/or CDM, e.g., as described above with respect to fig. 5A through 5D.

For both IFDMA and LFDMA, the number of subbands used for pilot transmission may be the same or different than the number of subbands used for data transmission. For example, a user may be assigned 16 subbands for data transmission and 8 subbands for pilot transmission. Another 8 subbands may be allocated to another user for data/pilot transmission. Multiple users may share the same subband group for the interleaved subband structure in fig. 1 or the same subband group for the narrowband subband structure in fig. 3.

For the interleaved subband structure of fig. 1, an FDM pilot may be transmitted on one or more subband groups to enable a receiver to perform various functions such as channel estimation, frequency tracking, time tracking, and so on. In a first staggered FDM pilot, pilot IFDMA symbols are transmitted on subband group p in some symbol periods and pilot IFDMA symbols are transmitted on subband group p + S/2 in other symbol periods. For example, if S-8, pilot IFDMA symbols may be transmitted using a {3, 7} staggered pattern, whereby pilot IFDMA symbols are transmitted on subband group 3, then on subband group 7, then on subband group 3, and so on. In a second staggered FDM pilot, the pilot IFDMA symbol is transmitted in symbol period t on subband group p (t) [ p (t-1) + Δ p ] modulo S +1, where Δ p is the difference between the subband group indices for two consecutive symbol periods, and +1 is an index scheme corresponding to starting from 1 instead of 0. For example, if S is 8 and Δ p is 3, pilot IFDMA symbols may be transmitted using a staggered pattern of {1, 4, 7, 2, 5, 8, 3, 6}, whereby pilot IFDMA symbols are transmitted on subband group 1, then on subband group 4, then on subband group 7, and so on. Other staggering patterns may also be used. The staggered FDM pilots enable the receiver to obtain channel gain estimates for more subbands, which improves channel estimation and detection performance.

Fig. 8 shows a process 800 performed by a receiver to estimate a response of a communication channel based on a TDM pilot or a CDM pilot sent by a transmitter. The receiver obtains one SC-FDMA symbol for each symbol period and removes the cyclic prefix in the received SC-FDMA symbol (block 810). For IFDMA, the receiver removes the phase ramp in the received SC-FDMA symbol. For both IFDMA and LFDMA, the receiver acquires K received data/pilot symbols from the SC-FDMA symbol.

The receiver then reverses the TDM or CDM performed on the pilot (block 812). For the TDM pilot scheme shown in FIG. 5A, K received pilot symbols r are obtained from each pilot SC-FDMA symbol_p(n), n is 1, … K. For the TDM pilot scheme shown in fig. 5B, multiple received pilot symbols are obtained from each SC-FDMA symbol that contains the TDM symbol.

For the CDM pilot scheme shown in FIG. 5C, the M received SC-FDMA symbols containing the CDM pilot are processed to recover pilot symbols as follows:

formula (11)

Wherein r (t)_iN) is the received sample in symbol period ti corresponding to sampling period n;

w_p，iis the ith chip of the orthogonal sequence corresponding to the pilot; and

r_p(n) is the received pilot symbol corresponding to sample period n.

Equation (11) assumes that the CDM pilot is in the symbol period t₁To t_MTransmitted, where M is the length of the orthogonal sequence. K received pilot symbols corresponding to the CDM symbol are obtained from equation (11).

For the CDM pilot scheme shown in FIG. 5D, each received SC-FDMA symbol containing the CDM pilot is processed to recover pilot symbols as follows:

formula (12)

Where r ((n-1). M + i) is the received sample corresponding to sample period (n-1). M + i in the received SC-FDMA symbol with the CDM pilot. K/M received pilot symbols corresponding to the CDM pilot are obtained from equation (12).

Frequency selective communication channels result in inter-symbol interference (ISI). However, because of the cyclic prefix, this ISI is confined within a single SC-FDMA symbol. Further, because of the cyclic prefix, the linear convolution operation due to the channel impulse response actually becomes a cyclic convolution, which is similar to OFDMA. Accordingly, channel estimation, equalization, and other operations may be performed in the frequency domain when pilot symbols and data symbols are not transmitted in the same SC-FDMA symbol.

For the TDM scheme shown in fig. 5A and the CDM scheme shown in fig. 5C, the receiver obtains K received pilot symbols from each pilot transmission. The K received pilot symbols r, which may be for n-1, …, K_p(n) perform a K-point FFT to obtain K received pilot values for K1, …, K in the frequency domain (block 814). The received pilot values may be given as follows:

R_p(k) h (K) · p (K) + n (K), K ═ 1, …, K, formula (13)

Where p (k) is the transmitted pilot value corresponding to subband k;

h (k) is the complex gain of the communication channel corresponding to subband k;

R_p(k) is the received pilot value corresponding to subband k; and

n (k) is the noise corresponding to subband k.

The K-point FFT provides K received pilot values corresponding to K total subbands. Only the N received pilot values corresponding to the N subbands used for pilot transmission (referred to as pilot subbands) are retained and the remaining K-N received pilot values are discarded (block 816). IFDMA and LFDMA use different pilot subbands and thus IFDMA and LFDMA retain different received pilot values. The reserved pilot value is denoted as R_p(k) K is 1, …, N. For simplicity, noise may be assumedMean of sound is zero and variance is N₀Additive White Gaussian Noise (AWGN).

The receiver may estimate the channel frequency response using various channel estimation techniques, such as MMSE techniques, Least Squares (LS) techniques, and so on. The receiver derives channel gain estimates for the N pilot subbands based on the N received pilot values and using MMSE or LS techniques (block 818). For MMSE techniques, an initial frequency response estimate for the communication channel may be derived based on the received pilot values as follows:

formula (14)

Wherein(k) Is the channel gain estimate corresponding to subband k and "x" denotes the complex conjugate. The initial frequency response estimate contains N channel gains corresponding to the N pilot subbands. The pilot symbol sequence may be generated based on a polyphase sequence having a flat frequency response. In this case, there is | p (k) | ═ 1 for all values of k, and equation (14) can be expressed as:

formula (15)

Removable constant factor 1/(1+ N)₀) To provide an unbiased MMSE frequency response estimate, which can be expressed as:

formula (16)

For the LS technique, an initial frequency response estimate may be derived based on the received pilot values as follows:

formula (17)

The impulse response of the communication channel may be characterized by L taps, where L may be much less than N. That is, if the transmitter applies an impulse to the communication channel, L time-domain samples (at the sampling rate of BS MHz) will be sufficient to characterize the response of the communication channel based on this impulse excitation. The number of taps (L) of the channel impulse response depends on the delay spread of the system, i.e. the time difference between the earliest and latest arriving signal instances with sufficient energy at the receiver. A longer delayed opening corresponds to a larger value of L and vice versa.

A channel impulse response estimate may be derived based on the N channel gain estimates and using LS or MMSE techniques (block 820). L taps with n-1, …, L may be derived based on the initial frequency response estimate(k) The least squares channel impulse response estimate of (a) is as follows:

formula (18)

WhereinContaining k 1, …, N(k) Or(k) N × 1 vector of (a);

W _N×Lis a Fourier matrixW _K×KA sub-matrix of (a);

containing n-1, …, L(k) L × 1 vector of (a); and

“^H"denotes conjugate transpose.

Fourier matrixW _K×KIs defined such that the (u, v) th element f_u，vGiven as:

u-1, …, K and v-1, …, K, formula (19)

Where u is the row index and v is the column index.W _N×LIncludedW _K×KN rows corresponding to the N pilot subbands.W _N×LEach row of (1) comprisesW _K×KThe first L elements of the corresponding line.L taps containing least squares channel impulse response estimates.

L taps with n equal to 1, …, LThe MMSE channel impulse response estimate of (n) may be derived based on the initial frequency response estimate as follows:

formula (20)

WhereinN _L×LIs the L x L noise and interference auto-covariance matrix. For Additive White Gaussian Noise (AWGN), the autocovariance matrix may be given asWherein sigma_n ²Is the noise variance. An N-point IFFT may also be performed on the initial frequency response estimate to obtain a channel impulse response estimate having N taps.

Filtering and/or post-processing may be performed on the initial frequency response estimate and/or the channel impulse response estimate as described below to improve the quality of the channel estimate (block 822). Final frequency response estimates corresponding to all K subbands may be obtained by (1) zero-padding the L-tap or N-tap channel impulse response estimates to length K, and (2) performing a K-point FFT on the extended impulse response estimates (block 824). The final frequency response estimate for all K subbands may also be obtained by (1) interpolating the N channel gain estimates, (2) performing a least squares approximation on the N channel gain estimates, or (3) using other approximation techniques.

The receiver may derive a longer channel impulse response estimate based on the staggered FDM pilot. Generally, based on L in one or more symbol periods_TPilot IFDMA symbols sent on different subbands may be obtained with L_TChannel impulse response estimation for each tap. For example, if L_T2N, an impulse response estimate having 2N taps may be obtained based on two or more pilot IFDMA symbols transmitted over two or more subband groups in two or more symbol periods. If the pilot is transmitted using a full staggering pattern across all S subband groups, a full-length impulse response estimate with K taps may be obtained.

The receiver may derive the longer length L by filtering the initial impulse response estimate of length N for a sufficient number of different subband groups_TThe impulse response estimation of (2). Each initial impulse response estimate may be derived based on pilot IFDMA symbols corresponding to a subband group. If the pilot is transmitted on a different subband group in each symbol period, filtering may be performed over a sufficient number of symbol periods to obtain a longer impulse response estimate.

For SC-FDMA, filtering may be performed on initial frequency response estimates, least squares or MMSE channel impulse response estimates, and/or final frequency response estimates obtained for different symbol periods to improve the quality of the channel estimates. The filtering may be based on a Finite Impulse Response (FIR) filter, an Infinite Impulse Response (IIR) filter, or some other type of filter. The filter coefficients may be selected to achieve a desired amount of filtering, and may be selected based on a tradeoff between various factors, such as a desired channel estimation quality, the ability to track rapid changes in the channel, filtering complexity, and so forth.

Other channel estimation techniques may also be used to obtain a frequency response estimate and/or a channel impulse response estimate for the communication channel.

Various post-processing may be performed to improve the quality of the channel estimate. In certain operating environments, such as multipath fading environments, a communication channel often has only a few taps in the time domain. The channel estimation described above may provide a channel impulse response with a large number of taps due to noise. Post-processing attempts to remove the taps due to noise and retain the taps due to the actual channel.

In one post-processing scheme, called truncation, only the first L taps of the channel impulse response estimate are retained and the remaining taps are replaced with zeros. In another post-processing scheme, called thresholding, the low energy taps are replaced with zeros. In one embodiment, thresholding is performed as follows:

n-1, …, K, formula (21)

Wherein(n) is the nth tap of the channel impulse response estimate, which may be equal to(n) or(n); and is

h_thIs a threshold used to zero out the low energy tap.

Threshold value h_thMay be calculated based on the energy of all K taps or only the energy of the first L taps of the channel impulse response estimate. The same threshold may be used for all tapsThe value is obtained. Alternatively, different thresholds may be used for different taps. For example, a first threshold may be used for the first L taps, and a second threshold (which may be lower than the first threshold) may be used for the remaining taps.

In yet another post-processing scheme, referred to as tap selection, the B best taps of the channel impulse response estimate are retained, where B ≧ 1, and the remaining taps are set to zero. The number of taps to be retained (denoted as B) may be a fixed or variable value. B may be selected based on a received signal-to-noise-and-interference ratio (SNR) of the pilot/data transmission, a spectral efficiency of the data packet using the channel estimate, and/or some other parameter. For example, if the received SNR falls within a first range (e.g., from 0 to 5 decibels (dB)), then the two best taps may be retained, if the received SNR falls within a second range (e.g., from 5 to 10dB), then the three best taps may be retained, if the received SNR falls within a third range (e.g., from 10 to 15dB), then the four best taps may be retained, and so on.

Channel estimation may be performed in the time domain for the TDM pilot scheme shown in fig. 5B, the CDM pilot scheme shown in fig. 5D, and other pilot schemes in which data and pilot symbols are transmitted in the same SC-FDMA symbol. A rake estimator may be used to identify strong signal paths by, for example, (1) correlating received symbols with a sequence of pilot symbols transmitted at different time offsets, and (2) identifying the time offset that provides the highest correlation result. The time domain channel estimate provides a set of taps for a channel impulse response estimate for the communication channel.

For all pilot schemes, the channel estimate provides an equalized channel impulse response estimate and/or frequency response estimate that may be used for the received data symbols. For the TDM pilot scheme shown in fig. 5A, a sequence of K received data symbols is obtained from each data SC-FDMA symbol, while for the CDM pilot scheme shown in fig. 5C, a sequence of K received data symbols is obtained from each set of M received SC-FDMA symbols. The sequence of K received data symbols may be equalized in the time or frequency domain.

Frequency domain equalization may be performed as follows. Firstly, the methodFor K received data symbols r of n-1, …, K_d(n) performing a K-point FFT to obtain K frequency-domain received data values R with K equal to 1, …, K_d(k) In that respect Only the N received data values corresponding to the N subbands used for data transmission are retained and the remaining K-N received data values are discarded. The retained data value is noted as R_d(n)，k＝1，…，N。

The N received data values may be equalized in the frequency domain using MMSE techniques as follows:

formula (22)

Wherein R is_d(k) Is the received data value corresponding to subband k;

(k) is the channel gain estimate corresponding to subband k, which may be equal to(k) Or(k) (ii) a And is

Z_d(k) Is the equalized data value for subband k.

Equalization may also be performed on the N received data values in the frequency domain using a zero-forcing technique as follows:

formula (23)

For both MMSE and zero-forcing equalization, N equalized data values Z with k equal to 1, …, N may be used_d(k) Transform back to time domain to obtain N data symbol estimates with N-1, …, N(N) that are estimates of the N data symbols in the original sequence.

Equalization may also be performed in the time domain on a sequence of K received data symbols as follows:

Z_d(n)＝r_d(n) * g (n), formula (24)

Wherein r is_d(n) represents a sequence of K received data symbols;

g (n) represents the impulse response of the time-domain equalizer;

z_d(n) represents a sequence of K equalized data symbols; and

* denotes a circular convolution operation.

The frequency response of the equalizer can be derived based on MMSE techniques as follows:k is 1, …, N. The frequency response of the equalizer may also be derived based on zero forcing techniques as follows:k is 1, …, N. The equalizer frequency response may be transformed to the time domain to obtain an equalizer impulse response g (N), 1, …, N, which is used for time domain equalization in equation (24).

The sequence of K equalized data symbols from equation (24) contains S copies of the transmitted data symbol. The S copies may be accumulated on a data symbol-by-data symbol basis to obtain N data symbol estimates as follows:

formula (25)

Alternatively, accumulation may not be performed and only the N equalized data symbols corresponding to one copy of the transmitted data are provided as N data symbol estimates.

The receiver may also estimate interference based on the received pilot values and channel estimates. For example, the interference corresponding to each subband may be estimated as follows:

formula (26)

Where I (k) is the interference estimate corresponding to subband k. The interference estimates i (k) may be averaged over all N subbands for each SC-FDMA symbol to obtain short-term interference estimates, which may be used for data demodulation and/or other purposes. The short-term interference estimates may be averaged over multiple SC-FDMA symbols to obtain long-term interference estimates, which may be used to estimate operating conditions and/or for other purposes.

Other techniques may also be used to improve the quality of the channel estimates derived from the TDM pilot or CDM pilot. These techniques include iterative channel estimation techniques and data-aided channel estimation techniques.

For iterative channel estimation techniques, an initial estimate of the communication channel is first derived based on the received pilot symbols using, for example, MMSE or least-squares techniques. The initial channel estimate is used to derive data symbol estimates as described above. In one embodiment, the estimation is based on data symbols(n) and an initial messageTrack estimation(n) to estimate the interference of the data symbols to the pilot symbols asWhereinAnd (n) represents interference estimation. In another embodiment, the data symbol estimates are processed to obtain decoded data. The decoded data is then processed in the same manner as performed at the transmitter to obtain remodulated data symbols, which are convolved with the initial channel estimate to obtain an interference estimate. For both embodiments, the interference estimate is subtracted from the received pilot symbols to obtain interference-canceled pilot symbolsThe pilot symbols are then used to derive an improved channel estimate. This process is then repeated for any number of iterations to obtain a progressively optimized channel estimate. This iterative channel estimation technique is more suitable for the TDM pilot scheme shown in fig. 5B, fig. 5C, andthe CDM pilot scheme shown in fig. 5D, and other pilot schemes in which data symbols may cause intersymbol interference to pilot symbols.

For data-aided channel estimation techniques, received data symbols are used for channel estimation along with received pilot symbols. A first channel estimate is derived based on the received pilot symbols and a data symbol estimate is obtained using the first channel estimate. A second channel estimate and a second symbol estimate are then derived based on the received data symbols. In one embodiment, a data symbol r is received_d(n) received data value R converted into frequency domain_d(k) And estimates the data symbolsConversion to frequency domain data values(k) In that respect The second channel estimate may be obtained by dividing R in equations (14) to (18)_d(k) Substitution into R_p(k) And will be(k) Substituting p (k) to obtain. In another embodiment, the data symbol estimates are processed to obtain decoded data, and the decoded data is processed to obtain remodulated data symbols D_rm(k) In that respect The second channel estimate may be obtained by dividing R in equations (14) to (18)_d(k) Substitution into R_p(k) And D is_rm(k) Substituting p (k) to obtain.

The two channel estimates obtained with the received pilot symbols and the received data symbols are combined to obtain an improved overall channel estimate. This combination may be performed, for example, as follows:

formula (27)

Wherein(k) Is a channel estimate obtained based on the obtained pilot symbols;

(k) is a channel estimate obtained based on the received data symbols;

C_p(k) and C_d(k) Weighting factors corresponding to pilot and data, respectively; and is

(k) Is the total channel estimate.

In general terms, the amount of the solvent to be used,(k) can be based on(k)、(k) A confidence in the reliability of the data symbol estimate, and/or any other factor. The processing described above may be performed in an iterative manner. Updating, for each iteration, based on channel estimates obtained from the data symbol estimates(k) And using the updated(k) To derive new data symbol estimates. The data-assisted channel estimation techniques may be used for all pilot schemes, including the TDM and CDM pilot schemes shown in fig. 5A through 5D.

Fig. 9 shows a block diagram of a transmitter 910 and a receiver 950. For the forward link, transmitter 910 is part of a base station and receiver 950 is part of a wireless device. For the reverse link, transmitter 910 is part of the wireless device and receiver 950 is part of the base station. A base station is generally a fixed station and may also be referred to as a Base Transceiver System (BTS), an access point, or some other terminology. A wireless device may be fixed or mobile and may also be referred to as a user terminal, a mobile station, or some other terminology.

At transmitter 910, a TX data and pilot processor 920 processes traffic data to obtain data symbols, generates pilot symbols, and provides the data symbols and pilot symbols. SC-FDMA modulator 930 multiplexes the data symbols with pilot symbols using TDM and/or CDM and performs SC-FDMA modulation (e.g., for IFDMA, LFDMA, etc.) to generate SC-FDMA symbols. A transmitter unit (TMTR)932 processes (e.g., converts to analog, amplifies, filters, and frequency upconverts) the SC-FDMA symbols and generates a Radio Frequency (RF) modulated signal, which is transmitted via an antenna 934.

At receiver 950, an antenna 952 receives the transmitted signal and provides a received signal. Receiver unit (RCVR)954 conditions (e.g., filters, amplifies, frequency downconverts, and digitizes) the received signal to generate a stream of received samples. SC-FDMA demodulator 960 processes the received samples and obtains received data symbols and received pilot symbols. A channel estimator/processor 980 derives channel estimates based on the received pilot symbols. SC-FDMA demodulator 960 performs equalization on the received data symbols with the channel estimates and provides data symbol estimates. A Receive (RX) data processor 970 performs demapping, deinterleaving, and decoding on the data symbol estimates and provides decoded data. In general, the processing by SC-FDMA demodulator 960 and RX data processor 970 is complementary to the processing by SC-FDMA modulator 930 and TX data and pilot processor 920 at transmitter 910, respectively.

Controllers 940 and 990 direct the operation of various processing units at transmitter 910 and receiver 950, respectively. Memory units 942 and 992 store program codes and data used by controllers 940 and 990, respectively.

Fig. 10A illustrates a block diagram of a TX data and pilot processor 920A, which is an embodiment of processor 920 in fig. 9 and may be used for a TDM pilot scheme. Within processor 920a, traffic data is encoded by an encoder 1012, interleaved by an interleaver 1014, and mapped to data symbols by a symbol mapper 1016. Pilot generator 1020 generates pilot symbols based on, for example, a polyphase sequence. A multiplexer (Mux)1022 receives and multiplexes the data symbols and pilot symbols based on TDM control and provides a stream of multiplexed data and pilot symbols.

Fig. 10B illustrates a block diagram of a TX data and pilot processor 920B, which is another embodiment of processor 920 in fig. 9 and may be used for a CDM pilot scheme. Within processor 920b, traffic data is encoded by an encoder 1012, interleaved by an interleaver 1014, and mapped to data symbols by a symbol mapper 1016. Riding deviceThe normalizer 1024a combines each data symbol with an orthogonal sequence { w } corresponding to the data_dMultiply the M chips of and provide M scaled data symbols. Similarly, multiplier 1024b combines each pilot symbol with an orthogonal sequence { w } corresponding to the pilot_pMultiply the M chips of and provide M scaled pilot symbols. An adder 1026 adds the scaled data symbols to the scaled pilot symbols and provides combined symbols, such as shown in fig. 5C or 5D.

Fig. 11A illustrates an SC-FDMA modulator 930a, which is one embodiment of the SC-FDMA modulator 930 in fig. 9, corresponding to IFDMA. Within modulator 930a, a repetition unit 1112 repeats the original data/pilot symbol sequence S times to obtain a spread sequence of K symbols. Phase ramp unit 1114 applies a phase ramp to the spread symbol sequence to generate a frequency-translated output symbol sequence. The phase ramp is determined by the subband u used for transmission. Cyclic prefix generator 1116 adds a cyclic prefix to the frequency-translated sequence of symbols to generate an IFDMA symbol.

Fig. 11B shows an SC-FDMA modulator corresponding to LFDMA, which is another embodiment of SC-FDMA modulator 930 in fig. 9. Within modulator 930b, an FFT unit 1122 performs an N-point FFT on the original data/pilot symbol sequence to obtain a sequence of N frequency-domain symbols. A symbol to subband mapper 1124 maps the N frequency-domain symbols to the N subbands for transmission and maps the K-N zero symbols to the remaining K-N subbands. IFFT unit 1126 performs a K-point IFFT on the K symbols from mapper 1124 and provides a sequence of K time-domain output symbols. Cyclic prefix generator 1128 adds a cyclic prefix to the sequence of output symbols to generate an LFDMA symbol.

Fig. 12A shows a block diagram of SC-FDMA demodulator 960a, which is one embodiment of demodulator 960 in fig. 9 and may be used for a TDM IFDMA pilot scheme. Within SC-FDMA demodulator 960a, a cyclic prefix removal unit 1212 removes the cyclic prefix for each received IFDMA symbol. A phase ramp removal unit 1214 removes the phase ramp in each received IFDMA symbol. Phase ramp removal may also be performed by down-conversion from RF to baseband. A demultiplexer (Demux)1220 receives the output of unit 1214 and provides received data symbols to equalizer 1230 and received pilot symbols to channel estimator 980. Channel estimator 980 derives channel estimates based on the received pilot symbols using, for example, MMSE or least-squares techniques. The equalizer 1230 performs equalization on the received data symbols in the time or frequency domain with the channel estimate and provides equalized data symbols. Accumulator 1232 accumulates equalized data symbols corresponding to multiple copies of the same transmitted data symbol and provides data symbol estimates.

Fig. 12B shows a block diagram of SC-FDMA demodulator 960B, which is another embodiment of demodulator 960 in fig. 9 and may be used for a CDM IFDMA pilot scheme. SC-FDMA demodulator 960b includes a data channelizer for recovering transmitted data symbols and a pilot channelizer for recovering transmitted pilot symbols. For a data channelizer, multiplier 1224a orthogonalizes the output of unit 1214 with the data sequence w_dThe M chips of the symbol are multiplied and provide a scaled data symbol. An accumulator 1226 accumulates the M scaled data symbols corresponding to each transmitted data symbol and provides a received data symbol. For the pilot channelizer, multiplier 1224b combines the output of unit 1214 with the pilot orthogonal sequence w_pMultiplies the M chips of and provides M scaled pilot symbols corresponding to each transmitted pilot symbol, which are accumulated by an accumulator 1226b to obtain a received pilot symbol corresponding to the transmitted pilot symbol. The processing by subsequent units within SC-FDMA demodulator 960b is the same as described above with respect to SC-FDMA demodulator 960.

Fig. 13A shows a block diagram of SC-FDMA demodulator 960c, which is yet another embodiment of demodulator 960 in fig. 9 and which may be used for a TDM LFDMA pilot scheme. Within SC-FDMA demodulator 960c, a cyclic prefix removal unit 1312 removes the cyclic prefix for each received LFDMA symbol. FFT unit 1314 performs a K-point FFT on the LFDMA symbol after the cyclic prefix is removed and provides K frequency-domain values. A subband-to-symbol demapper 1316 receives the K frequency-domain values, provides N frequency-domain values corresponding to the N subbands used for transmission, and discards the remaining frequency-domain values. IFFT unit 1318 performs an N-point FFT on the N frequency-domain values from demapper 1316 and provides N received symbols. A demultiplexer 1320 receives the output of unit 1318, provides received data symbols to equalizer 1330, and provides received pilot symbols to channel estimator 980. The equalizer 1330 performs equalization on the received data symbols in the time or frequency domain with the channel estimates from the channel estimator 980 and provides data symbol estimates.

Fig. 13B shows a block diagram of SC-FDMA demodulator 960d, which is yet another embodiment of demodulator 960 in fig. 9 and may be used for a CDM LFDMA pilot scheme. SC-FDMA demodulator 960d includes a data channelizer for recovering transmitted data symbols and a pilot channelizer for recovering transmitted pilot symbols. For the data channelizer, the multiplier 1324a combines the output of the IFFT unit 1318 with the data orthogonal sequence w_dThe M chips of the symbol are multiplied and provide a scaled data symbol. An accumulator 1326a accumulates the M scaled data symbols corresponding to each transmitted data symbol and provides a received data symbol. For the pilot channelizer, the multiplier 1324b combines the output of the IFFT unit 1318 with the pilot orthogonal sequence w_pMultiplies the M chips of and provides M scaled pilot symbols corresponding to each transmitted pilot symbol, which are accumulated by an accumulator 1326b to obtain a received pilot symbol corresponding to the transmitted pilot symbol. The processing by subsequent units within SC-FDMA demodulator 960d is the same as described above with respect to SC-FDMA demodulator 960 c.

The pilot transmission and channel estimation techniques described herein may be implemented by various means. For example, these techniques may be implemented in hardware, software, or a combination thereof. For a hardware implementation, the processing units at the transmitter used to generate and transmit pilots (e.g., each of the processing units shown in fig. 9-13B, or a combination of these processing units) may be implemented within one or more Application Specific Integrated Circuits (ASICs), Digital Signal Processors (DSPs), Digital Signal Processing Devices (DSPDs), Programmable Logic Devices (PLDs), Field Programmable Gate Arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, electronic devices, other electronic units designed to perform the functions described herein, or a combination thereof. The processing units used at the receiver to perform channel estimation may also be implemented within one or more ASICs, DSPs, electronic devices, and the like.

For a software implementation, the techniques may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The software codes may be stored in a memory unit (e.g., memory unit 942 or 992 in fig. 9) and executed by a processor (e.g., controller 940 or 990). The memory unit may be implemented within the processor or external to the processor.

The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims (59)

1. An apparatus, comprising:

a processor configured to form a first sequence of pilot symbols based on a polyphase sequence; and

a modulator to replicate the first sequence of pilot symbols a plurality of times to obtain a second sequence of pilot symbols for transmission via a communication channel.

2. The apparatus of claim 1, wherein the modulator is operative to apply a phase ramp to the second pilot symbol to obtain a third sequence of output symbols.

3. The apparatus of claim 2, wherein the modulator is further for using at least two different phase slopes for the phase ramp in at least two different symbol periods to transmit the first sequence of pilot symbols on at least two different sets of frequency subbands.

4. The apparatus of claim 1, wherein the modulator is operative to add a cyclic prefix to the second sequence of pilot symbols to obtain a third sequence of output symbols suitable for transmission in the time domain via the communication channel.

5. The apparatus of claim 1, wherein the polyphase sequence has a constant envelope in the time domain and a flat spectral response in the frequency domain.

6. The apparatus of claim 1, wherein the processor is operative to form a first sequence of data symbols, to multiplex the first sequence of data symbols in a first symbol period, and to multiplex the first sequence of pilot symbols in a second symbol period.

7. The apparatus of claim 1, wherein the processor is operative to form a first sequence of data symbols, to multiplex the first sequence of pilot symbols with the first sequence of data symbols, and to provide a sequence of multiplexed data and pilot symbols.

8. The apparatus of claim 1, wherein the processor is operative to form a first sequence of data symbols, to multiply the first sequence of data symbols with a first orthogonal sequence to obtain a plurality of sequences of scaled data symbols, to multiply the first sequence of pilot symbols with a second orthogonal sequence to obtain a plurality of sequences of scaled pilot symbols, and to combine the plurality of sequences of scaled data symbols with the plurality of sequences of scaled pilot symbols to obtain a plurality of sequences of combined symbols.

9. The apparatus of claim 1, wherein the processor is operative to form a first sequence of data symbols, to multiply the first sequence of data symbols with a first orthogonal sequence to obtain a scaled sequence of data symbols, to multiply the first sequence of pilot symbols with a second orthogonal sequence to obtain a scaled sequence of pilot symbols, and to combine the scaled sequence of data symbols with the scaled sequence of pilot symbols to obtain a combined sequence of symbols.

10. The apparatus of claim 1, wherein the first sequence of pilot symbols is sent on a first set of frequency subbands and the data symbols are sent on a second set of frequency subbands that contain more frequency subbands than the first set of frequency subbands.

11. A method of generating pilots in a communication system, comprising:

forming a first sequence of pilot symbols based on a polyphase sequence; and

the first sequence of pilot symbols is replicated a plurality of times to obtain a second sequence of pilot symbols for transmission via a communication channel.

12. The method of claim 11, further comprising:

a phase ramp is applied to the second pilot symbol sequence to obtain a third output symbol sequence.

13. The method of claim 11, further comprising:

adding a cyclic prefix to the second sequence of pilot symbols to obtain a third sequence of output symbols; and

transmitting the third sequence of output symbols in a time domain via the communication channel.

14. An apparatus, comprising:

means for generating a first sequence of pilot symbols based on a polyphase sequence; and

means for replicating the first sequence of pilot symbols a plurality of times to obtain a second sequence of pilot symbols suitable for transmission via a communication channel.

15. The apparatus of claim 14, further comprising:

means for applying a phase ramp to the second sequence of pilot symbols to obtain a third sequence of output symbols.

16. The apparatus of claim 14, further comprising:

means for adding a cyclic prefix to the second sequence of pilot symbols to obtain a third sequence of output symbols; and

means for transmitting the third sequence of output symbols in the time domain via the communication channel.

17. An apparatus, comprising:

a processor configured to form a first sequence of pilot symbols based on a polyphase sequence; and

a modulator for transforming the first sequence of pilot symbols to the frequency domain to obtain a second sequence of frequency-domain symbols, forming a third sequence of symbols by mapping the second sequence of frequency-domain symbols onto a group of frequency subbands used for pilot transmission, and transforming the third sequence of symbols to the time domain to obtain a fourth sequence of output symbols for transmission via the communication channel.

18. The apparatus of claim 17, wherein the modulator is operative to add a cyclic prefix to the fourth sequence of pilot symbols to obtain a fifth sequence of output symbols suitable for transmission in the time domain via the communication channel.

19. The apparatus of claim 17, wherein the polyphase sequence has a constant envelope in the time domain and a flat spectral response in the frequency domain.

20. The apparatus of claim 17, wherein data symbols are sent on a second group of frequency subbands that includes more frequency subbands than the group of frequency subbands used for pilot transmission.

21. An apparatus, comprising:

a processor configured to form a pilot symbol sequence, form a data symbol sequence, and time-division multiplex the data symbol sequence and the pilot symbol sequence; and

a modulator to generate at least one single-carrier frequency division multiple access (SC-FDMA) symbol based on the time-division multiplexed data symbols and pilot symbols.

22. The apparatus of claim 21, wherein the processor is operative to multiplex the sequence of data symbols in a first symbol period and multiplex the sequence of pilot symbols in a second symbol period, and the modulator generates a first SC FDMA for the sequence of data symbols in the first symbol period and generates a second SC-FDMA symbol for the sequence of pilot symbols in the second symbol period.

23. The apparatus of claim 21, wherein the processor is operative to multiplex the sequence of data symbols with the sequence of pilot symbols in different sample periods of a symbol period, and wherein the modulator is operative to generate SC-FDMA symbols for the multiplexed pilot and data symbols of the symbol period.

24. An apparatus, comprising:

means for forming a sequence of pilot symbols;

means for forming a sequence of data symbols;

means for time division multiplexing the sequence of data symbols with the sequence of pilot symbols; and

means for generating at least one single-carrier frequency division multiple access (SC-FDMA) symbol based on the time-division multiplexed data symbols and pilot symbols.

25. The apparatus of claim 24, wherein the means for time division multiplexing the sequence of data symbols with the sequence of pilot symbols comprises:

means for multiplexing said sequence of data symbols in a first symbol period, an

Means for multiplexing the sequence of pilot symbols in a second symbol period.

26. The apparatus of claim 24, wherein the means for time division multiplexing the sequence of data symbols with the sequence of pilot symbols comprises:

means for multiplexing the sequence of data symbols with the sequence of pilot symbols in different sample periods of a symbol period.

27. An apparatus, comprising:

a processor for forming a pilot symbol sequence and forming a data symbol sequence; and

a modulator to generate a broadband pilot based on the pilot symbols, to generate at least one single-carrier frequency division multiple access (SC-FDMA) symbol based on the sequence of data symbols, and to time-division multiplex the broadband pilot with the at least one SC-FDMA symbol.

28. The apparatus of claim 27, wherein the processor is operative to form the sequence of pilot symbols based on a pseudo-random number (PN) sequence.

29. The apparatus of claim 27, wherein the modulator is operative to generate at least one interleaved fdma (ifdma) symbol or at least one localized fdma (lfdma) symbol corresponding to the sequence of data symbols.

30. The apparatus of claim 27, wherein the broadband pilot is pseudo-random with respect to at least one other broadband pilot from at least one other transmitter.

31. The apparatus of claim 27, wherein the broadband pilot is time aligned with at least one other broadband pilot from at least one other transmitter.

32. An apparatus, comprising:

a demodulator to receive at least one single-carrier frequency division multiple access (SC-FDMA) symbol via a communication channel and process the at least one SC-FDMA symbol to obtain a time-domain received pilot symbol; and

a processor configured to transform the received pilot symbols to obtain frequency-domain pilot values and to derive a frequency response estimate for the communication channel based on the frequency-domain pilot values and using a Minimum Mean Square Error (MMSE) technique or a Least Squares (LS) technique.

33. The apparatus of claim 32, wherein the processor is operative to derive a channel impulse response estimate for the communication channel based on the frequency response estimate.

34. The apparatus of claim 32, wherein the processor is operative to filter the frequency response estimate.

35. The apparatus of claim 33, wherein the processor is operative to filter the channel impulse response estimate.

36. The apparatus of claim 32, wherein the processor is operative to derive frequency response estimates for SC-FDMA symbols sent on at least two sets of frequency subbands, to derive channel impulse response estimates based on the frequency response estimates, and to filter the channel impulse response estimates to obtain extended channel impulse response estimates having more taps than each of the channel impulse response estimates.

37. The apparatus of claim 33, wherein the processor is operative to reserve a predetermined number of taps in the channel impulse response estimate and to set remaining taps in the channel impulse response estimate to zero.

38. The apparatus of claim 37, wherein the processor is operative to select the predetermined number of taps based on a signal-to-noise-and-interference ratio (SNR) or spectral efficiency of a data transmission via the communication channel.

39. The apparatus of claim 33, wherein the processor is operative to reserve taps in the channel impulse response estimate that exceed a predetermined threshold and to set remaining taps in the channel impulse response estimate to zeros.

40. The apparatus of claim 33, wherein the processor is operative to reserve the first L taps in the channel impulse response estimate and to set the remaining taps in the channel impulse response estimate to zero, wherein L is an integer equal to or greater than 1.

41. The apparatus of claim 32, wherein the demodulator is operative to demultiplex received symbols of the at least one SC-FDMA symbol into received data symbols and received pilot symbols.

42. The apparatus of claim 32, wherein the demodulator is operative to process the at least one SC-FDMA symbol with an orthogonal sequence corresponding to a pilot to obtain the received pilot symbol.

43. The apparatus of claim 32, further comprising:

an equalizer for equalizing the received data symbols based on the frequency response estimate.

44. An apparatus, comprising:

means for processing at least one single-carrier frequency division multiple access (SC-FDMA) symbol received via a communication channel to obtain a received pilot symbol;

means for transforming the received pilot symbols to obtain frequency-domain pilot values; and

means for deriving a frequency response estimate corresponding to the communication channel based on the frequency domain pilot values and using a Minimum Mean Square Error (MMSE) technique or a Least Squares (LS) technique.

45. The apparatus of claim 44, further comprising:

means for deriving a channel impulse response estimate corresponding to the communication channel based on the frequency response estimate, an

Means for setting at least one tap of the channel impulse response estimate to zero.

46. The apparatus of claim 44, further comprising:

means for filtering at least two frequency response estimates derived from at least two SC-FDMA symbols corresponding to at least two symbol periods.

47. An apparatus, comprising:

a demodulator for receiving a pilot and processing the received pilot to obtain received pilot symbols, the pilot being comprised of multiple copies of a sequence of pilot symbols generated using a polyphase sequence; and

a processor configured to accumulate received pilot symbols corresponding to the plurality of copies of the pilot symbol sequence.

48. An apparatus, comprising:

a demodulator to receive at least one single-carrier frequency division multiple access (SC-FDMA) symbol via a communication channel and time division de-multiplex received symbols of the at least one SC-FDMA symbol into received data symbols and received pilot symbols; and

a processor configured to derive a channel estimate corresponding to the communication channel based on the received pilot symbols.

49. An apparatus, comprising:

a demodulator to receive at least one single-carrier frequency division multiple access (SC-FDMA) symbol via a communication channel and process the at least one SC-FDMA symbol to obtain a received pilot symbol; and

a processor configured to derive a channel estimate corresponding to the communication channel based on the received pilot symbols and using a Least Squares (LS) technique.

50. An apparatus, comprising:

a demodulator to receive at least one single-carrier frequency division multiple access (SC-FDMA) symbol via a communication channel and process the at least one SC-FDMA symbol to obtain a time-domain received pilot symbol; and

a processor configured to identify at least one tap corresponding to a channel impulse response estimate for the communication channel by correlating the received pilot symbols with transmitted pilot symbols at different time offsets.

51. The apparatus of claim 50, wherein the at least one SC-FDMA symbol comprises pilot symbols and data symbols multiplexed over multiple sample periods, and the demodulator is operative to demultiplex received pilot symbols and received data symbols in the at least one SC-FDMA symbol.

52. An apparatus, comprising:

a demodulator to receive at least one single-carrier frequency division multiple access (SC-FDMA) symbol via a communication channel, process the at least one SC-FDMA symbol to obtain received pilot symbols and received data symbols, and process the received data symbols with a first channel estimate corresponding to the communication channel to obtain data symbol estimates; and

a first processor configured to derive the first channel estimate based on the received pilot symbols, to estimate interference due to the received data symbols based on the first channel estimate and the data symbol estimates, to derive interference-canceled pilot symbols based on the received pilot symbols and the estimated interference, and to derive a second channel estimate based on the interference-canceled pilot symbols.

53. The apparatus of claim 52, further comprising:

a second processor for processing the data symbol estimates to obtain decoded data and processing the decoded data to obtain remodulated data symbols, and the first processor is for estimating the interference based on the remodulated data symbols.

54. The apparatus of claim 52, wherein the demodulator and the first processor are operative to, for a plurality of iterations: deriving the data symbol estimate, estimating the interference, deriving the interference canceled pilot symbol, and deriving the second channel estimate.

55. An apparatus, comprising:

a demodulator to receive at least one single-carrier frequency division multiple access (SC-FDMA) symbol via a communication channel and process the at least one SC-FDMA symbol to obtain received pilot symbols and received data symbols; and

a first processor configured to derive a first channel estimate corresponding to the communication channel based on the received pilot symbols, to derive a second channel estimate based on the received data symbols, and to derive a third channel estimate based on the first and second channel estimates.

56. The apparatus of claim 55, wherein the demodulator is operative to process the received data symbols with the first channel estimate to obtain data symbol estimates.

57. The apparatus of claim 56, wherein the first processor is operative to derive the second channel estimate based on the received data symbols and the data symbol estimates.

58. The apparatus of claim 56, further comprising:

a second processor for processing the data symbol estimates to obtain decoded data and processing the decoded data to obtain remodulated data symbols, and the first processor is for deriving the second channel estimates based on the received data symbols and the remodulated data symbols.

59. The apparatus of claim 55, wherein the first processor is operative to derive the third channel estimate based on a function of the first channel estimate, the second channel estimate, and an indication of confidence in reliability of the data symbol estimates.