HK1118137A - Method and apparatus for reducing interference with outer loop power control in a wireless communication system - Google Patents

Description

Method and apparatus for reducing interference in a wireless communication system

The present application is a divisional application of an invention patent entitled "method and apparatus for reducing interference in a wireless communication system" filed as 11/1/2004 and having an application number of 03809891.1.

Claiming priority in accordance with 35U.S.C. § 119(e)

This patent application claims priority from U.S. provisional application No. 60/364442, filed on 3, 14, 2002, which is assigned to the assignee of the present invention and incorporated herein by reference.

Technical Field

The present invention relates generally to wireless communication systems, and more particularly to methods and apparatus for reducing interference in wireless communications.

Background

There is an increasing need for packetized data services over wireless communication systems. In systems such as wideband code division multiple access (W-CDMA), the various channels are multiplexed together and transmitted on a single physical channel. Various other channels, such as the synchronization channel, are transmitted in parallel over the common air link. The channels may introduce interference to each other in a given situation. For example, the synchronization channel may cause interference to other channels since the synchronization channel is not constrained to be orthogonal to other physical channels.

There is therefore a need for a method to reduce inter-channel interference in a wireless communication system.

Drawings

FIG. 1 is a wireless communication system

Fig. 2 is a frame structure of a downlink physical channel in a wireless communication system using W-CDMA.

Fig. 3 is a frame structure of a downlink physical control channel in a wireless communication system using W-CDMA.

Fig. 4 is a Synchronization Channel (SCH) structure of a wireless communication system using W-CDMA.

Fig. 5 is a graph illustrating the signal-to-noise ratio (SNR) associated with various codes due to SCH interference.

Fig. 6 and 7 illustrate interleaving in a W-CDMA system.

Fig. 8A and 8B are Protocol Data Unit (PDU) formats in the W-CDMA system.

Fig. 9 and 10 are wireless devices that reduce interference.

Fig. 11 illustrates coding of an adaptive multi-rate/dedicated control channel (AMR/DCCH) transport channel.

Fig. 12 illustrates simulated SNR values for various transport format requirements.

Fig. 13 illustrates SNR requirements for various Transport Format Combinations (TFCs).

Fig. 14 illustrates a table of power offset values for corresponding TFCs.

Fig. 15 illustrates a flow diagram of a process for mapping TFCs to power offset values.

Fig. 16 illustrates a timing diagram for various scenarios of transmission of multiple transport channels (TrCHs) on a common physical channel.

Detailed Description

The word "exemplary" is used herein to mean "serving as an example, instance, or illustration. Any embodiment described herein as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments. While various aspects of the embodiments are illustrated in the accompanying drawings, the drawings are not necessarily drawn to scale unless specifically indicated.

It is worthy to note that the example embodiments are provided herein as examples discussed throughout; however, other embodiments may include various aspects without departing from the invention. In particular, the various embodiments may be applied to data processing systems, wireless communication systems, mobile IP networks, and any other system in which efficient use and management of resources is desired.

The exemplary embodiments provide a spread spectrum wireless communication system using wideband code division multiple access (W-CDMA). Wireless communication systems are widely deployed to provide various types of communication such as voice, data, and so on. These systems may be based on Code Division Multiple Access (CDMA), Time Division Multiple Access (TDMA), or some other modulation technique. CDMA systems offer advantages over other types of systems, including increased system capacity.

The System may be designed to support one or more CDMA standards, such as "TIA/EIA-95-B Mobile Station-Base Station Compatibility Standard for Dual-Mode Wireless broadband Spectrum Cellular System" (IS-95 Standard), by "3^rdThe standard provided by Generation Partnership Project (3GPP) is embodied in a set of documents including Nos.3G TS 25.211, 3G TS 25.212, 3G TS 25.213, and 3G TS 25.214(W-CDMA standard), by "3G TS 25.211^rdThe standard provided by Generation Partnership Project 2 "(3 GPP2), and TR-45.5, which IS known as the cdma2000 standard, was previously known as IS-2000 MC. The above criteria are specifically incorporated herein by reference.

Each standard specifically defines the processing of data for transmission from a base station to a mobile station and vice versa. As an exemplary embodiment, the following discussion considers a spread spectrum communication system that conforms to the cdma2000 protocol standard. Other embodiments may include other criteria.

The W-CDMA system is described in the set of specification documents defined by the 3GPP, which addresses ETSIMobile Competition Centre, 650, Route des Lucioles, 06921 Sophia-antipolisCedex, France.

Fig. 1 is an illustration of a communication system 100 that can support multiple users and that can implement at least some aspects of the embodiments discussed above. Any of a variety of algorithms and methods may be used to schedule transmissions within system 100. The system 100 provides communication for a plurality of cells 102A-102G, each respectively served by a corresponding base station 104A-104G. In the exemplary embodiment, some base stations 104 have multiple receive antennas, and others have only one receive antenna. Similarly, some base stations 104 have multiple transmit antennas, and others have only one transmit antenna. There is no limitation on the combination of the transmit and receive antennas. Thus, the base station 104 may have multiple transmit antennas and one receive antenna, or multiple receive antennas and one transmit antenna, or both single or multiple transmit antennas.

Terminals 106 within the coverage area may be fixed (i.e., stationary) or mobile. As shown in fig. 1, different terminals 106 are dispersed throughout the system. Each terminal 106 communicates with at least one and possibly more base stations 104 on the downlink and uplink at any given moment, depending on, for example, whether soft handoff is employed or whether the terminal is designed and used to receive multiple transmissions (concurrently or sequentially) from multiple base stations. Soft handoffs in CDMA communication systems are well known in the art and are described in detail in U.S. Pat. No. 5101501, entitled "Method and system for providing a Soft Handoff in a CDMA Cellular TelephoneSystems", assigned to the assignee of the present invention.

A W-CDMA system is provided as an example of the present discussion. It is noted that in a W-CDMA system, a base station is referred to as a node B and a mobile station is referred to as a User Equipment (UE). Other embodiments may use other communication systems where various channels may represent interference to other channels within the system. In particular in W-CDMA systems, channels are encoded and transmitted over the same air interface, with at least one channel being non-orthogonal to the other channels. In spread spectrum systems, the orthogonality of the channels avoids inter-channel interference. Thus, non-orthogonal channels present potential problems that lead to inter-channel interference. However, the invention is applicable to other systems in which one or more channels interfere with other channels.

Returning to fig. 1, the downlink refers to transmission from the base station 104 to the terminal 106, and the uplink refers to transmission from the terminal 106 to the base station 104. In the exemplary embodiment, some terminals 106 have multiple receive antennas, and others have only one receive antenna. In fig. 1, base station 104A transmits data to terminals 106A and 106J on the downlink, base station 104B transmits data to terminals 106B and 106J, base station 104C transmits data to terminal 106C, and so on.

Fig. 2 illustrates a downlink dedicated physical channel, a downlink dedicated physical channel (downlink DPCH) in a W-CDMA system. A plurality of logical channels called transport channels (TrCH) are multiplexed to form one physical channel, i.e., DPCH. In other words, dedicated data generated by a higher layer is multiplexed together within one downlink DPCH. A dedicated transport channel (DCH) is sent in a time-multiplexed manner along with control information such as pilot bits, Transmit Power Control (TPC) commands, and an optional Transport Format Combination Indicator (TFCI). The downlink DPCH may thus be seen as a time multiplex of a downlink Dedicated Physical Data Channel (DPDCH) and a downlink Dedicated Physical Control Channel (DPCCH).

Fig. 2 illustrates the frame structure of the downlink DPCH. Each frame of length 10ms is divided into 15 slots of length T_SLOT2560 chips, corresponding to one power control period. As illustrated, the DPDCH part alternates with the DPCCH part. In an example, the time slot includes N of DATA1_DATA1The first DPDCH part of the bit, followed by N with TPC_TPCBits and N of TFCI_TFCIThe DPCCH part of the bit. The next part is with N_DATA2The DPDCH portion of the DATA2 of the bits. The last part is with N_PILOTDPCCH part of the PILOT of a bit.

The parameter k determines the total number of bits per downlink DPCH slot. The parameter k is related to the Spreading Factor (SF) of the physical channel, where SF 512/2^K. The spreading factor ranges from 512 to 4.

Also, transmitted in the W-CDMA system is a synchronization sequence on a Synchronization Channel (SCH). Is worthy ofNote that the synchronization sequence may also be referred to as a synchronization message. As described in detail in 3GPP TS 25.211, section 5.3.3.5, the SCH is defined as non-orthogonal to other channels, in particular to the DPCH. The SCH is a downlink signal used by the UE for cell search. The SCH includes two sub-channels, a primary and a secondary SCH. The 10ms radio frame for the primary and secondary SCH is divided into 15 slots each of length 2560 chips. Fig. 3 illustrates the structure of an SCH radio frame. The primary SCH includes a modulated code of length 256 chips, denoted as c_pIs transmitted once per slot. The PSC is the same for every cell in the system.

The secondary SCH comprises a repeated transmission of 15 consecutive sequences of a modulated code of length 256 chips, with a Secondary Synchronization Code (SSC) transmitted in parallel with the primary SCH. SSC is denoted c in FIG. 3_s ^i，k(ii) a Wherein i is 0, 1.., 63 and denotes the number of the scrambling code group; and wherein k is 0, 1, 14 and denotes a slot number. Each SSC is selected from a set of 16 different codes of length 256. This sequence on the secondary SCH indicates to which code group the downlink scrambling code of the cell belongs. It is noted that the synchronization message is transmitted at a predetermined position within each time slot. Thus, the synchronization message has a known number of occurrences.

Various aspects of a WCDMA system are described below, which in combination may lead to the problems discussed above. In particular, the following discussion covers certain vulnerabilities in SCH and DPCH interaction, interleaving and channel mapping, reference configuration, power control, and layer 2(L2) Acknowledgement (ACK) messages. It is noted that similar inter-channel interference may come from other channels, but the SCH is described as an example.

DPCH and SCH interaction

Regarding DPCH and SCH interactions, SCH is a special signal consisting of two 256 chip sequences; primary sch (psch) and secondary sch (ssch). The two sequences are transmitted in parallel during each time slot on the downlink transmission. The SCH is transmitted with a 10% duty cycle in each slot. The SCH is used primarily by a terminal or UE to obtain system timing and to help identify cells available to the UE. In other words, the SCH reduces the number of assumptions made by the UE during cell identification.

Although the primary and secondary synchronization coding (PSC, SSC) structures described in 3GPP TS 25.213 are not orthogonal, they are designed to provide maximum isolation between the synchronization channel and other downlink channels. The isolation depends on the spreading factor of the DPCH under consideration to cover the scrambling code segmentation of a particular symbol.

Fig. 4 illustrates the signal-to-noise ratio (SNR) of the worst case coded bits due to SCH interference in dB. The result illustrated in fig. 4 is that the transmission of SCH and DPCH assumes equal power (different values reflect different SSC and DPCH OVSF code indices). The left column indicates the DPCH Spreading Factor (SF). The right column indicates the worst case SNR due to SCH interference on DPCH. Notably, when considering multipath or transmit diversity techniques, the SNR margin generally does not improve because SCH interference is deterministic and sufficiently correlated (i.e., not Average White Gaussian Noise (AWGN)). The median SNR for the non-orthogonal case (about 50% of the combinations result in SCH orthogonality with respect to DPCH) is typically 5dB higher than the worst case SNR case.

The SNR margin becomes a limiting factor when considering high geometries. High geometry refers to the ratio of the total power received from the target cell to the total power received from all cells. The total power includes interference, which is introduced by the environment and other transmission channels. The closer the UE is to the node B, the higher the geometry. It is noted that the channels within a typical cell are orthogonal (except for special cases such as SCH); however, from cell to cell, the channels are not necessarily orthogonal. Thus, for high geometry, the orthogonal channel seen by the UE is close to the node B. The interference presented by non-orthogonal channels such as SCH is large. In contrast, for low geometry positions, the non-orthogonal interference seen by the UE is far away from the node B. The SCH channel is not significant at low geometries because SCH power is added to other interference causing little difference. It is also considered that the SCH is constantly transmitted at the same power level, but the dedicated channel is transmitted according to the location of the UE, the SCH has a greater impact at higher geometries.

The international mobile telecommunications system (UMTS) multiplexing and interleaving scheme is described within 3GPP TS 25.212. The various transport channels are first encoded and separately encoded and interleaved on a per Transmission Time Interval (TTI) basis. The channels are then multiplexed on a radio frame basis, interleaved and mapped onto the physical channel payloads. The transport channel mapping may be according to fixed or flexible location rules.

Fig. 5 illustrates the mapping of multiple logical channels onto a common physical channel. The logical channels are illustrated as transport channel 1204A, transport channel 2204B, and transport channel 3204C. Three transport channels 204A, 204B, 204C are mapped to the physical channel 202. Each channel bit is separately interleaved. It is noted that in a W-CDMA system, each frame includes 15 slots, where each slot includes 2560 chips. The data information is time multiplexed with the control information provided in the known gaps.

Interleaving involves two steps, considering a first interleaver and a second interleaver. There is a potential problem when the structure of the first interleaver (described in detail below) causes a problem of repeating every radio frame within a TTI. In addition, the structure and associated periodicity of the second interleaver is similar to the slot period of the physical channel, and thus, the SCH periodicity is another potential factor that leads to SCH interference problems.

Fig. 6 illustrates the first interleaving procedure 300 in radio frame segmentation, assuming a 40ms TTI. The first interleaving procedure basically guarantees that TTI bits are spread over multiple radio frames. However, as can be seen at the location of the grey area, the first interleaving process does not affect the relative position of the bits within each radio frame compared to its relative position within the transport packet. As illustrated, a 40ms TTI includes four frames of 10ms each. The TTI is identified as TTI302, and the frames are identified as frames 304A, 304B, 304C, and 304D. Each frame is then divided into four parts. The frame division corresponds to the number of frames per TTI. The frame portions are then interleaved together to form a radio frame stream 306. The shaded portion identifies the process of the TTI frame 304A. The interleaving process 300 involves writing a TTI frame row by row and then reading out the frame portion column by column. The order of the interleaved portions is predetermined and predictable.

The interleaving process 350 of the second interleaver is illustrated in fig. 7 for three transport channels. The interleaver is based on a 30-column matrix, where the number of columns corresponds to the number of time slots per frame. From the radio frames 352, each frame is divided into 30 portions to form a matrix 354. The portions are then interleaved to form a final interleaved stream 356. The second interleaver ensures that some information from each transport channel is present in each slot. The second interleaver does not, however, change the relative position of each transport channel information within each slot, with the exception that each transport channel occurs with a period that is twice the slot period (1500Hz) (3000 Hz). However, if the periodic interference, such as SCH generation, lasts for as long or longer than the transmission gap of a given transmission channel within a given slot, the interference may affect half of the symbols of the particular transmission channel.

The order in which the transport channels are mapped onto the physical channels affects the absolute position of each transport channel, but not the periodicity of the data from each transport channel, which is always 3000 Hz. Furthermore, the mapping order does not affect the fact that the specific transport channel information always appears at the same position within the time slot.

Different position mappings result in that specific transport channel information is present at different positions (transport format combinations, or TFCs) of each combination of transport channel information elements, wherein a fixed position ensures that the transport channel information always appears at the same position within a time slot, regardless of the TFC. Thus, the mapping locations do not mitigate the periodic SCH interference problem.

Reference configuration for DCCH

In simulation results, a given transport channel may be affected by SCH interference, which is a function of channel position within the transport channel multiplexing structure. Up to half of the symbols within a given transmission channel may be affected by SCH interference. The worst-case condition occurs when the transport channel rate is low relative to other transport channels that are multiplexed together. In particular, if the relative size of the transport channel is less than 10% and the transport channel is an end channel, i.e. the first or last channel being multiplexed, half of the transport channel symbols may be affected by the SCH for some frame offsets.

Notably, for an interleaving process, where data is provided at variable locations within each slot, the problem of inter-channel interference is not severe. The variable location of the data means that not all data occurrences will interfere with the non-orthogonal channel. In W-CDMA, however, data is consistently transmitted at the same location within a slot. Therefore, inter-channel interference causes a large problem. Inter-channel interference can be a problem in another type of system, especially if data or control information is provided during a consistent location of each time slot.

Considering the reference channel configuration described in 3GPP TS 34.08, the Dedicated Control Channel (DCCH) is the lowest rate channel in most configurations and therefore may be most affected by SCH interference. Table 1 shows the relative size of the DCCH for various configurations on a per radio frame basis.

Table 1

Reference arrangement	Relative DCCH load occupancy
		DL 12.2+3.4(kbps)	～20.0％
DL 64+3.4(kbps)	～5.0％
		DL 384+3.4(kbps)	～0.9％

The reference configuration includes a first transmission rate of data followed by a second transmission rate of the DCCH. For example, in the first row, the Downlink (DL) defines a DCH data rate of 12.2kbps and a DCCH rate of 3.4 kbps. The first line refers to voice communication; the second line refers to video; the third row refers to packet data communication. The relative DCCH payload occupancy is calculated as the DCCH rate divided by the combined data plus DCCH rate. As an example, for voice communications defined in the first row, the occupancy is determined as:

occupancy ratio DCCH rate/(data rate + DCCH rate)%

Namely:

20％＝3.4/(12.2+3.4)

the relative DCCH payload occupancy is driven by the most required TFC within the TFCS rather than the instantaneous TFC. For example, for the 384+3.4 case, even if the instantaneous DTCH rate is 0kbps, the DCCH load is 0.9% of the total load, where the rate is not transmitted, i.e., discontinuous transmission mode or DTX.

SCH message transmission on non-orthogonal channels introduces interference to other channels such as the DPCH. The DPCH carries data and control information, and thus may cause any of various problems when the SCH interferes with the control information. As described below, a particular problem is introduced when the SCH interferes with the transmission of pilot bits.

The inter-channel interference occurs because data (or control information) has the same periodicity as synchronization information. This problem is true for any non-orthogonal channel transmitted within the system. The problem results in loss of data and control information, incorrect power control of the system, and/or increased use of power for transmissions within the system. These factors are discussed below. It is worth noting that many of these problems are mutually exclusive. For example, when the interference affects data, there is no impact on the control because the control is sent at different times.

The problem is not limited to SCH but may result in any non-orthogonal channel. The various solutions described below assume that the interfering channel is a deterministic component of the interference and that there is a known transmission period. In an example embodiment, the SCH transmission period is a multiple slot period. Additionally, in an example embodiment, the interfering channel is transmitted once per frame or slot, and the presence of the interfering channel may be identified by the receiver. In fact, any channel will severely cover other channels.

Various solutions are described below and include, but are not limited to, the following concepts:

1. an outer-loop target based on the weakest link;

2. data scrambling (improving coding robustness);

3. avoiding frame offsets that result in bit alignment of the SCH with the DCCH;

4. an aperiodic interleaver;

5. low weighting of symbols affected by SCH;

SCH suppression/cancellation; and

7. the power increases.

For certain factors discussed above, 1) and 2) may better ensure that the DCCH can be received by the UE at a desired error rate. However, 1) and 2) do not address the source of the problem, namely SCH interference.

Outer loop power control with highest quality of service

The end result associated with SCH interference is a radio link loss, i.e., a call loss. Especially when setting DL DPCH power, if the network is configured without regard to DCCH error rate. In this configuration, the network does not adjust the power allocation when the DCCH is subject to SCH interference and experiences a high error rate. Thus, higher error rate conditions will be maintained, higher layer protocols will not be able to exchange messages with the required reliability, and the radio link will eventually be dropped.

One embodiment attempts to address the problems associated with inter-channel interference by an outer loop power control mechanism for the quality of the DCCH. It is worth noting that each transport channel has a unique quality of service (QoS) criterion. In this example, QOS is defined by BLER. This allows for determining the most stringent QOS for all transport channels and for each channel to meet the most stringent requirements, regardless of individual requirements. In fact, the most stringent requirements apply to all transport channels.

It is worth noting that although each transport channel typically has a different QOS target (typically expressed in terms of data packet error rate), they may experience the same symbol error rate under given radio conditions. However, when SCH is added to other channels, DCCH symbols may be affected and the DCCH symbol error rate may be higher than the symbol error rate of other transport channels. This is true when the synchronization message is sent at the same location within the time slot as the control information. This results in a higher error rate for the DCCH.

In W-CDMA, even though the basic physical channel symbol error rate is the same for all transport channels, the system can achieve different quality of service for each transport channel by adjusting the weight of each transport channel to achieve its corresponding selected transport channel packet error rate.

To reduce the probability of radio link loss due to the DCCH experiencing high error rate conditions, the network may be configured such that the DCCH error rate is considered within the power control process. In particular, the network may set a BLER target for the DCCH; the Radio Network Controller (RNC) may set a DCCH packet error rate (BLER) target for a particular transport channel within the UE through Radio Resource Control (RRC) signaling as described in 3GPP TS 25.331. According to 3GPP specifications, the UE power control procedure ensures compliance with each BLER target set for each corresponding transport channel, including the BLER target for the DCCH. Assuming that the network has sufficient power to follow the power control commands received from the UE, outer-loop power control using DCCH will avoid the above described inter-channel interference effects. In general, to set a BLER target on a particular transport channel, the network guarantees that all conditions are met to enable BLER measurements on the transport channel, as described in 3GPP TS 5.125. For the specific case of DCCH, a Cyclic Redundancy Check (CRC) is added to all DCCH transmission modules, including when no data is being transmitted (i.e., a 0-bit packet should be defined for DCCH).

It is worth noting that while the use of a transmission channel affected by the DCCH or other interfering channel in the decision making process of power control may overcome the consequences of interference, such a solution may waste transmission power. The node B or transmitter may transmit using more power than necessary. The increased power may reduce the capacity of the system.

Time offset of SCH

When a particular channel or set of channels is affected by more SCH interference than other transport channels to which they are multiplexed, the system may adjust the SCH to cover the selected transport channel, which is considered to be insensitive or less sensitive to interference. There are several possible ways to implement this method to reduce SCH coverage. Each using a known frame offset which is the relative timing between the intra-cell DPCH radio frame boundary and the common pilot channel radio frame boundary.

In an embodiment, the system selects the DPCH frame offset such that the SCH does not interfere with the transport channel, which may be sensitive to the SCH, particularly the DCCH. Frame offset selection occurs within the RNC of each downlink.

Another embodiment may be used if the RNC has a limitation on the frame offset selection (e.g. due to the desire to allocate DPCCH transmissions in time), i.e. to change the order of the transport channel mapping onto the physical channels. This is also controlled by RRC in the RNC.

Each of these embodiments attempts to move the position of the transmission channel to coordinate the occurrence of the interfering channel with the channel that is predicted to be least sensitive to the interfering channel. It is noted that the node B sets the dedicated channel and thus controls the timing of the dedicated channel, i.e. the timing offset. Note that coordination of multiple node bs may be required during soft handoff. The node B shifts the offset for the dedicated channel based on the timing of the interfering channel, which in this case is the synchronization message channel SCH.

The transmission channel, which is less sensitive to the interfering channel, is generally the channel covering a larger part of the frame. Since the SCH uses only 10% of the frame, SCH transmissions can be completely covered using transmission channels less than or equal to 10% of the frame. In this case, the entire transport channel, i.e., information transmitted on the transport channel, is covered by the SCH. Information is easily lost due to interference from the SCH.

It is also possible to change the order of the transmission channels within the frame. Since the SCH is transmitted at the same location within each frame, changing the transmission order of the other transport channels over multiple frames ensures that the other transport channels do not cover every occurrence of the SCH.

Fig. 16 illustrates various solutions for reducing inter-channel interference. The original configuration maintains a predetermined order of transmission channels and applies an interleaver section for a number of slots per frame. The interference mechanism is identified as SCH. The first example illustrates a change in frame offset, where transport channel B associated with service B is transmitted when SCH occurs. Thus, the SCH has less impact on the transport channel B. In a second example, the transport channel TrCH) is mapped differently to reduce the impact of the SCH. In a third embodiment, the mapping of the transport channels is done on a per slot basis.

Aperiodic interleaver

As mentioned above, SCH interference appears to always affect the same transmission channel for a given downlink configuration. This is mainly due to the interleaver structure, which results in a period of a fully periodic transmission of the transport channel equal to the SCH period.

For example, in the above system, transmission is defined as 15 slots per frame. See fig. 2 and the discussion thereof. The second interleaver illustrated in fig. 7 defines the number of columns as a number of slots per frame. In particular, the number of columns is 30, which is a multiple of 15. It is noted that the order of interleaving of the constituent transport channels is constant. Thus, the periodicity of the interleaved channel is the same as the transmission periodicity. Thus, if a given transport channel has a portion of information (from one column of the interleaver) that is transmitted simultaneously with the SCH, this information will occur concurrently with the SCH each time.

Removing or reducing the persistent periodic interleaver structure can significantly reduce the effect of SCH interference on a particular transport channel. The result is that the SCH effect on all transport channels multiplexed on the same physical channel is shared. Notably, this assumes that multiple transport channels are mapped onto a common physical channel (which is typically the case). Some examples of interleavers that reduce intra-frame periodicity of SCH interference include:

● bit reversal interleaver

● packet interleaver with a number of columns different from a multiple of 15

● any aperiodic interleaver

Weighting received transmissions

The received symbols are typically scaled and combined with other multipath components prior to decoding. The scaling factor is typically a function of the Common Pilot (CPICH) signal-to-noise ratio. Since the SCH adds noise in a deterministic manner, this information may be used by the UE to weight the decoder input symbols affected by the SCH differently.

Consider a PSC that includes a repeat every 0.666.. millisecond and a SCH that includes a SSC that repeats every 10 millisecond radio frame. Unlike other downlink channels, PSCs and SSCs are not scrambled with a downlink scrambling code.

Thus, at the UE, after despreading the incoming signal with the complex conjugate of the downlink scrambling code and decovering the symbols with the OVSF code, the SNR per symbol at the decoder input is given by:

wherein

α ═ complex attenuation coefficient

Beta is a non-orthogonal factor

SF ═ spreading factor

E_ctEnergy per transmission channel chip

E_cschEnergy per SCH chip

I_ocThermal noise plus other cell interference power spectral density

The non-orthogonality factor varies as a function of time and the channelization code used in the downlink.

Once the UE obtains system timing, i.e., "knows" the SCH value and the location in time, i.e., the time occurrence, the UE can determine the weights of the different transmitters. It is noted that the indication and occurrence time of the SCH values imply that the beta values are known as a function of time for each channelization code. It is particularly noteworthy that as the value β increases, the SNR of the symbol deteriorates further.

The decoder input symbols are typically scaled by the common pilot strength before being combined with symbols from other multipath components. The UE may then interpret the common pilot strength from each finger as a time-varying weighting that may be applied to the symbols. Since the UE also knows the value of β, there are several ways in which the effect of additional interference from the SCH can be mitigated. For example, the UE may reduce the weight per symbol proportionally according to the β value. This assumes that:

a. the beta value differs from different multipath components from different node bs for the same symbol.

b. The value of beta differs from the same multipath component of the same node B for different symbols.

In a simpler implementation, the weighting would be zero if β is greater than a predetermined value, and a default value (pilot strength) otherwise. This is equivalent to assuming an erasure when the value of β is greater than a predetermined value.

In Soft Handover (SHO) mode, the symbol is affected by the SCH from one node B (assumed to be node B-1) and not affected by the SCH from another node B (assumed to be node B-2). In this case, the UE may assign a zero weight to the symbols affected by the symbols from node B-1 and a default weight to the symbols from node B-2 before combining the symbols from node B.

Suppression of interference channels

The UE receiver processes received signals, which typically include a composite signal received from one or more serving node bs as well as interference received from other node bs within the network and interference from other interference sources, such as thermal noise. Each serving node B transmits a composite signal that includes the UE-specific signals of all served UEs as well as some shared and overhead signals, such as the common pilot channel (CPICH). The composite signal for a particular node B is received at the UE receiver on a radio channel that introduces gross variations in signal phase and amplitude. If multiple radio propagation paths exist between the node B and the UE, multiple echoes of the transmitted composite signal may be received for different phase and amplitude changes for each echo. This effect is commonly referred to as multipath reception. Each propagation path characteristic within a multipath radio channel may be expressed in terms of a complex channel coefficient and a delay. For the signal component received through that particular propagation path, the channel coefficients define phase and amplitude changes relative to the transmitted signal. The delay defines the propagation delay required for a signal to propagate along a particular propagation path. The different propagation delays of the different propagation paths are also referred to as channel taps or delay taps. Within the UE receiver, the delay estimates (i.e., channel taps) of all relevant propagation paths of the radio channel used by all UE receivers for coherent demodulation, as well as the channel coefficients-or any other relevant information that adequately characterizes the radio channel, such as its complex frequency response-need to be able to be generated to achieve coherent demodulation. Typically, a UE receiver within a CDMA system uses a RAKE receiver to achieve coherent demodulation of the signal received on the associated propagation path. The RAKE receiver may use the phase, amplitude and delay estimates on each of the correlated propagation paths to time align, phase shift appropriately, and weight the signals received on the different propagation paths before combining into one signal. In addition to this coherent demodulation function, the RAKE receiver also enables UE-specific despreading of the CDMA signal. But other receiver architectures such as equalizers are also suitable for implementing coherent demodulation in a CDMA system.

SCH suppression/cancellation is a method for solving interference problems, such as SCH interference with PC bits (affecting UL power control). In an embodiment, this problem is addressed at the UE by identifying the location of the SCH message and canceling the interference caused by the SCH message to other DL channels during those slots in which the SCH message is transmitted.

Especially in third generation CDMA systems, there is potential for non-orthogonal channel component transmissions where other signal components from the same transmitter transmission are subject to increased interference. For example, when a time multiplexed Synchronization Channel (SCH) is transmitted or when a second scrambling code is used for transmission in a universal mobile telecommunications system terrestrial radio access (UTRA) frequency division multiplexed (FDD) system, mutual interference between different signal components is caused. As discussed above, under certain conditions, these non-orthogonal signal components cause significant interference to user or control data transmitted in parallel from the same transmitter. The effect of such interference may be a degradation of decoding performance within the receiver. This may also be caused under better radio conditions, such as when there is no multipath reception (i.e., single path reception), and when there is no or little fading. Decoding performance is greatly degraded especially when the user or control data being decoded in the receiver is transmitted as non-orthogonal signal components in close or identical time slots. The interference is large when there is time overlap of sufficient information with the interfering signal.

The interference contribution may be reduced by canceling the interference component, i.e. the non-orthogonal component. An embodiment of an apparatus for suppressing SCH is illustrated in fig. 9. The apparatus 400 includes a receiver front end 402 that includes an analog-to-digital converter, where the received signal is first processed by a UE within the receiver front end 402. Unit 402 is coupled to searcher 404, channel estimator 406, and estimator 408 for estimating interference components caused by non-orthogonal transmission signals. Searcher 404 uses a priori knowledge of the transmitted signal component, such as the common pilot channel (CPICH), to provide channel estimator 406 with information about significant delay taps. This can be achieved, for example, by using sliding correlation with a priori known symbols of the CPICH. Channel estimator 406 continuously outputs significant delay taps and corresponding channel tap coefficients, which are derived, for example, from the correlation of a priori known symbols of the CPICH for a given delay tap. Channel estimator 406 is coupled to demodulator 410, and demodulator 410 is further coupled to decoder 412. The output of estimator 408 is an estimated interference component, which is then subtracted from the output of unit 402. In this way, the interference component, i.e. the power of the interfering channel, is subtracted from the received signal. This is prior to demodulation. It is noted that the arrangement of fig. 9 corresponds to a rake receiver, i.e. a diversity receiver.

In operation, the received signal is considered to be affected by interference caused by non-orthogonal transmitted signals or "interference components". An interference component is estimated. The estimation of the interference component at the transmitter can be highly accurate when the relative strength of the interference component is sufficiently high. This condition is generally met for SCH in UTRA FDD systems, where-12 dB is a general power level compared to the total transmit power. Furthermore, when the data transmitted with the interference component is known at the receiver, this knowledge can be used to improve the quality of the estimated interference component at the receiver.

After the interference component is estimated, the total received signal is modified to reduce the effect of the interference component. In the ideal case, this interference component is cancelled. The modified received signal is then used to decode transmitted user and/or control data, such as an unmodified received signal in a conventional receiver. Due to the reduction of interference within the received signal, the decoding performance of the user and/or control data is improved. In particular, this decoding improvement may be desirable when user and/or control data contained within a transport packet is transmitted in parallel with non-orthogonal signal components. Various embodiments may be implemented to mitigate the effects of interference components.

A first embodiment subtracts a suitable digital representation of the estimated interference component at the input of each rake finger within the rake receiver. A second embodiment subtracts a suitable digital representation of the estimated interference component at the output of each rake finger within the rake receiver. A third embodiment subtracts the integrated digital representation of the interference component in the digital domain from the a/D converted received signal at the input of the digital receiver. A fourth embodiment subtracts a suitable digital representation of the estimated interference component at the output of the rake finger combiner in the rake receiver. The choice of which of the four embodiments is the most effective solution to the cancellation problem depends on design factors such as the sampling rate at the output of the a/D converter, the sampling rate at the input of the rake finger, the bit resolution at the output of the rake finger combiner, and others. For example, if the interference component is cancelled at the output of the a/D converter, the bit resolution of the estimated interference component is typically rather low, i.e. the accuracy of the estimated interference component does not need to be high. However, the sampling rate at the output of the a/D converter is typically much higher than the sampling rate at the input of the rake fingers.

As described above, when a time multiplexed Synchronization Channel (SCH) is transmitted on the downlink of a UTRA FDD system, mutual interference between different transmitted signal components may result. In particular, signals to be used as phase references, such as the common pilot channel (CPICH) in UTRA FDD systems, are subject to increased interference due to non-orthogonal transmission of other downlink signals such as the SCH. Consider a pilot signal, such as the CPICH, known a priori by the receiver, for generating phase and/or amplitude estimates of the channel coefficients to enable coherent demodulation. The quality of the phase and/or amplitude estimates deteriorates when the non-orthogonal signal components are transmitted in parallel to the phase reference signal. Deterioration of the phase and/or amplitude estimates of the channel coefficients can lead to deterioration of the demodulation and decoding performance of the receiver.

To follow the changes in the channel coefficients over time, the receiver using coherent demodulation continuously updates the phase and/or amplitude estimates of the channel coefficients. Since the temporal variation of the channel coefficients is limited by the maximum doppler shift, the implementation of the channel estimator uses low-pass filtering of a continuous channel estimator to improve the estimation quality by "averaging" over a reasonable period of time. This filtering is also referred to as "pilot filtering". The higher the expected maximum doppler shift, the shorter the selected "average" gap. In the case of time multiplexed transmission of non-orthogonal signal components, such as the SCH in a UTRA FDD system, the quality of the estimate of successive phase and/or amplitude estimates of the channel coefficients varies depending on the presence of non-orthogonal signal components and the relative power levels. In the conventional state of the receiver, the pilot filtering process does not take into account different levels of estimation quality of the channel coefficient estimates and uses all channel coefficient estimates generated in the same way, i.e. assuming that the estimation quality of successive estimates does not change.

If the time gap in which the non-orthogonal signal components are transmitted is known a priori, this information may be taken into account to reduce the correlation of the channel coefficient estimates generated during such time gaps. This can be done by introducing a weighting factor for the channel coefficient estimates that is proportional to their respective estimated quality, i.e. in the sense of the channel estimate signal-to-noise-and-interference ratio (SNIR). In an extreme case, the weights may be chosen such that estimates affected by interference from non-orthogonal transmitted signals are not used at all (weighting factors of zero). Since the channel coefficient estimates affected by such increased interference are less important in the pilot filtering process, the resulting channel estimate quality is thereby improved. Even if pilot filtering is not used, information about the presence of interference due to non-orthogonal transmitted signals can be used to skip the channel coefficient estimates generated in these time slots, which estimates and reuse the older estimates.

As described in the preceding paragraphs, the information about the transmitted time gap may be used in different ways to reduce channel estimation errors. Various embodiments and implementations include:

● skips channel coefficient estimates that are subject to increased interference due to the presence of non-orthogonal transmitted signals and reuses the most recently unaffected estimates.

● replace the channel coefficient estimates of increasing interference impact due to the presence of non-orthogonal transmitted signals with the average of the previous and current channel coefficient estimates.

● skips channel coefficient estimates that are subject to increased interference due to the presence of non-orthogonal transmitted signals and replaces them with an average of previous and next channel coefficient estimates.

● each channel coefficient estimate is weighted at the input of the pilot filter by a factor proportional to the estimated SNIR.

● each channel coefficient estimate is weighted at the input of the pilot filter by a factor of a monotonically increasing function of the estimated SNIR.

It is noted that the above-described embodiments and implementations are not a complete list, but illustrate various available methods that may be used to address the effects of interference. In particular, examples provide methods to leverage a priori knowledge of interference timing. In the case of SCH in UTRA FDD or WCDMA systems, the SCH time gap is known after the terminal successfully acquires the slot timing. Thus, improved channel estimation of the description is possible within such a system.

Fig. 10 illustrates an apparatus 500 comprising a receiver front end 502, the front end 502 comprising a digitizer coupled to a searcher 504, a channel coefficient estimator 506, and a demodulator and decoding unit 512. Searcher element 504 uses a priori knowledge of the transmitted signal component, such as the common pilot channel (CPICH), to provide significant delay tap information to channel coefficient estimator 506. This can be obtained, for example, by sliding correlation using a priori known symbols with the CPICH. Channel coefficient estimator 506 applies a sequence of channel coefficient estimates for a significant channel tap to pilot filter 510. The weighting factor generator 508 provides a sequence of weighting factors to the pilot filter 510 that is specific to each channel coefficient estimate. The weights should represent the quality of each channel coefficient estimate. The estimate of the CPICH pilot symbol SNIR can be used as a measure of the channel estimation quality. The pilot filter uses the weighting factors to generate a filtered version of the channel coefficient estimate. The pilot filter is further coupled to a demodulator and decoding unit 512 that performs coherent demodulation and decoding of the transmitted data. To achieve coherent demodulation, an estimate of the channel coefficients for all significant delay taps is required.

The described methods for canceling interference components and for mitigating the effects of interference components on channel estimation may also be combined and/or iteratively completed. For example: in a first step, the estimated interference component is subtracted from the digital representation of the received signal as described above. Then in a second step the channel coefficient estimation can be performed again, but this time based on the signal after subtraction of the estimated interference component. Some channel estimates may be better quality due to subtraction of the estimated interference components. This increased estimation quality may be taken into account when deriving weighting factors for the channel coefficient estimates filtered within the pilot filter as described above. A second iteration of interference cancellation can now be implemented. The output of the pilot filter may be used to derive a new and enhanced estimate of the interference component within the interference estimator. The new and enhanced estimate of the interference component may be subtracted from the original digital representation of the received signal. A second iteration of improved channel estimation may begin. And no further pilot weighting factors are added until further iterations. The received signal after the most recent channel coefficient estimation and the most recent interference cancellation is then used for coherent demodulation.

Power increase at a transmitter

In the 3GPP FDD downlink, SCH (synchronization channel) is transmitted in a non-orthogonal manner. The UE sees this to mean that other signals transmitted from the same cell/node B experience additional interference from the SCH. The interference is deterministic in nature, repeating every 10ms radio frame, degrading the received SNR at the terminal. The node B can mitigate this effect by increasing the transmit power of all channels for the duration of the SCH presence. The method can be generalized to any situation where the interference has a deterministic component known to the node B.

The SCH includes a PSC (primary scrambling code) repeated every 0.666.. ms and a set of SSCs (secondary scrambling codes) repeated every 10ms radio frame. Unlike other downlink channels, PSCs and SSCs are not scrambled with a Downlink Scrambling Code (DSC). Thus, at the UE, after despreading the incoming signal with the complex conjugate of the downlink scrambling code and decovering with the OVSF code (orthogonal variable spreading factor code), the SNR per symbol can be written as:

wherein

α ═ complex attenuation coefficient

Beta is a non-orthogonal factor

SF ═ spreading factor

E_ctEnergy per transmission channel chip

E_cschEnergy per SCH chip

I_ocThermal noise plus other cell interference power spectral density

In effect, the SCH (when present) degrades the SNR of the received symbol. These terminals close to the node B cannot receive symbols from the neighboring node B. Therefore, neighboring terminals generally cannot use diversity techniques. The non-orthogonal factor is a function of channelization code (OVSF code), DSC, SSC and time (modulo 10 ms). Therefore, the node B may unilaterally increase the transmit power of other channels, such as DPCH, when SCH is present. The transmit power increase is stored in a look-up table, which is pre-calculated using the parameters listed above.

In addition, the increase in power may be a function of the terminal geometry, which is a measure of the downlink C/I. If the terminal geometry is small, the increase in transmit power should be lower because the SCH has a smaller proportion of interference and negligible impact on received symbol SNR, and vice versa. The change in the non-orthogonality factor is a function of time and the channelization codes used in the downlink.

In an embodiment, the DPCCH includes dedicated pilot bits, uplink Transmit Power Control (TPC) bits, and Transport Format Combination Index (TFCI) bits. The dedicated pilot bits are used to calculate the downlink SNR. In the inner loop of DLPC (DL power control), the SNR is compared with a target SNR, which is set by the outer loop. If the calculated SNR is less than the target SNR, the UE sends a signal to the node B to increase the transmit power. The presence of the SCH at these bit positions degrades the SNR estimation. Therefore, the SNR computed at these locations is always low, resulting in the UE signaling to the node B to increase the transmit power.

To alleviate this problem, the following power control algorithm at the UE is proposed.

● calculate the SNR from the dedicated pilot bits.

● calculates the difference between the estimated SNR and the target SNR.

● if the difference is greater than zero (estimated SNR greater than target), signaling is sent to the node B to reduce the transmit power.

● if the difference is less than zero and less than the threshold, signaling is sent to the node B to increase the transmit power.

● if the difference is less than zero but greater than the threshold, signaling is sent to the node B to reduce the transmit power. Wherein the threshold may be a function of the UE geometry. In a simpler implementation, the threshold may be a constant.

The TPC bits are used to set the uplink transmit power. Any error in the estimated bit symbols results in a performance loss on the uplink. Typical TPC bit symbol estimation algorithms do not assume a deterministic component within the interference. If the SCH is present at the location of the TPC bits, then the threshold for determining the TPC bit symbol needs to be a function of the SCH, the channelization (OVSF) code, and the downlink scrambling code. The TFCI bits are used to calculate transport format combinations on a per slot basis. The TFCI bits are encoded. TCFI encoding is detailed within TS 25.211, 25.212, and 25.213.

Once the UE has obtained system timing, i.e., "knows" the SCH value and position in time, the UE knows that the beta value is a function of time for each channelization code. As the value of β increases, the symbol SNR deteriorates. The symbols are typically scaled by the common pilot strength before being combined with the multipath component symbols of other rake receivers. The UE may interpret the common pilot strength from each finger as a time-varying weighting applied to the symbol. Since the UE also knows that the value of β is a function of time and channelization code, there are several ways to mitigate the effects of additional interference from the SCH. Notably, the beta value is different for the same symbol than different multipath components from different node bs; the beta value is different for the same multipath component of the same node B for different symbols.

Interference: l2 example

In fact, service providers have observed reliability issues for the transmission of L2ACK/NACK messages sent on the downlink when operating at high geometries. For this case, the SCH transmission results in frequent loss of ACK/NACK messages at L2 (layer 2). Since ACK/NACK is used for acknowledgement transmission, loss interrupts a given communication and may result in a call being lost. Therefore, interference from the L2ACK is an unacceptable problem.

Based on the potential investigation and simulation resulting in the loss of the L2ACK message, the problem appears to be caused by multiple simultaneous conditions. When the conditions are simultaneously satisfied, the SCH channel has a significant impact on the transmission of the L2ACK/NACK message. The effect seems to prevent successful and reliable transmission of the message; the impact of missing L2 ACK/NACKs is described in the next section. Notably, inter-channel interference may affect other messages or the W-CDMA process.

The message may be particularly sensitive to SCH interference for a given set of cases. This was confirmed in the simulation results. There are a number of techniques and configurations that can mitigate the effects of SCH interference or other channel interference.

In UMTS, the L2 entity (radio link control, RLC) can be configured in three different ways:

● RLC transmission mode, most for voice services.

● RLC is not acknowledged mode, used for streaming services and some signaling messages.

● RLC acknowledges the mode, used for packet services and most signaling messages.

The inter-channel interference affects the acknowledgement pattern of L2 because L2ACK/NACK messages are used for this pattern. The SCH transmission introduces interference to the transmission of the ACK/NACK message within L2. The loss of these messages results in incorrect operation of the system. In one case, the L2ACK/NACK loss is limited to a specific configuration of measurement report messages, described in the measurement report message configuration (event 1B, in TS 25.331, part 14.1.2.2) for reporting that the primary CPICH is out of reporting range, i.e. that the radio link should be removed from the active set. The following reconstructs a similar case, i.e. the message may be lost:

1. the measurement report message is sent uplink and is not received correctly for the network. If all consecutive downlink ACK/NACK messages are lost, the RLC reset procedure is triggered.

As a result of the RLC reset, the contents of the retransmission buffer at the UE and network are refreshed, resulting in the loss of all messages that have not been successfully sent.

2. The measurement report message is in the RLC transmission buffer, waiting for the first time to be sent in the uplink, and the downlink ACK/NACK message for the previously sent message is lost. This triggers the RLC reset procedure during which the contents in the transmission buffer at the UE and the network are refreshed, resulting in the loss of all messages that have not yet been sent.

Once the measurement report for event 1B is lost, the network can no longer remove the radio link that triggered event 1B from the active set. The network is required to maintain communication to the UE over the radio link. As a result, the network will respond by increasing the radio link Tx power until synchronization is lost (possibly due to loss of uplink reception). This may cause a reduction in the capacity of the cell/sector from which the radio link is transmitted.

The loss of the L2ACK/NACK has a more general effect than the specific problem described above. For example, the Radio Resource Control (RRC) protocol relies heavily on the successful delivery of the L2 message. As an example, many RRC procedures are considered to terminate within a UE when a corresponding RRC message is delivered to lower layers of the transmission for transmission. These include:

RRC connection setup completion

RRC state

Signaling connection release indication

Counter check response

Radio bearer setup completion

Radio bearer reconfiguration completion

Radio bearer release completion

Transport channel reconfiguration complete

Physical channel reconfiguration complete

Radio carrier setup failure

Radio bearer reconfiguration failure

Radio bearer release failure

Transport format combination control failure

Physical channel reconfiguration failure

UTRAN mobility information failure

Active set update failure

Failed handover from UTRAN

Failed cell change command from UTRAN

Measurement reporting

If the L2ACK/NACK cannot be sent reliably, then the message may be lost, resulting in loss of synchronization between the UE and the network state machine. In many cases, the loss of synchronization will not be recoverable because most RRC messages are different messages, i.e. they convey only information about previous state changes, not a snapshot of the last state. As an example, the active set update message is used to add or remove radio links from the active set without including the current state of the active set. Also, the measurement control message only indicates a change in the neighbor list, not the last state of the neighbor list.

The RRC message may be lost whenever there is a reset of the L2 entity used by RRC (RB2, RB3, or RB 4). If the condition that caused the L2 entity to reset is not temporary, multiple L2 reset procedures may occur until L2 generates an unrecoverable error. L2 would then inform the RRC state machine and a message is sent by the UE to the network requesting the connection to be released. It is generally noted that this assumes that the maximum number of acceptable resets on the L2 entity used by RRC is one, even though higher values may be configured. This means that a single reset of the L2 entity used by RRC would generate unrecoverable errors.

Assume that L2 can be configured (RLC acknowledged mode) as the assumption of the lossless data transport layer of RRC messages is the backbone on which the entire RRC structure is based. In fact, according to the RRC protocol, the network should "initiate RRC connection release procedure" whenever the mobile station sends signaling "unrecoverable error within RB2, RB3 or RB 4". If the network is in accordance with the implementation currently specified in the RRC procedure, the call or packet session is lost whenever a problem occurs.

The problems caused by the L2ACK/NACK loss with the RRC protocol result in a loss of user data, since, in UMTS, the RLC acknowledged mode is also used to send user data for packet services. The consecutive loss of L2ACK/NACK may cause consecutive resets within the L2 entity being used, resulting in consecutive lost data. This loss of connection can be perceived by higher layers and eventually by the user, i.e. the data rate can be as low as 0 kb/s.

Even if only some L2 resets occur due to ACK/NACK message loss, the resulting data loss at L2 can cause many L3 retransmissions, resulting in a significant reduction in the data rate perceived by the application or user. In particular, the flow control mechanism (slow start) used by TCP/IP can significantly reduce the data rate whenever some data is lost by the power layer. The quality of service may then deteriorate and the application may terminate the packet session due to a timeout.

In W-CDMA, AM (acknowledged mode) L2 (layer 2) control data unit (PDU) has the structure illustrated in fig. 8A. PDU360 includes several fields. The 1-bit D/C field specifies whether the PDU carries control (value "0") or data information (value "1"). If the D/C field is set to "0", the following field is a 3-bit "PDU type" field. This field specifies which type of control information the PDU carries. For release 99, this field may have three possible values, 000, 001, and 010, indicating that the PDU is a status PDE, a reset PDU, or a reset acknowledgement PDU, respectively. Reset and reset acknowledge PDUs are only needed in very extreme cases, whereas status PDUs are necessary for the operation of the basic RLC protocol. The basic format of the status PDU is illustrated in fig. 8. In the figure, each SUFI (super field) carries RLC protocol status information.

The AM (acknowledged mode) RLC protocol is designed around a selective repetition scheme with explicit acknowledgement. In such a scheme, reliable transmission (negative or positive) of acknowledgements is critical in order to preserve synchronization of the receive and transmit windows and avoid delay out of synchronization. These acknowledgements are sent in SUFI as part of the status PDU. In its simplest form, the acknowledgement message will indicate a positive acknowledgement for all PDUs up to a certain number within the transmitter window. The SUFI that makes up the message consists of three fields:

1. a four bit field indicating a status super field (SUFI) type, which takes the value 0010 in case all PDUs up to a certain sequence number are acknowledged

2. The twelve bit field 370 indicates the sequence number reached by all PDUs that are positively acknowledged.

This may take different values, although always representing the value of the counter, which starts at a value of 0 when the RRC connection is established.

3. Padding field 372 pads the remainder of PDU 360. The padding value is not specified by the standard and will be discarded by the receiver of the status PDU. However, it is generally assumed that padding is set by default to all zeros by most L2 implementations.

It is noted that typically the PDU size is of the order of 150 bits. Thus, within a message such as that described above, an average of 7 bits is set to 1 and 143 bits are set to 0 (at least 130 of which are consecutive). It is worth noting that scrambling the data can mitigate the problem when using coding. However ciphering is not applied to the status PDU.

Scrambling code

The effect of inter-channel interference depends on the bit sequence being transmitted. Some sequences are more affected than others. W-CDMA specifies a way that the same bit sequence is sent on the physical layer during transmission and retransmission of the same data set, even when ciphering is configured. To minimize the impact of this problem, systems generally expect (1) to avoid transmitting some bit sequences more frequently than others (e.g., all 0 sequences), and (2) to change the pattern of transmissions on the physical layer during retransmissions of the same set of bits.

The following provides a set of solutions that attempt to obtain (1) and (2). The L2ACK reliability problem is particularly exacerbated by the long zero sequence introduced by padding. The status PDU may be "piggybacked" on a data PDU sent on the downlink. This leads to the introduction of additional (random) bits within the payload, reducing the chance of seeing a full 0-length sequence, avoiding repetition of a specific bit sequence on the physical layer. The main problem with this solution is that there is not always downlink data sent with the status PDU.

Explicit scrambling at the RLC layer can only be done with a specific scrambling sequence that is known to the network and the mobile unit (which is potentially negotiable during connection setup). This method can only replace one bit sequence with another and is therefore not recommended, although it is easier to implement. In the case of intra-MAC scrambling, the scrambling code is time-specific (based on, e.g., CFN). This removes data-dependent errors and ensures that the likelihood of successfully completing a certain number of retransmissions is the same for any data sequence. This is the best solution in the long term and (1) and (2) above can be implemented for ACK/NACK messages as well as for data PDUs.

The padding field in the message does not necessarily need to set any specific value for the peer entity, as the padding value may be ignored for the protocol. The padding values are thus specified to set the bits to some non-zero value (for the network side this does not necessarily require any standard changes).

There are also two possibilities to generate the padding. One is to use a special non-zero octet repeat padding. The second is to generate pseudo-random bits for padding. The latter is the best solution because the method enables the system to obtain the above given (1) and (2) at least for ACK/NACK messages. However, the problem still exists for certain higher layer data sequences.

From a standards and implementation point of view this solution is non-intrusive, not perfect even from a user data point of view, but it is a short term optimal solution.

Interference: power control

W-CDMA supports fast downlink power control. In principle, if the DCCH is subject to a particular interference, the power control mechanism should adapt to the environment and adjust the power control loop parameters to meet the target DCCH packet error rate. This assumes, however, that the power control loop is set up so as to take into account the actual DCCH performance.

According to the W-CDMA specification, the system may set up a power control procedure such that only the performance of a single transport channel is monitored (even when multiple transport channels are multiplexed together). The other transport channels are controlled by relative weighting within the rate matching/multiplexing process. Within 3GPP TS 34.108, all reference configurations may be configured with or without explicit possibility of power control of the DCCH.

When the power control process is driven solely by the data channel performance, any abnormal performance of the DCCH is not corrected by the power control loop process. In particular, if the DCCH experiences some interference that does not affect (or has less of a) the dominant channel (e.g., DTCH), then the power control outer loop does not increase the inner loop set point and a problem exists. In one case, the system does not run an explicit power control outer loop on the DCCH. This is the most severe case and can lead to a constant degradation of DCCH performance.

When the power control procedure takes into account the DCCH error rate, the system should be able to adapt to the conditions and reliably carry signaling messages over the air. However, the power required to overcome the SCH interference may become impractical or the power requirement may be greater than the upper power limit allowed for a particular RL within the node B. It is noted that this configuration should be considered as a related invention to solve the specific problems described above.

There are a number of situations that when they occur simultaneously can potentially result in significant impact on link performance and stability; the main source of the problem is interference associated with non-orthogonal SCH channels. The SCH is always present, however several factors can degrade the inter-channel interference. First, high geometry affects the impact of inter-channel interference. As described above, interference is isolated when the UE approaches the node B, thus resulting in information loss. Second when the outer-loop control mechanism is based on a transport channel that is not affected by the SCH. In this case, the power control is not adjusted to overcome the effect of the SCH. Third, diversity, including multipath, transmit diversity, amplifies this effect. In addition, when there is no diversity gain on the SCH interference, it is due to 100% correlation in addition to STTD. This is only a degradation factor if and when the power outer loop is based on channels that benefit from diversity. Fourth, when the low rate transport channel is multiplexed with the higher rate transport channel. Fifth, messages with bit length sequences of the same polarity result susceptible to inter-channel interference.

The L2ACK case includes all of the deteriorating factors. The L2ACK message is a zero-length sequence mapped on the DCCH within a higher rate packet configuration. The SCH covers the DCCH bits. The power control set point is driven by the DCH quality at high geometry.

Interference: AMR voice service

Inter-channel interference may create problems associated with adaptive multi-rate (AMR) speech services within W-CDMA, resulting in unacceptably high error rates on the DCCH, i.e., the radio signal bearer. It is noted that AMR voice services are provided as an example, however, inter-channel interference may have an effect on any of a variety of other services. High error rates can result in delays in signaling, including signaling associated with handover procedures. In certain cases, this can greatly increase the number of dropped calls. The SNR requirement of AMR DTCH depends on the used transport format. The silent frames require much less base station DPCH transmit power than the full rate frames. Longer periods of silence on the downlink may result in a significant reduction in the base station DPCH transmit power. The reduced transmit power is insufficient to reliably communicate on the DCCH signaling channel. Since the DCCH channel does not carry CRC on all transport formats and therefore cannot be power controlled. There is no way for the transmitter to know that the error rate on the signaling channel is unacceptably high. Therefore, the transmitter will not take the correct action.

Part of the solution is to always transmit at least one transport packet with zero bits (1x0 format) on the DCCH instead of no data packet (0x148 format). The 1x0 format includes CRC bits (as opposed to 0x148 which results in no transmission or DTX) which enables the DCCH to be power controlled. CRC errors on the DCCH may force the base station to increase transmit power and further retransmissions are more likely to be successful.

It is worth noting that this is a partial solution, since the signalling messages that are subsequently muted for a long time may be delayed by retransmissions. This also does not solve the problem of using messages in unacknowledged mode. A better solution to this problem is to use different DPDCH/DPCCH offsets for each transport format combination on the DPCH.

In a W-CDMA system, voice services are provided using adaptive multi-rate (AMR) voice services. The AMR source encoder generates either a full rate frame, SID frame or no data (NULL frame) every 20 milliseconds (i.e., TTI ═ 20 ms). SID frames are typically transmitted every 160 milliseconds during any silence period.

There are many modes for the AMR codec, but the most widely used mode is 12.20kbps with Unequal Error Protection (UEP). Each of the 244 bits per TTI is divided into 81 class a bits; 103 bits of class B and 60 bits of class C. Operating point is at most 10 for full rate class A bits^-4BER or 8.1X 10^-3BLER。

Along with the AMR channel, the DCCH of the radio signal bearer message is multiplexed on the same CCTrCH. The accuracy of rate matching is not defined within the standard. However, based on some published information, the industry accepted standard is to use a rate matching attribute, as indicated in the table of fig. 11, which indicates a typical coding of AMR/DCCH transport channels.

Based on simulations of closed loop power control for different propagation conditions, the DCCH BLER is from 2% to 8% when the full rate class a bits are operated at 0.7% BLER. This seems reasonable. There are problems when maintaining the same RM properties for SID and NULL frames. The SNR requirement is much lower for such class a frames because SID and NULL frames contain much fewer bits than 81 bits for full rate frames. The table of fig. 12 illustrates simulation results of SNR for various transport format requirements for class a and DCCH channels.

Notably, outer-loop power control cannot be run on the DCCH because there is a 0x148 bit transmission format on the DCCH, which has no CRC. So during silent periods, when the outer loop is driven only by class a frames containing most NULL frames, then a 1x148 frame on DCCH will be received at 3.3-0.5-2.8 dB for a BLER less than 1%. In the laboratory, simulations observed that the actual BLER of DCCH at this low SNR could be as high as 60%.

In addition, there is a possibility that a not well-selected offset of the SCH relative to the data channel may result in "collisions" which may increase the SNR requirement of the DCCH and thus exacerbate the problem.

Notably, the basic problem here is the different SNR requirements inherent to class a NULL frame DCCH signaling frames. This SNR requirement is a function of the transmission packet length, coding and propagation conditions. Since the UE has no control of these parameters, the better approach comes from the base station side. (note that one approach is that the UE always requests the necessary power for the weakest channel, e.g., DCCH 1x 148. so then a class a frame will always be received at a BLER much better than 1% BLER. however, this defeats the purpose of the overall power control and is not considered an acceptable solution.)

It is possible to adjust the rate matching properties to balance the SNR requirements for class a NULL frames and DCCH 1x148 frames. In this case, the system would expect to increase repetition on the DCCH, while increasing class A, B and puncturing on the C bits to maintain its relative level of protection.

This approach actually sacrifices transmit power because the SNR requirements of class A, B and C both rise for DCCH 1x148 frames due to truncation. This is not a good compromise as the duty cycle of the DCCH is lower compared to classes A, B and C and therefore power may be wasted most of the time.

Another approach is to have the UE run outer-loop power control on the DCCH by using a 1x0 transmission format instead of a 0x148 format on the DCCH. In the DCCH outer loop, once the UE detects a packet error on the DCCH, the UE requests more power and eventually there will be enough power to pass 1x 148.

Sending 1x0 instead of 0x148 means that there may be some transmit power overhead on the DCCH. In addition, in this scheme, the first frame within the 1x148 frame sequence on the DCCH may always experience a higher error rate than the successive frames before the outer loop target has time to "catch up". This may be acceptable if there are some acknowledgement/retransmission schemes on the DCCH.

Here, a scheme is proposed that can operate without transmitting any more power than necessary, and that does not require the outer loop to "catch up" on the delay. This is based on the idea that the base station applies a variable DPDCH-DPDCH power offset based on the TFC (transport format combination) of the instantaneous transmission.

Assume that the base station is provided with a table of SNR requirements for all transmission formats in table 2. Then the base station may calculate the total SNR requirement as the maximum of all the individual SNR requirements for all possible TFCs, as illustrated in fig. 13. For each frame, the base station may then automatically adjust the transmit power based on the TFC to be transmitted without waiting for the transmit power control instructions of the UE.

Conceptually, this splits the transmit power into two components, one of which may be adjusted by the base station via the TFC table (based on transport format, coding, etc.) and the other by the UE via inner loop power control (based on instantaneous channel conditions).

For example, assume there is a period of silence and the base station is sending NULL frames (1x0 on class a) and no signaling (0x148 on DCCH). This corresponds to a SNR of 0.5dB according to table 3. At a later time, the signaling message 1x148 is multiplexed with the NULL frame, whose corresponding requirement is 3.3dB according to table 3. The base station would automatically apply an additional power of 2.8dB higher than that used for the no signaling message case to compensate for the different SNR requirement.

In this scheme, if the UE inner loop detects a sudden change in received power without the UE sending any corresponding power control commands, the UE assumes that the channel conditions have changed and may attempt to reverse the power adjustment provided by the base station. This is because the UE must wait until the frame is received to know the TFC for the frame and thus know that the change in received power is due to the new transport format and not due to a change in channel conditions.

To address this problem, according to an embodiment, only the DPDCH transmit power is adjusted per TF, while the DPCCH power remains constant over the TF. In other words, the base station would transmit the DPCCH at a reference power level and would adjust the power level relative to the DPDCH depending on the transport format. At the same time, the DPCCH reference level is adjusted as usual according to the normal up/down commands determined by the inner loop power control.

At the node B, a table is stored to map the TF combinations to power offset values. An example of such a table is illustrated in fig. 14. Fig. 15 illustrates a process for power control using a table with a mapping of TF combinations to power offset values. Process 600 begins with receiving power control feedback from a UE at step 602. The power control feedback may be in the form of up/down commands. In an embodiment, the instructions are based on comparing a measured signal-to-interference ratio (SIR) to a target SIR. The node B transmits the transmit power of the DPCCH based on the power control feedback from the UE in step 604. The transmit power of the DPDCH is then calculated by applying a power offset to the power of the DPCCH, step 606. The channel is transmitted at step 608.

Notably, the method can be simply generalized to support different BLER targets for each transport format. The base station would simply consider each separate BLER target when deriving the transport format dependent power offset table.

The key requirement for implementing the method is the capability of the base station to set the power ratio of the DPDCH to the DPDCH based on the TFC. Moreover, the exact SNR requirements within table 2 may vary depending on implementation or propagation conditions. The more accurate the requirements are in terms of transmit power, the more efficient the system is. It is worth noting that in the case where all SNR requirements are set identically to 0dB, the scheme is simply reduced to the original scheme with no transmit power offset.

Those of skill in the art would understand that information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may be represented by voltages, circuits, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof.

Those of skill would further appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the present invention.

The various illustrative logical blocks, modules, and circuits described in connection with the embodiments disclosed herein may be implemented or performed with: a general purpose processor, a Digital Signal Processor (DSP) or other processor, an Application Specific Integrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof, to implement the functions described herein. A general purpose processor is preferably a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor may be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.

The steps of a method or algorithm described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such the processor can read information from, and write information to, the storage medium. In the alternative, the storage medium may be integral to the processor. The processor and the storage medium may reside in an application specific integrated circuit, ASIC. The ASIC may reside in a user terminal. In the alternative, the processor and the storage medium may reside as discrete components in a user terminal.

The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims (30)

1. A method in a wireless communication system supporting wideband code division multiple access (W-CDMA), comprising:

preparing a message for transmission;

determining a filling amount of the message; and

a non-uniform bit sequence for padding is generated.

2. The method of claim 1, wherein the non-uniform bit sequence is a predetermined sequence set by a central controller.

3. The method of claim 1, wherein the non-uniform bit sequence is a dynamic sequence that adjusts as a function of time.

4. The method of claim 1, wherein the non-uniform bit sequence is a pseudorandom sequence.

5. An apparatus in wireless communications supporting wideband code division multiple access (W-CDMA), comprising:

means for preparing the message for transmission;

means for determining a fill level of the message; and

means for generating a non-uniform bit sequence for padding.

6. User equipment in a wireless communication system supporting wideband code division multiple access (W-CDMA), comprising:

the receiver is used for receiving the transmission; and

a processor to:

determining a padding amount for the received transmission; and

a non-uniform bit sequence for padding is identified.

7. A node B in a wireless communication system supporting wideband code division multiple access (W-CDMA), comprising:

a transmitter for preparing and sending a transmission; and

a processor to:

preparing a message for transmission;

determining a filling amount of the message; and

a non-uniform bit sequence for padding is generated.

8. The node B of claim 7, wherein said padding is applied to a medium access control layer.

9. The node B of claim 7, wherein said padding is applied by a central controller to a radio link control layer.

10. The node B of claim 17, wherein the non-uniform bit sequence is a pseudo-random sequence.

11. A method in a wireless communication system, comprising:

determining a first timing of a non-orthogonal channel; and

a second timing of the second channel is adjusted based on the first timing.

12. The method of claim 21, wherein the wireless communication system is a wideband code division multiple access (W-CDMA) system.

13. The method of claim 22, wherein the non-orthogonal channel is a synchronization channel.

14. The method of claim 21, wherein the second channel is a physical channel comprising a plurality of transport channels, and wherein adjusting the second timing comprises changing a mapping of the plurality of transport channels to the second channel.

15. An apparatus in a wireless communication system, comprising:

means for determining a first timing of a non-orthogonal channel; and

means for adjusting a second timing of the second channel based on the first timing.

16. A method in a wireless communication system supporting transmission of a plurality of transport channels on a common physical channel, the method comprising:

preparing for transmission including a plurality of transport channels combined over a plurality of frames, wherein each of the plurality of transport channels is active for an associated percentage of frames;

determining at least one of a plurality of transport channels, the transport channel being active at a higher percentage than the other plurality of transport channels;

determining a transmission timing of a non-orthogonal channel, wherein the non-orthogonal channel is non-orthogonal with respect to the transmission;

the transmission of at least one of the plurality of transmission channels that conforms to the non-orthogonal channel timing is adjusted.

17. The method of claim 26, wherein the non-orthogonal channel is a synchronization channel.

18. The method of claim 27, wherein the common physical channel is a dedicated physical channel.

19. The method of claim 26, wherein adjusting transmissions comprises setting an offset for at least one of a plurality of transmission channels.

20. The method of claim 26, wherein adjusting transmissions comprises changing a mapping of a plurality of transport channels to physical channels.

21. A method in a wireless communication system, comprising:

identifying a first channel that causes inter-channel interference to other channels;

determining a transmission slot location on a first channel; and

the transmit power of at least one of the other channels is increased during the slot position.

22. The method of claim 31, wherein the wireless communication system is a wideband code division multiple access (W-CDMA) system.

23. The method of claim 32, wherein the first channel is a synchronization channel and a synchronization message is sent at a slot position.

24. The method of claim 31, wherein at least one of the other channels comprises a dedicated physical channel.

25. The method of claim 31, wherein at least one of the other channels comprises a control channel.

26. The method of claim 31, wherein increasing the transmit power comprises determining an increased power level from a look-up table.

27. The method of claim 31, wherein increasing the transmit power comprises determining an increased power level based on terminal geometry.

28. An apparatus in a wireless communication system, comprising:

means for identifying a first channel that causes inter-channel interference to other channels;

means for determining a transmission slot location on a first channel; and

means for increasing the transmit power of at least one of the other channels during the slot position.

29. The apparatus of claim 38, wherein the wireless communication system is a wideband code division multiple access (W-CDMA) system and the apparatus is a node B.

30. An apparatus in a wireless communication system, comprising:

a memory storage unit storing a power level adjustment value;

a processor to:

identifying a first channel that causes inter-channel interference to other channels;

determining a transmission slot location on a first channel; and

the transmit power of at least one of the other channels is increased during the slot position.