HK1108489B - Multiple transmitter and receiver well logging device with error calibration system - Google Patents

Description

Multiple transmitter and receiver logging device with error calibration system

This application is a partial continuation of my currently co-pending application serial No.10/237,439, filed on 9/2002.

Technical Field

The present invention relates to the field of well logging. More particularly, the present invention relates to novel apparatus and techniques for eliminating data acquisition errors inherent to electromagnetic propagating wave devices. The present invention also relates to an apparatus and method for measuring the resistivity of the geological formation surrounding a borehole during logging and logging while drilling operations.

Background

Formation resistivity is commonly used to evaluate the geological formations surrounding a borehole. Formation resistivity indicates the presence of hydrocarbons in the geological formation. Porous formations with high resistivity generally indicate that they are primarily saturated with hydrocarbons, while porous formations with low resistivity indicate that such formations are primarily saturated with water.

Devices have been developed in the past for measuring the resistivity of earth formations. Many of these devices measure formation resistivity by measuring properties of transmitted electromagnetic waves. For example, FIG. 1 shows an early uncompensated propagating wave resistivity tool that included one transmitter and two receivers to measure electromagnetic wave properties on two propagation paths. Property P11 represents the electromagnetic propagation property of the propagation path from the transmitter (Tx) to the first receiver (Rx1), while P12 represents the electromagnetic propagation property of the same propagation path as for P11 but from the transmitter to the second receiver (Rx 2). Typically, the measured propagation properties are attenuation and phase. The differential measurement (M) is formed by taking the difference between P12 and P11. This difference allows any errors associated with the transmitter unit of the system to be removed from the final measurement (M). The measurement (M) is then converted to formation resistivity (R) via a function (f) that provides a relationship between the differential propagation property (M) and the surrounding formation resistivity.

FIG. 2 shows another propagating wave resistivity tool described in U.S. Pat. No.4,949,045 to Clark et al (1990) and U.S. Pat. No.4,968,940 to Clark et al (1990). This provided tool improves measurement accuracy and reduces susceptibility to borehole irregularities when compared to the "uncompensated" tool shown in FIG. 1. Such tools include two transmitters and a receiver pair located between the two transmitters and are referred to as borehole compensated tools. M_URepresents a differential measurement of an electromagnetic wave transmitted upward from a transmitter (Tx1), and M_DRepresenting a differential measurement of the electromagnetic wave transmitted down from a transmitter (Tx 2). Borehole compensated measurement M_BHCCan be measured by pair up-transfer_UAnd transmitting the measurement M downward_DAnd averaging for calculation. Propagating the attribute (M) through a function (f)_BHC) Converted to resistivity and the formation resistivity is determined in a manner similar to an uncompensated tool. By averaging measurements from the upward and downward propagating electromagnetic waves, the effect of borehole buckling on the measured formation resistivity may be reduced. This average value also removes the error corresponding to the two receiver units Rx1 and Rx2 of the system. Like uncompensated equipment, which also measures M by using a differential receiver_UAnd M_DTo eliminate errors associated with the transmit unit of the system.

While borehole compensated tools provide more accurate formation resistivity measurements than conventional uncompensated tools, such techniques require tools that are approximately twice as long as uncompensated tools. The tool length of an uncompensated tool having a radial depth of investigation is directly related to the spacing between the transmitter and receiver pairs. Longer spacing between transmitter and receiver pairs provides greater depths of investigation than shorter spacing and therefore requires longer tool masters. The tool length of a borehole compensated tool having an equivalent radial probe depth to the uncompensated tool as described in the '045 and' 940 patents would be approximately twice as long because of the need for the upper and lower two transmitter units.

Another compensated tool is described in U.S. Pat. No.5,594,343 to Clark et al (1997) in which the transmitters are located asymmetrically on both sides of the receiver pair. Similar to the previously described '045 and' 940 patents, such tools also require at least one transmitter on each side of the receiver pair, and also require a long tool master.

Compensated tools as described above require long tool bodies in the borehole to properly position the transmitter and receiver. Long logging tools not only require additional material and more manufacturing costs, they are likely to pin or jam in narrow or deviated boreholes. This problem is particularly acute in multilateral wellbores having reduced entry radii and in highly deviated wellbores. Accordingly, there is a need for an improved system with reduced cost that can also facilitate movement of the tool within the borehole while collecting useful information about the formation characteristics, such as resistivity and other formation indicators.

Disclosure of Invention

The present invention provides a system for evaluating a property of a geological formation proximate a borehole through the formation. The system includes a tool body movable through the borehole, a first transmitter engaged with the tool body for generating a signal into the formation, a second transmitter engaged with the tool body proximate the first transmitter for generating a signal into the formation, a first receiver engaged with the tool body for receiving signals generated by the first and second transmitters, and a second receiver engaged with the tool body proximate the first receiver for receiving signals generated by the first and second transmitters.

Another embodiment of the present invention provides an apparatus comprising a tool body movable through a borehole, a first transmitter engaged with the tool body for generating electromagnetic waves into a geological formation, a second transmitter engaged with the tool body proximate the first transmitter for generating electromagnetic waves into the geological formation, a first receiver engaged with the tool body for receiving electromagnetic wave energy generated by the first and second transmitters and generating electrical signals representative of the electromagnetic wave energy, a second receiver engaged with the tool body proximate the first receiver for receiving electromagnetic wave energy generated by the first and second transmitters and generating electrical signals representative of the electromagnetic wave energy, and a controller for processing the electrical signals generated by the first and second receivers.

The method of the invention comprises the following steps: deploying a tool body in a borehole, generating electromagnetic wave energy from a first transmitter at a selected location in the borehole, generating electromagnetic wave energy from a second transmitter at a selected location in the borehole, operating first and second receivers in response to the electromagnetic wave energy generated by the first and second transmitters to generate electrical signals representative of the electromagnetic wave energy, and transmitting the electrical signals to a controller.

In a second preferred embodiment disclosed herein, an apparatus and method for evaluating geologic formation properties proximate a borehole traversing such a formation is disclosed. The method includes providing a device within the borehole, and the device includes a first transmitter disposed on the device for transmitting a signal into the geological formation; first and second receivers disposed on the device for receiving the transmitted signal; and processor means for processing the receiver signal.

The method includes generating a signal from a transmitter into the geological formation and receiving the transmitted signal at a first and second receiver. The method further includes injecting a calibration signal into the first receiver and the second receiver by means of the calibration circuit, and processing the received signal from the geological formation and the calibration signal within the processor means. The method includes correcting errors associated with first and second receiver units of the system and determining a resistivity measurement.

In a preferred embodiment, the step of processing the uncalibrated receiver signal and the calibrated signal comprises measuring the phase difference of the uncalibrated signals provided by the first receiver and the second receiver. Further, in a preferred embodiment, the step of processing the uncalibrated receiver signal and the calibration signal includes calculating a phase difference free of data acquisition errors by calculating the phase difference as follows:

PD＝(θ_M1-θ_M2)-(θ_MC1-θ_MC2)

-wherein PD is the phase difference without acquisition error;

-θ_M1is the measured phase of the uncalibrated signal from the first receiver;

-θ_M2is the measured phase of the uncalibrated signal from the second receiver;

-θ_MC1is the measured phase of the calibration signal from the first receiver;

-θ_MC2is the measured phase of the calibration signal from the second receiver.

In one embodiment, the calibration signal is at a first frequency and the receiver signal is at the first frequency, and the method further comprises time-multiplexing the uncalibrated receiver signal with the calibration signal. The step of time multiplexing may be achieved by sequentially activating the transmitters and then activating the calibration circuit.

Additionally, the second embodiment includes a frequency domain multiplexing scheme, and the method may further include separating the calibration signal from the received signal by providing a frequency difference value by means of a frequency domain multiplexing circuit operatively associated with the processor means.

In a preferred embodiment, the frequency difference is selected as follows:

ΔF＝N/t_a，

where Δ F is the frequency difference between the receiver and the calibration signal;

n is an integer;

t_ais the acquisition time interval.

The calibration signal may be injected into the front end of the receiver unit to add the calibration signal to the receiver unit in series in accordance with the teachings of the present invention.

In a preferred embodiment, the calibration signal is injected into the receiver front-end using a current loop, and wherein the current loop has a current transformer placed therein, and the method further comprises sampling the current in the loop by means of a current sampling resistor.

Furthermore, the device may be provided with a third receiver, and the method further comprises injecting the calibration signal into the third receiver. The method may further include calculating differential calibration values for the first receiver and the second receiver, calculating differential calibration values for the second receiver and the third receiver, and calculating differential calibration values for the first receiver and the third receiver.

An apparatus for obtaining resistivity measurements of a subsurface geological formation is also disclosed. The subsurface geological formation is traversed by a borehole. The apparatus includes a transmitter for transmitting a signal into the geological formation, first and second receivers for receiving the transmitted signal, and means for injecting a calibration signal into the first receiver and the second receiver. The apparatus may further include means for processing the uncalibrated receiver signal and the calibration signal to obtain a resistance measurement.

The processor means may comprise receiver data acquisition circuitry for correcting data acquisition errors associated with the first receiver and the second receiver.

In one embodiment, the signal injection means comprises means for applying a calibration signal to the first receiver and the second receiver in series. The apparatus may be provided with a third receiver and further comprising means for injecting the calibration signal into the third receiver. The apparatus may further comprise means for calculating a differential calibration value for the first receiver and the second receiver; means for calculating differential calibration values for the second receiver and the third receiver; and means for calculating differential calibration values for the first receiver and the third receiver.

In a preferred embodiment, the processing means further comprises means for measuring the phase difference between the first receiver and the second receiver without errors. The phase difference measuring apparatus calculates the phase difference as follows:

PD＝(θ_M1-θ_M2)-(θ_Mc1-θ_MC2)

-wherein PD is an error-free phase difference;

-θ_M1is the measured phase of the uncalibrated signal from the first receiver;

-θ_M2is the measured phase of the uncalibrated signal from the second receiver;

-θ_Mc1is the measured phase of the calibration signal from the first receiver unit;

-θ_MC2is the measured phase of the calibration signal from the second receiver unit.

In one embodiment, the calibration signal is at a first frequency and the receiver signal is at the first frequency, and the apparatus further comprises means operatively associated with the processing means for time-multiplexing the uncalibrated receiver signal with the calibration signal. The time division multiplexing means comprises means for sequentially activating the transmitters and the calibration signal injection circuit operatively associated with the processing means.

The apparatus may further comprise frequency domain multiplexing means for separating the calibration signal from the received formation signal by frequency difference. The frequency difference of the separating means is selected as follows:

ΔF＝N/t_a，

wherein Δ F is the frequency difference;

n is an integer;

t_ais the acquisition time interval.

In a preferred embodiment, the frequency domain multiplexing device cancels the received formation signal when processing the calibration signal and cancels the calibration signal when processing the received formation signal.

Additionally, the calibration signal may be injected into the first receiver and the second receiver front-ends using a current loop. The current loop may contain a current transformer placed therein and the apparatus further comprises means for sampling the current in the loop by means of a current sampling resistor.

Advantages of the present invention include that the apparatus and method provide a way to correct data acquisition errors in real time for propagating wave devices that use multiple receivers to measure propagation parameters such as attenuation and phase difference.

Another advantage of the present invention is that the disclosed calibration method is less complex than other methods that use depth calibration like propagation measurements to determine errors introduced by receiver data acquisition electronics.

Yet another advantage is a simpler calibration method that does not require borehole depth information. Another advantage is the elimination of errors associated with the time drift of the electronic parameters. Yet another advantage is calibrated dual receiver propagation measurements real-time downhole computation.

Drawings

FIG. 1 shows a conventional uncompensated propagating wave resistivity tool.

FIG. 2 shows one form of a conventional compensated propagating wave resistivity tool.

FIG. 3 shows a schematic diagram of a propagating wave resistivity system.

Fig. 4 shows a schematic of two depth positions resulting in compensated measurements.

FIG. 5 illustrates reducing the effects of borehole irregularities by means of a compensated tool.

FIG. 6 shows an asymmetric longitudinal response of an uncompensated tool.

FIG. 7 shows a symmetric longitudinal response of a compensated tool.

FIG. 8 shows a general arrangement of a depth offset compensated tool.

Figures 9 to 12 show different transmitter and receiver configurations for a depth offset compensated propagating wave resistivity tool.

FIG. 13 shows a block diagram for a depth offset compensated tool.

Fig. 14 shows a depth adjustment process for electronic error compensation.

Fig. 15 shows a block diagram of a propagating wave resistivity device incorporating an auto-calibration feature that provides receiver data acquisition errors.

Fig. 16 shows a block diagram of a propagating wave resistivity device incorporating a calibration current loop and a current transformer to inject a receiver calibration signal into the receiver front-end.

Detailed Description

The present invention provides a unique propagating wave resistivity system. The system can provide two depths of investigation as shown in figure 3. The tool 10 includes a tool body 12 carrying two transmitters identified as a transmitter pair 14 and two receivers identified as a receiver pair 16. The first transmitter Tx1 is disposed immediately adjacent to the second transmitter Tx 2. Disposed at selected distances from the transmitter pair 14 are a first receiver Rx1 and a second receiver Rx 2.

The property P11 represents the electromagnetic property of the propagation path from the first transmitter Tx1 to the first receiver Rx 1. The property P12 represents the electromagnetic property of the propagation path from the first transmitter Tx1 to the second receiver Rx 2. Similar attributes are represented for the second transmitter Tx2, where attribute P21 represents the propagation path from the second transmitter Tx2 to the first receiver Rx1, and P22 represents the propagation path from the second transmitter Tx2 to the second receiver Rx 2.

The tool 10 provides two differential measurements (M) from a receiver pair 16_RSAnd M_RL). Obtaining M from a receiver pair 16 using a short range transmitter TX2_RSAnd M is derived from the receiver pair 16 using a long-spaced transmitter TX1_RL. These two measurements can be converted to resistivity using the functions f and g.

Except for two dual receiver measurements (M)_RSAnd M_RL) In addition, two additional differential measurements (M) may be generated from the transmitter pair 14_TSAnd M_TL). Obtaining M from transmitter pair 14 using short-spaced receiver Tx2_TSM is obtained from transmitter pair 14 using a long-spaced transmitter Tx1_TL. If the spacing between transmitter pair 14 is equal to the spacing of receiver pair 16, then the dual transmitter will measure M_TSAnd M_TLThe function converted to resistivity may be the same function (f and g) as used for the dual receiver measurement from the receiver pair 16.

One advantage of this embodiment of the invention over the standard borehole compensation apparatus shown in figure 2 is that the invention provides two different spacings and two different depths of investigation. By using receiver pair 16 measurements and transmitter pair 14 measurements, borehole anomaly effects are reduced in this new embodiment. This new compensation method is implemented by depth shifting to align data from transmitter pair 14 with data from receiver pair 16 in terms of depth, as shown in fig. 4.

As shown in fig. 4, an average of the receiver pair 16 measurements may be generated with the transmitter pair 14 measurements to obtain two compensated resistivity measurements. This compensated measurement will have reduced borehole irregularity effects similar to conventional borehole compensation equipment. In this manner, the present invention achieves borehole compensation in an apparatus having a tool body that is significantly shorter than standard borehole compensation equipment. Furthermore, two measurements with different probing depths are provided instead of the single probing depth that the conventional borehole compensation apparatus shown in fig. 2 has. As used herein, the term "depth offset compensation" is used to describe a compensation method using depth adjusted receiver pair 16 and transmitter pair 14 data.

Figure 4 shows how the measurements from the receiver pair 16 and the transmitter pair 14 are averaged to produce a compensated resistivity. This process may be accomplished by averaging the dual transmitter and dual receiver propagation measurements and then converting the average measurement to resistivity, or alternatively, by averaging the resistivity calculated from the dual transmitter measurements with the resistivity calculated from the dual receiver measurements.

The reduction of the effect of borehole irregularities with this compensation is shown in fig. 5. As shown in the figure, the effects from a hole that enlarges the borehole diameter by two inches or borehole washout result in very little drift in the measured resistivity when comparing compensated measurements to uncompensated measurements.

In addition to the effects of borehole irregularities, the compensated arrangement shown in FIG. 4 also removes the longitudinal response asymmetry typically associated with uncompensated equipment. This asymmetry manifests itself in the formation interface when compared to a well log of a tool exiting the conductive layer, wherein the well log of measured resistivity as a function of depth from the uncompensated tool will have different characteristics than when the tool entered the conductive layer. This effect is shown in fig. 6.

This asymmetric longitudinal response result can be interpreted by examining the uncompensated measurements from the receiver pair 16. When the receiver pair 16 of the device enters the resistive formation interface from above, the corresponding transmitter has penetrated the formation. At this location, a larger portion of the propagating electromagnetic waves are contained in the resistive earth formation. The opposite occurs at the bottom bed boundary as the receiver pair 16 traverses the lower bed boundary because the corresponding transmitter is no longer embedded in the resistive bed. In this way, a smaller portion of the transmitted electromagnetic wave is contained in the resistive layer at this location, and the effect of this geometry results in the resistivity log having different shapes on the top and bottom of the formation interface. By using receiver pair 16 and transmitter pair 14 measurements, the effects of this longitudinal response asymmetry are averaged to provide a measurement responsive to the formation interface in a consistent symmetric manner regardless of the tool geometry as it traverses the formation interface. The symmetric response provided by this compensation scheme is shown in fig. 7.

Receiver and transmitter errors (eliminated with standard borehole compensated tools) still exist. The dual receiver measurements from receiver pair 16 contain receiver errors and the dual transmitter measurements from transmitter pair 14 contain transmitter errors, but the errors described above can be compensated by means of electronic features incorporated into the device design, as described later in this document.

One embodiment of a depth-offset compensated propagating wave resistivity tool 20 is shown in FIG. 8. Four parameters may describe the location and overall placement of the tool 20 antenna. The four parameters are the total number of transmitters (J), the total number of receivers (K), the upper transmitter TxJ and the lower receiver Rx1 (Z)_min) And between, and adjacentThe spacing between the transmitter and receiver antennas. Four different tool overall placement examples are shown in figures 9 through 12, in which different combinations of receivers and transmitters are shown.

As described above, the transmitter is disposed below the receiver. However, a configuration may be used in which the receiver is placed below the transmitter and will have the same response as a device in which the transmitter is placed below the receiver. The placement of the transmitter and receiver above or below the other depends on the desired implementation.

The previously described method of depth offset compensation can be extended to other allowable tool layouts by appropriately adjusting the equally spaced dual receiver and dual transmitter data in depth. The total number of different sounding depths provided by this method is equal to the total number of unique transmitter-to-receiver pair spacings (N)_TRR). A block diagram of a 3-transmitter 3-receiver version of the depth offset compensated propagating wave resistivity tool 30 is shown in fig. 13. The tool 30 includes a transmitter circuit 32 that drives one of three different transmitter antennas Tx1, Tx2, and Tx3 via an electronic switch 34. These transmitters are typically selected in turn under the control of an acquisition program programmed in an acquisition controller and processor 36. Three similar sets of receiver electronics 38 simultaneously acquire the data from three receive antennas Rx1, Rx2, and Rx 3. The local oscillator provides a frequency reference for the transmitter and receiver mixing circuit 40. After the mixer 40, the receiver signal is passed to a low pass filter 42 and then to a multi-channel analog-to-digital converter 44.

The acquisition controller and processor module 36 directs the sequence and timing of acquisition of the electronics and also acquires and processes measurement data. An interface is also provided by module 36 to accept commands from the user and to transmit data to the user. Such an interface may be connected to a telemetry system (not shown) to provide a means of acquiring and transmitting data in real time, such as in determining formation resistivity while drilling.

Although depth offset compensation reduces the effects of borehole irregularities and provides a symmetric longitudinal response, electronic errors associated with the transmitter and receiver preferably use additional compensation methods. The electronic compensation method includes measuring transmitter errors directly with electronic circuitry, calculating receiver errors through depth adjustment and comparing equivalent propagation measurements from different transmitter-receiver pairs. This process is illustrated in FIG. 14 for a three transmitter and three receiver tool. In this example, the electronic calibration requires only four values, the differential propagation measurement error of two close receiver pairs, and the differential propagation measurement error of two close transmitter pairs. Typically, these differential propagation measurements are attenuation and phase difference, but the calibration process can also be applied to other measurements. Referring to fig. 14, at depth position a, the transmitter propagation measurements using Tx1 and Tx2 of receiver Rx1 may be written as:

A₁₁＝E_Tx1+P11_A+E_RX1

A₂₁＝E_Tx2+P21_A+E_RX1

wherein E_Tx1And E_Tx2Is the error associated with the transmitters Tx1 and Tx2, respectively, and E_RX1Is connected to a receiver R_x1The associated error. The dual transmitter propagation measurement using Tx1 and Tx2 of Rx1 can be written as:

M_TM＝A21-A11＝(E_Tx2+P21_A+E_Rx1)-(E_Tx1+P11_A+E_Rx1)

or

M_TM＝(P21_A-P11_A)+(E_Tx2-E_Tx1). (1)

Numerical value (P21)_A-P11_A) Is the error-free differential propagation characteristic to be measured. The error associated with receiver Rx1 cancels, while the error (E) remains_Tx2-E_Tx1) Due to transmitters Tx1 and Tx 2. In this example, M_TMIs a medium-spaced differential propagation measurement. Can proceedShort spacing M_TSAnd a long spacing M_TLSimilar derivation of dual transmitter propagation measurements. M_TSAnd M_TLCan be written as:

M_TS＝A31-A21＝(P31_A-P21_A)+(E_Tx3-E_Tx2)(2)

and

M_TL＝A23-A13＝(P23_A-P13_A)+(E_Tx2-E_Tx1).(3)

as shown in equations 1, 2 and 3, at M_TS、M_TMAnd M_TLThe errors in are all differential transmitter errors.

These differential transmitter errors are measured directly in the tool by sampling the transmitter current 46 and transmitter voltage 48 and by deriving correction factors for the data acquired by each transmitter pair 14. The output from the transmitter sensing circuit 50 is processed in a manner similar to the receiver signal and passed to the analog-to-digital converter 44. The differential transmitter error is thus calculated by forming a difference of certain characteristics of the sampled transmitter signal. For example, the differential transmitter phase error may be calculated by the phase difference of the sampled signals, and the differential transmitter attenuation error may be calculated by the amplitude difference of the sampled transmitter signals. Since the transmitter sense outputs are processed by the same circuitry, any systematic errors associated with the acquisition circuitry are eliminated when calculating the differential corrections. After the differential transmitter error has been calculated from the sampled transmitter signals, the error can be calculated from M_TS、M_TMAnd M_TLTo eliminate errors associated with the transmitter units of the system.

In a similar manner, an expression for dual receiver propagation measurements can be derived. Referring to fig. 14, at depth position a, the receiver propagation measurements using Rx1 and Rx2 of receiver Tx2 may be written as:

A₂₂＝E_Tx2+P22_A+E_Rx2，

A₂₁＝E_Tx2+P21_A+E_Rx1，

dual receiver propagation measurement data M_RMThus can be written as:

M_RM＝A21-A22＝(E_Tx2+P21_A+E_Rx1)-(E_Tx2+P22_A+E_Rx2)

or

M_RM＝(P21A-P22A)+(E_RX1-E_RX2).(4)

Can perform short spacing M_RSAnd a long spacing M_RLSimilar derivation of dual receiver propagation measurements. M_RSAnd M_RLCan be written as:

M_RS＝(P31A-P32A)+(E_Rx1-E_RX2). (5)

and

M_RL＝(P12A-P13A)+(E_RX2-E_Rx3). (6)

at M, as shown in equations 4, 5 and 6_RS、M_RMAnd M_RLThe errors in are all differential receiver errors.

The differential receiver error can be determined by means of transmitter error measurements as described above and by means of a process comprising depth adjustment and comparison of equivalent propagation measurements from different transmitter-receiver pairs. Referring again to FIG. 14, the tool body has moved an amount Δ Z from depth position A to depth position B. As shown in fig. 14, many propagation paths at position a are equivalent to other propagation paths at position B. For example, P11_AEquivalent to P22_B，P12_AEquivalent to P23_BAnd so on. If there is no transmitter or receiver error in the system, for propagation path P11_AMeasurement A11 of will be equal to that for propagation path P22_BMeasurement B11. Tong (Chinese character of 'tong')Having introduced the transmitter error above, the measurement a11 can be written as:

A₁₁＝E_Tx1+P11_A+E_RX1

similarly, the expression of B22 can be written as:

B22＝E_Tx2+P22_B+E_RX2，

forming the difference between a11 and B22 results in:

A11-B22＝(E_Tx1+P11_A+E_RX1)-(E_TX2+P22_B+E_RX2)

it is simplified as follows:

A11-B22＝(E_RX1-E_RX2)+(E_TX1-E_TX2)+(P11_A-P22_B).

since the propagation paths are the same, the entry (P11)_A-P22_B) Equal to zero. This allows the differential receiver error to be expressed as:

(E_RX1-E_Rx2)＝(A11-B22)+(E_TX2-E_Tx1)

similarly, the remaining differential receiver errors can be expressed as:

(E_Rx2-E_Rx3)＝(A22-B33)+(E_Tx3-E_Tx2).

the present invention allows for the determination of all four required differential measurement errors, including two differential transmitter measurement errors and two differential receiver measurement errors. M can be measured from a suitable dual transmitter by means of a defined differential error_TS、M_TMAnd M_TLSubtract the differential transmitter error and measure M from the appropriate dual receiver_RS、M_RMAnd M_RLThe differential receiver error is subtracted to provide a propagation measurement free of errors associated with the transmitter and receiver units of the system.

Other differential errors may be identified, such as differential receiver error (E) derived from measurements A11 and B22_RX1-E_RX2). An alternative relationship using A21 and B32 can be used to obtain the following for (E)_RX1-E_RX2) Another expression of (a):

(E_Rx1-E_Rx2)＝(A21-B32)+(E_Tx3-E_Tx2).

it is also possible to deduce (E)_Rx2-E_Rx3) Similar alternatives, thereby reducing the noise in the differential errors by averaging all possible determinations of each differential error. Furthermore, since they do not vary directly as a function of depth, noise in the differential error can be further reduced by averaging the measurements over depth. This occurs because the primary mechanism that causes drift in the differential error is time, temperature, or pressure, rather than depth.

The controller is capable of generating compensated resistivity measurements of the geological formation by averaging uncompensated dual receiver resistivity measurements taken from the geological formation at two selected locations within the borehole with uncompensated dual transmitter resistivity measurement data. This averaging provides compensated resistivity measurements with symmetric longitudinal response and reduces the effects of borehole irregularities. The controller is also capable of generating compensated resistivity measurement data for the geological formation by averaging dual receiver propagation measurements (such as attenuation and phase difference) taken from the geological formation at two selected locations within the borehole with the dual transmitter propagation measurement data. This averaging results in compensated resistivity measurements with symmetric longitudinal response and reduces the effects of borehole irregularities.

The compensation of errors from the transmit and receive units of the system may be made by measuring the current and voltage produced by the first and second transmitters, by measuring the current and voltage of the electrical signals produced by the first and second receivers, and by operating the controller to derive a correction for transmitter propagation errors from the difference between the current and voltage measurements described above. Furthermore, the controller may be operable to derive the receiver propagation error from a correction in the transmitter propagation error and from the depth adjusted receiver propagation measurement.

Compared with the logging instrument in the prior art, the invention has great advantages. These advantages include shorter tool length, multiple depths of investigation with fewer antennas, compensation for electromagnetic wave tool asymmetric longitudinal response, compensation for borehole irregularities, and compensation for errors caused by the transmitter and receiver units of the apparatus.

Thus, compensation for data acquisition errors associated with dual transmitter measurements is eliminated using electronic circuitry that measures transmitter current and voltage. Data acquisition errors associated with dual receiver measurement data are eliminated by utilizing electronic transmitter error compensation and by deriving correction factors from data acquired by each receiver. This receiver error compensation process, as previously described, requires a depth adjustment technique similar to propagation measurement to determine the errors introduced by the receiver data acquisition electronics.

In a second embodiment, which is a preferred embodiment of the present application, an apparatus and method for correcting data acquisition errors in real time for a propagating wave device that measures propagation parameters, such as attenuation and phase difference, using multiple receivers will be described with reference to fig. 15 and 16. This second preferred embodiment does not require the use of depth-adjusted propagation measurements to remove errors in the dual-receiver measurements introduced by the receiver data acquisition system. This simplifies the correction of receiver unit errors and allows these corrections to be performed in real time as the processor drills down.

Referring now to fig. 15, a block diagram of a second preferred propagation wave resistivity device 60 will now be described, the propagation wave resistivity device 60 incorporating features that provide for automatic calibration of receiver data acquisition errors.

Fig. 15 summarizes the basic concept of the new receiver calibration scheme. As shown in fig. 15, the calibration signal is injected into the front end of each receiver (Rx1 and Rx2) via calibration signal injection circuits 62, 64. This calibration signal is applied in series to the receive antennas Rx1 and Rx2 and is thus added electronically to the receive signal. The received signal and the injected calibration signal are processed using receiver data acquisition electronics and amplifiers 66, 68, respectively. By measuring the receive signal and the calibration signal with the same acquisition electronics, errors in the acquisition electronics can be eliminated.

The operation of the resistivity device 60 is similar to the operation of the resistivity tool 30 discussed with respect to FIG. 13. The preferred embodiment of fig. 15 depicts the receiver pair Rx1, Rx2 and calibration signal injection circuits 62, 64. The transmitter electronics 70 generate a signal that causes the transmitter antenna Tx to generate an electromagnetic propagating wave. The transmitter electronics 70 are controlled by a transmitter and calibration signal source device 74, which transmitter and calibration signal source device 74 is in turn controlled by an acquisition controller and processor 76.

The receivers Rx1, Rx2 receive signals that have been transmitted from the transmitter Tx. The signal is then passed to amplifier and data acquisition electronics 66, 68, respectively. As previously described, the calibration signal injection circuitry 62, 64 has injected a calibration signal into the front end of the receivers Rx1, Rx 2. The data acquisition electronics 66, 68 will measure the receive signal and the calibration signal, which will then be passed to the acquisition controller and processor 76. The acquisition controller and processor 76 will calculate the phase difference and then the resistivity.

To illustrate the method for removing this data acquisition error, a measurement of the phase difference of the dual receivers will be given. Theta_Rx1And theta_Rx2Representing the true phase of the received signals from antennas Rx1 and Rx 2. Furthermore, phi_E1Represents the phase error introduced by the acquisition electronics of Rx1, and phi_E2Representing the phase error introduced by the acquisition electronics of Rx 2. The resulting measurement phase can thus be expressed as:

θ_M1＝θ_Rx1+φ_E1

θ_M2＝θ_Rx2+φ_E2

the measured value of the phase difference of the dual receivers is calculated by forming the following difference:

PD_UC＝θ_M1-θ_M2

PD_UC＝(θ_Rx1+φ_E1)-(θ_Rx2+φ_E2)

rewriting PD_UCWe get:

PD_UC＝(θ_Rx1-φ_RX2)+(θ_E1-φ_E2) (7)

the measured phase difference PD is shown in the above equation_UCContains an error term associated with the phase error introduced by the acquisition electronics of Rx1 and Rx 2. PD (photo diode)_UCIs an uncalibrated phase difference measurement.

Let theta_Ca1Representing the phase of the calibration signal, the measured phase of the injected calibration signal can be expressed as:

θ_MC1＝θ_Ca1+φ_E1

θ_MC2＝θ_Ca1+φ_E2

we can thus use the measured calibration phase to correct for errors φ introduced into the dual receiver phase difference measurements_E1And phi_E2。

PD＝PD_UCz-(θ_MC1-θ_MC2)

PD＝(θ_Rx1-φ_RX2)+(θ_E1-φ_E2)-(θ_MC1-θ_MC2)

PD＝(θ_Rx1-φ_RX2)+(θ_E1-φ_E2)-((θ_Ca1+φ_E1)-(θ_Ca1+φ_E2))

PD＝(θ_Rx1-φ_RX2) (8)

As shown in equation (8) above, the PD does not contain errors associated with the acquisition electronics. Furthermore, the above formula shows how to cancel the value θ_Ca1This indicates that the known theta is not required to remove the acquisition electronic error from the measured value of the phase difference_Ca1The value of (c).

Although the examples described above show how the apparatus and method can be applied to phase difference propagation measurements, the same techniques can be applied to other propagation measurements. For example, the same process may be used to calibrate dual receiver attenuation measurements. This method is the same as the phase difference calibration method, except that the signal phase is replaced by a corresponding signal amplitude level in decibels. The relationship for the attenuation example is as follows:

AT＝(A_M1-A_M2)-(A_MC1-A_MC2)，

AT＝(A_Rx1-A_Rx2)+(A_E1-A_E2)-(A_MC1-A_MC2)

AT＝(A_Rx1-A_Rx2)+(A_E1-A_E2)-((A_Ca1+A_E1)-(A_Ca1+A_E2))

AT＝(A_RX1-A_Rx2) (9)

-wherein AT is error-free attenuation in decibels;

-A_M1measured amplitude of the uncalibrated signal from the first receiver in decibels;

-A_M2measured amplitude of the uncalibrated signal from the second receiver in decibels;

-A_MC1the measured amplitude of the calibration signal from the first receiver unit in decibels;

-A_MC2from the second connection in decibelsMeasuring amplitude of a calibration signal of the receiver unit;

-A_Rx1is the true amplitude of the received signal in decibels at the first receiver;

-A_Rx2is the true amplitude of the received signal in decibels at the second receiver;

-A_Ca1true amplitude of the calibration signal in decibels;

-A_E1error in amplitude measurements introduced by the units of the first receiver in decibels;

-A_E2is the error in the amplitude measurement introduced by the unit of the second receiver in decibels.

There are at least two methods of injecting and measuring the calibration signal in accordance with the teachings of the present invention. The first is time multiplexing. If the frequency of the calibration signal is chosen to be exactly the same as the frequency of the received signal, the two signals will interfere with each other if the calibration signal is injected in the presence of the received signal. The acquisition controller may overcome signal interference by time multiplexing the received signal with the calibration signal. This is accomplished by having the acquisition controller sequentially activate the transmitter and then the calibration signal circuit. This provides a time multiplexed sequence of receive data and calibration data.

A second method that can be used is frequency domain multiplexing. The method separates the calibration signal from the received signal by a small frequency difference. As long as the frequency difference is chosen to be:

ΔF＝N/t_a， (10)

wherein N is an integer and t_aIs the acquisition time interval, the two signals can be processed independently. Applying Δ F by the above limitation ensures that the received signal can be completely cancelled when the calibration signal is processed and the calibration signal can be completely cancelled when the received signal is processed. Make Δ F small to ensureTo accurately measure the acquisition electronic error affecting the received signal by the calibration signal. For example, if the received signal is assumed to be at 2.00MHz and the acquisition time interval is 1.0 seconds, then af may be equal to 10 Hz. This places the calibration signal at 2.000010MHz and the receive signal at 2.000000 MHz. At this relatively small frequency separation, the electronic error measured by means of the calibration signal accurately reflects the error introduced into the received signal.

An important aspect of the means for this calibration is the differential accuracy of the injected calibration signal. That is, very little difference, or known and robust difference, is required in the calibration signal injected into each receiver. Any unaccounted for differences between the two calibration signals will result in errors in the final propagation measurement. For example, in the case of phase difference measurement, the phase of the calibration signal injected into Rx1 must be equal to the phase of the calibration signal injected into Rx 2. If there is a phase difference in the two calibration signals, these differences must be known and constant. Therefore, the implementation of the calibration device is important.

Referring now to FIG. 16, a preferred embodiment of an apparatus for minimizing differential calibration errors is shown. It should be noted that like numbers refer to like parts throughout the different figures. In this embodiment, the calibration signal is injected into the receiver front-end of Rx1, Rx2 using current loops 78, 80. A calibration current source device 86, electronically connected to the transmitter and the calibration signal source 74, generates a current I that is delivered to the current loops 78, 80. Current transformers 82, 84 are placed in the current loops 78, 80 and are disposed near the front end of the receivers, sampling the current in the loops of the respective receivers Rx1/Rx 2. Via low-value resistors 88, 90 on the secondary of the current transformer, the sampled current is converted into a small voltage, which is added in series with the received signal. Since the amplitude and phase of the current flowing in the current loop are substantially equal at any point in the loop, the calibration signal generated at each receiver will track and minimize any potential differential calibration signal errors. The current transformer also provides voltage isolation between the receivers and prevents any mutual coupling of the receiver signals.

The present disclosure has discussed the application of the present invention to devices having two receivers. However, the apparatus and method may be extended to any device having more than two receivers. For example, the present invention may be applied to a device having three receivers by injecting a calibration signal into a third receiver in a manner similar to the first two receivers (as shown in fig. 11). The method can thus calculate three differential calibration quantities: one for Rx1/Rx2 pairs, one for Rx2/Rx3 pairs and one for Rx1/Rx3 pairs. For a four receiver device, the method may calculate calibration values for six different pairs, Rx1/Rx2, Rx2/Rx3, Rx3/Rx4, Rx1/Rx3, Rx2/Rx4, and Rx1/Rx 4. In general terms, the invention can be used to calibrate K total differential receiver pairs, where K is defined as

And N equals the total number of receivers.

While the invention has been described in terms of certain preferred embodiments, it will be apparent to those of ordinary skill in the art that variations and modifications of the inventive concept herein may be made without departing from the scope of the invention. The embodiments shown herein are merely illustrative of the concept of the present invention and should not be construed as limiting the scope of the present invention.

Claims (35)

1. A method of obtaining electromagnetic propagation measurements of a subsurface geological formation, the formation being traversed by a borehole, the method comprising:

-providing a device within a borehole, the device comprising: a transmitter disposed on the device for transmitting a signal; a first receiver and a second receiver disposed on the device for receiving the transmitted signal; and processor means for processing the received signals;

-generating a signal from a transmitter;

-receiving the transmitted signal at a first receiver and a second receiver;

-injecting a calibration signal into the first receiver and the second receiver by means of the calibration circuit;

-processing the receiver signal and the calibration signal within the processor means;

-correcting data acquisition errors associated with the first receiver and the second receiver;

-determining an electromagnetic propagation measurement result,

wherein the step of processing the receiver signal and the calibration signal comprises calculating a phase difference free of data acquisition errors by calculating the phase difference as follows:

PD＝(θ_M1-θ_M2)-(θ_MC1-θ_MC2)

-wherein PD is the phase difference without data acquisition error;

-θ_M1is the measured phase of the uncalibrated receiver signal from the first receiver;

-θ_M2is the measured phase of the uncalibrated receiver signal from the second receiver;

-θ_MC1is the measured phase of the calibration signal from the first receiver; and

-θ_MC2is the measured phase of the calibration signal from the second receiver.

2. The method of claim 1, wherein the step of processing the receiver signal and the calibration signal comprises calculating an attenuation without data acquisition error by calculating the attenuation as follows:

AT＝(A_M1-A_M2)-(A_MC1-A_MC2)

-wherein AT is the attenuation in decibels without data acquisition error;

-A_M1measured amplitude of the uncalibrated receiver signal from the first receiver in decibels;

-A_M2measured amplitude of the uncalibrated receiver signal from the second receiver in decibels;

-A_MC1calibration signal from first receiver expressed in decibelsThe measured amplitude of (d);

-A_MC2the measured amplitude of the calibration signal from the second receiver is expressed in decibels.

3. The method of claim 1, wherein the calibration signal is at a first frequency and the receive signal is at the first frequency, and further comprising:

-time-multiplexing the receiver signal with the calibration signal.

4. The method of claim 3, wherein the step of time multiplexing comprises sequentially activating the transmitters and then activating the calibration circuit.

5. The method of claim 1, wherein the step of injecting the calibration signal comprises separating the calibration signal from the receiver signal by providing a frequency difference for frequency domain multiplexing circuitry operatively associated with the processor device.

6. The method of claim 5, wherein the frequency difference is selected as follows:

ΔF＝N/t_a

wherein Δ F is the frequency difference;

n is an integer;

t_ais the acquisition time interval.

7. The method of claim 1, wherein the step of injecting a calibration signal comprises:

-adding the calibration signal in series with the transmitted signal received by the first receiver and in series with the transmitted signal received by the second receiver.

8. The method of claim 1, wherein the calibration signal is injected into the receiver front-end using a current loop.

9. The method of claim 8, wherein the current loop has a current transformer placed therein, and the method further comprises sampling the current in the loop by means of a current sampling resistor on the current transformer.

10. The method of claim 1, wherein the device is provided with a third receiver, and the method further comprises injecting a calibration signal into the third receiver.

11. The method of claim 10, further comprising:

-calculating differential calibration values for the first receiver and the second receiver;

-calculating differential calibration values for the second receiver and the third receiver; and

-calculating differential calibration values for the first receiver and the third receiver.

12. An apparatus for obtaining electromagnetic propagation measurements of a subsurface geological formation traversed by a borehole, comprising:

-a transmitter arranged on the device for transmitting an electromagnetic signal;

-a first receiver and a second receiver arranged on the device for receiving the transmitted electromagnetic signals;

-means for measuring and correcting errors associated with the transmitter;

-means for injecting a calibration signal into the first receiver and the second receiver;

-means for processing the receiver signal and the calibration signal to obtain electromagnetic propagation measurements,

wherein the processing device comprises:

-a receiver data acquisition circuit for correcting data acquisition errors associated with the first receiver and the second receiver,

wherein the processing means further comprises means for calculating the attenuation without data acquisition error as follows:

AT＝(A_M1-A_M2)-(A_MC1-A_MC2)

-wherein AT is the attenuation in decibels without data acquisition error;

-A_M1measured amplitude of the uncalibrated receiver signal from the first receiver in decibels;

-A_M2measured amplitude of the uncalibrated receiver signal from the second receiver in decibels;

-A_MC1the measured amplitude of the calibration signal from the first receiver in decibels;

-A_MC2the measured amplitude of the calibration signal from the second receiver is expressed in decibels.

13. The apparatus of claim 12, wherein the signal injection means comprises means for applying the calibration signal in series with the transmitted signal received by the first receiver and in series with the transmitted signal received by the second receiver.

14. The apparatus of claim 13, wherein the apparatus is provided with a third receiver, and the apparatus further comprises means for injecting the calibration signal into the third receiver.

15. The apparatus of claim 14, further comprising:

-means for calculating differential calibration values for the first receiver and the second receiver;

-means for calculating differential calibration values for the second receiver and the third receiver; and

-means for calculating differential calibration values for the first receiver and the third receiver.

16. The apparatus of claim 12, wherein the processing means further comprises means for calculating an error-free phase difference as follows:

PD＝(θ_M1-θ_M2)-(θ_MC1-θ_MC2)

-wherein PD is an error-free phase difference;

-θ_M1is the measured phase of the uncalibrated receiver signal from the first receiver;

-θ_M2is the measured phase of the uncalibrated receiver signal from the second receiver;

-θ_MC1is the measured phase of the calibration signal from the first receiver;

-θ_MC2is the measured phase of the calibration signal from the second receiver.

17. The apparatus of claim 12, wherein the calibration signal is at a first frequency and the receiver signal is at the first frequency, and further comprising:

-means operatively associated with the processing means for time-multiplexing the uncalibrated receiver signal with the calibration signal.

18. The apparatus of claim 17, wherein the time division multiplexing means comprises means for sequentially activating the transmitter and the calibration injection means.

19. The apparatus of claim 12, further comprising frequency domain multiplexing means operatively associated with the processing means for separating the calibration signal from the receiver signal by frequency difference.

20. The apparatus of claim 19, wherein the frequency difference of the frequency domain multiplexing means is selected as follows:

ΔF＝N/t_a

wherein Δ F is the frequency difference;

n is an integer;

t_ais at the time of acquisitionAnd (4) spacing.

21. The apparatus of claim 20, wherein the frequency domain multiplexing means cancels the receiver signal when processing the calibration signal and cancels the calibration signal when processing the receiver signal.

22. The apparatus of claim 12, wherein the calibration signal is added in series with the transmitted signal received by the first receiver and in series with the transmitted signal received by the second receiver.

23. The apparatus of claim 22, wherein the calibration signal is injected into the first receiver and the second receiver front-ends using a current loop.

24. The apparatus of claim 23, wherein the current loop has a current transformer disposed therein, and the apparatus further comprises means for sampling the current in the loop by means of a current sampling resistor.

25. A method of obtaining electromagnetic propagation measurements of a subsurface geological formation traversed by a borehole, the method comprising:

-providing a device within a borehole, the device comprising: a transmitter disposed on the device for transmitting an electromagnetic signal; a first receiver and a second receiver disposed on the device for receiving the transmitted signal;

-transmitting a signal from a transmitter;

-receiving signals at a first receiver and a second receiver;

-measuring a current and a voltage associated with the transmitter, and operating the processor to derive a correction amount for the transmitter error from a difference between the current and voltage measurements;

-injecting a calibration signal into the first receiver and the second receiver;

-processing the receiver signal and the calibration signal in a processor for obtaining electromagnetic propagation measurements of the subsurface geological formation,

wherein the step of processing the receiver signal and the calibration signal comprises removing data acquisition errors associated with the first receiver and the second receiver,

wherein the step of processing the receiver signal and the calibration signal comprises obtaining a phase difference measurement, and wherein the phase difference measurement is calculated as follows:

PD＝((θ_M1-θ_M2)-(θ_MC1-θ_MC2)

-wherein PD is the phase difference without acquisition error;

-θ_M1is the measured phase of the receiver signal from the first receiver;

-θ_M2is the measured phase of the receiver signal from the second receiver;

-θ_MC1is the measured phase of the calibration signal from the first receiver;

-θ_MC2is the measured phase of the calibration signal from the second receiver.

26. The method of claim 25, wherein the step of processing the receiver signal and the calibration signal comprises obtaining an attenuation measurement, and wherein the attenuation measurement is calculated as follows:

AT＝(A_M1-A_M2)-(A_MC1-A_MC2)

-wherein AT is the attenuation in decibels without acquisition error;

-A_M1measured amplitude of the uncalibrated receiver signal from the first receiver in decibels;

-A_M2measured amplitude of the uncalibrated receiver signal from the second receiver in decibels;

-A_MC1the measured amplitude of the calibration signal from the first receiver in decibels;

-A_MC2the measured amplitude of the calibration signal from the second receiver is expressed in decibels.

27. The method of claim 25, wherein the calibration signal is at a first frequency and the receiver signal is at the first frequency, and further comprising:

-time-multiplexing the receiver signal with the calibration signal.

28. The method of claim 27, wherein the step of multiplexing is accomplished by sequentially activating the transmitters and then activating the calibration circuit for injecting the calibration signal.

29. The method of claim 25, further comprising separating the calibration signal from the receiver signal by frequency difference by means of a frequency domain multiplexing device operably associated with the processor.

30. The method of claim 29, wherein the frequency difference of the frequency domain multiplexing means is selected as follows:

ΔF＝N/t_a

wherein Δ F is the frequency difference;

n is an integer;

t_ais the acquisition time interval.

31. The method of claim 30, wherein the frequency domain multiplexing means cancels the receiver signal when processing the calibration signal and cancels the calibration signal when processing the receiver signal.

32. An apparatus for obtaining resistivity measurements of a subsurface geological formation traversed by a borehole, comprising:

-means for generating a transmission signal source and a calibration signal source;

-a transmitter arranged on the device for transmitting a source of electromagnetic signals;

-a first receiver and a second receiver arranged on the device for receiving the transmitted signal;

-a first calibration signal injection circuit for injecting a calibration signal into the first receiver;

-a second calibration signal injection circuit for injecting a calibration signal into the second receiver;

-first data acquisition electronics for digitizing the uncalibrated and calibrated signals from the first receiver;

-second data acquisition electronics for digitizing the uncalibrated and calibrated signal from the second receiver;

an acquisition processor for receiving digitized data from the first data acquisition electronics and the second data acquisition electronics and obtaining resistivity measurements,

wherein the step of processing the receiver signal and the calibration signal comprises calculating a phase difference free of data acquisition errors by calculating the phase difference as follows:

PD＝(θ_M1-θ_M2)-(θ_MC1-θ_MC2)

-wherein PD is the phase difference without data acquisition error;

-θ_M1is the measured phase of the uncalibrated receiver signal from the first receiver;

-θ_M2is the measured phase of the uncalibrated receiver signal from the second receiver;

-θ_MC1is the measured phase of the calibration signal from the first receiver; and

-θ_MC2is the measured phase of the calibration signal from the second receiver.

33. The apparatus of claim 32, wherein the injected calibration signal is added in series with the transmitted signal received by the first receiver and in series with the transmitted signal received by the second receiver.

34. The apparatus of claim 32, wherein the injected calibration signal is injected into the first receiver and the second receiver front ends using a current loop.

35. The apparatus of claim 34, wherein the current loop has a current transformer disposed therein, and the apparatus further comprises means for sampling the current in the loop by means of a current sampling resistor.