HK1186308A - Dual-polarity multi-output synchronous boost converter, operating method therefor, and voltage regulators - Google Patents

Description

Bipolar multi-output synchronous boost converter, operation method thereof and voltage regulator

The present application is a divisional application of an invention patent application having an application date of 2008/8/5, an application number of 200880102314.5 (international application number of PCT/US 2008/072257), and an invention name of "bipolar multi-output DC/DC converter and voltage regulator".

Technical Field

The invention relates to a bipolar multi-output synchronous boost converter, a method of operating the same and a voltage regulator.

Background

Voltage regulation is often required in order to prevent variations in the supply voltage that powers various microelectronic components, such as digital ICs, semiconductor memories, display modules, hard disk drives, RF circuits, microprocessors, digital signal processors and analog ICs, especially in battery powered applications like cellular phones, notebook computers and consumer products.

Such regulators are referred to as DC-DC converters because the battery or DC input voltage of the product often has to be boosted to a higher DC voltage or reduced to a lower DC voltage. When the voltage of the battery is greater than the desired load voltage, a step-down converter is used. The buck converter may include an inductive switching regulator, a capacitive charge pump, and a linear regulator. Conversely, when the voltage of the battery is lower than the voltage required to power the load, a step-up converter is required, which is commonly referred to as a boost converter. The boost converter may comprise an inductive switching regulator or a capacitive charge pump.

In the aforementioned voltage regulator, the inductive switching converter can achieve excellent performance in the widest range of currents, input voltages and output voltages. The underlying principle of a DC/DC inductive switched-mode converter is based on the following simple premise: the current in the inductor (coil or transformer) does not change immediately and the inductor will develop an opposite voltage to resist any change in its current.

The basic principle of an inductor-based DC/DC switched-mode converter is: the DC power supply is switched or "ramped" into pulses or bursts and those bursts are filtered out using a low pass filter comprising an inductor and a capacitor to produce a well behaved time varying voltage, i.e. changing DC to AC. Inductors can be used as inputs to step-up or step-down converters by switching the inductor repeatedly at high frequency with one or more transistors to produce a different output voltage than its input. After the AC voltage is changed up or down with magnetization, the output is then rectified back to DC and filtered to remove any ripple.

Transistors are typically implemented with MOSFETs having a low on-state impedance (commonly referred to as "power MOSFETs"). With feedback control of the switching conditions from the output voltage of the converter, a constant well-regulated output voltage can be maintained, regardless of whether the input voltage of the converter or its output current changes rapidly.

In order to remove any AC noise or ripple generated by the switching action of the transistors, output capacitors are provided across the output of the switching regulator circuit. The output capacitor together with the inductor form a "low pass" filter that can remove most of the switching noise of the transistors to prevent them from reaching the load. The switching frequency (typically 1MHz or greater) must be "high" relative to the resonant frequency of the "LC" tank (tank) of the filter. Averaging over multiple switching cycles, the switched inductor behaves like a programmable current source with a low changing average current.

Because the average inductor current is controlled by the transistor being biased as an "on" or "off" switch, the power dissipation in the transistor is theoretically small and high converter efficiencies in the range of eighty percent to ninety percent can be achieved. In particular, when a power MOSFET is biased as an on-state switch with a "high" gate bias, it exhibits a low R_DS(on)Linear I-V drain characteristics of impedance (typically 200 milliohms or less). For example, at 0.5A, such a device will exhibit a maximum voltage drop I of only 100mV_D·R_DS(on)Regardless of its drain current. Power consumption during its on-state conduction time is I_D ²·R_DS(on). In this example, the power consumption during the conduction of the transistor is given as (0.5A)²·(0.2Ω)=50mW。

In its off stateThe power MOSFET has its gate biased to its source, i.e. so that V_GSAnd = 0. Even if the drain voltage V is applied_DSEqual to the battery input voltage V of the converter_battDrain current I of power MOSFET_DSSAnd is also very small, typically under 1 microamp and more typically a few nanoamps. Current I_DSSMainly including junction leakage (junction leakage).

Therefore, a power MOSFET used as a switch in a DC/DC converter is efficient, since in its off-condition it exhibits a low current at a high voltage, and in its on-state it exhibits a high current at a low voltage drop. In addition to switching transients, I in power MOSFETs_D·V_DSThe product remains small and the power consumption in the switch remains low.

Power MOSFETs are not only used to convert AC to DC by ramping the input power supply, but may also be used to replace the rectifying diodes needed to rectify the resultant AC back to DC. The operation of a MOSFET as a rectifier is often achieved by: the MOSFET is arranged in parallel with the schottky diode and is turned on when the diode is conducting, i.e. in synchronism with the conduction of the diode. In such applications, the MOSFET is therefore referred to as a synchronous rectifier (synchronous rectifier).

Because the synchronous rectifier MOSFET can be sized to have a low on-resistance and lower voltage drop than Schottky (Schottky), the conduction current is transferred from the diode to the MOSFET channel, so that the overall power consumption in the "rectifier" is reduced. Most power MOSFETs include parasitic source-drain diodes. In a switching regulator, the inherent P-N diode orientation must be the same as the polarity of the schottky diode, i.e., cathode to cathode, anode to anode. Since this parallel combination of silicon P-N diode and schottky diode only carries current for a brief interval of time called "break before make" before the synchronous rectifier MOSFET turns on, the average power consumption in these diodes is low and schottky is usually completely removed.

Assuming that the transistor switching event is relatively fast compared to the oscillation period, the power loss during switching can be negligible or alternatively considered a fixed power loss in the circuit analysis. In general, power losses in low voltage switching regulators can then be estimated by taking into account conduction and gate drive losses. However, switching waveform analysis becomes important when there are multiple megahertz switching frequencies, and must be considered by analyzing the device's drain voltage, drain current, and gate bias voltage drive versus time.

Based on the above principles, today's inductor-based DC/DC switching regulators are implemented with a wide range of circuits, inductors and converter technologies. Broadly, they are divided into two main technology types, non-isolated converters and isolated converters.

The most common isolated converters include flyback and forward converters and require either a transformer or a coupled inductor. When the power is high, a full bridge converter is also used. Isolated converters are able to step up or step down their input voltage by adjusting the ratio of the primary and secondary windings of the transformer. A transformer with multiple windings can produce multiple outputs simultaneously, including a higher voltage than the input and a lower voltage than the input. The transformer has the following disadvantages: they are large and suffer from undesirable stray inductance compared to single winding inductors.

The non-isolated power supply includes a Buck-type Buck converter, a boost-type boost converter, and a Buck boost converter. Buck converters and boost converters are particularly efficient and compact in size, especially operable in the megahertz frequency range where inductors of 2.2 muH or less may be used. Such techniques produce a single regulated output voltage for each coil and require a dedicated control loop and separate PWM controllers for the respective outputs in order to constantly adjust the switch on-time to regulate the voltage.

In portable and battery-powered applications, synchronous rectification is often employed to improve efficiency. A Buck converter employing synchronous rectification is referred to as a synchronous Buck converter. Boost type boost (boost) converters employing synchronous rectification are known as synchronous boost converters.

Synchronous boost converter operation: as shown in fig. 1, a prior art synchronous boost converter 1 includes a low side power MOSFET switch 2, a battery connected to an inductor 3 of the battery, an output capacitor 6, and a "floating" synchronous rectifier MOSFET4 in parallel with a rectifier diode 5. Driven by break-before-make circuitry (not shown) and responsive to voltage feedback V from the output of the converter by a PWM controller 7_FBThe gate of the control MOSFET is connected across the filter capacitor 6. BBM operation is required to prevent shorting the output capacitor 6.

The source and drain terminals at the synchronous rectifier MOSFET5 (which may be N-channel or P-channel) are not permanently connected to either supply rail (i.e., not connected to ground nor to V)_batt) In the sense that it is considered to be floating. Diode 5 is a P-N diode inherent to synchronous rectifier MOSFET4 regardless of whether the synchronous rectifier is a P-channel device or an N-channel device. A schottky diode may be included in parallel with MOSFET4 but the series inductance may not operate fast enough to divert current from the forward biased intrinsic diode 5. Diode 8 comprises a P-N junction diode inherent to N-channel low side MOSFET2 and remains reverse biased under normal boost converter operation. Diode 8 is shown as a dashed line because it does not conduct under normal boost operation.

If the duty cycle D of the converter is defined as the time at which energy flows from the battery or power source to the DC/DC converter, i.e. during which time the low-side MOSFET switch 2 is conducting and the inductor 3 is being magnetized, the output-to-input voltage ratio of the boost converter is proportional to 1 minus the inverse of its duty cycle, i.e.,

although this equation describes a wide range of conversion ratios, boost converters do not successfully achieve a single transfer characteristic without extremely fast device and circuit response times. For high duty cycles and switching ratios, the inductor conducts large current spikes and thus reduces efficiency. In view of these factors, the boost converter duty cycle is practically limited to the range of 5% to 75%.

The need for bipolar regulated voltages: today's electronic devices require a large number of regulated voltages to operate, some of which may be negative with respect to ground. Some smart phones may utilize more than twenty-five individually regulated power supplies in a single handheld host (hangdheld), including some organic light emitting diodes or LEDs, negative bias power supplies required for displays. Space limitations prevent the use of so many switching regulators each having a separate inductor.

Unfortunately, a multi-output non-isolated converter capable of generating both negative and positive supply voltages requires a multi-winding or tapped inductor. Although smaller than isolated converters and transformers, tapped inductors are also substantially larger and taller in height than single-winding inductors, and suffer from increased parasitics and radiated noise. As a result, multi-winding inductors are not typically employed in any space sensitive or portable devices and portable consumer electronics such as headsets.

As a compromise, today's portable devices employ only a small number of switching regulators and a large number of linear regulators to generate the necessary number of independent supply voltages. While low dropout linear regulators (or LDOs) are generally less efficient than switching regulators, they are smaller and less costly because no coil is needed. As a result, efficiency and battery life are sacrificed for lower cost and smaller size. Negative supply voltages require dedicated switching regulators that cannot be shared with positive voltage regulators.

There is a need for a switching regulator implementation that is capable of producing both positive and negative outputs (bipolar outputs) from a single-winding inductor to reduce cost and size.

Disclosure of Invention

The present disclosure describes an inventive boost converter that is capable of producing two independently regulated outputs of opposite polarity from a single-winding inductor, namely an output above a positive ground potential and an output below a negative ground potential. A representative embodiment of a dual output, bipolar inductive boost converter includes an inductor; a high side gate coupled between a voltage source and a first terminal of the inductor; a low side gate coupled between ground and a second terminal of the inductor; a first output node coupled to a first output gate coupled to a second terminal of the inductor; a second output node coupled to a second output gate coupled to a first terminal of the inductor; and a controller configured to activate or deactivate the high side gate, the low side gate, the first output gate and the second output gate, the controller further configured to provide a first mode, a second mode and a third mode of circuit operation.

The first mode of operation charges the inductor to a voltage equal to the input voltage. The second mode of operation transfers charge to the first output node and the second output node simultaneously. Once the first output node reaches the target voltage, the second mode ends. The third mode of operation continues to charge the second output node until it reaches its target voltage. In this manner, the boost converter provides two regulated outputs from a single inductor.

For the second embodiment, the same basic components are used. However, in this case, the switching network provides the following modes: 1) a first mode in which a positive electrode of the inductor is connected to an input voltage and a negative electrode of the inductor is connected to ground; 2) a second mode in which a positive electrode of the inductor is connected to the input voltage and a negative electrode of the inductor is connected to the second output node; and 3) a third mode in which a positive electrode of the inductor is connected to the first output node and a negative electrode of the inductor is connected to ground.

The first mode of operation charges the inductor to a voltage equal to the input voltage. The second mode of operation transfers charge to the first output node and ends when the first output node reaches the target voltage. The third mode of operation charges the second output node and ends when the second output node reaches its target voltage. In this manner, the boost converter provides two regulated outputs from a single inductor.

According to one embodiment of the present invention, a method is provided for operating a bipolar dual output synchronous boost converter including an inductor, a high-side gate coupled between a voltage source and a first terminal of the inductor, a low-side gate coupled between ground and a second terminal of the inductor, a first output node coupled to a first output gate, a second output node coupled to a second output gate, and a controller, the first output gate coupled to the second terminal of the inductor, the second output gate coupled to the first terminal of the inductor, the controller configured to activate or deactivate the high-side gate, the low-side gate, the first output gate, and the second output gate. The method comprises the following steps: configuring the controller to cause the boost converter to operate in a first mode in which the controller activates the high-side gate to connect a first terminal of the inductor to the voltage source to receive an input voltage and the controller activates the low-side gate to connect a second terminal of the inductor to ground; configuring the controller to cause the boost converter to operate in a second mode in which the controller deactivates the high-side gate and the low-side gate, the controller also activating the first output gate and the second output gate to connect a first terminal of the inductor to the first output node and a second terminal of the inductor to the second output node; and configuring the controller to cause the boost converter to operate in a third mode in which the controller activates the high side gate and deactivates the second output gate to connect the first terminal of the inductor to the input voltage, and the controller deactivates the low side gate and activates the first output gate to connect the second terminal of the inductor to the second output node.

According to yet another embodiment of the present invention, a method is provided for operating a bipolar dual output synchronous boost converter including an inductor, a high-side gate coupled between a voltage source and a first terminal of the inductor, a low-side gate coupled between ground and a second terminal of the inductor, a first output node coupled to a first output gate, a second output node coupled to a second output gate, and a controller, the first output gate coupled to the second terminal of the inductor, the second output gate coupled to the first terminal of the inductor, the controller configured to activate or deactivate the high-side gate, the low-side gate, the first output gate, and the second output gate. The method comprises the following steps: configuring the controller to cause the boost converter to operate in a first mode in which the controller activates the high-side gate to connect a first terminal of the inductor to the voltage source to receive an input voltage and the controller activates the low-side gate to connect a second terminal of the inductor to ground; configuring the controller to cause the boost converter to operate in a second mode in which the controller activates the high side gate and deactivates the first output gate to connect a first terminal of the inductor to the voltage source to receive an input voltage, and the controller activates the second output gate and deactivates the low side gate to connect a second terminal of the inductor to the second output node; and configuring the controller to cause the boost converter to operate in a third mode in which the controller activates the first output gate and deactivates the high side gate to connect the first terminal of the inductor to the first output node, and the controller activates the low side gate and deactivates the second output gate to connect the second terminal of the inductor to ground.

Drawings

Fig. 1 is a schematic diagram of a prior art single output synchronous boost converter.

Fig. 2 is a schematic diagram of a bipolar dual output synchronous boost converter provided by the present invention.

Fig. 3A to 3C illustrate an operational sequence performed by the boost converter of fig. 2 to implement a mode called synchronous transfer. The synchronous transfer mode comprises the following successive operating phases: the inductor is magnetized (3A) and the charge is transferred synchronously to + V_OUT1and-V_OUT2(3B) Charge is passed on exclusively to + V_OUT1（3C）。

Fig. 4 is a plot of switching waveform characteristics for the boost converter of fig. 2 operating in synchronous transfer mode.

FIG. 5 shows the boost converter of FIG. 2 exclusively towards V_OUT2An alternate operational phase of transferring charge.

Fig. 6 is a flow chart of the boost converter of fig. 2 utilizing a synchronous transfer mode.

Fig. 7A to 7C show the boost converter of fig. 2 performing an operational sequence implementing a mode known as time-division multiplexed transfer. The time-division multiplex transmission mode comprises the following successive operating phases: the inductor is magnetized (7A) and the charge is transferred exclusively to + V_OUT1(7B) Charge is transferred exclusively to + V_OUT2（7C）。

Fig. 8 is a flowchart illustrating an operation sequence in which the boost converter of fig. 2 operates in a time division multiplexing transfer mode.

Fig. 9 is a block diagram illustrating the boost converter of fig. 2 instead utilizing digital control with multiplexed feedback.

Detailed Description

As previously described, conventional non-isolated switching regulators require a single winding inductor and corresponding dedicated PWM controller for each regulated output voltage and polarity. In contrast, the present disclosure describes an inventive boost converter that is capable of producing two independently regulated outputs of opposite polarity, i.e., a positive above ground output and a negative below ground output, from a single-winding inductor.

As shown in FIG. 2, a dual output bipolar inductive boost converter 10 includes a low side N-channel MOSFET11, an inductor 12, a high side P-channel MOSFET13, a floating positive output synchronous rectifier 14 with an intrinsic source-drain diode 16, a floating negative output synchronous rectifier 15 with an intrinsic source-drain diode 17, a pair output + V_OUT1and-V_OUT2Filtered output filter capacitors 18 and 19. The regulator operation is controlled by a PWM controller 20, the PWM controller 20 including break-before-make gate buffers (not shown) that control the on-time of the MOSFETs 11, 13, 14 and 15. The PWM controller 20 may operate at a fixed frequency or a variable frequency.

Closed loop regulation by self-adaptationIs fed back to_FB1And V_FB2V of_OUT1and-V_OUT2Feedback of the output. The feedback voltage may be scaled by a resistive divider (not shown) or other level shift circuit, as desired. The low side MOSFET11 includes an intrinsic P-N diode 21 shown in dashed lines, which remains reverse biased and non-conductive under normal operation. Similarly, the high-side MOSFET13 includes an intrinsic P-N diode 22, shown in dashed lines, that remains reverse biased and non-conductive under normal operation. The gate drive circuit may be suitably adapted to implement the high-side MOSFET13 with either a P-channel MOSFET or an N-channel MOSFET.

Unlike conventional boost converters, in bipolar boost converter 10, magnetizing the inductor requires turning on both high-side MOSFET13 and low-side MOSFET 11. Inductor 12 is therefore not hardwired to Y_battOr ground. As a result, node V_xAnd V_yThe terminal voltage of the inductor is not permanently fixed or limited to any given voltage potential unless intrinsic P-N diodes 21 and 22 are forward biased and the avalanche breakdown voltage of the device is employed.

Specifically, there is no forward bias at the P-N diode 22 and it is clamped to a voltage (V)_batt+V_f) In case of (2), node V_yMust not exceed battery input V_battA forward biased diode voltage drop V_f. In the disclosed converter 10, the inductor 12 cannot couple V_yNode voltage driven to V_battAbove, only switching noise can cause the diode 22 to become forward biased.

However, within a specified operating voltage range of the relevant device, V_yMay be less than V_battOperating at positive voltage, even at a voltage below ground, i.e. V_yMay be operated at a negative potential.

Most negative V_yBV with potential subjected to high side MOSFET_DSS1The breakdown voltage, i.e., the voltage limit corresponding to the reverse bias avalanche of the intrinsic P-N diode 22. In order to avoid a breakdown,the breakdown voltage of the MOSFET must exceed V_y(which may be negative) and V_battThe largest difference between them, i.e. BV_DSS1>(V_batt-V_y)。V_yIs bounded by the breakdown voltage and the forward bias voltage of diode 22 given by the relationship

(V_batt+V_f)>V_y>(V_batt–BV_DSS1)。

Similarly, there is no forward bias at P-N diode 21 and it is clamped to voltage V_x=-V_fIn case of (2), node V_xCannot be biased below ground by more than one forward-biased diode drop V_f. However, in the disclosed converter 10, the inductor 12 cannot couple V_xThe node voltage is driven below ground so only switching noise can cause the diode 21 to become forward biased.

However, within a specified operating voltage range of the relevant device, V_xCan be operated at a voltage above ground potential and is typically of ratio V_battThe corrected voltage is operated. Most positive V_xPotential from low side MOSFET BV_DSS2The breakdown voltage, i.e. the voltage corresponding to the reverse bias avalanche of the intrinsic P-N diode 21. BV of MOSFET to avoid breakdown_DSS1The breakdown voltage must be V_xShould exceed V_battI.e. BV_DSS2>V_x。V_xIs bounded by the breakdown voltage and the forward bias voltage of diode 21 given by the relationship

BV_DSS2>V_x>(-V_f)。

Due to V of inductor 12_yThe terminals being capable of operating below ground potential and V of the inductor 12_xThe terminal can be at V_battAbove, the circuit technology of the disclosed bipolar boost converter 10 is greatly different from that of the conventional boost converter 1, and the conventional boost converter 1 can only be usedOperate above ground potential and hard-wire its inductor to its positive input voltage. Because inductor 12 is not hardwired to any supply rail, the disclosed boost converter can therefore be considered a "floating inductor" switched mode converter. Conventional boost converters are not floating inductor technology.

Operation of the disclosed bipolar boost converter involves alternating between magnetizing the inductor and then transferring energy to the output before magnetizing the inductor again. The energy from the inductors may be delivered to the output simultaneously as described in algorithm 120 in fig. 6, or time multiplexed as described in algorithm 180 in fig. 8. Regardless of the algorithm employed, however, the first step in the operation of the disclosed bipolar boost converter is to store energy to an inductor, or referred to herein as "magnetizing" the inductor, in a process similar to charging a capacitor, except that the energy is stored in a magnetic field rather than an electric field.

Magnetizing the inductor: fig. 3A illustrates operation 25 of converter 10 during magnetizing of inductor 12. Since the inductor 12 is connected to the battery input V by not one but two series connected MOSFETs_battTherefore, both low-side MOSFET11 and high-side MOSFET13 must be turned on simultaneously to allow current I_L(t) rises. At the same time, the synchronous rectifier MOSFETs 14 and 15 remain off and non-conductive. The current-voltage relationship for the inductor is given by a differential equation

For small time intervals, the equation can be approximated as a differential equation as follows

V assuming minimum voltage drop across the on-state MOSFETs 11 and 12_L≈V_battAnd the above formula can be reset to

This describes the current I in the inductor 12 for a short magnetization time interval_L(t) may be approximated as a linear rise in current over time. For example, as shown in graph 70 of FIG. 4, at t₀To t₁During a time interval of (D), current I_LLinearly from time t₀Some non-zero current of time to time t₁The peak 71 at (end of the magnetization operation phase) rises. The energy stored in the inductor 12 at any time t is given by

This energy reaches a peak E only before its current is interrupted by turning off one or both of the MOSFETs 11 and 13_L(t₁). As shown in the graphs 70, 80, and 90 of FIG. 4, during magnetization, the current I in the low side MOSFET11₁And current I in high-side MOSFET13₂Equal and equal to the inductor current I_LSo that at time interval t₀To t₁During the period of time in which the air is being discharged,

I₁(t)=I₂(t)=I_L(t)。

when the current is I₂(t), a voltage drop V occurs across the series-connected low-side N-channel MOSFET11_DS2(on). In its linear region and carrying a current I_L(t) and on-state resistance R_DS2(on)In operation, voltage V_xIs given by

V_x=V_DS2(on)=I_L·R_DS2(on)，

As shown by line 51 in graph 50 of fig. 4. For low on-resistance (typically hundreds of microohms or less), V_xIs approximately equal to ground potential, i.e., Vx ≈ 0. Similarly, a voltage drop V occurs across the series-connected high-side P-channel MOSFET13_DS1(on). In its linear region and carrying a current I_L(t) and an on-state resistance of R_DS2(on)In operation, voltage V_yIs given by the following formula

V_y=V_batt-V_DS1(on)=V_batt-I_L·R_DS1(on)，

As shown by line 52 in graph 50 of fig. 4. For low on-resistance, V_yIs approximately equal to the cell potential, i.e. V_y≈V_batt。

If Vx ≈ 0 and V_y≈V_battThen is approximately V_L=(V_y-V_x)≈V_battValid assumptions. Thus, as indicated previously, shown in graph 70The rise in inductor current is therefore approximately a slope of (V)_battL) straight line segment. Further, assume the voltage + V across capacitor 18_OUT1Above ground potential and the voltage-V across capacitor 19_OUT2Below ground potential, then + V_OUT1>V_xAnd V_y>-V_OUT2Thereby making both P-N diodes 16 and 17 reverse biased and non-conductive.

Synchronous energy transfer to dual outputs: after magnetizing the inductor 12, the low-side MOSFET and the high-side MOSFET are simultaneously turned off in the synchronous transfer algorithm 120, as at time t in the graph 50 of fig. 4₁As shown. Interrupting the current I in the high-side MOSFET13₁And current I in low side MOSFET11₂So that V of the inductor_xTerminal fly-up to ratio V_OUT1A large positive voltage 53, forward biasing the diode 16 and outputting + V to the first voltage_OUT1Energy is transmitted. This also results in V of the inductor_yTerminal flying to ratio V_OUT2A more negative sub-ground voltage 54, thereby forward biasing diode 17 and simultaneously outputting-V to the second voltage_OUT2Energy is transmitted.

During the transition, the break-before-make circuit prevents the synchronous rectifier MOSFETs 14 and 15 from turning on and momentarily shorting the filter capacitors 18 and 19. In the case of non-conduction of the MOSFET, the diodes 16 and 17 carry the inductor current I_LAnd exhibits a forward bias voltage drop V_f. Then, V_xInstantaneous voltage on is equal to (V)_OUT1+V_f). Similarly, V_yInstantaneous voltage on is equal to (-V)_OUT2-V_f)。

According to Kirchoff's current law, at I_LTime t of peak value₁While interrupting the current I of the high side MOSFET13₁Causes the current to be redirected to the synchronous rectifier MOSFET and diode, so at node V_y（node V_y) At the position of the air compressor, the air compressor is started,

wherein, I₃Including the current in diode 17 and any junction capacitance associated with the turn-off MOSFET 15. Refer to graph 80 of FIG. 4 because of inductor current I_LCannot be changed immediately, so its current is changed from I₁Is rerouted to I₃As indicated by point 81.

At the same time, interrupting the current I in the low side MOSFET11₂Redirecting current to the synchronous rectifier diode and MOSFET to provide a voltage at node V_x（node V_x) To

Wherein, I₄Including the current in the diode 16 and any junction capacitance associated with the turn-off MOSFET 14. Refer to graph 80 of FIG. 4 because of inductor current I_LCannot be changed immediately, so its current is changed from I₁Is rerouted to I₃As indicated by point 81. At one sectionPoint V_xAt current of I₂And I₄Are "passed" between and at node V_yAt current from I₁To I₃Means V_xAnd V_yAct independently as uncorrelated circuits sharing a common energy storage element (i.e., inductor 12). In other words, the inductor 12 is substantially opposite to the node V_xAnd V_yThe voltages at (a) are decoupled to allow them to act independently during the time that energy is being delivered to the load and thus to the output capacitors 18 and 19.

As shown in circuit 30 of fig. 3B, at break-before-make time interval t_BBMThe synchronous rectifier MOSFETs 14 and 15 then turn on and divert current out of the diodes 16 and 17. With the MOSFET conducting, the voltage drop across the parallel combination synchronous rectifier and P-N diode is forward biased by a voltage drop V_fTransition to the on-state voltage V of the MOSFET_DS(ON)=I_L·R_DS(on). This changes the voltage V shown in curves 54 and 55, respectively, of graph 50_xAnd V_yIs clearly shown therein

V_x=V_OUT1+I_L·R_DS4(on)And is and

V_y=-V_OUT2+I_L·R_DS3(on)。

during this energy transfer phase, the current in inductor 12 simultaneously charges capacitors 18 and 19. In this manner, the positive polarity outputs + V_OUT1And negative polarity output-V_OUT2While being charged from a single inductor. According to the algorithm 120, the conditions shown in the schematic 30 should continue until one of the capacitors reaches a specified tolerance range. The tolerance range of the target voltage is responded by the controller to the feedback signal V_FB1And V_FB2To be determined. With analog control, the PWM controller 20 includes an error amplifier, a ramp generator, and a comparator for determining when to shut down the synchronous rectifier. With digital control, this decision can be made by logic or software according to the algorithm 120.

Synchronous energy transfer to one output: depending on the load conditions, either output may reach its target voltage first, as indicated by condition logic 121 and 122 in algorithm 120. Once any output reaches its specified output voltage, the converter is again reconfigured to stop charging the fully charged output capacitor, but continue to charge output capacitors that are not yet within the tolerance range until their specified voltage target.

For example, if at time t₂Negative output-V_OUT2At + V_OUT1Before reaching its target voltage, the first action is to turn off the synchronous rectifier MOSFET15, referred to herein as the "negative synchronous rectifier," and stop charging the capacitor 19. Since Δ Q = C · Δ V, the charge refreshed on each output capacitor during the charge transfer period is given by

Wherein, C₂Is the capacitance of the negative output filter capacitor 19.

Instant at which the synchronous rectifier is turned off and for t_BBMThe entire break-before-make time interval 59 of the period, P-N diode17 must carry the full inductor current I_LAnd inductor node voltage V_yBack to the value (-V)_OUT2-V_f). After the end of BBM time interval 59, high-side MOSFET13 is turned on and V is turned on in step 124_yJump to voltage V_batt-I_L·R_DS1(on)As shown by line 56 in graph 50. At time t₂During the transfer of (2), the inductor current I_LFrom I in the transition shown by point 82 in graph 80₃Steering I₁. However, the current I₄Remain unchanged.

This condition is shown in circuit 35 of FIG. 3C, where I_LCurrent path from V_battThrough the conducting high side MOSFET13, inductor 12 and on-state positive synchronous rectifier 14, and so on_L=I₁=I₄. Therefore, although the charging of the capacitor 19 has been stopped, the capacitor 18 continues to be charged. With V_yRise to near V_battand-V_OUT2Below ground potential, the P-N diode 17 remains reverse biased and non-conductive.

The operational phase of circuit 35 is maintained by conditional logic 126 according to algorithm 120, which continues until + V_OUT1Until it reaches its target voltage. Once + V_OUT1Upon reaching its target voltage, the positive synchronous rectifier MOSFET14 is turned off and for break-before-make period t_BBMDiode 16 carries the inductor current 60. During this time interval, V_xIncrease to a voltage V_OUT1+V_f.

However, once BBM time interval 60 ends, low-side MOSFET11 is turned on, as shown in graph 90 of FIG. 4, with current flowing from I₄Turn to I₂And inductor 12 begins a new cycle in which inductor 12 is magnetized back to the state shown in circuit 25. After this period is over, the total time is described as a time period T, which will vary depending on the load current. This time period is determined by the duration of the magnetization and the positive or negative charge transfer phase, which is even longer.

At a time from t₁The charge transferred to the capacitor 18 during the time interval to T is given by

Wherein, C₁Is the capacitance of the positive output filter capacitor 18.

The example given in FIG. 3C depicts the negative output-V_OUT2At positive output + V_OUT1Before reaching its target voltage. Algorithm 120 shows that the converter is also adapted to the opposite situation, i.e. the positive voltage reaches its specified point first. If the result of conditional 121 is "yes," then positive synchronous rectifier MOSFET14 is first turned off, thereby turning off for time interval T_BBMDiode 16 continues to supply current to capacitor 18. At step 123, the low side MOSFET is turned on, thereby turning V on_xForcing it to near ground potential causes the diode 16 to reverse bias and stop charging the capacitor 18.

At the same time, the negative synchronous rectifier MOSFET15 continues to be paired-V_OUT2The capacitor 19 is conductively charged. This situation (shown in circuit 110 of fig. 5) continues until condition 125 in the algorithm is satisfied, in which case negative synchronous rectifier 12 is turned off and after the BBM time interval, high-side MOSFET13 is turned on, thereby turning V on_yForced approach to V_battDiode 17 is reverse biased and charging of capacitor 19 is stopped.

Voltage regulation of the bipolar floating inductor regulator: operation of the bipolar boost converter requires turning on both the high side MOSFET13 and the low side MOSFET11 to magnetize the inductor 12, and then turning off these MOSFETs to transfer energy to the converter output. In this synchronous energy transfer algorithm 120, both the aforementioned high-side and low-side MOSFETs are turned off at the same time, thereby simultaneously beginning the transfer of energy from the inductor to both outputs.

Independent adjustment of the positive and negative outputs, whether charged simultaneously or not, is determined by the duration of energy transfer to the respective outputs. In particular by via feedback V_FB1And V_FB2Controlling the turn-off time of the low-side MOSFET11 and the high-side MOSFET14 allows the positive output voltage + V to be adjusted independently from the single inductor 12_OUT1And a negative output voltage-V_OUT2。

The on-time of the synchronous rectifiers 14 and 15, while affecting the efficiency of the converter, does not determine the charging time of the output capacitor. For example, when the positive synchronous regulator MOSFET14 is turned off, the diode 16 delivers charge to the capacitor 18 until the low side MOSFET11 is turned on. Turning on the low-side MOSFET11 and not turning off the synchronous rectifier MOSFET14 terminates charging the capacitor 18 and thus determines its voltage. Similarly, when the negative synchronous regulator MOSFET14 is turned off, the diode 16 delivers charge to the capacitor 18 until the low side MOSFET11 is turned on.

In this converter, the maximum voltage condition occurs when the diode conduction occurs, i.e., when both MOSFETs are turned off. For example, V_xThe maximum voltage at the node occurs when both the low-side synchronous rectifier MOSFET11 and the high-side synchronous rectifier MOSFET14 are turned off. In such a case, this voltage is derived from the output voltage + V_OUT1Plus a forward bias voltage V across the clamping diode_fTo determine, i.e. V_x(max)≤(V_OUT1+V_f). The MOSFET11 needs to be able to block V in its off state_x(max)。

Similarly, V_yThe maximum negative voltage at the node occurs when both the low-side synchronous rectifier MOSFET13 and the high-side synchronous rectifier MOSFET15 are turned off. In such a case, this voltage is derived from the output voltage-V_OUT2Subtracting the forward bias voltage-V across the clamping diode_fTo determine, i.e. V_y>(-V_OUT2-V_f). The MOSFET13 needs to be able to block V in its off state_y。

One feature of the disclosed converter 10 is: because the inductor is floating, i.e., not permanently connected to the supply rail, turning on either of the high-side and low-side MOSFETs 11 and 13 without fully turning on may force V_yOr V_xThe voltage at does not magnetize the inductor 12 or increase the current in the inductor 12. For a conventional boost converter like the converter in fig. 1, it is not possible for a single MOSFET to control both V and V_xThe voltage in turn causes the current to conduct and magnetize the inductor. In other words, in conventional converters, controlling the inductor voltage also causes additional (or sometimes undesirable) energy storage. In the disclosed converter, V_xAnd V_yEither of which can be forced to supply voltage without magnetizing the inductor.

Another consideration is the output voltage range of the conventional converter 1. If a P-N diode 5 is connected across the synchronous rectifier MOSFET, the minimum output voltage must be V for the output of the boost converter_battSince the diode is forward biased immediately when power is applied to the input of the regulator, thereby boosting the output to V_batt. In the disclosed dual output converter, from V_battTo + V_OUT1Comprises two switches with diodes of opposite polarity P-N to allow the + V to be made_OUT1Adjusted to the ratio V_battSmall voltage, a feature that cannot be achieved by conventional boost converter technology.

Thus, although the boost circuit can only boost the voltage, the disclosed converter produces a voltage that may be less than the battery voltage, may be less than the battery voltageA positive output voltage equal to or greater than the battery voltage and therefore not limited to only above V_battThe operation of (2). Application of the boost converter technology to buck regulation is one object of a Related patent application entitled "High-Efficiency Up-Down and Related DC/DCConverters" by Richard k.

The use of time-division multiplexed inductors in positive and negative Output boost converters is described in a related patent application entitled "Dual-Polarity Multi-Output DC/DCConverters and Voltage Regulators" by Richard K.Williams, filed on even date herewith, and is incorporated herein by reference.

Time division multiplexing bipolar floating inductance regulator: as described previously, the preferred embodiment of the present invention charges both the positive output and the output, and continues charging the other output while stopping charging the output whose output reaches the target regulation voltage.

Fig. 7 shows an alternative sequence using time division multiplexing. In the circuit 140 of fig. 7A, the low side MOSFET and the high side MOSFET are turned on to magnetize the inductor 12. In fig. 7B, only the low-side MOSFET11 is turned off, so that V is_xFlying and rising combined + V_OUT1Capacitor 18 is charged up to V_OUT1Until it reaches its target value. The synchronous rectifier MOSFET is turned on as the diode 16 conducts to improve efficiency. In this period, the output capacitor q9 is not charged.

Once V is_OUT1To reach its target voltage, the synchronous rectifier 14 is turned off and the low side MOSFET11 is turned on, thereby turning V on_xForcing to ground potential and stopping the charging of capacitor 18. At the same time, the high-side MOSFET13 is turned off to allow V_yThe positively biased diode 17 is turned to negative and-V is output to negative_OUT2The capacitor 19 is charged. The synchronous rectifier MOSFET15 is turned on to improve efficiency. once-V_OUT2Upon reaching its specified voltage target, the synchronous rectifier 15Is turned off. The high-side MOSFET13 is then turned on and the inductor 12 is magnetized again. This cycle is then repeated in a time division multiplexed order. An algorithm for time division multiplexing is shown in the flow chart 180 of fig. 8.

While this algorithm may be implemented using analog circuitry, an alternative approach utilizes a digital controller or microprocessor 220 as in diagram 200. As shown, an analog feedback V from the output_FB1And V_FB2May be multiplexed by MOSFETs 226A and 226B and put in digital format using a single a/D converter 225. Voltages below ground require level shifting circuit 227 to convert the voltage to a positive potential.

As shown, the positive output of microcontroller 220 can directly drive MOSFETs 211 and 213, but level shifting circuits 223 and 224 are required to drive floating synchronous rectifier MOSFETs 214 and 215.

Claims (25)

1. A bipolar dual output synchronous boost converter, comprising:

an inductor;

a high side gate coupled between a voltage source and a first terminal of the inductor;

a low side gate coupled between ground and a second terminal of the inductor;

a first output node coupled to a first output gate coupled to a second terminal of the inductor;

a second output node coupled to a second output gate coupled to a first terminal of the inductor; and

a controller configured to activate or deactivate the high side gate, the low side gate, the first output gate, and the second output gate, the controller further configured to provide a first mode, a second mode, and a third mode of circuit operation.

2. The converter of claim 1 wherein for the first mode, the controller activates the high side gate to connect a first terminal of the inductor to the voltage source to receive an input voltage and the controller activates the low side gate to connect a second terminal of the inductor to ground.

3. The converter of claim 1 wherein for the second mode, the controller deactivates the high side gate and the low side gate, the controller also activating the first output gate and the second output gate to connect a first terminal of the inductor to the first output node and a second terminal of the inductor to the second output node.

4. The converter of claim 1 wherein for the second mode, the controller activates the high side gate and deactivates the first output gate to connect the first terminal of the inductor to the voltage source to receive an input voltage, and the controller activates the second output gate and deactivates the low side gate to connect the second terminal of the inductor to the second output node.

5. The converter of claim 1, wherein for the third mode, the controller activates the high side gate and deactivates the second output gate to connect the first terminal of the inductor to the voltage source to receive an input voltage, and the controller is further implemented to deactivate the low side gate and activate the first output gate to connect the second terminal of the inductor to the second output node.

6. The converter of claim 1 wherein for the third mode, the controller activates the first output gate and deactivates the high side gate to connect the first terminal of the inductor to the first output node, and the controller activates the low side gate and deactivates the second output gate to connect the second terminal of the inductor to ground.

7. The converter of claim 1 wherein the control circuit causes the first, second and third modes to be selected in a repeating sequence.

8. The converter of claim 7, wherein the repeating sequence is a first pattern, a second pattern, a first pattern, a third pattern.

9. The converter of claim 7, wherein the repeating sequence is a first pattern, a second pattern, a third pattern.

10. The converter of claim 1 wherein the controller is further configured to provide a fourth mode in which the controller deactivates the high side gate and activates the second output gate to connect the first terminal of the inductor to the first output node, and the controller activates the low side gate and deactivates the first output gate to connect the second terminal of the inductor to ground.

11. The converter of claim 1 further comprising a feedback circuit having a first port coupled to the first output node and a second port coupled to the controller, the controller configured to utilize voltage feedback sensed from the first port to adjust a duration of the second mode to control the voltage of the first output node.

12. The converter of claim 11, wherein the feedback circuit includes a third port coupled to the second output node and a fourth port coupled to the controller, the controller configured to adjust a duration of the third mode to control a voltage of the second output node.

13. The converter of claim 1, further comprising a feedback circuit including a third port coupled to the second output node and a fourth port coupled to the controller, the controller further configured to adjust a duration of the third mode to control a voltage of the second output node.

14. A method for operating a bipolar dual output synchronous boost converter including an inductor, a high-side gate coupled between a voltage source and a first terminal of the inductor, a low-side gate coupled between ground and a second terminal of the inductor, a first output node coupled to a first output gate, a second output node coupled to a second output gate, and a controller, the first output gate coupled to the second terminal of the inductor, the second output gate coupled to the first terminal of the inductor, the controller configured to activate or deactivate the high-side gate, the low-side gate, the first output gate, and the second output gate, the method comprising:

configuring the controller to cause the boost converter to operate in a first mode in which the controller activates the high-side gate to connect a first terminal of the inductor to the voltage source to receive an input voltage and the controller activates the low-side gate to connect a second terminal of the inductor to ground;

configuring the controller to cause the boost converter to operate in a second mode in which the controller deactivates the high-side gate and the low-side gate, the controller also activating the first output gate and the second output gate to connect a first terminal of the inductor to the first output node and a second terminal of the inductor to the second output node; and

configuring the controller to cause the boost converter to operate in a third mode in which the controller activates the high side gate and deactivates the second output gate to connect the first terminal of the inductor to the input voltage, and the controller deactivates the low side gate and activates the first output gate to connect the second terminal of the inductor to the second output node.

15. The method of claim 14, wherein the first, second, and third modes are selected in a repeating sequence.

16. The method of claim 15, wherein the repeating sequence is a first pattern, a second pattern, a first pattern, a third pattern.

17. The method of claim 15, wherein the repeating sequence is a first pattern, a second pattern, a third pattern.

18. The method of claim 14, further comprising adjusting a duration of the second mode to control a voltage of the first output node.

19. The method of claim 18, further comprising adjusting a duration of the third mode to control a voltage of the second output node.

20. A method for operating a bipolar dual output synchronous boost converter including an inductor, a high-side gate coupled between a voltage source and a first terminal of the inductor, a low-side gate coupled between ground and a second terminal of the inductor, a first output node coupled to a first output gate, a second output node coupled to a second output gate, and a controller, the first output gate coupled to the second terminal of the inductor, the second output gate coupled to the first terminal of the inductor, the controller configured to activate or deactivate the high-side gate, the low-side gate, the first output gate, and the second output gate, the method comprising:

configuring the controller to cause the boost converter to operate in a first mode in which the controller activates the high-side gate to connect a first terminal of the inductor to the voltage source to receive an input voltage and the controller activates the low-side gate to connect a second terminal of the inductor to ground;

configuring the controller to cause the boost converter to operate in a second mode in which the controller activates the high side gate and deactivates the first output gate to connect a first terminal of the inductor to the voltage source to receive an input voltage, and the controller activates the second output gate and deactivates the low side gate to connect a second terminal of the inductor to the second output node; and

configuring the controller to cause the boost converter to operate in a third mode in which the controller activates the first output gate and deactivates the high side gate to connect the first terminal of the inductor to the first output node, and the controller activates the low side gate and deactivates the second output gate to connect the second terminal of the inductor to ground.

21. The method of claim 20, wherein the first, second, and third modes are selected in a repeating sequence.

22. The method of claim 21, wherein the repeating sequence is a first pattern, a second pattern, a first pattern, a third pattern.

23. The method of claim 21, wherein the repeating sequence is a first pattern, a second pattern, a third pattern.

24. The method of claim 20, further comprising adjusting a duration of the second mode to control a voltage of the first output node.

25. The method of claim 20, further comprising adjusting a duration of the third mode to control a voltage of the second output node.