HK1175250B - System and method for acoustic doppler velocity processing with a phased array transducer - Google Patents

Abstract

Systems and methods for measuring velocity in fluid are disclosed. In one aspect, a method (900) comprises transmitting a first set of signals of a bandwidth broader than the measuring system, receiving echoes from the first set of signals, obtaining a first velocity estimate based on the echoes, transmitting a second set of signals of a bandwidth narrower than the measuring system, receiving echoes from the second set of signals, obtaining velocity estimates based on the echoes from the second set of signals, selecting one of the velocity estimates based on the first velocity estimate. In another aspect, a method (280) comprises removing substantially a bias related to a first velocity from raw velocity estimates. In another aspect, a method (1900) comprises obtaining a velocity estimate for each of a set of transmitted pings, calculating a velocity based on the sum of the velocity estimates.

Description

System and method for acoustic Doppler velocity processing with phased array transducers

The present application is a divisional application of the invention patent application entitled "system and method for acoustic doppler velocity processing with phased array transducer" filed on 27/9/2007 with application number 200780044377.5.

Technical Field

The present invention relates to velocity measurement systems, and in particular to acoustic doppler current profilers, other underwater devices such as doppler logs, and radar applications.

Background

A current profiler is a sonar system for remotely measuring the velocity of water over varying distances. Current profilers are used in freshwater environments such as rivers, lakes and estuaries, and also in saltwater environments such as oceans to study the effect of current velocity. Accurate measurement of water flow velocity is important in various fields, such as weather forecasting, biological research of nutrients, environmental research of sewage dispersion, and commercial development of natural resources including petroleum.

Typically, a water flow profiler measures the flow rate of a vertical volume of water for each depth "cell" of water up to a maximum distance, thus producing a "profile" of the water's velocity. A typical profiler system includes a transducer to generate pulses of sound (which sound like "pings" when down-converted to human auditory frequencies) that are backscattered as echoes from plankton, small particles, and small heterogeneous species in the water. Similarly, a bottom tracking doppler velocity log receives backscattered echoes from the bottom surface. The received sound has a doppler shift proportional to the relative velocity between the scatterer and the transducer.

The physical phenomenon of determining the component Vx of a single velocity vector from this doppler shift can be briefly explained by the following equation:

equation 1

In equation 1, c is the speed of sound in water, which is about 1500 m/s. Thus, by knowing the frequency of the transmitted sound, f_TAnd the angle of declination of the transmitter transducer, theta, and by measuring the frequency received from a single narrow-band pulse, the Doppler shift, f_DThe components of a velocity vector can be determined. By subtracting the measured value of vessel ground reference velocity, Ve, the relative velocity of the measured horizontal "slice" or depth cell can be determined. The ground reference velocity is measured by echo ranging whenever the seafloor reaches within sonar range, or by a navigation system such as LORAN or GPS. Fig. 1a and 1b show exemplary water flow profiles, where the water flow velocities (Vx, Vy) for the north and east faces are shown as a function of depth cell.

In some configurations, the water flow profiler is configured as an assembly of four discrete transducers spaced 90 ° azimuthally from each other around the electronics housing. This transducer arrangement is known in the art as the Janus configuration. Assuming that the water flow is uniform in a plane perpendicular to the interaction axis of the transducer, a system with three beams allows the measurement of three velocity components, Vx, Vy and Vz (identified in the oceanographic literature as u, v, w, respectively). However, the use of four beams is often for redundancy and reliability. The current profiler system may be attached to the hull of the vessel, held on a stationary buoy, or moored to the sea floor like the current profiler 10 shown in fig. 2.

The current profiler is controlled by a combination of factors including maximum profile distance and time, space (the size of the depth cell), and velocity resolution. Temporal resolution refers to the time required to achieve a velocity estimate with the required accuracy. In a typical application, the current profiler will make a series of measurements, which are then averaged together to produce a single velocity estimate, and which has an acceptable variance, or squared error, of velocity. In some applications, the variance is more of a concern than the variance in the observations. The deviation is the difference between the measured speed and the actual speed. For example, it is caused by the asymmetry of the band-limited system components. Measurement bias is present even when long-term averaging has reduced the variance to a predetermined acceptable limit. For example, bias dominance typically occurs on measuring large-scale features, such as those occurring on temperature and salinity interfaces.

In addition to the current profiler, there are many other velocity measurement systems. Some examples are radar systems, air flow measurement systems, and other underwater instruments, such as doppler meters that measure the velocity of a car or boat relative to the surface or bottom of a body of water. All of these speed measurement systems have a wide range of applications, and it would be beneficial in the art to utilize and/or modify the characteristics of these types of devices so that their features can be developed to improve existing products and create new products that have not yet been developed.

Disclosure of Invention

The system, method, and apparatus of the present invention each have several aspects, none of which can solely afford the desirable attributes. Without limiting the scope of the invention, its more prominent features will now be discussed briefly.

In one aspect, there is a method of measuring velocity of a fluid medium with a measurement system, wherein the measurement system includes a transducer. The method includes transmitting a first set of one or more signals, wherein the signals have a bandwidth substantially wider than a bandwidth of the measurement system, receiving echoes from the first set of signals; obtaining a first velocity estimate based at least in part on echoes of the first set of signals; transmitting a second set of one or more signals, wherein the signals have a bandwidth substantially equal to or narrower than a bandwidth of the measurement system; receiving echoes from the second set of signals; obtaining two or more possible velocity estimates based at least in part on the echoes of the second set of signals; and selecting one of the possible velocity estimates based on the first velocity estimate.

In another aspect, there is a system configured to measure velocity. The system includes a transducer configured to transmit a first set of one or more signals and a second set of one or more signals, and to receive echo signals from the first and second sets of signals, wherein a bandwidth of the first set of one or more signals is substantially wider than a bandwidth of the system; the bandwidth of the second set of one or more signals is substantially equal to a bandwidth narrower than the measurement system. The system further includes a processing module to obtain a first velocity estimate based in part on the echoes of the first set of signals and to obtain two or more possible velocity estimates based at least in part on the echoes of the second set of signals, and to select one of the possible velocity estimates based on the first velocity estimate.

In another aspect, there is a system configured to measure velocity. The system includes means for transmitting a first set of one or more signals, wherein the signals have a bandwidth substantially wider than a bandwidth of the measurement system; means for receiving echoes from the first set of signals; means for obtaining a first velocity estimate based at least in part on echoes of the first set of signals; means for transmitting a second set of one or more signals, wherein the bandwidth of the signals is substantially equal to or narrower than the bandwidth of the measurement system; means for receiving echoes from the second set of signals; means for obtaining two or more possible velocity estimates based at least in part on the echoes of the second set of signals; and means for selecting one of the possible velocity estimates based on the first velocity estimate.

In another aspect, there is a method of measuring velocity of a fluid medium using a phased array transducer. The phased array transducer includes a plurality of transducer elements arranged to form a single two-dimensional array. The method includes receiving echoes of a plurality of beams generated by a transducer, calculating a preliminary velocity estimate based at least in part on the echoes; and substantially removing the bias associated with the first velocity from the preliminary velocity estimate. The first velocity is perpendicular to the surface of the two-dimensional array.

In another aspect, there is a system configured to measure velocity. The system includes a phased array transducer comprising a plurality of transducer elements arranged to form a single two-dimensional array, wherein the transducer is configured to generate a plurality of beams and to receive echoes of the beams. The system further includes a processing module configured to calculate a coarse velocity estimate based at least in part on the echoes, and to substantially remove a bias associated with the first velocity from the preliminary velocity estimate. The first velocity is perpendicular to the surface of the two-dimensional array.

In another aspect, there is a system configured to measure velocity. The system includes means for generating a plurality of beams and receiving echoes of the beams, where the means includes a phased array transducer including a plurality of transducer elements arranged to form a single two-dimensional array. The system further includes means for calculating a preliminary velocity estimate based at least in part on the echo; and means for substantially removing the bias associated with the first velocity from the preliminary velocity estimate. The first velocity is perpendicular to the surface of the two-dimensional array.

In another aspect, there is a method of measuring velocity of a fluid medium using a transducer. The method includes transmitting an acoustic signal, wherein the acoustic signal includes N (where N is an integer and N > 1) pulse signals for each of a plurality of beams, receiving an echo from each pulse signal, obtaining a velocity estimate for each of the N pulse signals based on the echoes of the pulse signals, and calculating a velocity based on a sum of the N velocity estimates such that there is substantially no error in the velocity due to cross-coupling between the beams.

In another aspect, there is a system configured to measure velocity. The system includes a transducer for transmitting an acoustic signal and receiving an echo of each pulse signal, wherein the acoustic signal includes N (N is an integer and N > 1) pulse signals for each of a plurality of beams. The system further includes a processing module configured to obtain a velocity estimate for each of the N pulsed signals based on the echoes of the pulsed signals, and to calculate a velocity based on a sum of the N velocity estimates to substantially eliminate errors due to cross-coupling between the beams.

In another aspect, there is a system configured to measure velocity. The system includes means for transmitting an acoustic signal, wherein the acoustic signal includes N (where N is an integer and N > 1) pulse signals for each of a plurality of beams, means for receiving echoes from each pulse signal, means for obtaining a velocity estimate for each of the N pulse signals based on the echoes of the pulse signals, and means for calculating a velocity based on a sum of the N velocity estimates such that there is substantially no error in the velocity due to cross-coupling between the beams.

Drawings

FIG. 1a is a scatter plot of an exemplary water flow profile showing east velocity vectors plotted as a function of depth, and FIG. 1b is a scatter plot of an exemplary water flow profile showing north velocity vectors plotted as a function of depth;

FIG. 2 is a perspective view of one example of a current profiler moored to the sea floor;

FIG. 3 is a pulse diagram illustrating pulses transmitted by different embodiments of a water flow profiler, including a pulse incoherent Doppler system, a pulse coherent Doppler system, a wideband Doppler system, and a coded pulse Doppler system;

4a, 4b, 4c are diagrams of sets of code pulses illustrating exemplary transmit codes for a wideband Doppler system and a code pulse Doppler system;

FIG. 5 is a block diagram illustrating one embodiment of a two-dimensional transducer array, which is part of one embodiment of the water flow profiler 10 of FIG. 2;

FIGS. 6a and 6b illustrate the operation of the two-dimensional array with phase-shifted beamformer of FIG. 5 described above;

FIG. 7 shows a detailed view of the "Y-axis transmit, transmit beamformer" of FIG. 6b for illustrating how the beamformer transmits two beams simultaneously;

FIG. 8 is a perspective view illustrating an example of the structure of four acoustic beams that are tilted with respect to the array normal (i.e., Z-axis) and lie in two planes perpendicular to the array surface plane (i.e., the X-Y plane) of the transducer array of FIG. 5;

FIG. 9 illustrates a top view of one embodiment of the transducer array of FIG. 5;

FIG. 10 is a three-dimensional view of one embodiment of the transducer array of FIG. 5 illustrating a multi-layer structure;

FIG. 11 is a functional block diagram illustrating one embodiment of ADCP 10, which includes the two-dimensional transducer array of FIG. 5;

FIGS. 12a and 12b illustrate a comparison of two examples of coded sequences to be transmitted in measuring velocity;

FIGS. 13a and 13b illustrate a comparison of two examples of coding elements in the time and frequency domains;

figures 14a and 14b illustrate examples of signals to be transmitted for wide and narrow bandwidth velocity estimates, respectively;

FIGS. 15a and 15b illustrate the process of wide and narrow bandwidth velocity estimation and blur resolution, respectively;

FIG. 16 is a flow chart illustrating an embodiment of a velocity processing method suitable for use with phased array transducers that uses wide bandwidth transmission to resolve ambiguities in estimating narrow bandwidth velocities;

FIG. 17 is a flow chart for explaining an example of a velocity processing method which substantially removes a deviation caused by a vertical component of a velocity estimation value;

FIGS. 18a and 18b illustrate the operation of extrapolating the phase function of the received signal to the nominal lag time of each beam;

FIG. 19 illustrates one embodiment of a velocity processing method that substantially removes cross-coupled side lobe errors from the velocity estimates;

20a, 20b and 20c show three examples of coded pulses that may be used in the velocity processing method;

FIG. 21 is a table illustrating one example of a set of signal codes that may be transmitted by the method of FIG. 19;

fig. 22 illustrates the format of signal coding associated with the pulse signals 1-4 of beam 1 in fig. 21.

Fig. 23a and 23b illustrate two examples of a scheme to generate 90 phase increments/decrements between successive code sequences.

Detailed Description

Various aspects and features of the present invention will become more fully apparent from the following description and appended claims, taken in conjunction with the accompanying drawings. In the drawings, like reference numbers can indicate identical or functionally similar elements. In the following description, a detailed description is given to provide a thorough understanding of the disclosed methods and apparatus. However, it will be understood by those of ordinary skill in the art that the disclosed systems and methods may be practiced without these specific details. For example, electrical components are shown in block diagram form in order not to obscure certain aspects with unnecessary detail. In other instances, the components, other structures and techniques may be shown in detail to further explain certain aspects.

It is also noted that some aspects may be described as a process which is depicted as a flowchart, a flow diagram, a structure diagram, or a block diagram. Although a flowchart may describe the operations as a sequential process, many of the operations can be performed in parallel or concurrently, and the process can repeat. In addition, the order of the operations may be rearranged. A process is terminated when its operations are completed. A process corresponds to a method, function, procedure, subroutine, etc. When a procedure corresponds to a function, its termination corresponds to a return of the function to the calling function or the main function.

The description will be given for the case of a water flow profiler, but other velocity measurement systems, such as a doppler velocity log, also have these general characteristics. The various embodiments of the velocity processing method described below can be applied to flow profilers and other velocity measurement applications.

Water flow profile

FIG. 1a is a scatter plot of an exemplary water flow profile, illustrating east-facing velocity vectors plotted as a function of depth. FIG. 1b is a scatter plot of an exemplary water flow profile, illustrating velocity vectors for the north plane plotted as a function of depth. The exemplary water velocity profile depicted in the scatter plots of fig. 1a and 1b is one type of information that is also the target of the water profiler.

FIG. 2 is a perspective view of one example of a current profiler moored to the sea floor. The current profiler 10 is semi-permanently moored to the seafloor 12. In such profiler deployments, the recording of the water flow profile is typically stored in a non-volatile memory (not shown) located inside the water flow profiler 10.

The water flow profiler 10 as shown in fig. 2 generates a set of acoustic beams 14a, b, c, d emanating from the transducers. The current profiler 10 is looking up, i.e., the acoustic beam 14 is directed vertically at the ocean surface. Each beam 14 "illuminates" a water column, which may be broken down into horizontal slices known as distance or depth elements, such as the elements indicated at 16. By properly sending and receiving the acoustic pulses, the phase shift between the pulse echoes is calculated. The phase shift is then translated stepwise to a doppler frequency, velocity along the beam 14, followed by one or more orthogonal water velocity components, such as indicated by 18a, b.

The transducer of the water flow profiler 10 may be implemented in various ways. In one embodiment, the water flow profiler 10 includes an assembly of four discrete transducers spaced 90 ° azimuthally apart from each other around the electronics housing. This transducer arrangement is known in the art as the Janus configuration. In some embodiments, the water flow profiler 10 includes a two-dimensional transducer array, which will be described in detail in FIG. 5. The current profiler 10 may be deployed in other ways than that shown in fig. 2, including, for example, various combinations of downward, upward, or other angled views on a fixed or moving platform, or at a surface, bottom, or mid-depth berth.

Various Doppler measurement techniques

Figure 3 is a pulse diagram illustrating pulses transmitted by different embodiments of a water flow profiler, including a pulse incoherent doppler system, a pulse coherent doppler system, a wideband doppler system, and a coded pulse doppler system. Figure 3 provides in schematic form a number of different doppler measurement techniques for use in Acoustic Doppler Current Profilers (ADCP).

In a first technique, a pulse incoherent ADCP transducer 20 is shown producing a pulse 22 at time t. The size of a single pulse 22 to be transmitted is tailored to fit the associated depth cell. After traversing the depth cell, at time t plus a time equal to the length of the pulse (Lpulse), pulse 22 is shown, which moves to a new position as indicated at 24.

Depending on the density of scattering at each depth, the pulse 22 may produce an echo (not shown) at each depth cell. The water flow rate is measured at the ideal depth cell based on a predetermined lag time between sending the pulse and receiving the ideal echo. Pulse incoherent ADCP measures water flow velocity by measuring doppler shift in the frequency of the returned echoes. The echo of each pulse is used independently. The doppler frequency is indirectly calculated from the difference between two different samples of the received signal. The term "incoherent" refers to the fact that coherence does not have to be maintained between pulses.

In fig. 3, a pulsed coherent ADCP transducer 26 is shown as transmitting pulses 28. Pulse 28 is shorter in duration (greater depth resolution) than pulse 22 in a pulse incoherent system. As with pulse incoherent doppler systems, the echo of each individual pulse is allowed to return to the transducer 26 before the next pulse 30 is transmitted. However, unlike pulse-incoherent systems, the fundamental measure of a pulse-coherent system is the phase change between two successive echoes at the same depth. The term "coherent" refers to the fact that coherence needs to be maintained between pulses. In some embodiments, pulse-coherent ADCP transmits a series of short pulses, where phase coherence is maintained over the transmitted sequence.

Figure 3 also illustrates the pulses generated by the broadband ADCP transducer 32. The wideband method differs from the pulse incoherent or pulse coherent method in that the wideband method utilizes two (or more) pulses in a beam (or equivalent) simultaneously, such as the pulses indicated at 34a and 34 b. In fig. 3, the pulses are separated by a lag time, L1, equal to the pulse interval. After a distance has passed and the echoes returned to the transducer 32, the phase change between the pulsed echoes at the same distance is measured using the autocorrelation function.

Unlike the pulse-coherent approach, the maximum profile distance of a broadband current profiler is not limited to the pulse repetition interval. The pulse length, or width, is typically much shorter than the depth cell size, which results in a large temporal bandwidth product (the term "wideband"). The time-bandwidth product is the product of the average time and the pulse bandwidth.

Fig. 3 further illustrates the pulses generated by the encoded pulse wideband ADCP. The transducer 38 generates pulses 40a, b which propagate through the water, as shown by the subsequent pulses 41a, b. Each pulse 40 includes four equal-sized code elements 42a, b, c, d, each code element including one or more periods (or portions thereof) of the acoustic waveform to be transmitted. The code elements 42 represent phase codes such that each element is either 0 degrees phase or 180 degrees phase. Although only two encoding pulses are shown in fig. 3, the method can be generalized to include more than two pulses.

For the coded pulse ADCP, the phase change is measured the same as for the wideband method discussed earlier. In addition, however, pseudo-random phase encoding is suitable for pulses that allow longer pulses to be used without reducing bandwidth. Longer pulses increase echo power, thus delaying the signal associated with greater range dissociation and extending the useful profile distance of the system. The encoding pulse may be as large as the size of the depth cell. If the pulse separation or lag time L1 is equal to the pulse length, then the pulses are combined into a single continuous coded transmission.

Fig. 4 shows three examples of "ideal" code pulses with different lengths, wherein the code pulses are generated by a code pulse wideband ADCP. Each graph (fig. 4a, b, c) corresponds to a pulse, or pulse signal. The actual waveform injected into the water is slightly different than that depicted in fig. 4 due to the limited bandwidth of the transducer and power amplifier. Therefore, there is a short recovery time after the phase reversal in the corresponding actual waveform.

Fig. 4a includes three different representations of a series of coding elements, generally indicated by 44 a-j. The first coded representation is a transmit waveform indicated generally at 46. Each code element 44 is a set of four cycles of the carrier signal. A 180 degree phase shift may occur between code elements 44, as shown, for example, by the transition between code elements 44a and 44 b. The exemplary pulse of fig. 4a has M-10 code elements 44, with the first five code elements 44a-e inverted and repeated by the last five code elements 44f-j to essentially combine the two pulses into a continuous waveform 46. Inverting the second pulse, e.g., encoding elements 44f-j, is useful to reduce noise variance.

Thus, for waveform 46, the autocorrelation function (discussed further below) is performed on the first five elements 44a-e and the last five elements 44f-j after inversion with a lag time equal to the time to transmit the five code elements. In some cases, the number of coding elements in a particular application will match the size of the depth cell.

The pulse code may also be represented in binary form, which is illustrated by a code sequence generally indicated by 47 in fig. 4 a. The coding sequence 47 is based on each coding element 44 defined by two bits. The most significant bit (B1) indicates whether the transmitter is on (1) or off (0) for the duration of the coding element 44. The least significant bit (B0) indicates the phase of the encoding element 44, "0" indicates 0 °, and "1" indicates 180 °. When B1 is the value "0", the value of B0 does nothing.

The code sequence 47 shows the decimal equivalent of the binary code. For example, the code element 44a, defined as "2" in the code sequence 47, means that the transmitter is on and the code element 44a is 0 degrees phase. The phase waveform 48 provides the same basic information as the transmit waveform 46 and the code sequence 47, but it is expressed in the form of a square wave.

Fig. 4b shows an encoded pulse, which differs from fig. 4a in that the pulse is twice as long (M ═ 20). The first ten code elements 44 in fig. 4b are identical to the code elements 44 in fig. 4 a. The last ten coding elements 44' are repetitions of the first ten. Thus, the two pulses 44, 44' combine into a single transmit waveform having a lag time equal to the time that ten code elements are transmitted.

Fig. 4c shows a coded pulse which differs from fig. 4b in that the pulse is longer (M ═ 30), since a dead zone of ten code elements is placed between two groups of ten code elements 44, 44' to be transmitted. Thus, the lag time is equal to twenty code elements. The short term error (i.e., variance) in the doppler frequency is inversely proportional to the pulse spacing. The distance resolution is determined by the length of the encoding pulse.

In some embodiments, the coding is carefully chosen to eliminate bias from center peak and side lobe noise in the autocorrelation function. By inverting the second pulse, e.g., as shown in fig. 4a, to half of the pulse signal to be transmitted, the center peak noise is effectively eliminated. The following steps are taken to cancel sidelobe noise: (1) using a code with zero autocorrelation at each side of the side peak at a lag time (where the phase measurement is taken), (2) using a code with minimal side lobes around the side peak, where the side lobes are symmetrically arranged around the side peak, and (3) using a code on successive pulse signals such that the deviations cancel by averaging.

The pulse spacing, or lag time L1, determines the accuracy of the distance-velocity resolution, where shorter lag times mean greater resolution. It is even possible to make the lag time smaller than the length of a single encoding pulse by sending pulses that overlap in one or more encoding elements. For example, using letters of the alphabet to represent code elements, the sequence "ABABA" would allow two pulses "ABA" where the two pulses have a length of three code elements to be transmitted and a lag time equal to the time to transmit two code elements.

Those skilled in the art will understand and appreciate that a balance is required in selecting the appropriate code, code length, and pulse separation for the multipulse waveform, depending on the particular application of the invention. Hereinafter, the wideband ADCP and coded pulse wideband ADCP systems and methods will be collectively referred to as wideband ADCP, unless otherwise indicated.

Structure and operation of phased array transducer

FIG. 5 is a block diagram illustrating one embodiment of a two-dimensional transducer array, which is part of one embodiment of the water flow profiler of FIG. 2. A typical structure 100 of an array of planar acoustic wave transducers is depicted. The individual array elements 102 are electrically interconnected along front side columns 104 and back side rows 106. The array elements 102 are interconnected to associated beamformers 108, 110 by a two-axis transmit/receive (T/R) switch 118. The transmit 108 and receive 110 beamformers may be phase or time delay beamforming networks. In an exemplary embodiment, the beamformer is a phase beamforming network.

The coordinate system used for this description is as follows, with rows 106 running on the X-axis, columns 104 running on the Y-axis, and the Z-axis perpendicular to plane 116.

The array surface 116 is circular, but other forms of factors, such as ellipses or polygons that are generally two-dimensional symmetric, are also suitable for forming narrow, tilted beams in the form of generally circular cones. The array is made up of a large number of small elements 102 having a plane of symmetry, typically square, circular, or rectangular (i.e., their surface cross-section). In one embodiment, the surface width of each element is approximately 0.5 λ, where λ is the acoustic wavelength in water of the desired center frequency. To form a beam of 4 ° beam width requires an array diameter of approximately 16 λ, consisting of a 32 × 32 element array of approximately 800 elements. The rear row 106(X direction) and front column 104(Y direction) of the array elements are electrically connected together along parallel lines of elements with a thin acoustically transparent material as shown in fig. 5. The rows and columns are typically, but not necessarily, orthogonal.

Each of the X-axis rows 106 and Y-axis columns 104 of the array is connected to a T/R switch 118 that electrically connects the X and Y sets of lines to the respective X and Y receive beamformers 110 in the receive mode and to the X and Y transmit beamformers 108 in the transmit mode. In some embodiments, T/R switch 118 is controlled by T/R logic signal 120 to switch between transmit and receive modes. In other embodiments, the T/R switch includes passive components that operate by detecting whether a transmit signal is applied by the transmit beamformer 108. The T/R switch switches to a transmit mode if a transmit signal is detected and to a receive mode if a transmit signal is not detected.

When in transmit mode, the array lines are connected through the T/R switch 118 to the transmit beamformer 108, which provides electrical transmit drive signals (relative to the electrical impedance of the lines of transducer elements) from a low impedance power supply. When in receive mode, the array lines are connected through the T/R switch to a receive beamformer 110, which receives electrical signals from the transducer lines.

The low power/load impedance on each of the X and Y lines (low power impedance during transmission) allows simultaneous and independent access to each of the X rows 106 and Y columns 104 to apply the transmitted electrical drive signal to each of the X rows and Y columns. Also, parallel sets of X and Y axis lane arrays may be formed simultaneously and independently. The X-axis transmit and receive line array is formed by parallel electrical connections along the rows 106 of the back side and providing low impedance ground signals on all the Y-axis columns 104 of the front side.

During the transmit mode, a transmit drive signal is applied to the parallel X-axis back side electrical interconnect lines through the T/R switch from a transmit amplifier having a low output impedance relative to signal ground. When the X-axis drive signals are applied to the individual X-axis line arrays, the entire Y-axis 32 parallel line array surface remains a low impedance path to signal ground (through which it passes through the Y-axis T/R switch 118a to the low impedance Y-axis driver of the Y-beamformer 108 a) to ensure that the X-axis drive signals are applied only across the rows of the X-axis and are not coupled to the Y-axis side of the array. Similarly, when a Y-axis drive signal is applied to the Y-axis line array, the entire X-axis array surface remains a low impedance path to signal ground to allow the signal to be applied independently to the Y-axis without coupling to the X-axis. Thus, by superimposing the X and Y axis transmit drive signals, the low impedance associated with the transmit beamformer source allows the X and Y axis line transmit arrays to be independently formed simultaneously.

During receive mode, the presence of an electrical signal on each X-axis row 106 represents the sum of the electrical signals received by all elements of each row. When a signal is received from a column, the column signal is independent of the simultaneously received row signals. Similarly, when a signal is received from a row, the row signal is independent of the column signal being received at the same time.

Independent and simultaneous electrical access of the X rows and Y columns via the X and Y signal lines during transmit and receive modes allows the array to be used as a two-dimensional array to simultaneously and independently form multiple tilted groups of acoustic beams in the X-Z and Y-Z planes. The beamforming operation in each plane is the same as a conventional one-dimensional phased and/or time delay array. Thus, a two-dimensional beamforming operation is generally equivalent to two overlapping one-dimensional arrays, where one array is rotated by 90 °.

During transmit mode operation, the phase or time delay signals applied to the X rows form a tilted acoustic transmit beam in the Y direction (Y-Z plane). Simultaneously and independently, the phase or time delay signals applied to the Y columns produce a tilted acoustic transmit beam in the X direction (X-Z plane). During receive mode operation, the electrical signals received at the X rows are phase or time delayed, which are combined in the X row receiver beamformer to produce a tilted received acoustic beam in the Y direction. Simultaneously and independently, the signals received in the Y column and combined in the Y-side beamformer produce a tilted received acoustic beam in the X direction. Thus, by superimposing the X and Y axis electrical and acoustic signals, two dimensional acoustic beam formation from a single planar array in both transmit and receive modes is achieved.

Fig. 6a and 6b illustrate the operation of the two-dimensional array with a phase-shifted beamformer described previously in fig. 5. To understand the basic principles of operation of how these two-dimensional transmit and receive acoustic beams are formed, we consider the operation of a subset of a 16 element array of a 32 x 32 element two-dimensional array transducer.

A long burst of sound is received at a single frequency (narrow band), f, and at a wavelength λ ═ c/f, where c is the speed of sound propagation in the fluid medium, and the incoming acoustic radiation wave front 200, which proceeds in the X direction and at an angle θ 202 to the Z axis (Z orthogonal to the plane of the array, or to the plane of the figure), proceeds to a different distance from each of the column line arrays 204 of the Y axis (front side), thus passing through each of the line arrays at different times, and typically at different phases. As illustrated in fig. 6a, the path length difference between adjacent line arrays (α)206 is related to the center-to-center separation distance (d) of the elements by the following equation:

equation 2 of a ═ dsin θ

The wavefront arrival time difference (τ) between adjacent line arrays is:

τ ═ α/c ═ d/c sin θ equation 3

If the elements are spaced apart by a distance corresponding to, for example, one-half wavelength (d ═ λ/2) of the arriving narrowband signal, then the path length difference expressed in terms of the wavelength of the arriving signal is:

α ═ λ/2 sin θ equation 4

For an angle of arrival of e.g. 30,

α ═ (λ/2) sin30 ═ λ/4 equation 5

This corresponds to a 90 ° angular phase shift between the elements of the arriving narrowband signal. Thus, when a narrowband pulse is received by all Y-axis line arrays, the phases of the received electrical signals along the group of four Y-axis line arrays are 0, 90, 180, and 270 degrees, respectively, with the back side coupled to the low impedance virtual ground 208, as previously described.

Consider first the receive operation of the front side (Y) columns, where the rear side rows 106 are all coupled to ground signals in the X-axis receive beamformer 110 b. Each set of four X-axis electrical signals (in the 4X 4 array for illustration) is connected to the virtual ground node 208 in the receive preamplifier of the receive beamformer 110a to form reference signals for the rows on the back side and phase shifted by-90 (0, -90, -180, and-270 degrees) between the connected line arrays as shown. The applied phase shift is used to compensate for the phase shift in path length between different elements in the narrowband acoustic wave pulse, which is associated with the line array, as illustrated in fig. 6 a. The resulting four signals 210 will be in phase and when summed will form the maximum acoustic interference pattern when receiving a wavefront arriving at an angle of incidence of 30. The maximum corresponds to the central axis of one main lobe of the formed beam.

A second receive beam may be formed by inverting the 90 deg. sign of the applied phase shift on the four signals and summing the signals for an incoming acoustic ray wavefront that proceeds in the-X direction and makes an angle theta (e.g., at an angle of incidence of-30 deg.) with the Z direction. Since the phase groups of four signals are repeated for additional groups of four line arrays, a larger array can be achieved by summing the signals of all four line arrays to further enhance the interference pattern at ± 30 °. When an additional set of four line array segments is utilized, as described above, the acoustic signal gain along the ± 30 ° direction increases, or correspondingly, the beam width in that direction decreases when additional array sets are added.

Another method of beamforming is to first sum all the equiphased signals from different array groups and then apply the applied 90 ° phase shift in the summed group of four signals. This can be achieved by simply electrically connecting each fourth array of lines in parallel. The effective beam width in the X direction is determined by the number of line array groups in the array. In the Y direction, the beam width is determined by the beam pattern of the line array, which is inversely proportional to the length of the array lines (acoustic wavelength). In some embodiments, narrow slanted acoustic beams with similar widths in both planes are desirable, with the X and Y plane sizes remaining the same.

During transmit mode, the two-axis array operates similarly to the receive mode described above, except that the signal flow is reversed, as illustrated in fig. 6 b. First consider a transmit operation of a front-side column, where the rear-side rows are all coupled to a ground signal. The long tone burst carrier frequency 300 is applied to the phase shifted transmit beamformer 108a, producing four drive signals having relative phases of 0, 90, 180 and 270 degrees. These signals are applied from the low impedance driver to four parallel wire sets 302 of the Y column. This applied phase shift will compensate for the phase shift resulting from the different path lengths between the line arrays and will form a transmitted acoustic signal interference pattern at an angle of incidence of-30 deg., which corresponds to the center of one main beam lobe. As previously described, by reversing the sign of the 90 ° to which the phase shift is applied, another transmit beam can be formed at an angle of incidence of-30 °.

The receive and transmit operations are the same in the Y-axis. When considering signals applied and received from the rows of the rear side, the columns of the front side are coupled to the ground signal through a low impedance. The presence of a low transmit drive load impedance to ground on each side results in completely independent X and Y axis operation. From the superposition of the X and Y axis signals, it can also be seen that both axes (i.e., rows and columns) can be operated simultaneously.

Fig. 7 shows a detailed view of the "Y-axis transmit beamformer" of fig. 6b for illustrating how the beamformer transmits two beams simultaneously. The transmit beamformer of fig. 7 includes two additional inputs (in addition to the transmit signals) to the beamformer which control the temporal and spatial phase shifts, respectively. As illustrated, these phase shifts are applied to the transmit signal to produce four different drive signals.

The spatial phase shift control signal controls two switches of the converter. Each switch may be located in one of two settings: 0 ° or 180 °. In an exemplary embodiment, the two switches are in the "0 °" setting without the use of a spatial phase shift control signal.

The time phase shift control signal is configured to control whether the left beam, the right beam, or the two beams are generated on one plane. The left beam refers to the beam that is in the-X direction and is traveling at an angle to the Z direction. A right beam refers to a beam that is in the X direction and that is traveling at an angle to the Z direction. The two switches are controlled by a time phase shift control signal to switch to one of three settings.

The left or right beam can be generated by controlling the phase shift of the four drive signals illustrated in fig. 6B. By superposition, the beamformer can simultaneously generate two beams by adding together the drive signals required to generate each beam.

The table at the top of fig. 7 illustrates four drive signals that produce a left beam, a right beam, and two beams. Each drive signal is represented by a vector. The vector of each of the four drive signals used to generate the left and right beams is the vector sum of the drive signals used to generate each beam. For example, in the first column, the driving signals for generating the left beam, the right beam, and the two beams are a vector having a unit magnitude and a 315 ° phase, a vector having a unit magnitude and a 45 ° phase, and a vector having a magnitude and a 0 ° phase, respectively. Similarly, the receive beamformer of fig. 6a may be employed to receive two beams simultaneously.

The two-axis beamforming technique described above, which utilizes fixed phase delays to form narrow transmit and receive beams, is referred to as a "two-dimensional phased array" transducer. It can be used in narrowband applications that transmit long bursts of essentially a single frequency or narrow bandwidth. Four tilted narrow beams, positioned in the X-Z (beams 3 and 4) and Y-Z planes (beams 1 and 2), each at an angle to the Z direction, are formed from a single planar array of apertures, as shown in fig. 8.

In other embodiments, phased array transducers are used in broadband applications. As seen in the sound ray diagram of fig. 6a, for a fixed element spacing d, the angle of each beam is related to the acoustic frequency by the following equation:

θ＝sin^-1(λ/4d)＝sin^-1(c/4fd) equation 6

Thus, the beam angle is frequency dependent, and if the incoming or outgoing wave has a broad frequency spectrum, the mainlobe beam pattern will be correspondingly broadened in the angular domain. Due to the beam spreading caused by this bandwidth, phased array technology cannot work with either wideband ADCP, which transmits signals with a very wide spectrum (typically 20-50% of the carrier frequency), or narrowband applications.

It will be appreciated from the foregoing that certain inventive aspects may be included to produce many combinations of two-axis tilted beams having different carrier frequencies, beam characteristics, and signal bandwidth capabilities.

Figure 9 illustrates a top view of one embodiment of the transducer array of figure 5. The exemplary embodiment is configured to produce two narrow beam-wide beams at a carrier frequency of 150kHz on each of two axes for use in ADCP applications.

The exemplary embodiment includes a circular transducer array and two substantially identical beam forming networks, each providing drive signals for forming two tilted transmit/receive beams. For example, the diameter D600 of the array is approximately 160 mm. A150 kHz piezoceramic element 102 having 800 individual square faces closely spaced and having a center-to-center distance 604 of 5mm (at 150kHz, at a wavelength of about 1/2, based on a propagation velocity of about 1536 m/s). The exemplary embodiments can be modified to meet the specific requirements of an application.

Figure 10 is a three-dimensional view of one embodiment of the transducer array of figure 5 illustrating a multi-layer structure. In this view, the thickness dimension is exaggerated to show the stacked structure. Ceramic array elements 700, such as the 800 elements 102 shown in fig. 9, are electrically and mechanically connected by two very thin acoustically transparent Flexible Printed Circuits (FPCs) 702, 704 on the top and bottom surfaces of the ceramic. The circuit may be made of kapton (tm) or other suitable material. Electrical connection to each ceramic element 700 is achieved, for example, by mounting and bonding (or alternatively, soldering) printed electrical leads to the conductive surfaces of the array elements. The bonding may be accomplished with a suitable adhesive or glue, although other forms of bonding are also suitable. The connection is along the columns of elements on the front side and along the rows on the back side and into the columns on one side (Y wires 705) and the rows on the other side (X wires 707). A fiberglass material 706, for example 1/8 inches (3.18mm) thick (for example having the trade mark "G-10" or other similar material) having surface dimensions matching the ceramic is bonded to the front end of the top flex circuit on each 150kHz transducer array. The glass fiber (G-10 or equivalent) sheet is an acoustic quarter wave transformer to improve impedance coupling between the array and the water and significantly increase the bandwidth of the transducer elements. In certain embodiments, a significant increase in transducer bandwidth is desirable for broadband ADCP applications. A urethane layer 708 bonded to the front of the fiberglass sheet seals the surface to the water in front. A gas-filled cardboard layer 710 is placed between the back plane of the housing 712 and the back of the bottom flex circuit to reflect the acoustic wave energy transmitted backwards and provide the necessary mechanical support to resist the water pressure at the front of the transducer array face 714. It will be appreciated that other front and back matching layers may be used depending on the particular application.

Exemplary ADCP Using phased array transducers

Figure 11 is a functional block diagram illustrating one embodiment of an ADCP 10 that includes the two-dimensional transducer array of figure 5. The electronics may be functionally divided into a front end transducer assembly 160 that receives acoustic signals, and an electronics assembly 162 that is used to coordinate transmission and reception, as well as perform signal processing.

As discussed with reference to fig. 15, the rows 106 in the X-axis and columns 104 in the Y-axis of the array are each connected to a T/R switch 118 that electrically connects the X and Y line sets to respective X and Y receive beamformers 110 in the receive mode and to the X and Y transmit beamformers 108 in the transmit mode.

In transmit mode, the coded pulse transmission is initialized by the digital signal processor 196. The digital signal processor may be a digital signal processor, or any other appropriate signal processing circuit, including any general purpose single-or multi-chip microprocessor, such as an ARM,PentiumPentiumPentiumPro，8051，Poweror any special purpose microprocessor such as microcontrollers and programmable gate arrays. In some embodiments, the digital signal processor 196 may be configured to execute one or more software modules.

A set of user-specified parameters, including the number of cycles per code element and the code length, are stored in ROM of the digital signal processor 196. The digital signal processor 196 communicates the waveform specific parameters to the timing generator 170 via the digital bus 168. Under the control of the digital signal processor 196, the timing generator 170 controls the code transmitter 172 to generate the appropriate code pulse pairs, including the dead band. The coded pulses are amplified by the power amplifier 174 and ultimately transmitted as coded acoustic waveforms into the water by the transducer array 100 (see fig. 5).

During some user-specified blanking intervals, when no pulses are transmitted, echo pulses received from the transducer array 100 are passed from the T/R switch circuits 118a and 118b to a set of receive beamformers 110a and 110b, as discussed with reference to FIG. 5.

In one embodiment, the receive amplifiers 180 each comprise a Signetics SA604A semiconductor chip. Although designed for intermediate frequency conversion applications, the two amplifiers of the SA604A chip occasionally operate over the desired frequency range of the water flow profiler. The amplifiers are connected in series to the outputs of the beamformers 110a and 110 b. The signal strength of the echo can also be obtained by the system through the receive amplifier 180, e.g., from pin 5, the RSSI output of the SA604A chip. In one embodiment, the signal strength is digitized and recorded for subsequent processing.

The signal intensity signal can be calibrated when used to measure backscatter intensity, particle concentration and particle flux. For example, one application of such measurements is in fishing operations, where signal strength is used to determine silt concentration and vertical flux in a plume produced by dumping a heap of debris.

The output signal of the receive amplifier 180 is fed to a set of in-phase mixers 182a, b, c, d and a set of quadrature mixers 183a, b, c, d. The mixers 182, 183 form the product of the received signal and the carrier signal. More particularly, the mixers 182, 183 are used to heterodyne the received signal to translate the carrier signal into a DC signal (where the carrier signal includes in-phase [ cosine ] and quadrature [ sine ] signals, collectively referred to as quadrature signals). In an exemplary embodiment, the mixers 182, 183 are implemented by two 74HC4053 triple two-channel analog multiplexer/demultiplexer chips, such as the chips provided by Signetics. Mixers 182, 183 receive quadrature signals from quadrature generator 184.

In one embodiment, the quadrature generator 184 includes a pair of series-connected D flip-flops (not shown). The inverted output Q' of the second flip-flop is fed back to the input D of the first flip-flop. In operation, quadrature generator 184 receives an oscillating signal from timing generator 170. The oscillation signal is sent to the clock input ends of the two D flip-flops. Thus, the in-phase signal is sampled from the inverting output Q' of the second flip-flop and the quadrature signal is sampled from the non-inverting output Q of the first flip-flop. The quadrature signal is then sent from the quadrature generator 184 to the mixers 182, 183.

The mixers 182, 183 pass their respective amplified quadrature signals to a bank of programmable low pass filters 188a, b, c, d and 189a, b, c, d. The low pass filter 188 is programmed by the controller 192 to pass sideband frequencies, e.g., carrier frequencies up to 20%, which correspond to phase modulation of the encoding pulses. The filtered quadrature signal outputs from the low pass filters 188, 189 (labeled cosine and sine channels) are fed into a sampling module 194.

The controller 192 and the timing generator 170 control the functions of the sampling module 194. The receive period is initialized by the timing generator 170 after the last element of the encoded sequence has been transmitted. After a user programmable delay to allow the receive electronics in the transducer assembly 160 to recover, the timing generator 170 generates a sampling strobe chain for triggering the analog-to-digital converter in the sampling module 194. Thus, each sample bit corresponds to one sample of one of the quadrature components of one of the four waveforms received by the transducer array 100. The digital data is sent over the digital bus 168 to a Digital Signal Processor (DSP) 196. In an exemplary embodiment, the digital bus 168 is a custom, asynchronous bus having 16 data lines (BD0-BD15) and 12 address lines (BA1-BA 12). In some embodiments, the digital bus 168 may transmit data at speeds up to 400ns per character.

In some embodiments, sampling module 194 includes a multi-bit analog-to-digital converter (ADC) configured to sample each quadrature component of the four waveforms, rather than the single bit samples previously discussed. This approximately linearly samples the waveforms.

The DSP 196 calculates the autocorrelation function (r (h)) of the received signal at a predetermined lag time corresponding to the number of code elements of the first pulse. The autocorrelation function is used to measure the correlation of the waveform received at time t and the waveform received at the delay lag time. In an exemplary embodiment, the received signal is a series of samples. Thus, r (h) is used to measure the correlation of the series of samples and the series of samples delayed by h (a predetermined lag time represented by an integer number of samples). To calculate this function, DSP 196, through sampling module 194, applies the following equations independently to each of the four cosine-sine pair outputs:

equation 7

Wherein:

h is a predetermined lag time represented by an integer number of samples;

j is an integer number of samples within the depth cell of interest;

cosine and sine are data sampled from the cosine and sine channels (e.g., from low pass filters 188, 189 in FIG. 11)

i＝(-1)^1/2；

S_j＝cos_j+sin_ji; and

s denotes the complex conjugate of S.

In an exemplary embodiment, resolution has been sacrificed for speed, and each sample value is represented by one bit. However, it can be shown that only half of the available information in the cosine-sine information is lost with this method.

Thus, DSP 196 may perform fast multiplication by XORing two 16-bit data characters received from the cosine-sine channel via sampling module 194. The digital representation (0, 1) is interpreted by the DSP 196 as (-1, + 1). Once the multiplication is performed, the summation of the products can be done using a look-up table stored in the EPROM. In an exemplary embodiment, DSP 196 uses a Texas Instruments TMS320vc 3332 bit, digital signal processing chip.

Once a complex representation of each autocorrelation result is obtained, the DSP 196 calculates the Doppler frequency f_D. For a linear system, the following calculation can be followed:

equation 8

Wherein:

f_Dis the doppler frequency of the echo;

i is the imaginary part of the complex number;

r is the real part of the complex number;

h is the lag time for calculating the autocorrelation; and

t is the time between samples.

For a hard limit system, such as the one described and illustrated herein, the digital signal processor 196 utilizes the following doppler frequency equation:

equation 9

In addition, the digital signal processor 196 uses the normalized values of I and R in equation 9, i.e., the normalized autocorrelation function, by dividing each by the autocorrelation at zero lag time. Note that for a linear system, the normalization step is eliminated in the division I/R, so it is not necessary.

In an alternative embodiment, the digital signal processor 196 calculates the vertical velocity components based on equation 1 and then converts these velocities to a reference to ground, e.g., subtracting the velocity component generated by the chip. In another embodiment, the Doppler frequency and/or other intermediate calculations may be sent to the transport vessel via the I/O port 156. The I/O port 156 is configured to connect to a transmission cable (not shown) for measurement, where real-time post-processing of the flow profiler is desirable. In another embodiment of the water flow profiler electronics, the results of the Doppler frequency are stored in a recording medium, such as an EEPROM or flash nonvolatile memory added to the digital bus 168.

In some embodiments, the DSP 196 further generates a time phase shift control signal for each beamformer (see fig. 7). In some embodiments, the timing generator 170 further generates a spatial phase shift control signal for each beamformer (see fig. 7).

Blurred resolution

The above-described wideband velocity processing method may suffer from bandwidth starvation when used with phased array transducers due to the narrower bandwidth of the beamformer in the phased array transducer and beam propagation of the array for the wideband signal. A balance needs to be made among the design choices to meet the requirements of a particular application. These design choices include maximum operating speed, short term noise, and zero speed performance, also referred to as position maintenance performance. Short-term noise refers to the variance in velocity due to the random effect of eventually reaching equilibrium in time. Position holding performance is the accuracy of the measurement when at rest. Since a balance of maximum operating speed and short term noise in the speed processing algorithm is necessary, the phased array speed processing is limited to lower speed operation or higher short term noise and may be a non-zero output when at rest. As described below, certain embodiments utilize wide bandwidth transmissions to resolve ambiguities in narrow bandwidth velocity estimates to set resolution. It should be noted that "wide bandwidth" and "narrow bandwidth" are used to indicate that "wide bandwidth" basically has more frequency components or frequency components of a wider spectrum than "narrow bandwidth".

Fig. 12a and 12b illustrate a comparison of two examples of code sequences to be transmitted in measuring velocity. The two code sequences 802 have 7 code elements 804 of equal size (i.e., the same length), where each code element has one or more periods (or portions) of the acoustic waveform to be transmitted. In some embodiments, each encoding element is the same except that the phase of the encoding element is different. The bandwidth of each code element 804 is inversely proportional to the length of each code element or the number of carrier cycles per code element, as illustrated in fig. 13a and 13 b.

In some embodiments, the encoding elements represent phase encodings such that each element is 0 (indicated by a "1") or 180 (indicated by "-1") degrees phase. A phase encoding (e.g., a pseudo-random phase encoding) is applied to the code elements such that a code sequence comprising a variable number of code elements has the same bandwidth as each code element.

The code sequence in fig. 12a and the code sequence in fig. 12b are the same, except that the length of each code element in fig. 12a is twice the length of each code element in fig. 12 b. Thus, the bandwidth ratio of the pulses in fig. 12a to the pulses in fig. 12b is 2 to 1.

Fig. 13a and 13b illustrate a comparison of two examples of time and frequency domain coded elements. The left graph is a time domain representation of the coding element, while the right graph is a frequency domain representation of the coding element. As illustrated, the coding elements in fig. 13a and 13b comprise 8 and 16 carrier periods, respectively. Thus, the coding element of fig. 13a has a bandwidth (about 12% of the carrier frequency) that is twice the bandwidth (about 6% of the carrier frequency) of the coding element of fig. 13 b.

Figures 14a, 14b, 15a and 15b illustrate the operation of one embodiment of a velocity processing method that uses wide bandwidth transmission to resolve ambiguities in narrow bandwidth velocity estimates. Figures 14a and 14b illustrate examples of signals to be transmitted for wide and narrow bandwidth velocity estimates, respectively. Fig. 15a and 15b illustrate the process of wide and narrow bandwidth velocity estimation and blur resolution, respectively.

Figure 14a is a two-dimensional graph of a signal to be transmitted, i.e., a pulsed signal, during wide bandwidth speed processing and reception of an echo signal. The vertical axis represents the amplitude (or power) of the signal, while the horizontal axis represents time. In an exemplary embodiment, the signal 810 to be transmitted (also referred to as a pulse or pulse signal) includes three code sequences 812, all of which are identical (different shading is used only to identify the three code sequences). Each code sequence 812 has a length TL, S and includes one or more equal-sized code elements (not shown). An appropriate phase encoding (e.g., pseudo-random phase encoding) is applied to the code elements of each code sequence 812 so that the code sequences and code elements have the same bandwidth. Similarly, a transmitted signal 810 comprising three consecutive encoded sequences 812 has the same bandwidth as the encoding elements. Thus, the bandwidth of the transmitted signal 810 is inversely proportional to the length of each coding element, wherein all coding elements of the transmitted signal 810 have the same length.

After transmitting the signal, receiving an echo signal814 and range gates the quadrature signal (described with reference to fig. 11). For a depth cell at a distance R from the transducer, the time between sending a signal and beginning to receive an echo signal is approximately 2R/c as illustrated, where c is the speed of sound in water. The received echoes are placed in a storage receiver defined by a "range-gated" orthogonal signal, i.e. the echoes received at time tn come from a receiver located at a distance R ═ ct_nA diffuser of/2. The width of the strobe signal matches the transmitted signal 810, which is 3 × T_L，S. The phase change between the range-gated signal and the range-gated signal delayed by the processing lag time is calculated. The velocity is then estimated based on the phase change.

In an exemplary embodiment, the transmitted signal 810 needs to be designed such that its bandwidth is substantially greater than the nominal bandwidth of the phased array beamformer. For example, the bandwidth of the pulse signal may be twice the nominal bandwidth of the phased array beamformer. The nominal bandwidth of the beamformer is the bandwidth beyond which unacceptable errors in phase and gain will occur, resulting in long-term and short-term inaccuracies. In an exemplary embodiment, the nominal bandwidth is approximately 6% of the carrier frequency. The desired bandwidth is achieved by varying the length of the coding elements of the transmitted signal 810.

Also, the processing lag time needs to meet the maximum speed requirement, i.e. the maximum speed for which the speed measurement method is designed to process. The processing lag time is inversely proportional to the maximum speed. The selection processing lag time is limited by the transmitted signal 810. In an exemplary embodiment, for example, the processing lag time is either T_L，SOr 2T_L，S. Therefore, the signal 810 to be transmitted needs to have an appropriate T_L，SThe value is such that the selected processing lag time meets the maximum speed requirement. As described above, the code sequence length T is determined by the desired bandwidth of each code element_L，SCan be adjusted by the number of coding element primes within each coding sequence. It should be noted that other factors need to be further considered when designing the signal 810 to be transmitted.

Figure 14b is a two-dimensional graph of a signal to be transmitted during narrow bandwidth speed processing and echo signals. The transmitted signal includes two pairs (820a and 820b) of two code sequences (822), where the two pairs are separated by a time period during which no pulses are transmitted.

In an exemplary embodiment, all four encoded sequences 822 of the transmitted signal are the same. Each encoding sequence 822 includes one or more equal-sized encoding elements (not shown). An appropriate phase encoding (e.g., pseudo-random phase encoding) is applied to the code elements of each code sequence 822 such that the code sequences have the same bandwidth as the code elements. Similarly, the signal to be transmitted, or the pulse signal, has the same bandwidth as the coding element. Thus, the bandwidth of the signal to be transmitted is inversely proportional to the length of each coding element, wherein all coding elements of the signal to be transmitted have the same length. By varying the length of the coding elements of the transmitted signal 822, the desired bandwidth is achieved.

After transmitting the signal, receiving the echo signal and processing the delayed time T_L，LAt, the range gates the quadrature signal. The phase change between the echo signal and the echo signal delayed by the processing lag time is calculated. The velocity is then estimated based on the phase change.

Fig. 15a is a two-dimensional graph illustrating the relationship between the phase of the autocorrelation function and the estimated physical velocity value for narrow bandwidth velocity processing. The phase of the autocorrelation function of the echo signal and the echo signal delayed by the processing lag time can be calculated based on equation 7. The vertical axis represents the autocorrelation function ρ_LPhase phi of_LAnd the horizontal axis represents the estimated value V of the physical velocity_Physical. The relationship is described by the following equation:

Φ_L＝Ang(ρ_L)＝π*V_Physical/U_A，L+ k 2 pi equation 10

U_A，L＝c/(4*N_C，L) Equation 11

Wherein N is_C，LIs the number of carrier cycles in the processing lag time, k can be any positive or negative integer such that Φ_SIn the range of-pi to pi. Since the maximum value of the physical speed is greater than U_A，LThen, as illustrated, multiple velocities are possible for the detected phase change. This is called "speed blur". The velocity ambiguity is caused by the fact that samples separated in phase by 2 pi radians cannot distinguish this phenomenon.

Fig. 15b is a two-dimensional graph illustrating the relationship between the phase of the autocorrelation function and the estimated value of the physical velocity for wide bandwidth velocity processing. The vertical axis represents the autocorrelation function ρ_SPhase phi of_SAnd the horizontal axis represents the estimated value V of the physical velocity_Physical. The relationship is described by the following equation:

Φ_L＝Ang(ρ_S)＝π*V_Physical/U_A，Sequation 12

U_A，S＝c/(4*N_C，S) Equation 13

Wherein N is_C，SIs the number of carrier cycles in the processing lag time, phi_SIn the range of-pi to pi. Since the maximum value of the physical speed is not more than U_A，SThus, as illustrated, only one velocity corresponds to a detected phase change.

As illustrated, the wide bandwidth velocity processing method may obtain an estimate of velocity from the detected phase change. However, the estimate includes a small bias, a higher short term and a higher position hold error. In some embodiments with lower performance requirements, it is possible to use the wide bandwidth estimate directly.

In some embodiments where higher performance is desired, the wide bandwidth velocity estimate is used to resolve the velocity ambiguity in fig. 15a by determining which channel can be used to determine the narrow bandwidth velocity estimate. Once the channel is determined, there is only one narrow bandwidth velocity estimate corresponding to the phase change. The narrow bandwidth velocity estimate is more accurate than the wide bandwidth velocity estimate because it removes substantially all small deviations in the wide bandwidth estimate. Further, the narrow bandwidth velocity estimate has lower short term noise and position hold error due to the long lag time used in it. The process of selecting one from a set of possible values based on another factor may also be referred to as ambiguity resolution.

The blur resolution process can be mathematically described as follows. First, the value of k (which may be a positive or negative integer) is determined such that Φ is based on equations 10 and 11_LFalling within the range-pi to pi, wherein V_Physical＝V_Broad。V_BroadIs a wideband velocity estimate. The selection of k corresponds to the above description of the selection of one channel. Second, once k is determined, there is a one-to-one correspondence between the phase of the autocorrelation function and the narrow bandwidth velocity estimate based on equations 10 and 11, where k is a determined constant value. The narrow bandwidth velocity estimate obtained is the velocity to be selected.

The choice of processing lag time and exact transmission used in narrow bandwidth transmission will depend on the particular application. In an exemplary embodiment, the transmission is designed as follows. Referring again to the narrow bandwidth transmission depicted in FIG. 14b, the lag time T between the first and second pairs_L，LIs the processing lag time used in the subsequent processing of the received echo signal. The processing lag time needs to be long enough to meet the short-term noise requirements and position-keeping requirements. Also, the processing lag time needs to be short enough to provide a sufficiently large blur speed for avoiding blur errors due to short-term noise of the wideband speed estimate. Typically, if the short-term noise of the wide bandwidth speed estimate is a fraction of the blur speed of the narrow bandwidth blur speed, then the blur error can be avoided.

Fig. 16 is a flow diagram illustrating an embodiment of a velocity processing method suitable for use with phased array transducers that uses wide bandwidth transmission to resolve ambiguities in estimating narrow bandwidth velocities. Depending on the embodiment, certain steps of the method may be deleted, combined together, or reordered.

The method 900 begins at block 902 where a first set of signals are transmitted via a phased array transducer. The set of signals may include one or more signals depending on the particular application. In the exemplary embodiment, four beams are transmitted. Each signal to be transmitted has a bandwidth substantially wider than the nominal bandwidth of the measuring device. For example, the transmit signal may have a bandwidth that is twice the bandwidth of a phased array beamformer.

The measurement device includes a transducer. In some embodiments, the bandwidth of the measurement device may be determined by the bandwidth of the transducer, or the bandwidth of the beamformer.

And the transmit signal is designed such that an echo signal of the transmit signal can be processed at a processing lag time that meets maximum speed requirements. There may be many signals that meet these requirements. In one embodiment, the signal may comprise a plurality of consecutive code sequences, each code sequence being identical. Each code sequence further comprises a plurality of consecutive code elements, each code element being identical except that phase encoding is applied to the code elements such that the bandwidth of the code sequence and the code elements are identical.

Next, at block 904, a first velocity estimate V is obtained by the DSP 196 by processing echo signals of the first set of signals at a processing lag time that meets maximum velocity requirements_Broad. In one embodiment, as discussed, a velocity estimate may be obtained with reference to FIG. 11. First, based on equation 7, the phase of the autocorrelation function between the echo and the echo delayed by the processing lag time is calculated. Then, based on equations 12 and 13, a velocity estimate is obtained from the phase.

Moving to block 906, a second set of signals are transmitted outward via the phased array transducer. The set of signals includes one or more signals, depending on the particular application. In the exemplary embodiment, four beams are transmitted. The bandwidth of each signal to be transmitted is substantially equal to, or narrower than, the nominal bandwidth of the phased array beamformer. And the transmit signal is designed such that the echo signal of the signal to be transmitted can be processed at a processing lag time that is in accordance with the short-term noise requirement and the position-keeping requirement and that is capable of avoiding ambiguity errors.

Next, at block 908, the DSP 196 obtains a set of possible velocity estimates comprising two or more estimates by processing the echo signals of the second set of signals at processing lag times that meet the short-term noise requirements and the position-keeping requirements and avoid ambiguity errors. In one embodiment, the phase of the autocorrelation function between the echo signal and the echo signal delayed by the processing lag time is calculated based on equation 7. A set of velocity estimates is then determined based on equations 10 and 11, where each velocity estimate corresponds to a different value of k.

Moving to block 912, a selection is made from a set of possible velocity estimates based on the first velocity estimate. The selected velocity estimate is more accurate than the first velocity estimate. In one embodiment, the closest one to the first velocity estimate (i.e., wideband velocity) is selected from the set of possible velocity estimates. In another embodiment, the velocity estimates are selected as follows. The value of k (which may be a positive or negative integer) is determined such that Φ L falls within the range of- π to π, based on equations 10 and 11, where V_BroadAs a V_PhysicalAn estimate of (d). A narrow bandwidth velocity estimate is then selected that corresponds to the determined value of k.

It should be noted that although the exemplary embodiments are described in conjunction with phased array transducers, the method may be equally applicable to other transducers. When other transducers are used, the bandwidth of the phased array beamformer referred to in the method may be replaced by the bandwidth of the other transducers or the devices used with the transducers. Many signals may be used in this exemplary embodiment, one of which is illustrated in fig. 14a and 14 b.

Removing deviations caused by vertical velocity components

When the acoustic Doppler velocity processing method described above is used in a water flow profiler that includes a phased array transducer, certain deviations caused by the velocity component normal to the array surface (also referred to as the "vertical component") may not be accounted for. This deviation is the result of two discrete effects: the acoustically dependent uncompensated velocity of this velocity component and the error that is not related to the doppler effect but is caused by the phase gradient. Other speed processing applications, such as radar applications, are also prone to similar deviations.

In some applications, the array elements as illustrated in FIG. 6B are spaced nominally one-half wavelength apart and one-quarter period in phase. A quarter period in phase corresponds to a quarter wavelength of the wavefront displacement at the actual (not nominal) sound velocity and frequency. Thus, the phased array geometry gives the following relationship:

equation 14

Equation 15

Wherein:

d is the spacing of the elements of the array,

λ₀is the nominal wavelength of the light at which the light is emitted,

the x is the actual wavelength of the light,

c₀1536m/s is the nominal speed of sound,

c is the actual speed of sound (at the transducer),

f₀is the carrier center frequency and is,

f_cis the centroid frequency of the received frequencies (due to Doppler shift, receiver bandpass tilt, water absorption, etc.)

θ₀30 ° is the nominal beam Janus angle, and

θ is the actual beam Janus angle (at the transducer).

In some applications, it is assumed that the appropriate scaling factor for the doppler shift is determined by the speed of sound of the array, rather than the speed of sound of the diffuser, since in practice it is the array, rather than the diffuser, that moves relative to the water. According to this assumption, the Doppler shift f of one beam is measured_DComprises the following steps:

equation 16

Where u is the velocity component of x or y (parallel to the array surface), and w is the velocity component of z (perpendicular to the array surface).

The vertical velocity can be determined from the sum of the doppler shifts of the relative beams, with the u velocity component being accurately eliminated. However, the scaling factor depends on the speed of sound c and the centroid frequency f_c. If the scaling factor is not calculated correctly, the measurement of vertical velocity will be biased. The horizontal velocity is determined from the difference in doppler shift relative to the beam. If the centroid frequency f of the different beams_cIn contrast, the w velocity component will not be accurately eliminated. If this phenomenon cannot be properly explained, there is a deviation in the measured u, where measured u is approximately proportional to w instead of u.

Fig. 17 is a flowchart for explaining an example of a velocity processing method that substantially eliminates a deviation caused by a vertical component in a velocity estimation value. Depending on the embodiment, certain steps of the method may be deleted, combined together, or reordered. In an exemplary embodiment, the method is performed by the DSP 196 (see FIG. 11). The method is applicable to quadrature phase signals of returned acoustic energy received after transmission.

The method 280 begins at block 2802 where an autocorrelation function of the received orthogonal signals for each beam is calculated. As described above with reference to fig. 11, the received quadrature signal is passed from the sampling module 194 to the DSP 196. As described above, referring to fig. 11, the autocorrelation function is calculated by equation 7, except that now h can represent any lag time by an integer number of samples.

Next, at block 2804, the phase of the autocorrelation function for each beam of the received signal is calculated. For the autocorrelation result, the phase of beam n, can be calculated as follows:

φ_n＝tan^-1(I/R) equation 17

Where I and R are the imaginary and real parts of the complex autocorrelation result, respectively.

Moving to block 2806, the phase function of the received signal is extrapolated to a predetermined lag time (also referred to as a nominal lag time) for each beam according to equation 18 below. If a sample is at the nominal lag time, the phase for that sample input to equation 18 is simply the phase for that beam less than the phase offset to be transmitted. The nominal lag time depends on the emitted encoding pulse. In an exemplary embodiment, an encoding pulse as illustrated in fig. 4a, 4b and 4c is emitted. In this case, the predetermined lag time corresponds to the number of coded symbols prime in the first pulse.

Equation 18

Wherein:

T_Lis the nominal lag time of the time at which,

φ_n(T_L) Is the sampling phase at the nominal lag time for beam n,

φ_n，Tis the phase of the beam n to be transmitted, which is determined by the coded pulse to be transmitted (which, in the exemplary embodiment, is equal to zero),

f₀is the carrier frequency to be transmitted and,

τ₁，τ₂respectively the sample points immediately before and after the autocorrelation peak,

Δτ₁，Δτ₂respectively, from tau in lag time₁，τ₂Distance to nominal lag time.

The operation of this extrapolation will be described in further detail below with reference to fig. 18.

Next, at block 2808, a calculation is made of the nominal lag time u based on the phase function extrapolated at block 2806_raw(T_L)，v_raw(T_L) And w_raw(T_L) The preliminary velocity estimate of (a).

Equation 19

u_raw(T_L)＝u₁(T_L)-u₂(T_L) Equation 20

v_raw(T_L)＝u₄(T_L)-u₃(T_L) Equation 21

Equation 22

Equation 23

Wherein

d is the spacing of the elements of the array,

e_raw(T_L) Is an error estimate of the preliminary velocity indicating the quality of the velocity estimate; it may optionally include or exclude the error estimate in the calculation.

Moving to block 2810, a plurality of correction factors are determined based on the speed of sound and the centroid frequency shift. Estimating the centroid frequency f of a beam n as follows_n，low：

Equation 24

Equation 25

A plurality of correction factors are then calculated:

equation 26

Equation 27

Next, at block 2812, the preliminary velocity estimate is corrected based on the correction factor such that deviations caused by the vertical velocity component are substantially removed. This vertical velocity component w is first corrected as follows:

equation 28

The horizontal velocity estimation values u and v are then corrected. Calculating the error velocity estimate e is optional and in some embodiments excluded.

Equation 29

Equation 30

Equation 31

When at least four beams are correctly received, equations 28-30 above may be employed. When only three beams are correctly received, the preliminary velocity estimate is corrected based on the following correction factors:

equation 32

Equation 33

Equation 34

FIGS. 18a and 18b illustrate the phase function of the signal to be received in FIG. 17 according to equation 15Operation to push to the nominal lag time of each beam. Fig. 18a is a two-dimensional graph of the autocorrelation function. The vertical axis represents the magnitude of the autocorrelation between samples, while the horizontal axis represents the lag time between samples. As illustrated, τ₁，τ₂Are the sample points immediately before and after the autocorrelation peak, respectively. Fig. 18b is a two-dimensional graph of the sampling phase of the autocorrelation between samples. The vertical axis represents the phase of the sampling and the horizontal axis represents the time delay t between samples. It should be noted that the phase phi of the samples in figure 18b has been adjusted_n(t) is such that phi_n，TI.e. the phase of the beam n to be transmitted has been removed. The extrapolation operation is simply to draw a junction a (representing τ)₁Phase of the sample) and point b (representing τ)₂Phase of the sample) and finds the intersection c of the lines a-b and is defined by the function T ═ T_LThe straight line depicted (where t represents the lag time). The sampling phase of point c is phi (T)_L)。

Removing side lobe errors

The velocity estimates produced by the velocity processing methods described above tend to be higher than ideal side lobes. The higher side lobes than ideal are due to cross-coupling between the beams, which produces a velocity that depends on the bias of the velocity estimate.

The cross-coupling mechanism between beams can be understood from the following description. When multiple beams are transmitted simultaneously, each beam transmits power along its axis and receives this energy backscattered by the water or bottom suspension material. However, it also receives energy from other beam directions as a result of backscatter of the energy transmitted along the beams. Although the energy from other beam directions is often reduced by the beam pattern of the receive beam, the energy is still a serious bias in some applications. Certain embodiments described below disclose a method of removing cross-coupled side lobe errors, i.e., errors due to cross-coupling between beams.

FIG. 19 illustrates one embodiment of a velocity processing method that substantially removes cross-coupled side lobe errors in the velocity estimates. This embodiment can be applied to different types of transducers, such as pistonic transducers and phased array transducers.

The method begins at block 1902 with a transducer (e.g., a phased array transducer or a set of pistonic transducers) transmitting a signal for each beam that includes N pulsed signals (N is a predetermined integer and N > 1). In one example, N is equal to 4. Here, a pulse signal refers to an encoded pulse further comprising one or more encoded sequences. Each coding sequence includes one or more coding elements. In an exemplary embodiment, the signals to be transmitted are designed such that the factors of side lobe cross-coupling between any two beams of all N pulsed signals add together to cancel each other out. In some embodiments, the pulse signals of each beam include coded sequences having substantially the same length.

Next, at block 1904, a velocity estimate is obtained for each pulse signal based on the echoes of the pulse signal. Since there are N pulse signals, N sets of velocity estimates are obtained. Each velocity estimate includes a deviation from a side lobe coupling between a pair of beams. As discussed with reference to fig. 11, the phase of the autocorrelation function between the echo and the echo delayed by a predetermined lag time is first calculated based on equation 7. The velocity estimate is then obtained from the phase based on equations 1 and 8.

In some embodiments, a velocity estimate is obtained for each pulse signal within each beam. For example, when there are 4 beams each including 4 pulse signals, a total of 16 velocity estimates are obtained.

In some embodiments, when a pulse signal is transmitted, the pulse signal includes increments/decrements from one code sequence to the next. When the phase of the autocorrelation function between the echo and the echo delayed by a predetermined lag time is calculated, it is necessary to remove the phase increment/decrement to be transmitted. In one embodiment, the removal of the phase to be transmitted is achieved by subtracting or adding one quarter of the phase period to the autocorrelation function before calculating the final velocity. For example, if the phase to be transmitted is increased by 90 ° between sequences in the pulse signal (as depicted in the second half of pulse signal 2 of fig. 22), then one-quarter of the period of ambiguity of the phase needs to be subtracted from the autocorrelation phase.

Moving to block 1906, a velocity is calculated based on the sum of the N velocity estimates. For example, the speed is calculated by averaging the N speed estimates. By summing the N velocity estimates, the bias with respect to sidelobe cross-coupling between any two beams is substantially removed from the velocity.

The above method is designed to remove the bias from velocity with respect to sidelobe cross-coupling between any two beams. In some applications that allow for low precision velocity estimates, a revised approach may be used. This revised approach is designed to remove only the bias with respect to cross-coupling between beams in the same plane, i.e., X-Z (beams 1 and 2) or Y-Z (beams 3 and 4), instead of cross-coupling between any two beams. In a revised approach, the signals to be transmitted are designed such that the cross-coupling factors between beams in the same plane for all of the N pulsed signals cancel each other when added together.

The above description illustrates how a velocity is determined. However, the method may be extended to determine multiple speeds. In this configuration, the process of blocks 1904 and 1906 is repeated for each speed.

Exemplary signals to be transmitted in a speed processing method

Fig. 20a, 20b and 20c show three examples of encoding pulses used in the velocity processing method. Each graph includes three different representations of a coding sequence, where the coding sequence includes one or more coding elements. The representation of these coding sequences is similar to that of FIGS. 4a, 4b, and 4 c. The phase encoding definition of the code sequence is illustrated in numerical form. "0" means that no signal is transmitted in this period, and "1", "2", "3" and "4" represent transmission of carrier signals with a particular phase shift.

Fig. 20a shows a biphone code sequence with 90 ° elements. Has two diphone coding sequences: diphone I and diphone II. Diphone I is the illustrated diphone coding sequence with 90 ° elements. By 90 ° elements it is meant that the second and fourth encoding elements each have a 90 ° phase. Diphone II refers to a diphone code sequence having-90 ° elements, which is identical to diphone I, except that the second and fourth code elements in diphone II each have-90 ° phase.

Fig. 20b shows a pentatonic coded sequence with 90 ° elements. Similar to the coded diphone classification, there are two pentatonic coding sequences: fifth-degree sound I and fifth-degree sound II. The pentatonic I is the illustrated pentatonic coding sequence with 90 ° elements. In this case, the fourth encoding element has a phase of 90 °. The pentatonic II refers to a pentatonic coding sequence having-90 ° elements, which is the same as the pentatonic I except that the fourth coding element in the pentatonic II has-90 ° phase. Fig. 20c shows a barker code sequence, which is well known in the literature.

Fig. 21 is a table illustrating an example of a set of signal codes that may be transmitted by the method of fig. 19. This example is illustrated with a phased array transducer that produces the four beams shown in fig. 8, although it may be implemented with other types of transducers.

A signal having four pulse signals for each beam is transmitted. Each pulse signal is composed of one or more coded sequences. The number of code sequences per pulse signal is selected to meet the distance resolution requirement.

The signals transmitted for each pulse signal of each beam are shown in fig. 21. For example, fig. 21 shows "0 ° biphone I" for pulse signal 1 of beam 1. The diphone I indicates the type of code sequence to be transmitted and 0 ° indicates the phase increment between successive code sequences to be transmitted. Thus, a signal of the pulse signal 1 for the beam 1 is transmitted, which signal comprises a plurality of biphone I-coded sequences.

For the pulse signal 2 of beam 1, fig. 21 shows that "90 ° fifths I" are transmitted. The 90 ° indication has a 90 ° phase increment between a given pentatonic I code sequence and the next, as will be further described with reference to fig. 22.

Fig. 21 also illustrates side lobe coupling between beams of each pulse signal in the last column. For example, for pulse signal 1, the sidelobe coupling between beam 1 and beam 2 is represented by a 0 ° phase difference, indicating a sidelobe coupling factor of 1. Similarly, a phase difference of 180 ° indicates a sidelobe coupling factor of-1. The scheme is designed such that the sidelobe coupling factors between any two beams of all 4 pulse signals cancel each other out when summed. For example, for the side lobe coupling between beam 1 and beam 2 shown at the far left of the three columns, the coupling factors for all four pulse signals are 1, -1, 1, -1, respectively, with the sum equal to zero.

The scheme illustrated in fig. 21 can be varied in many different ways. The order of the pulse signals, each represented by a row, may be rearranged. For each pulse signal, the signals transmitted on the two beams in the same plane (plane 1-2 or plane 3-4) can be switched. In some embodiments, different codes may be used as an alternative to bark coding sequences.

It should be noted that other types of coded pulses following the above-described operating principles may be used for the speed measurement method.

Fig. 22 illustrates a form of signal coding, wherein the signal coding is associated with the pulse signals 1-4 of beam 1, the pulse signals 1-4 adopting the structure of the pulse signals shown in fig. 21. For example, the second code sequence in the pulse signal 2 of beam 1 is shown as "five degree tone I90 °" which represents a five degree tone I code sequence with a 90 ° phase shift.

The pulse signal 1 comprises a plurality of diphone I-code sequences. The pulse signal 2 has a plurality of pentatonic I code sequences. Each code sequence is transmitted with a phase shift. The first coded sequence of pulse signals 2 is transmitted with no phase shift and the second coded sequence with a 90 ° phase shift. The pulse signals 3 and 4 are phase shifted by 180 ° or 90 °, respectively, as shown in fig. 22.

Fig. 23a and 23b illustrate two examples of a scheme to produce 90 ° phase increase/decrease between successive code sequences using a phased array transducer. This scheme shows, for example, how the pulse signal 2 of figure 22 for beams 1 and 2 is generated. The pulse signal 2 of beam 1 comprises code sequences, wherein each code sequence has a phase increment of 90 ° on the basis of the previous code sequence. The pulse signal 2 of the beam 2 comprises code sequences, wherein each code sequence has a 90 ° phase reduction on the basis of the previous code sequence. The phase change of each beam as a function of time may be represented by a phase vector rotating in opposite directions.

In the table of fig. 23a, each row is four drive signals for a time corresponding to the start of a series of coded sequences, which drive signals are used for the elements of the phased array (see fig. 7). The drive signal for each beam is represented by a vector having a unit amplitude and phase as shown. The drive signals for the two beams are represented by vectors of magnitude. Each row shows at time T_0，S0-T_0，S3Applied drive signal, wherein the time T_0，S0-T_0，S3The start times of the coding sequences 0-3, respectively. As illustrated, the array elements to which the four drive signals are applied are half a wavelength from each other. Each column represents one drive signal for one array element at the beginning of four consecutive code sequences.

As discussed in connection with fig. 7, to produce beam 1 (the beam on the right), the drive signals are configured to decrease in phase by 90 ° from one signal to the next along the X direction. Similarly, to produce beam 2 (the beam on the left), the drive signals are configured to increase in phase by 90 ° from one signal to the next along the X direction. As discussed in connection with fig. 7, the beamformer simultaneously generates two beams by summing together the drive signals required to generate each beam. The driving signals for generating the two beams are illustrated in the table at the bottom of fig. 23 a. The "-1" in the table represents a vector of unit amplitudes and 180 ° phase. It should be noted that the vector of drive signals that produce the two beams is the vector shown in the table multiplied by a factor. As shown, some of the drive signals for the two beams have null outputs. This can be problematic for high power transmission using transducer arrays due to cavitation in the water at the face of the transducer array. See fig. 23b for an alternative method in which the method distributes power evenly across the array.

Fig. 23b illustrates another example of a scheme to produce 90 phase increments/decrements between successive encoded sequences, where the example can distribute power evenly across the transducer array. By virtually adding 45 ° and-45 ° spatial phase shifts to the drive signals of beam 1 and beam 2, respectively, in fig. 23b, uniform power distribution across the transducer array can be achieved.

By using a spatial phase shift control signal, a phase shift between the coded sequences of drive signals can be achieved (see fig. 7). For sequence 0, switches A and B are at the 0 setting. For sequence 1 an additional 180 deg. phase shift needs to be applied to the first and third drive signals in the X-direction. Therefore, switches a and B are to be in the 0 ° and 180 ° settings, respectively. Similarly, for sequence 2, switches A and B are both at the 180 setting. The phase shift in fig. 23a can be achieved following the same principles of operation.

As explained above, by reversing the polarity of the drive signal on successive code sequences, an increase/decrease of the phase between one code sequence and the next can be achieved. The drive signals can be divided into two groups: group I (first and third) and group II (second and fourth). Reversing the polarity of one set may produce a ± 90 ° phase shift in the half wavelength direction of each beam. Reversing the polarity of the two sets may produce a 180 ° phase shift along the direction of the two beams.

Conclusion

The velocity processing methods described herein may be used to measure various velocities depending on the particular application. Some examples include, but are not limited to, measuring the velocity of a vehicle or vessel relative to the bottom or surface of a fluid, measuring the velocity of an air flow in an air medium, and measuring the velocity of a target (e.g., in radar applications).

Further background information on the present invention may be found in U.S. Pat. Nos. 5483499 and 5808967, each of which is incorporated herein by reference in its entirety.

The foregoing description details certain embodiments of the invention. It will be appreciated, however, that no matter how detailed the foregoing appears in text, the invention can be practiced in many ways. It should be noted that no particular term used when describing certain features or aspects of the invention should be taken to imply that the term is again defined herein to be restricted to including any specific feature of the feature or aspect of the invention with which that term is associated.

Claims (12)

1. A method of measuring velocity in a fluid medium using a transducer, the method comprising:

transmitting an acoustic signal, wherein the acoustic signal comprises N pulsed signals for each of a plurality of beams, where N is an integer and N >1, wherein the beams are transmitted simultaneously;

for each beam, receiving echoes from each pulse signal, the each pulse signal including energy from the pulse signal for at least one other beam;

obtaining a velocity estimate for each of the N pulse signals based on the echoes of the pulse signals; and

the velocity is calculated based on a sum of the N velocity estimates such that there is substantially no error in the velocity due to receiving energy from the pulsed signal for at least one other beam.

2. The method of claim 1, wherein the transducer comprises a plurality of piston transducers.

3. The method of claim 1, wherein the transducer comprises a phased array transducer comprising a plurality of transducer elements arranged to form a single two-dimensional array.

4. The method of claim 1, wherein the measured velocity is the velocity of water flow in the fluid medium.

5. The method of claim 1, wherein the measured velocity is a velocity of the vehicle or vessel relative to the bottom or surface of the fluid medium.

6. The method of claim 1, wherein the measured velocity is a velocity of the target.

7. The method of claim 1, wherein the transmitted signals are selected such that, for each pair of beams, a sum of N cross-coupling factors between the pair of beams is substantially 0, wherein each cross-coupling factor corresponds to a pulse signal.

8. The method of claim 3 wherein the phased array transducer is configured to form at least a first beam plane and a second beam plane, wherein the transmitted signals are selected such that a sum of N cross-coupling factors between each pair of beams generated within the same plane is substantially 0, wherein each cross-coupling factor corresponds to an impulse signal.

9. The method of claim 1, wherein the plurality of beams comprises at least 4 beams, where N is an integer no less than 4, wherein the signals to be transmitted comprise signals shown in the following table:

10. the method of claim 9, wherein the transducer comprises a phased array transducer configured to produce a first beam plane and a second beam plane, wherein one of the planes comprises beams 1 and 2 and the other plane comprises beams 3 and 4.

11. A system configured to measure velocity, comprising:

a transducer for transmitting an acoustic signal and receiving echoes of each pulsed signal, wherein the acoustic signal comprises pulsed signals for each of a plurality of simultaneously transmitted beams, N being an integer and N >1, the each pulsed signal comprising energy from a pulsed signal for at least one other simultaneously transmitted beam; and

a processing module configured to obtain a velocity estimate for each of the N pulsed signals based on echoes of the pulsed signals, and to calculate a velocity based on a sum of the N velocity estimates to substantially eliminate errors due to receiving energy from the pulsed signals from beams for at least one other simultaneous transmission.

12. A system configured to measure velocity, comprising:

means for transmitting an acoustic signal, wherein the acoustic signal comprises a pulsed signal for each of a plurality of simultaneously transmitted beams, where N is an integer and N > 1;

means for receiving echoes from each pulsed signal comprising energy of pulsed signals from beams for at least one other simultaneous transmission;

means for obtaining a velocity estimate for each of the N pulse signals based on the echoes of the pulse signals; and

means for calculating a velocity based on a sum of the N velocity estimates such that there is substantially no error in the velocity due to energy received from the pulsed signal for at least one other simultaneously transmitted beam.