HK1141085B - Multichannel absorberless near field measurement system - Google Patents

Description

Multichannel absorber-free near-field measurement system

Technical Field

The invention relates to the measurement, testing and inspection of performance parameters of electromagnetic radiation equipment.

Background

The performance parameters of the electromagnetic radiation device include Effective Isotropic Radiated Power (EIRP) and Effective Radiated Power (ERP), radiation pattern, directivity, RF current distribution of the mounting surface, and magnetic near-field distribution. These radiating devices may include multi-mode, multi-band, or multiple-input/multiple-output (MIMO) radiating devices, such as cellular telephones, as well as wireless transceivers including WiFi devices (WiFi gears), as well as wireless PDAs and laptop computers.

When manufacturing cellular phones or other radiating devices, they have to be calibrated in order to transmit a known RF power (galvanic power) from the transmitter to the antenna structure and in order to radiate a known RF power (EIRP/ERP) from the antenna structure. Power measurements and checks must also be performed at various levels throughout the operating range of the radiating device. Such measurements and checks ensure that, for a given EIRP/ERP limit, the highest power transmitted to and from the antenna results in a legal and acceptable Specific Absorption Rate (SAR). In addition, power measurement and verification helps to maintain an efficient radio connection in cellular communications while minimizing power usage, thereby extending battery life and maximizing cell sector coverage and capacity.

Conventionally, samples of each cell phone model that will be released into the retail market are tested in the laboratory for a maximum EIRP/ERP level for several hours with a considerable measurement uncertainty of greater than 2.0 dB. Before performing such a test, the galvanic power of the cellular phone has to be calibrated and the cellular phone is set to radiate at maximum galvanic power.

Conventionally, cellular telephone RF power is delivered to cellular telephone test equipment using a physical hard-wired connector just before the antenna portion of the RF circuit, and is regulated by a cable connection between the RF connector on the cellular telephone and the test equipment. Once the maximum EIRP/ERP level is adjusted to or found to meet regulatory limits for a given galvanic power, only SAR level measurements are performed for regulatory compliance.

In order to measure and verify the RF power of cellular telephones or other radiating devices having more than one antenna, as well as devices having MIMO architectures, manufacturers typically provide a single RF connector for each antenna circuit, as well as RF switches, filters, and impedance matching. Since the RF connector is located entirely before the RF switch, filter and matching circuit, the performance of each antenna circuit is unknown even after all manufacturing tests of the cellular telephone have been successfully completed using conventional methods.

In the execution of the SAR measurement, the maximum galvanic power level obtained in the first step is used as a starting level. If the dc power needs to be adjusted in order to meet SAR limits, this adjusted dc power level will be considered as the maximum power that can be fed to the antenna, and the EIRP/ERP level must then be re-evaluated.

Most manufactured cell phone (or radiating device) samples of the same model are calibrated using this new galvanic power level as the maximum power to the antenna. Once this maximum level is measured and verified, up to 20 intermediate power levels are set and measured throughout the dynamic range. To perform these measurements, a galvanic RF link is established using the cable between the cellular telephone RF connector and the test equipment. The RF connector of the cable for cellular telephone connection wears out over time and is replaced based on the estimated maximum number of interpolations in the manufacturing test cycle of all production units (which is usually very large). The production test is stopped and a new cable must be introduced and recalibrated before manufacturing testing can continue. This introduces delay and expense.

After each cell phone is measured and checked for proper galvanic RF power levels to reach legal EIRP/ERP and SAR levels, each cell phone is further tested for Tx and Rx performance. To perform this test, the cell phone is connected to the cell phone tester using the RF cable between its RF connector and the test equipment as described above. In most cases, RF power measurements and checks are made at one location and Tx/Rx parametric tests are made at another location. In the case where these tests are performed at different locations, the RF cable connected between the cellular phone and the test equipment must be frequently replaced with a new RF cable due to a large number of interpolations. Recalibration of the RF cable must be performed before continuing the manufacturing Tx/Rx parameter test of the cellular phone, which introduces further delay and expense.

These measurements are performed with galvanic RF connections during board level manufacturing or designer testing to optimize the RF parameters of the cellular phone. This method does not provide all the necessary measurements to know the integrity performance of the RF circuit.

During the design and development of radiating devices, designers typically make a series of iterations in order to improve the radiation performance of the antenna model(s) in order to achieve a larger usable range in both frequency and sensitivity while targeting low SAR levels and low galvanic RF power. Each time the radiation performance of a radiation device is measured, it is necessary to go to a test laboratory that can optimize the EIRP/ERP level through a series of measurements. There currently exists no tool for finding an accurate spatial distribution of RF radiation in the near field in order to minimize unwanted radiation. Designers rely on conventional testing methods for far field radiation patterns in a test laboratory, and then debug at the circuit board level, which is a very tedious and complex process.

To measure antenna properties such as radiation pattern, gain and directivity, accurate amplitude and phase data is collected using a near-field scanner and then equivalent far-field values are calculated using one of many transformations known and available in the art. To accurately estimate the far field, those skilled in the art consider that the measurement distance between the probe and the antenna under test should be greater than or equal to one wavelength. Current near field testing is performed using a mechanical scanner with a single compensation probe that can detect both polarizations. Typically these measurements take many hours in order to complete the scan of the entire radiation surface.

When measuring near field radiation, the array elements and the conductive planes and the dielectric surrounding them greatly influence the near field distribution of the radiation source and its far field properties. In the prior art, using multi-axis near-field measurement systems, measurements are performed at more than one wavelength from the antenna under test in order to minimize the effect of the ground plane, which is then relatively easy to consider. The array sensitivity is reduced and the dynamic range of the measurement is limited. In addition, the measurement speed and physical size make this approach impractical in high-speed production testing environments or in traditional development laboratories where real-time feedback and efficient use of physical laboratory space are highly appreciated.

In another approach, perfect near-field absorbers such as those described in U.S. patent No. 6,762,726B2, issued on 7/13/2004, are used in order to increase the isolation between the radiating surface and the array surface, thereby reducing the mutual coupling effect that distorts the measured field strength of the electromagnetic radiation emitted from the signal-emitting circuit. The array sensitivity is significantly reduced and the dynamic range of the measurement is limited. In addition, the described probe density, as well as the required attributes and performance of the increased physical absorber solution, add significant complexity, challenges to sustainable yield, and expense to deploying a physically realizable solution. With the addition of a physical absorber, there is still an interaction between the radiation source and the absorber surface and a modified near field representation of the radiation source results.

There is a need in the art for a method and apparatus for measuring performance, such as EIRP and ERP from electromagnetic radiation devices, using near field measurement techniques that addresses the limitations of the referenced solutions.

Disclosure of Invention

The present invention comprises a novel multi-channel near-field scanning system for measuring performance parameters, such as EIRP/ERP, and generating a far-field map of an electromagnetic radiation device through a series of input power levels. Preferably, the scanning system is also transparent to accurate and repeatable measurements of round trip Tx/Rx performance. In at least one embodiment, the system can be implemented without a galvanic RF connection. A radiating device such as a multi-mode, multi-band, or MIMO (or combination thereof) mobile or cellular telephone is placed on the limited area scanner at a distance equal to or less than about 1/1.8 of the wavelength of the operating frequency of the radiation source. Preferably, the distance is from about 1/1.8 to about 1/88 wavelengths in the frequency range of about 8GHz to about 170 MHz. A multi-channel electromagnetic scan is performed in real time using an electrically switched array of probes, and the near-field amplitude and phase of both the x-and y-components of the radiation source are measured, corrected, re-measured and displayed. Using the corrected near-field data, far-field transformation, and radiation source model, performance parameters of the radiation device, such as EIRP/ERP, directionality, and radiation pattern, are estimated and displayed.

Embodiments of the present invention may be used to perform rapid testing in a production environment, or to determine characteristics of radiating devices, to measure RF current distribution on the mounting surface of an antenna, to improve RF circuitry, to debug and locate faulty antennas or sub-arrays or arrays, and to optimize antenna performance, due to their real-time scanning speed and accurate near-field and far-field measurement capabilities.

Embodiments of the present invention can also measure the Tx/Rx performance of the radiation source without the need for galvanic RF connections. The radiating device, which may be a multi-mode and/or multi-band and/or MIMO mobile or cellular phone, is placed on the scanner at a distance of less than one wavelength, preferably at 1/1.8-1/88 of one wavelength of the operating frequency of the radiation source. Two different and optimal RF channels of the near-field scanner are selected and assigned to Tx and Rx modes of the transceiver. Tx and Rx performance of the transceiver is estimated using external test equipment.

Accordingly, in one aspect, the invention may comprise an absorber-free multi-channel near-field microwave scanning system comprising:

(a) a switched array of dielectric-embedded antenna elements for sensing electromagnetic field components at predetermined locations and forming an array surface, wherein the array outputs a raw uncorrected signal, the raw uncorrected signal being representative of the electromagnetic field and including mutual coupling effects;

(b) a scan surface for placement of a Device Under Test (DUT), wherein the scan surface is generally parallel to the array surface and separated by a distance of less than about 1/1.8 of a wavelength of a measured frequency;

(c) a processing engine operatively connected to the switched antenna array for obtaining and processing array outputs, said processing engine adapted to correct for mutual coupling effects at the individual probe level.

In one embodiment, the mutual coupling effects include reflection and dynamic coupling effects between individual antenna elements on the array, as well as effects of proximity of the array to the DUT. In addition, the finite scanner size may also affect the far-field transform and is considered in the processing engine.

In one embodiment, a processing engine comprises:

i. a controller for controlling the operation of the electronic device,

a channel selector and a sampler,

a channel corrector for fine tuning for differential path loss and delay;

iV. data converter and interpolator;

v. an amplitude and phase detector;

near field corrector for correcting at single probe level for reflections and dynamic coupling between individual antenna elements on an array

A transformer for transforming near field data into far field patterns and design performance parameters, an

A user interface.

In another aspect, the invention may include an absorberless method of measuring EIRP/ERP or Tx/Rx performance of RF and microwave transceivers, the method comprising the steps of:

(a) a switching array using antenna elements forming an array surface;

(b) using a scanning surface, wherein the scanning surface is substantially parallel to the array surface and separated by a distance of less than about one-half of a wavelength of the measured frequency;

(c) generating near field data by receiving an output from each antenna, the near field data representing an electromagnetic field but including mutual coupling effects and effects due to limited scanner size;

(d) correcting the near field data to correct for reflection and mutual coupling effects at the single probe level on the array; and

(e) near field data is transformed into far field data.

In one embodiment, the mutual coupling effects include reflection and dynamic coupling effects between individual antenna elements on the array, as well as effects of proximity of the array to the DUT.

Drawings

The invention will now be described by way of example embodiments with reference to the accompanying simplified, diagrammatic, not-to-scale drawings. In these drawings:

fig. 1 is a schematic diagram of an antenna array of a scanner;

FIG. 2 is a side view of an antenna array and scan plane;

FIG. 3 shows an alternative arrangement of a half-loop antenna array;

FIG. 4 shows a two-layer switch array;

fig. 5A shows a schematic diagram of a processing engine, and fig. 5B shows a schematic diagram of a controller function. FIG. 5C shows a schematic flow chart depicting near field correction;

fig. 6 shows a schematic diagram of the external field of a radiating antenna;

FIG. 7 shows a schematic of the geometry of a planar near field measurement;

8A-8E illustrate different screen shots of various displays produced by a graphical user interface of a processing engine.

Detailed Description

The invention provides a method and apparatus for measuring the radiation power of a radiation source in the near field. When describing the present invention, all terms not defined herein have meanings generally recognized in the art. When the term "about" is used in connection with a numerical value, it is meant that the value includes a range of 10% above and below the stated value, or within a known tolerance of the method by which the value is measured. The term "near field" means a field within a distance from the antenna of less than or equal to about one wavelength of the radiated radio frequency. Where permitted, the references listed herein are incorporated by reference as if reproduced in full.

The present invention includes an absorber-free microwave near-field scanner. In one embodiment, a scanner (100) includes a plurality of antennas (101) arranged in a two-dimensional array capable of transmitting and receiving electromagnetic radiation. The antenna is preferably, but not necessarily, a half-loop antenna. As shown in fig. 1, the array may have m elements in the x-axis and n elements in the y-axis. In one embodiment, the loop size length (L) and depth are optimized to provide sufficient discrimination between H and E field strength. For a given scan area and radiation power accuracy, the inter-element spacing (d) and the total number of array elements are determined. In one embodiment, m may be 24 and n may be 16, and d may be equal to about 10 mm. In one embodiment, d may be equal to about 5L. A higher number of antennas (i.e. a value smaller than d) in a given area will provide better accuracy, however, at the expense of an increased mutual coupling effect between the antennas and their feed structures.

As shown in FIG. 2A, the scan plane (102) is positioned at a distance (D) from the array surface (103), the distance (D) preferably having a range of about 1/88 to about 1/1.8 of the wavelength, and the corresponding inter-array element distance (D) preferably ranges from about 1/176 to about 1/3.6 of the wavelength. If D brings the scan plane too close to the array surface, as shown in FIG. 6, the array surface may be in the absolute reactive near field region, which has a negative effect. However, as D becomes larger, the size of the array must be increased in order to obtain the same scan energy. In one embodiment, D/D may be approximately 2.0.

The illustrated embodiment shows a generally flat scan plane and antenna array that are generally parallel to each other. Alternative embodiments may include spherical, cylindrical, or other geometric shaped scanning surfaces.

The typical arrangement of the half-rings (101) is such that successive array elements transmit or receive orthogonal polarizations of the H field strength. This scanner or array may also use alternative layout arrangements, including those shown in fig. 3.

The output from the array antenna is fed through the backplane to an auxiliary surface of a multilayer Printed Circuit Board (PCB). The PCB layer stack and layout should preferably be able to provide better than 20dB of inter-component isolation over the frequency range of interest. In one embodiment, one end of the half-loop antenna (101) is connected to a ground plane next to the antenna layer and the other end of the half-loop antenna is connected to the microstrip line layer through a feed via without matching.

The output from a particular antenna (101) is selected by means of a switch (110) which can select the output from any one of the antenna elements (101). Since a large number of antenna elements makes it difficult to implement a single switch for each antenna, one embodiment of the present invention includes a system that allows for hierarchical switching using a relatively small number of switches. In one example, using a 3-tier SP4T switch, the number of signals can be reduced to 1/64. Thus, the 384 element array can be reduced to 6 RF outputs. In fig. 4 is a 16 element module with two layers of switches.

Along with the switch matrix, the second channel is connected to an antenna element to provide a reference signal. This reference signal is necessary for making relative phase measurements. The antenna array is constructed so that it can be extended to simultaneously make radiated power measurements of more than one radiating surface or device by appropriately selecting a pair of channels.

The selected and unselected antennas should preferably be sufficiently isolated from each other. Poor isolation is typically due to leakage in the cavity formed by the abutting ground planes, so that the antenna feed via extends conduction of antenna current through the inner layer to the component layer, making the feed via an effective radiator.

Measurements and simulations of antenna isolation in two-layer boards show very good isolation. As a result, we believe that coupling is not due to the antenna structure and that only one ground plane does not occur. Significant leakage was observed when the simulation model was changed to include more than two ground planes. Electrical energy will flow with very little attenuation from one via to the next within the substrate between the ground layers.

Coaxial feed can theoretically provide excellent isolation, however, coaxial feed can be difficult to manufacture. A more practical solution may be realized with ground vias or ground strips. Thus, in one embodiment, the PCB includes isolation devices consisting of ground vias (or ground strips) that connect all ground layers together. We have found that locating the ground strip closer to the feed-through produces better isolation and that using multiple ground strips also produces better isolation.

The processing engine accepts the antenna signals (referred to herein as near field data) from the scanner PCB and processes them to provide useful information. The antenna signals include mutual coupling effects such as reflections and dynamic coupling between individual antenna elements on the array, as well as the proximity of the Device Under Test (DUT) to the array, and effects related to both the physical and virtual size of the finite scanner. Thus, in one embodiment, the processing engine provides a means for removing or minimizing mutual coupling effects at the individual probe level. The processing engine also accounts for the effects of proximity of the array near the DUT, and also accounts for limited virtual scanner size by transforming to the far field using a Plane Wave Spectroscopy (PWS) model.

In one embodiment, as shown schematically in fig. 5, the processing engine (10) includes a controller (12), channel selectors and samplers (14), a channel corrector (16) that fine-tunes the differential path loss and delay, a data converter and interpolator (18), an amplitude and phase detector (20), a near-field corrector (22), a transformer (24) for transforming near-field data into far-field data, and a user interface (26) including a graphics card or other means for driving a display. The processing engine may also include a post-processor (28) and means (30) for determining the EIRP. The traceability module (32) is optional. The components of the processing engine may be implemented by software, firmware, hardware, or a combination thereof, and are well known in the art.

As shown in fig. 5B, the controller (12) functions primarily to power the rest of the system and to control the switches and attenuators on the PCB. The controller (12) receives commands from an operating processor, which may be a desktop or laptop computer, and converts the data into signals required to operate the antenna panel and signal conditioning portion. The inputs of the control board are connected to the I/O on the computer. The input lines are used for data transmission of the state control signals as well as routing signals, which causes the appropriate state control to enter the appropriate output data lines.

In order to fully control the state of the antenna board and signal conditioning system and obtain accurate measurements within the required dynamic range, two sets of inputs are required. One bit on the input is dedicated to group selection of the input. The feedback and delay portions are necessary to handle the handshake requirements of the I/O card. This portion is also used to generate the CLK signal.

An ACK signal is sent from the I/O card and the REQ needs to be returned before the card will output the next set of data. The REQ signal must have some minimum delay and duration. This handshake requirement is met by a simple feedback and delay circuit. The REQ delay is introduced by having the ACK signal pass through two inverters implemented using NAND gates. The CLK signal is also introduced into the system using a two inverter approach.

Since there are many total output data lines required from the power supply and control board (38 lines in the embodiment shown), some form of demultiplexing or decoding is required and in the preferred embodiment, two strategies are used on the board. Suitable demultiplexing and decoding strategies are well known to those skilled in the art.

As is well known in the art, the user interface and display (22) may display data on a conventional computer monitor and accept user input through a computer keyboard and mouse. In one embodiment, the user interface is a Graphical User Interface (GUI) and the display framework is designed to provide flexibility in feeding test parameters, such as selection of scan regions, reference probes, scan types, model selection, frequency ranges, and loading data such as raw data, DAQ corrected data, probe corrected data, converted data, path corrected data, and reference far field data. Once all the test parameters are loaded, the GUI and display portion (22) of the processing engine (10) interprets the loaded test parameters and creates test sequences and, with the aid of the controller, begins execution of each test sequence while the scan data is measured/registered into the computer memory. Additionally or alternatively, the scan data may be written to a hard disk drive or other data storage device for further processing.

The scan data is then further processed to determine in real time at least one performance parameter such as near field distribution specific to the 2D and 3D components, total near field distribution, amplitude and phase distribution, far field map of the main slice and any desired slices, as well as ERP, EIRP and directionality.

In a two channel system, the channels are designated as reference channels and measurement channels, respectively. In one embodiment, the reference channel is connected to a unique element in the array, however it may also be reconfigured to connect to a different element in the array that is dynamically determined by the controller based on the scanned information or input parameters. In one embodiment, the system architecture enables a pair of antenna elements in the array to be selected and connected to both the reference and measurement channels.

In one embodiment, both the reference channel and the measurement input channel are mixed down to an Intermediate Frequency (IF). The IF signal is further amplified and processed by a bandpass filter. These filters will determine the frequency range of the IF, so in order to cover the full measurement frequency range, the Local Oscillator (LO) needs to be programmed in order to produce the correct IF range. The complete input frequency range is decomposed into N segments with a width equal to the IF filter bandwidth. Preferably, the LO is designed to cover only the frequency region of interest, i.e., the cellular band. For the reference channel, the logarithmic amplifier determines the peak or average peak amplitude. The slicer output from the logarithmic amplifier is passed through a comparator and fed into a counter which determines the signal frequency. An additional switchable attenuator is used after the amplifier on the measurement channel in order to increase the range of allowed input signal strengths. The RMS detector measures the amplitude of the measurement channel. Alternatively, the same detector may also be used to determine the peak amplitude. Using two detectors, the signal strength of received modulated RF energy having various modulation formats can be detected and measured.

For phase measurement, two phase detectors may be used. One directly inputs the reference and measurement channels from the IF filter, and the other has a 90 degree phase delay filter on the reference channel.

The microprocessor controls and reads the measured values from the associated a/D converter and counter. The microprocessor communicates with the processing engine to determine the input frequency band and other necessary information and transmits the signal measurements to the processing engine. To achieve the required accuracy, the a/D converter should preferably have a minimum resolution of 10 bits. The sampling rate is preferably at least 1MSPS, although faster sampling rates may reduce the time required to make all the required measurements while also allowing some averaging of the data.

The amplitude and phase measured by the RF sampler are in the raw state and various corrections are applied to them in order to create an accurate data set of scan planes. Initially, RF sampler amplitude and phase corrections are applied at a given frequency and for a given temperature. Subsequently, both amplitude and phase are corrected for path loss at a given frequency and for a given temperature. Finally, the corrected amplitude and phase are converted into field quantities by using antenna factor correction.

Because each element of the antenna array measures only one magnetic field component orthogonal to the magnetic field components of its neighboring elements, interpolation is applied to obtain two transverse components at each sampling point of the scan plane. For the amplitude, the interpolation is performed by averaging its 4 neighboring measurement points. For an edge element, the data is interpolated from its neighboring 3 elements. For corner elements, the data is interpolated from its neighboring 2 elements. In one embodiment, the phase interpolation may be achieved by a three-point method. First, 4 adjacent data points are ordered from smallest to largest. If the phase difference between the sorted adjacent data points is greater than a predetermined threshold, the most specific one is discarded and the remaining 3 points are averaged. Otherwise, 4 adjacent measurement points are averaged. Preferably, special treatments for the edge points and 4 corners may be used in order to obtain better results. Alternatively, extrapolation of interior points is employed for those points.

In the amplitude and phase detection module (20), after the raw data has undergone correction and interpolation stages, the amplitude and phase of the near field data is available for further processing, display and storage.

The methods described herein may be used to account for mutual coupling effects, which may include reflections between individual antenna elements on the array, dynamic coupling, and DUT proximity effects. In addition, the effect of limited scanner size is taken into account using methods known in the art. Calculations are performed to calculate the various models and their NF corrections. The far field radiation pattern and radiated power of an antenna can be measured and studied by measuring Near field radiation [ Johnson J.H.Wang, "analysis of the Theory and practice of Near-field measurements," IEEE transactions, antennas Propagat, Vol.36 pp.746-753, January 1986 ].

Fig. 6 shows the external field of a radiating antenna, which is generally divided into three regions: a reactive near field region, a radiating near field region, and a far field region. The reactive near field region is excited in a small volume just outside the antenna and results in electrical and magnetic energy stored around the antenna and decays very rapidly. The reactive Near field region is generally considered to extend approximately λ/2 π from the surface of the antenna, although conventional Near field Measurements use a distance of one wavelength (λ) or more in order to minimize system uncertainty [ Arthur D. Yaghjian, "An Overview of Near-field antenna Measurements," IEEE trans. antennas Propagat, Vol. AP-34 pp.30-45, January 1986 ].

Conventional scanning techniques for near-field measurements of antennas are based on publications found in whitetaker and Watson [ g.t. whitetaker and g.n. Watson, modern analysis, 4th ed.london: plane Wave Spectrum (PWS) representation of the field in Cambridge Unit v.Press, 1927, ch.XVIII ].

A planar near field measurement system is shown in fig. 7. The aperture of the radiation antenna is in the x-y plane with z less than or equal to 0. The plane for near-field measurement is inIn the x-y plane of (a). Considering that the region where z > 0 is passive, the solution of the time-harmonic electromagnetic field before the antenna aperture can be expressed as:

and satisfy k_xA_x(k_x，k_y)+k_yA_y(k_x，k_y)+k_zA_z(k_x，k_y)＝0 (3)

Wherein k is_xAnd k_yIs a real variable, and

k may be referred to as a wavenumber vector, and A (k)_x，k_y) Is called plane spectrum, because of the expression A (k)_x，k_y)e^-jk·rIn this expression, a uniform plane wave propagating in the direction k is represented.

Transforming and rearranging these equations to express PWS A (k) from the near field using the components H (x, y, z)_x，k_y)。

In The far field region of The antenna (kz > 1), based on The steepest descent method, it can be confirmed that equation (1) [ p.c. clemmow, The Plane Wave spectral representation of Electromagnetic fields.london: pergamon, 1966 ].

When performing a planar near field scan on a radiation surface, the scan must be limited to a limited area in the x-y plane for practical reasons and limitations. A planar spectral transform may be applied to this scan data to determine the far field properties of the radiating surface. The accuracy of the far-field transform data for a given frequency is limited by the limited area used for scanning. The data may be further processed in a post-processing module to improve accuracy.

Conventional radiated power measurements are performed in free space or in the presence of a large ground plane. Far-field data estimated using the PWS provides an estimate in free space. The data set is corrected, if necessary, to take into account the effects of ground plane interaction.

The calculations of the power density map or radiation map, directional gain, radiation power, and EIRP may be performed as follows:

Z²is taken as P_offsetRemoval of P_offsetOther coefficients are also contemplated. PDS → U in Matlab. The radiation power is obtained by integrating the power density over a hemisphere. The hemisphere was divided into 50X 100 parts. And, likewise, by focusing the power density over a hemisphereAccumulation is performed to perform integration.

This value can be obtained for one complete scan. If the scanning is continued one after the other, a quasi-real time curve can be provided.

In the current implementation, d θ is 1.8 ° and d Φ is 3.6 °

From the coordinates of a sphereThe power gain of the antenna in the specified direction is defined as:

wherein the intensity of the radiationIs defined as a directionThe upper "power radiated from the Antenna per unit solid angle" [ c.a. balanis, "Antenna Theory: analysis and design', Second Edition，John Wiley & Sons，1997]And P is_inIs the total power that the antenna accepts from the source. P is calculated from the voltage and current at the source as follows_in：

And is

In the direction of rOn the premise that R ═ R, E is obtained from equation (28). Directionality is simply defined as:

wherein P is_radIs the total power radiated by the antenna,

and P is_lossIs the total ohmic loss in the antenna.

If no direction is specified, the direction implying the maximum radiation intensity (maximum directionality) is expressed as

If it is assumed that the signal is radiated equally in all directions, i.e. spherical waves emanating from a point source, the Effective Isotropic Radiated Power (EIRP) is the apparent power transmitted towards the receiver. This power is given by the following equation:

EIRP＝G_t·P_t

＝D·P_rad

wherein:

G_twhich is the gain of the transmitter antenna,

P_tpower of transmission

Examples of the present invention

The following examples illustrate the proposed invention but do not limit it. Typical accuracies that are industrially achievable using far-field measurement techniques for gain and directivity are on the order of +/-0.25dB over the cellular telephone operating frequency range. To achieve traceability, a number of electromagnetic numerical simulations were performed to achieve similar far-field accuracy by implementing and adjusting the numerical model parameters of the reference source at predefined cellular telephone band frequencies. Using these simulations, the EIRP for the reference source was found to be 29.66dBm and 24.95dBm at 1880MHz and 836.4MHz, respectively, with an accuracy of +/-0.3 dB. Near field amplitude and phase accuracy at very close distances were estimated from near field data sets derived from far field simulations and found to be on the order of about 0.30dB and +/-5 degrees. The scanner system was calibrated with amplitude and phase data from the simulation with an amplitude of +/-0.3dB and a phase accuracy of +/-5 degrees using frequency and model sensitivity NF correction factors.

Fig. 8A shows a 3D near-field total amplitude distribution of the radiation device under test. This is the resultant amplitude of the x and y magnetic field strengths of the irradiation device measured by each probe positioned at a predetermined physical location.

Fig. 8B shows the 2D near-field amplitude distribution of the x and y components of the radiation device under test. This is the amplitude of the x and y components of the magnetic field strength of the irradiating device measured by each probe positioned at a predetermined physical location.

Fig. 8C shows the estimated values of EIRP, directivity and radiated power (shown in real time) of the radiating device. The radiated power is calculated from the corrected near-field amplitude and phase distribution and applying an appropriate near-field to far-field transformation. The directivity and the EIRP are also calculated from the radiation power of the radiation device and the calculated radiation pattern.

Fig. 8D shows a 3D hemispherical radiation pattern of the radiating device and is calculated after applying a near-field to far-field transformation to the corrected near-field amplitude and phase distribution.

FIG. 8E shows an integrated GUI that combines FIGS. 8A, 8B, 8C, and 8D. Any of these figures may be exaggerated to clearly illustrate the parameters displayed. The displays shown in fig. 8A and 8B may be interchanged by selecting the appropriate option in the menu bar. The upper right quadrant shows a polar representation of the radiation pattern in which a standard pattern of the device under test obtained from any test laboratory may be superimposed on the calculated radiation pattern of the scanner system.

As will be apparent to those skilled in the art, various modifications, adaptations, and variations of the specific disclosure set forth above may be made without departing from the scope of the invention as set forth herein. The various features and elements of the described invention may be combined in other combinations than those described or claimed herein without departing from the scope of the invention.

Claims (13)

1. An absorber-free near-field microwave scanning system, comprising:

(a) a switch array of dielectric-embedded antenna elements for sensing electromagnetic field components at predetermined locations and forming an array surface, wherein the switch array outputs a raw uncorrected signal that is representative of the electromagnetic field and that includes mutual coupling effects;

(b) a scanning surface for placing a Device Under Test (DUT), wherein the scanning surface is substantially parallel to the array surface, and the scanning surface and the array surface are separated by a distance that is less than 1/1.8 of a wavelength of a measured frequency;

(c) a processing engine connected to the switch array for obtaining and processing switch array outputs, the processing engine adapted to correct for mutual coupling effects at a single probe level, and the processing engine comprising:

i. a controller for controlling the operation of the electronic device,

a channel selector and a sampler coupled to the controller,

a channel corrector coupled to the channel selector and sampler for adjusting for differential path loss and delay,

a data converter and interpolator coupled to the channel corrector,

v. an amplitude and phase detector coupled to the data converter and interpolator,

a near field corrector coupled to the amplitude and phase detector,

a transformer coupled to the near field corrector for transforming near field data into far field data, and

a user interface coupled with the controller, the amplitude and phase detector and the transducer.

2. The system of claim 1, wherein the antenna element is included in an antenna layer of a multi-layer printed circuit board, the printed circuit board including a microstrip line layer and a ground layer immediately adjacent the antenna layer.

3. The system of claim 2, wherein the multi-layer printed circuit board includes a plurality of ground layers, and the selected and unselected antenna elements are isolated from each other by ground vias that connect all of the ground layers together.

4. The system of claim 1, wherein the distance (D) between the scanning surface and the array surface is between 1/88 for the wavelength and 1/1.8 of the wavelength.

5. A system as claimed in claim 1, 2 or 3, wherein the inter-antenna element distance (d) of the switch array ranges between 1/176 for the wavelength and 1/3.6 for the wavelength.

6. The system of claim 4, wherein the distance (D) between the scan surface and the array surface/distance (D) between antenna elements of the switch array is 2.0.

7. A method of measuring and testing performance parameters of an electromagnetic radiation apparatus, the method comprising the steps of:

(a) a switching array of dielectric-embedded antenna elements for sensing electromagnetic field components at predetermined locations and forming an array surface;

(b) placing a Device Under Test (DUT) on a scanning surface, the scanning surface being generally parallel to the array surface and the scanning surfaces being separated by a distance of 1/2 less than a wavelength of a measured frequency;

(c) generating near field data by receiving output from the switch array, the near field data being representative of an electromagnetic field but including mutual coupling effects;

(d) correcting the near field data to correct for mutual coupling effects at a single probe level; and

(e) the corrected near-field data is converted into far-field data.

8. The method of claim 7, wherein the antenna element is included in an antenna layer of a multi-layer printed circuit board, the printed circuit board including a microstrip line layer and a ground layer immediately adjacent the antenna layer.

9. The method of claim 8, wherein the multi-layer printed circuit board includes a plurality of ground layers, and the selected and unselected antenna elements are isolated from each other by ground vias that connect all of the ground layers together.

10. The method of claim 7, wherein the distance (D) between the scanning surface and the array surface is between 1/88 for the wavelength and 1/1.8 of the wavelength.

11. The method of claim 7, wherein the inter-antenna element distance (d) of the array ranges between 1/176 for the wavelength and 1/3.6 for the wavelength.

12. The method of claim 10 or 11, wherein the distance (D) between the scanning surface and the array surface/distance (D) between antenna elements of the switch array is 2.0.

13. The method of claim 7, wherein the mutual coupling effects targeted for correction include reflections and dynamic coupling effects between individual antenna elements on a switch array, and proximity of a Device Under Test (DUT) to a limited scanner surface.