HK1025461B - High frequency heating equipment - Google Patents

Description

High-frequency heating apparatus

Technical Field

The present invention relates to the field of high-frequency heating apparatuses, and more particularly, to dielectric heating using a magnetron such as a microwave oven. In particular, a circuit structure of a power supply unit for driving a magnetron is disclosed.

Background

A power supply unit is installed in the range of appliances including a high-frequency heating apparatus for home use. The conventional power supply unit is generally heavy and bulky, so that a small and light power supply unit is increasingly required. Accordingly, research and development of a small, lightweight, and inexpensive power supply unit employing a switching power supply is actively being conducted to be used in a wide range. There is also a need for a small and lightweight power supply unit for driving a magnetron in a high-frequency heating apparatus for cooking food using microwaves generated from the magnetron.

The switching power supply converts alternating current or direct current into alternating current having different frequencies and voltages or into direct current having different voltages. This can be achieved using semiconductor switching devices such as transistors and thyristors. In other words, the switching power supply is generally used for electric power conversion. Since the switching power supply operates the semiconductor switching device at a high frequency, reducing the switching loss is a key technical focus. In particular, a switching power supply unit used in a high-frequency heating apparatus for home use can convert electric power at a rate higher than 1 kW. It is also important to reduce switching losses with respect to energy conversion.

Therefore, the structure employing the resonance circuit system in the high-frequency heating apparatus is advantageous in reducing the switching loss. This circuit system is called a single switching element voltage resonance circuit, and it is a system designed to reduce the influence of switching loss by using the resonance circuit, and to alleviate the voltage skew by applying a voltage of a sinusoidal waveform to the semiconductor switching device.

However, the single switching element voltage resonance circuit of the related art has the following disadvantages.

First, the voltage applied to the semiconductor switching device rises due to the influence of the resonant circuit. This may require that the semiconductor switching device or related electrical components be configured to have a higher withstand voltage, resulting in a large and expensive power supply unit.

Second, although the ON time of the semiconductor switching device can be set, the OFF time is a function of the operation of the resonant circuit and cannot be adjusted as desired. This reduces the flexibility of control of the single switching element voltage resonant circuit. This causes a lot of inconvenience in that it will be described below while describing in detail the single switching element voltage resonance circuit of the prior art.

Fig. 21 shows a circuit diagram of a power supply unit for driving a magnetron in a high-frequency heating apparatus of the related art.

Fig. 21 shows a power supply circuit for driving a magnetron supplied with alternating current. Looking at fig. 21 from the left, a full wave rectifier converts an ac power supply from an ac voltage to a dc voltage and a voltage V_DCAdded to the circuit in which the semiconductor switching device is connected in series with a parallel circuit comprising a capacitor and a leakage transformer (transformer with a magnetic connection between the primary and secondary coils and the tertiary coil below 1 due to the leakage flux). This semiconductor switching device operates at high frequencies. Here, an IGBT (insulated-gate bipolar transistor) is used as the semiconductor switching device. The leakage transformer and the parallel capacitor form a resonant circuit.

If the drive signal V is to be driven_GAnd is applied to the gate of the IGBT to turn on the IGBT, so that the current I flows to the IGBT and passes through the primary coil of the leakage transformer. This is the period T of the waveform shown in FIG. 22A₁. When at time T_ONWhen the IGBT is then turned off, current begins to flow to the capacitor and resonance is established. This is the period T₂. Fig. 22B shows the waveform of the drive signal for the IGBT. Can convert the energy W of the leakage transformer_LIs defined as:

W_L＝(L1²) /2 (equation 1)

Where I is the current and L is the inductance of the leakage transformer.

The current I can be defined as:

I＝V_DCT_ON/L (equation 2)

Wherein V_DCIs the voltage of the capacitor, i.e. the dc supply voltage.

When resonance begins, the energy is transferred to the capacitor, establishing the following equation:

W_L＝(CV²)/2+W_MG(equation 3)

Where C is the capacitance of the capacitor, V is the voltage of the capacitor, W_MGIs the energy dissipated in the rectifier and magnetron connected to the secondary winding of the leakage transformer.

After energy is transferred to the capacitor, energy starts to be supplied from the capacitor to the leakage transformer, and resonance continues while the period T is the same₃Shown therein, produce attenuation. In order to maintain stable resonance, it is desirable to replace the energy consumed by the magnetron. Therefore, VG is applied to the IGBT gate to turn on the IGBT again for a period T₄The secondary coil is supplied with energy. The resonant circuit is characterized by a voltage V between the collector and emitter of the IGBT_CETurning on the IGBT again reduces switching losses when going down to zero. FIG. 22C shows the primary coil voltage waveform V_PThe resonant waveform of (1).

V of IGBT can be converted_CEIs defined as:

V_CE＝V_DC-V_P(equation 4)

Wherein V_PIs the voltage of the primary coil.

Thus, due to the influence of resonance, V_CEHas a high voltage peak as shown in fig. 22D. The circuit constant of the magnetron and the total amount of energy supplied to the leakage transformer in the circuit shown in FIG. 21 are determined by the capacitor, the leakage transformer, the rectifier connected to the secondary coil, and the circuit constant of the magnetron₂To T₃Time T of_OFF. It is composed ofMiddle V_P≥V_DCPeriod T of₃For permission V_CEIt is desirable to drop to zero or lower. In the period T₄In which the IGBT is turned on again to replace the energy consumed by the magnetron, thereby allowing a stable resonance to be established.

By the ON time T of the IGBT_ONDetermining the energy supplied to the leakage transformer and a shorter ON time T_ONInto a smaller amount of power. The drive frequency f of the IGBT can be defined:

f＝1/(T_ON+T_OFF)。

due to T_OFFMost are fixed, so when T_ONAs f becomes shorter, f rises, i.e. less total power.

V is given by equation 4_CEAnd V is_CEDoes not become zero or lower unless in the period T₃Middle satisfaction relationship V_P≥V_DC. If the power is reduced, the energy supplied to the leakage transformer, i.e., the energy supplied to the resonance, becomes smaller, and this relationship may not be satisfied. This prevents switching on the IGBT at zero voltage, resulting in switching losses.

Furthermore, it is also possible to use the supply voltage V according to equations (2) and (3)_DCThe energy of the resonance is determined. A smaller voltage means a lower energy, resulting in a more difficult to satisfy relationship V_P≥V_DC. This is a third disadvantage of the prior art.

A brief explanation of the magnetron is given below.

The magnetron is a vacuum tube for generating microwaves, and two conditions are required for driving the magnetron. The first condition is that the cathode temperature needs to be raised to about 2,100K. The second condition is to use a high negative voltage between the anode and the cathode. To satisfy the first condition, a current from the third-stage coil of the leakage transformer is supplied to the cathode to increase the cathode temperature. In order to satisfy the second condition, the high voltage output of the secondary winding of the leakage transformer is converted into high voltage DC by a rectifier, and the DC is supplied to the anode and the cathodeA high dc voltage is applied across the cathodes. When the cathode temperature is about 2,100K, the voltage V across the anode and cathode of the magnetron_ACAnd anode current I_AThe relationship therebetween is shown in fig. 23.

VBM in FIG. 23 is referred to as the onset voltage, and typically will be V of-3.8 kV_BMFor use in a domestic microwave oven. The power P of the magnetron can be adjusted_MGIs defined as:

P_MC＝V_ACl a (equation 5),

approximately 70% of the power is emitted as microwaves.

The frequency of the generated microwave is 2.45GHz, but unnecessary radio waves of other frequencies of low level are also generated. To eliminate these radio waves, the magnetron may require a noise filter including a capacitor and a coil.

In the circuit diagram shown in fig. 21, the third-stage coil of the leakage transformer is connected to the cathode of the magnetron. The power is controlled by the ON time of the IGBT, and shortening the ON time reduces the power as described above. This reduces the voltage generated in the tertiary coil, resulting in a reduction in the current through the cathode. The frequency f also increases. Impedance Z of a coil defining a noise filter provided in a magnetron_L：

Z_L＝2πfL_N(equation 6)

Wherein L is_NIs the inductance of the coil of the noise filter.

Since the frequency f also rises, the cathode current is suppressed as the impedance becomes high, resulting in a further decrease in the cathode current.

This is a fourth disadvantage of the prior art single switching element voltage resonant circuit.

A fifth drawback is associated with the starting of the magnetron.

The magnetron is inoperable unless the cathode temperature reaches a temperature of about 2,100K. At start-up, it takes some time for the cathode temperature to rise. Since one advantage of microwave ovens is fast cooking, it is important that the magnetron in a microwave oven can be started as fast as possible. For this purpose, as much current as possible is applied to the cathode when starting up to produce a rapid temperature rise. However, if a large current is applied to the cathode at the time of starting, since the third-stage coil for supplying a current to the cathode and the secondary coil for supplying a high voltage to the magnetron are constructed using a single leakage transformer, the voltage of the secondary coil is simultaneously increased. Further, since the magnetron is started from low power by rapidly shortening the ON time of the IGBT, the coil impedance supplied to the cathode of the magnetron rises to a high level, thereby enhancing suppression of the cathode current. In order to provide sufficient cathode current under these conditions, the voltage of the secondary coil needs to be further increased. FIG. 24 shows the voltage V between the anode and cathode_ACThe characteristic of the steady state from the starting point to the normal oscillation of the magnetron varies with time. At time T_SMeanwhile, since a large current is supplied to the cathode, the voltage across the secondary coil is high. Then, after time TS, the magnetron starts to operate, and V_ACDown to V_BM。V_BMIs about-3.8 kV and the voltage generated at the starting point is about-7 kV. Therefore, in consideration of this voltage, it is desirable to design the withstand voltage of the diode and the capacitor constituting the rectifier. This is the fifth disadvantage.

The performance requirements for the power supply used to drive the magnetron are described below. First, a high voltage may be required for driving the magnetron. Therefore, if any foreign matter such as dust reaches the high-voltage portion, a spark may be generated. If this occurs, the circuit operation must be immediately stopped to avoid fire or smoke due to the continued sparks generated by the components forming the power circuit.

In addition, since the magnetron is a vacuum tube, gas can be generated from copper and tungsten constituting it. If such gas is generated in the electric field portion concentrated in the vacuum tube, a spark may occur in the tube. If a spark occurs, the impedance between the anode and cathode of the magnetron changes rapidly, and this may affect the operation of electrical components such as IGBTs. Furthermore, in this case, it is necessary to ensure operation without causing malfunction of the electrical element. This is the second requirement.

Summary of The Invention

The present invention relates to a power supply for driving a magnetron in a high-frequency heating apparatus.

The present invention relates to a power supply for driving a magnetron of a high-frequency heating apparatus, which is made to solve the problem of a single-switching-element type voltage resonance circuit which is a circuit type used for a conventional magnetron driving power supply, and is made to solve the problem of an increase in voltage applied to a semiconductor switching element due to the action of the resonance circuit, which is the first problem; and the second problem is that the ON time of the semiconductor switching element can be arbitrarily set, but the OFF time is determined by the resonant circuit and cannot be arbitrarily adjusted, and therefore the following configuration is used.

A high-frequency heating apparatus of the present invention includes a direct-current power supply, a leakage transformer connected to the direct-current power supply, a first semiconductor switching device connected in series with a primary coil of the leakage transformer, a first capacitor, a series circuit including a second capacitor and a second semiconductor switching device, a drive circuit having an oscillator for driving the first semiconductor switching device and the second semiconductor switching device, and a magnetron connected to a secondary coil of the leakage transformer. This structure allows the OFF time of the first semiconductor switching device (main switching element) to be adjusted by the second semiconductor switching device (auxiliary switching element). Meanwhile, the voltage applied to the main first semiconductor switching device can be reduced by the auxiliary second capacitor having a larger capacitance than the first capacitor forming the resonance circuit together with the leakage transformer.

In order to solve the third problem, that is, the problem that the voltage applied to the semiconductor switching element is not zero or less and the loss increases due to the switching operation when the power supply voltage for supplying power to the power supply circuit for driving the magnetron is decreased or when the power is decreased, the following configuration is adopted in addition to the configuration for solving the first and second problems.

An AC power supply is rectified to a DC power supply by a full-wave rectifier, and a frequency modulation circuit for modulating a frequency in accordance with a signal obtained from a voltage of the DC power supply is provided in a drive circuit. Further, a pulse width modulation circuit is provided in the drive circuit. This allows the range of the supply voltage or power used to implement the switching operation to be extended in the case where the voltage applied to the main first semiconductor switching device is zero or lower.

In order to solve the fourth problem that the impedance ZL of the coil of the noise filter provided in the magnetron is increased and the cathode current is decreased when the power is decreased and the ON time of the semiconductor switching element is shortened and the frequency f is increased, the following configuration is adopted in addition to the configuration for solving the first and second problems.

The constituted electric power is controlled by varying the proportion of the ON-time of the pulses for driving the first semiconductor switching device and driving the second semiconductor switching device, while at the same time keeping the frequency constant. This prevents the impedance of the coil of the noise filter from increasing, resulting in a decrease in the rate of cathode current drop during power reduction.

In order to solve the fifth problem that the secondary coil voltage of the leakage transformer increases when the cathode current is increased at the time of starting the magnetron, the following configuration is adopted in addition to the configuration for solving the first and second problems.

A leakage transformer having a three-stage coil for supplying a cathode current to a magnetron, and a magnetron having a cathode with a filter composed of a capacitor and a coil are provided. Further, a start frequency setting circuit for the device frequency to reduce the start impedance of the coil is provided to the drive circuit. This can provide sufficient current to the cathode while preventing an increase in voltage across the secondary coil.

In order to satisfy the first requirement, that is, to satisfy the requirement that sparks may be generated when dust or the like adheres to a high-voltage portion, and to quickly stop the operation of the circuit without causing danger in such a case, the following configuration is adopted in addition to the configuration for solving the first and second problems.

The invention further comprises a direct current power supply, a coil connected to the direct current power supply, a capacitor connected to the coil, a voltage detector for detecting a change in the voltage of the capacitor, and a comparator for comparing a reference level with the voltage level of the capacitor detected by the voltage detector. The drive circuit is designed to stop its operation if the detected level of the capacitor voltage rises above or falls below a reference level. It allows immediate interruption of the circuit in the event of a spark.

Drawings

Fig. 1 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a first exemplary embodiment of the present invention. The rectifier connected to the secondary winding of the leakage transformer uses a full-wave voltage doubler rectifier system.

FIG. 2 is a primary coil voltage V of a leakage transformer for explaining the operation of the circuit of FIG. 1_PThe waveform of (2).

Fig. 3 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a second exemplary embodiment of the present invention. A series circuit of a second semiconductor switching device and a second capacitor is connected in series with the primary coil of the leakage transformer.

Fig. 4 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a third exemplary embodiment of the present invention. The first capacitor is connected in series with the primary coil of the leakage transformer. In other words, the first capacitor is connected in parallel with the first semiconductor switching device.

Fig. 5 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a fourth exemplary embodiment of the present invention. The series circuit of the second semiconductor switching device and the second capacitor and the first capacitor are connected in series with the primary coil of the leakage transformer, in other words, in parallel with the first semiconductor switching device.

Fig. 6 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a fifth exemplary embodiment of the present invention. The rectifier connected to the secondary coil of the leakage transformer uses a half-wave voltage doubler rectifier system.

Fig. 7 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a sixth exemplary embodiment of the present invention. The rectifier connected to the secondary coil of the leakage transformer uses a full-wave rectifier system.

Fig. 8 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a seventh exemplary embodiment of the present invention. The secondary coil of the leakage transformer is divided into two parts and is provided with a center tap.

FIG. 9 shows a voltage waveform V of a DC power supply as a rectified AC power supply_DC。

Fig. 10 is a block diagram of a driving circuit of an electric power converter for driving a magnetron according to an eighth exemplary embodiment of the present invention.

Fig. 11A to 11D are waveforms related to the drive circuit in fig. 10, in which the abscissa represents time t. FIG. 11A is V_DCThe output voltage waveform of the detector 65, fig. 11B is the output waveform of the frequency modulator 29, fig. 11C is the waveform of the sawtooth wave generated by the oscillator 30, and fig. 11D is the output waveform of the pulse width modulator 28.

Fig. 12A to 12D are waveforms in the drive pulse signal generator 31 shown in fig. 10.

Fig. 13A to 13C show the case when pulse width modulation and frequency modulation occur. Fig. 13A is a voltage waveform of the dc power supply, fig. 13B is an anode current waveform of the magnetron, and fig. 13C is a current waveform of the dc power supply.

Fig. 14A to 14C show the case when there is no pulse width modulation and frequency modulation. Fig. 14A is a voltage waveform of a dc power supply, fig. 14B is an anode current waveform of a magnetron, and fig. 14C is a current waveform of the dc power supply.

Fig. 15 is a circuit diagram of a filter used in the magnetron.

FIG. 16 is a diagram for detecting the voltage V of the first semiconductor switching device 3_CEV of_CEA circuit diagram of the detector 45.

FIG. 17A is a voltage V of the first semiconductor switching device 3_CEFIG. 17B is V_CEThe waveform of the output voltage of the detector 45.

Fig. 18 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a ninth exemplary embodiment of the invention.

Fig. 19A to 19D show example waveforms of an operation when spark occurs between secondary coil terminals of the leakage transformer. Fig. 19A is a current waveform of the first semiconductor switching device 3, fig. 19B is a voltage waveform of the capacitor 51, fig. 19C is an output signal waveform of the comparator 54, and fig. 19D is a voltage waveform of the capacitor 60 (shown in fig. 20) forming the stop determination circuit 19.

Fig. 20 is a circuit configuration of an electric power converter for driving a magnetron used in a high-frequency heating apparatus according to a tenth exemplary embodiment of the invention.

Fig. 21 is a circuit configuration of a power supply for driving a magnetron of a high-frequency heating apparatus of the related art.

Fig. 22A to 22D are waveforms for explaining the operation of the power supply for driving the magnetron of the high-frequency heating apparatus of the related art. Fig. 22A is a current waveform of the IGBT, fig. 22B is a drive signal waveform of the IGBT, fig. 22C is a voltage waveform of the primary coil of the leakage transformer, and fig. 22D is a voltage waveform of the collector of the IGBT.

FIG. 23 shows the voltage V between the anode and cathode of a magnetron_ACAnd anode current I_AThe relationship characteristic(s).

FIG. 24 shows the voltage V between the anode and cathode_ACCharacteristic of the transition with time from the start-up of the magnetron to the normal oscillation.

Description of the preferred embodiments

First exemplary embodiment

Fig. 1 shows a circuit diagram of an electric power converter for driving a magnetron used in a high-frequency heating apparatus in a first exemplary embodiment of the present invention. The high-frequency heating apparatus of the first exemplary embodiment includes a direct-current power supply 1, a leakage transformer 2, a first semiconductor switching device 3, a first capacitor 4, a second capacitor 5, a second semiconductor switching device 6, a drive circuit 7, a full-wave voltage doubler rectifier 8, and a magnetron 9. Conversion of AC power from AC voltage to DC voltage V by full-wave rectifier_DCAnd is provided to a circuit in which the first semiconductor switching device 3 is connected in parallel with a circuit including the first capacitor 4 and the primary coil of the leakage transformer 2. A series circuit of a second capacitor 5 and a second switching device 6 is connected in parallel with the first capacitor 4. The high voltage output generated in the secondary coil of the leakage transformer 2 is converted into a high direct current voltage by a full-wave voltage doubler rectifier 8, and is applied between the anode and cathode of the magnetron 9. The third stage coil of the leakage transformer 2 supplies current to the cathode of the magnetron 9.

The first semiconductor switching device 3 includes an IGBT10 and a diode 11 connected in parallel with the IGBT 10. The second semiconductor switching device 6 further includes an IGBT13 and a diode 12.

The drive circuit 7 has an oscillator for generating drive signals for the first semiconductor switching device 3 and the second semiconductor switching device 6. This oscillator generates a signal with a specific frequency and power and supplies a drive signal to the first semiconductor switching device 3. The second semiconductor switching device 6 is supplied with a delayed and inverted (inverted) signal of the drive signal for the first semiconductor switching device 3.

Reference is made to the primary winding voltage V of the leakage transformer 2 as shown in FIG. 2_PThe waveforms of (a) explain the operation of the circuit as shown in figure 1. Using the primary winding voltage V_PAnd the voltage V of the DC power supply 1_DCThe voltage V of the first semiconductor switching device 3 can be adjusted_CEThe definition is as follows:

V_CE＝V_DC-V_P(equation 7)

First, when the IGBT10 is turned on, the collector current I_CFlows through the primary coil of the leakage transformer 2. Voltage V of the direct current power supply 1 when the IGBT10 is turned on_DCCompared to the voltage V of the first semiconductor switching device 3_CEIs very small, so that the primary coil voltage V_PVoltage V to dc power supply 1_DCAre almost equal. This is the period T in FIG. 2₁. Here, the secondary winding output of the leakage transformer 2 starts to charge the capacitor 15 of the full-wave voltage doubler rectifier 8. When V is₂+V₃The following steps are achieved:

V₂+V₃＞V_BM(equation 8)

Wherein V₂: the initial voltage of the capacitor 14 is,

V₃: voltage of capacitor 15, and

V_BM: oscillating starting voltage

The magnetron 9 starts oscillating and the anode current starts flowing to the magnetron 9.

When the first semiconductor switching device 3 is turned off, the current in the primary coil of the leakage transformer 2 starts to flow to the first capacitor 4. Then, the secondary winding output of the leakage transformer 2 starts to charge the capacitor 14, and the primary winding voltage V starts_PThe period T in fig. 2 is reduced as shown₂. When equation 8 is satisfied, the magnetron 9 starts to oscillate again. When the voltage of the first capacitor 4 reaches the initial voltage V of the second capacitor 5_SAt this time, the diode 12 in the second semiconductor switching device 6 is turned on, and the second capacitor 5 starts to be charged. Primary winding voltage V_PBecomes a period T as shown in FIG. 2₃。

The second capacitor 5 is designed to have a larger capacitance than the first capacitor 4, and thus the voltage drop is suddenly alleviated during the period T3. Conversely, when the current flowing from the primary coil of the leakage transformer 2 to the second capacitor 5 starts to flow from the second capacitor 5 to the primary coil, the primary coil voltage V_PMovement in FIG. 2 for a period T₄And (4) the following steps. At this point, it may be necessary to turn on the IGBT13 in the second semiconductor switching device 6. When at a specific time T₄When the IGBT13 is turned off, a current starts to flow from the first capacitor 4 to the primary coil of the leakage transformer 2, and the primary coil voltage V_PIs increased to be in period T of FIG. 2₅Moving in the inner direction. In the period T₅In this case, the voltage rises rapidly. When the voltage reaches V_DCWhen the voltage of the first semiconductor switching device 3 becomes zero according to equation 7. By driving the first semiconductor switching device 3 again at this point, from the period T₁The same operation starts to be repeated.

This realizes a switching operation, which can reduce switching loss. By the period T during which the second semiconductor switching device 6 is switched on₄At a specific time T₄Determining the initial voltage V of said second capacitor 5_S. In particular, due to the extension of the ON-time of the second semiconductor switching device 6, the initial voltage V of the second capacitor 5_SLowering results in a decrease of the voltage of the first semiconductor switching device 3.

As described above, the OFF time of the first semiconductor switching device 3, or the ON time of the second semiconductor switching device 6 can be freely set, which is not possible with the circuit configuration of the related art. Further, by making the second capacitor 5 have a larger capacitance than that of the first capacitor 4, the voltage of the first semiconductor switching device 3 can be fixed (clamp).

Second example embodiment

Fig. 3 shows a circuit diagram of a power converter (electric power supply converter) for driving a magnetron used in the high-frequency heating apparatus according to the second exemplary embodiment of the present invention.

In fig. 3, the same elements as those in fig. 1 are denoted by the same reference numerals, and their explanation is omitted here. The difference from the configuration in fig. 1 is that a second semiconductor switching device 6 and a second capacitor 5 connected in series are connected in series with the primary coil of the leakage transformer 2 and in parallel with the first semiconductor switching device 3.

In this example embodiment, in comparison with the operation of the circuit in fig. 1, during the period in which the current flows from the primary coil of the leakage transformer 2 to the second capacitor 5 and vice versa from the second capacitor 5 to the primary coil, the current flowing between the primary coil of the leakage transformer 2 and the second capacitor 5 passes through the direct current power supply 1 in the circuit. Other features are the same as those of the circuit in the first exemplary embodiment.

In the case of the circuit configuration in this exemplary embodiment, if, for example, a bipolar transistor is used as the second semiconductor switching device 6, then a pnp type transistor is preferably used. If MOS transistors or IGBTs are used, p-channel IGBTs are preferably used. However, this circuit configuration allows the emitters of the first semiconductor switching device 3 and the second semiconductor switching device 6 to be kept at the same potential. The pair of second semiconductor switching devices 6 prevents a high emitter potential, allowing the drive circuit to be designed to have a lower withstand voltage, as compared with the structure shown in fig. 1.

Third exemplary embodiment

Fig. 4 shows a circuit diagram of a power converter for driving a magnetron used in a high-frequency heating apparatus in a third exemplary embodiment of the invention.

In fig. 4, the same elements as those in fig. 1 are denoted by the same reference numerals, and thus their explanation is omitted. The difference from the structure in fig. 1 is that a first capacitor 4 is connected in parallel with the first semiconductor switching device 3.

This allows the current flowing to the first capacitor 4 to pass through the direct current power supply 1. If an IGBT or MOS transistor is employed in the first semiconductor switching device 3, a certain capacitance exists between the collector and the emitter in the case of the IGBT, and between the drain and the source in the case of the MOS transistor. Therefore, by connecting the first capacitor 4 in parallel with the first semiconductor switching device 3, the capacitance of the first capacitor 4 can be designed to be smaller, the value of which is equal to the capacitance existing in the first semiconductor switching device 3.

Fourth exemplary embodiment

Fig. 5 shows a circuit diagram of a power converter for driving a magnetron used in a high-frequency heating apparatus in a fourth exemplary embodiment of the invention.

In fig. 5, the same elements as those in fig. 1 are denoted by the same reference numerals, so that explanation thereof is omitted here. The difference from the configuration in fig. 1 is that a second semiconductor switching device 6 and a second capacitor 5 connected in series are connected in series with the primary coil of the leakage transformer 2, and the first capacitor 4 is connected in parallel with the first semiconductor switching device 3. In other words, this is a combination of the structures shown in fig. 4 and 5 to take advantage of both.

With this structure, a high potential can be not applied to the second semiconductor switching device 6, thereby allowing the drive circuit to be designed to have a lower withstand voltage. In addition, the capacitance of the first capacitor 4 can be designed to be smaller, with a value equal to the capacitance in the first semiconductor switching device 3.

Fifth exemplary embodiment

Fig. 6 shows a circuit diagram of a power converter for driving a magnetron used in a high-frequency heating apparatus in a fifth exemplary embodiment of the invention.

In fig. 6, the same elements as those in fig. 1 are denoted by the same reference numerals, and explanation thereof is omitted here. The difference from the configuration in fig. 1 is that the rectifier 16 connected to the secondary coil of the leakage transformer 2 includes a capacitor 17 and a diode 18, and a half-wave voltage doubler rectifier system is employed. The capacitor 17 is charged by the voltage generated in the secondary coil while the first semiconductor switching device 3 is turned off, and the magnetron 9 is driven by the sum of the voltage generated in the secondary coil of the leakage transformer 2 and the voltage of the capacitor 17 while the semiconductor switching device 3 is turned on.

A supplementary description will be given below of full-wave voltage doubler rectification as a rectifier system used in the configuration shown in fig. 1. In fig. 1, the capacitor 15 is charged by the voltage generated in the secondary coil of the leakage transformer 2 while the first semiconductor switching device 3 is turned on, and the magnetron 9 is driven by the sum of the voltage and the voltage of the capacitor 14. The voltage generated in the secondary coil of the leakage transformer 2 charges the capacitor 14, and the sum of the voltage and the voltage of the capacitor 15 drives the magnetron 9 while turning off the first semiconductor switching device 3.

Sixth exemplary embodiment

Fig. 7 shows a circuit diagram of a power converter for driving a magnetron used in a high-frequency heating apparatus in a sixth exemplary embodiment of the invention.

In fig. 7, the same elements as those in fig. 1 are denoted by the same reference numerals, and explanation thereof is omitted here. The difference from the configuration in fig. 1 is that the rectifier 19 connected to the secondary coil of the leakage transformer 2 comprises 4 diode bridges and employs a full-wave rectifier system. The number of coils of the secondary coil of the leakage transformer 2 in this system may be required to be about twice the number of coils of the secondary coil of the leakage transformer in the structure of fig. 1. However, this allows the magnetron 9 to be driven in both cases, simultaneously turning on and off the first semiconductor switching device 3, which is the same as the structure of fig. 1.

Seventh example embodiment

Fig. 8 shows a circuit diagram of a power converter for driving a magnetron used in a high-frequency heating apparatus in a seventh exemplary embodiment of the invention.

In fig. 8, the same elements as those in fig. 1 are denoted by the same reference numerals, and a description thereof will be omitted. The difference from the structure in fig. 1 is that the secondary coil of the leakage transformer 20 in fig. 8 is divided into two parts, and a center tap 21 is provided. The two separate coils are tightly coupled. The diode 24 is turned on while the first semiconductor switching device 3 is turned on to supply a voltage for driving the magnetron 9.

The diode 25 is turned on while the first semiconductor switching device 3 is interrupted to supply a voltage for driving the magnetron 9.

A common feature in the configurations shown in fig. 1, 6, 7 and 8 is that the energy in the leakage transformer 2 or 20 is dissipated by driving the magnetron 9 or by charging the capacitor of the rectifier connected to the secondary winding while switching the first semiconductor switching device 3 on and off.

The dc power supply of fig. 1 is generated by rectifying an ac power supply. Voltage waveform V of dc power supply 1_DCAs will be shown in fig. 9. The period T in fig. 9 is the period of the ac power source, and the peak voltage VM_AXIs the effective voltage of the AC power supplyAnd (4) doubling. Since this voltage waveform changes from about 0V to V_MAXSo that the resonance energy also depends on the voltage V of the power supply 1_DCAs described in the prior art. In particular, when the supply voltage becomesAs smaller, the resonance energy becomes smaller, and it becomes more difficult to reduce the voltage applied to the semiconductor switching device to zero. Therefore, by extending the ON time of the first semiconductor switching device 3, the voltage applied to the semiconductor switching device can be more easily reduced to zero, thereby increasing the resonance energy of the leakage transformer 2 due to the reduction of the power supply voltage.

Eighth example embodiment

Fig. 10 shows a block diagram of the drive circuit 7 for explaining the eighth exemplary embodiment of the present invention. Inverting and amplifying circuit 26 and amplifier 27 receive V_DCAn output signal of the detector 65, and the detector 65 detects the voltage V of the DC power supply 1_DC. A pulse width modulator 28 receives the output of the inverting and amplifier circuit 26 and a frequency modulator 29 receives the output of the amplifier 27. The oscillator 30 generates a sawtooth wave whose frequency is set by the frequency modulator 29. The drive pulse signal generator 31 generates a drive pulse signal for driving the first semiconductor switching device 3 and the second semiconductor switching device 6. The drive pulse signal generator 31 is designed to generate a drive pulse signal with a pulse width set by the pulse width modulator 28 and according to a frequency set by the oscillator 30.

Next, it is checked that a constant current I is maintained in the leakage transformer 2 without receiving the power supply voltage V_DC(voltage of the dc power supply 1). Can change the power supply voltage V_DCIs defined as:

wherein E is₀Is the effective value of the ac power source, ω is the angular frequency of the ac power source, and t is time.

Since the current I of the leakage transformer can be defined according to equation 2, it is necessary to be in equation 2T_ONThe following equation is satisfied to maintain a constant current in the leakage transformer without being affected by the voltage:

T_ON＝T₀/SIN(ωt)

wherein, T₀Is a specific value.

Thus, V is inverted and amplified by the inverting and amplifying circuit 26_DCThe output signal of the detector 65 is then input to the pulse width modulator 28. By making the ON time T of the semiconductor switching device_ONThe above conditions are satisfied, and the current I in the leakage transformer 2 is kept constant with the power supply voltage V_DCIs irrelevant.

Since when the power supply voltage V is applied_DCBecomes lower, the ON time T of the drive pulse_ONAnd extended, it is necessary to make the period T longer. In other words, it may be required to reduce the frequency f. Thus, V is amplified by amplifier 27_DCThe output signal of the detector 65 is then input to the frequency modulator 29 so as to be supplied with the power supply voltage V_DCThe frequency increases as the height increases.

FIG. 11A shows V_DCOutput voltage waveform V of detector 65_DC. It is based on the envelope of the waveform produced by full-wave rectification of an alternating current power supply. Fig. 11B shows an output waveform V of the frequency modulator 29₂₉. Fig. 11C shows the sawtooth wave after frequency modulation generated by the oscillator 30 based on the signal shown in fig. 11B. FIG. 11D shows the output waveform V of the pulse width modulator 28 as the inverted waveform of FIG. 11A_ON。

Fig. 12A to 12D show part of the waveform processing performed in the drive pulse signal generator 31. FIGS. 12A and 12C show how the sawtooth waveform V after frequency modulation in FIG. 11C will be_NCAnd the output waveform V of the pulse width modulator 28 in FIG. 11D_ONAnd (6) comparing. Fig. 12A shows an enlarged view of the bottom of the envelope, which is the low supply voltage portion, while fig. 12C is the top portion of the envelope, which is the high supply voltage portion. Compare FIGS. 12A and 12C with the sawtooth waveform V in FIG. 12C_NCPeriod T of_TIn contrast, the sawtooth wave V in FIG. 12A_NCPeriod T of_BIs T_B＞T_T. With respect to the output V of the pulse width modulator 28_ONWhen V in FIG. 12A is considered_ONViewed as V_ONBAnd V in FIG. 12C_ONViewed as V_ONTWhen, V_ONB＞V_ONT. FIG. 12B shows V in FIG. 12A_ON＞V_NCThe signal during this period is output, and this is a pulse signal for driving the first semiconductor switching device 3. In a portion where the power supply voltage is low, the ON time of the semiconductor switching device 3 is extended, and also the period is extended proportionally to increase the energy supplied to the leakage transformer so that the voltage supplied to the semiconductor switching device becomes zero. In a portion where the power supply voltage is high, the ON time of the first semiconductor switching device 3 is shortened, and the period is also shortened proportionally to reduce the energy supplied to the leakage transformer to a certain extent, thereby applying a zero voltage to the semiconductor switching device.

Even if the power supply voltage varies greatly, ON and OFF operations of the semiconductor switching device can be performed at zero voltage by the above-described control.

Next, with reference to fig. 13, an anode current of a magnetron which is one of the advantages of the present invention is explained. FIG. 13A shows the voltage waveform V of the DC power supply 1_DCFIG. 13B shows the anode current I of the magnetron in proportion to the output_AAnd FIG. 13C shows a current waveform I of the DC power supply 1_INIt has the same shape as the envelope of the anode current waveform represented by the broken line of the anode current waveform in fig. 13B. Fig. 13B is characterized in that it suppresses the output around the top of the envelope where the frequency is high due to the influence of pulse width modulation and frequency modulation, and the ON time of the first semiconductor switching device is short, and it increases the output around the bottom of the envelope where the frequency is low, and the ON time of the first semiconductor switching device is long. This approximates the envelope of the anode current waveform to a trapezoid. For comparison, fig. 14A-C show waveforms when the pulse width and frequency are not modulated. FIG. 14A shows the voltage waveform V of the DC power supply 1_DCFIG. 14B shows the anode current I of the magnetron_AFIG. 14C shows a current waveform I of the DC power supply 1_IN. Compare each of the positives in FIGS. 13B and 14BPolar current waveform I_APeak value of (1)_AP1And I_AP2Realizing I at the same power level_AP1＜I_AP2。

The degradation of the magnetron is closely related to the peak value of the anode current. Higher peaks tend to accelerate degradation of the magnetron. Therefore, if the peak value of the anode current is reduced by the action of the pulse width modulation and the frequency modulation as shown in fig. 13B, the degradation of the magnetron can be suppressed.

Further, in the input current waveform as shown in fig. 13C, in the period T₁Meanwhile, when the current does not flow, the power supply voltage is so low that the voltage applied between the anode and the cathode of the magnetron becomes higher than the oscillation start voltage V of the magnetron_BMAnd lower. Thus, the anode current does not flow, and the input current does not flow. And a period T as shown in FIG. 14C₂In contrast, the period T in which the input current does not flow₁Is T₁＜T₂. As a result, the high frequency component of the input current waveform shown in fig. 13C becomes smaller than the input current waveform shown in fig. 14C, so that the power factor can be improved.

Next, a method of removing noise to operate a large cathode current at the time of start-up in the present invention is explained. Since the magnetron generates noise in the TV band, a filter is provided to remove such noise. As shown in fig. 15, the filter includes coils 35 and 36 inserted in series to the cathode, capacitors 37 and 38 connected between the anode and the cathode, and a capacitor 139 connected in parallel to the cathode. Impedance Z of cathode_CIs about 0.3 omega and the impedance Z of the coils 35 and 36 can be defined by equation 6_LWhere f is the frequency of the cathodic current. If the frequency is 40kHz, the impedance Z_LWill be about 0.5 omega. This value is approximately equal to the impedance Z of the cathode_CAnd is the main factor determining the cathode current level. As described in the prior art, a large cathode current is required for rapid starting of the magnetron. It is therefore clear that the impedance of the coils 35 and 36 can be reduced ideally by reducing the frequency for this purpose. For example, if the frequency drops to 20kHz, then the coil impedance may be according to equation 6Which is half at about 40KHz, which results in an increase in cathode current.

In fig. 10, the frequency of the sawtooth wave generated by the oscillator 30 at the time of startup is set to the drive circuit 7 by the minimum frequency setting circuit 39. In other words, by operating at a lower frequency at the time of startup than at the time of normal operation, it is possible to reduce the impedance of the coils 35 and 36 forming the filter of the magnetron, thereby increasing the cathode current.

Further, in fig. 10, the frequency modulation switch circuit 40 removes the signal from the frequency modulator 29 at startup to prevent modulation. This enables operation at a fixed low frequency set by minimum frequency setting circuit 39 regardless of the voltage level of the dc power supply at startup.

The frequency modulation switching circuit 40 includes a current detector 41, a judgment circuit 42, and switches 43 and 44. When the magnetron oscillates, an anode current flows through a capacitor or a diode of the full-wave voltage doubler rectifier 8 in fig. 1. Therefore, the current detector 41 can detect the current of the diode, the capacitor, or the anode using the current transformer. The judgment circuit 42 judges whether or not the magnetron starts oscillation from the output level of the current detector 41. Even if the magnetron is not oscillated, a certain dark current flows. Since the magnetron is provided with a filter as shown in fig. 15, a current also flows through its capacitor. In addition, a charging or discharging current of the capacitor also flows in the diode and the capacitor forming the full-wave voltage doubler rectifier 8 in fig. 1. Since such a current level is extremely low, the determination circuit 42 can distinguish it from a current flowing at the time of magnetron oscillation. When the magnetron is not oscillated, the judgment circuit 42 opens the switch 43 to remove the signal of the frequency modulator 43 to the oscillator 30, and at the same time, closes the switch 44 to send the signal of the minimum frequency setting circuit 39 to the oscillator 30. When the judging circuit 42 judges that the magnetron is oscillating, the switch 43 is closed and the switch 44 is opened simultaneously to perform frequency modulation. The minimum frequency setting circuit 39 has a function of setting a minimum frequency or audio frequency at which the transition cathode current is prevented from flowing. The minimum frequency is set to 20kHz or higher.

By reducing the frequency at the time of start-up, a sufficiently large current can be applied without applying a large voltage to the secondary coil of the leakage transformer 2 at the time of start-up as in the prior art. The voltage of the secondary winding is proportional to the voltage of the primary winding, and the voltage V of the DC power supply_DCA voltage V applied to the first semiconductor switching device_CEAnd primary winding voltage V_L1The relationship of (a) to (b) is as follows:

V_DC＝V_L1+V_CE(equation 9)

Thus, in order to apply an appropriate secondary coil voltage, V needs to be controlled_CETo a suitable voltage. Due to V_CELower than the secondary coil voltage, it has the advantage of being easy to detect. In addition, the leakage transformer is also insulated between the primary coil and the secondary coil. When the secondary coil voltage is detected, its signal needs to be sent to a driving circuit connected to the primary coil side using an insulating means such as a photocoupler (photocoupler). Such a structure may be complicated. Detecting a voltage V of a first semiconductor switching device_CEA method of (2) also has the advantage of a simplified structure.

The drive circuit 7 shown in fig. 10 is provided with a voltage V for detecting the first semiconductor switching device 3_CEV of_CEA detector 45. Handle V_CEOutput signal V of detector 45_OCT(refer to fig. 10 and 16) to the pulse width modulator 28. V_CEThe detector 45 has an effect on the pulse width modulator 28 shown in figure 11D at start-up. In particular, when V_CEWaveform V of solid line in FIG. 11D when detector 45 is not in operation_ONIs the output waveform of the pulse width modulator 28. When V is_CEIn operation, the dashed line is the output waveform. It is obvious that V_CEDetector 45 reduces V_ONThe voltage level of (c). Thus, by reducing the output waveform V of the pulse width modulator 28_ONTo the voltage V to be applied in the first semiconductor switching device 3_CEThe time width of the pulse signal for driving the first semiconductor switching device 3 can be narrowed by controlling at an appropriate level. As a result, the secondary winding voltage is controlled to be highThe minimum level necessary to supply the proper cathode current.

Fig. 16 shows a diagram for detecting the voltage V of the first semiconductor switching device 3_CEV of_CEAn example of the structure of the detector 45. The voltage V is divided by a resistor_CETo drive the transistor to charge the capacitor 46. Capacitor 46 discharges at resistor 47 in parallel. The voltage V of this capacitor 46_OUTIs the signal input to the pulse width modulator 28. Fig. 17A shows the voltage V of the first semiconductor switching device 3_CEThe waveform of (2). The period T is a period of the ac power source, and it is about 16 msec, for example. Since the first semiconductor switching device 3 operates in a period of 50 seconds or less, the collector voltage V in fig. 17A_CEComprises 320 half cycles. However, the waveforms in fig. 17A are simplified. Due to the collector voltage V of the first semiconductor switching device 3_CEHas a pulse shape, the control may become unstable if this information is directly input to the pulse width modulator 28. Thus, the signal is represented by V as shown in FIG. 16_CEDrives the transistor and charges a capacitor 46 through a resistor 48 connected to the power supply. Capacitor 46 discharges through resistor 47. The time constant of charging and discharging is set to be shorter than the period of the alternating-current power supply and longer than the drive period of the first semiconductor switching device 3. This enables output of the voltage V shown in FIG. 17B_OUTIs made to approach the voltage V of the first semiconductor switching device 3 as shown by the broken line in fig. 17B_CEThe envelope of (c). Thus, by providing this V to the pulse width modulator 28_OUTStable operation can be ensured.

As described above, the magnetron is provided with a filter, and the coil impedance in the filter varies with the frequency, resulting in variation of the cathode current. With the single switching device voltage resonant circuit structure of the prior art, the output is reduced by shortening the ON time of the semiconductor switching device, resulting in a significant reduction in cathode current due to the increase in frequency. Therefore, the drive circuit of the present invention adjusts the ON time of the first semiconductor switching device 3 to control the output, and keeps the signal of the frequency modulator 29, which is changed in response to the above-described power voltage waveform, constant regardless of the output. Thus, the frequency at a specific point of the envelope is not changed in controlling the output, suppressing a decrease in the cathode current at low output. This enables to maintain an appropriate cathode temperature at a low output, thereby achieving a lower output than the prior art. This greatly improves the cooking performance, in particular, the food thawing performance.

Ninth exemplary embodiment

Referring to fig. 18, a method (measure) of resisting sparks and lightning in the ninth exemplary embodiment is described. A high voltage is required to drive the magnetron, and sparks are released if dust or soot reaches a portion carrying the high voltage. If sparking occurs, it is preferable to stop the operation of the circuit immediately. The electric power converter of the present invention for driving a magnetron used in a high-frequency heating apparatus is provided with a voltage detector 52 for detecting a voltage variation of a capacitor 51, wherein the capacitor 51 forms a filter with a coil 50 to prevent a high-frequency component from being transmitted to a direct-current power supply 1, a reference voltage source 53 and a comparator 54. The comparator 54 compares the output of the voltage detector 52 with the voltage level of the reference voltage source 53. If the output of the voltage detector 52 is larger, the comparator 54 outputs a stop signal to the drive circuit 7 to stop the circuit operation. For example, if the voltage of the voltage detector 52 is measured by analog-to-digital conversion, the comparator 54 may replace the reference voltage source 53 with a constant as a reference level, and may compare the value with the reference level after the analog-to-digital conversion.

If a spark occurs between the secondary coil terminals of the leakage transformer, an overcurrent flows when the IGBT10 of the first semiconductor switching device 3 is turned on because the inductance of the leakage transformer is reduced. At this time, electric charge is supplied from the capacitor 51, and by releasing the electric charge, the voltage of the capacitor 51 suddenly drops. Next, when the IGBT10 is turned off, the current in the primary coil of the leakage transformer 2 flows to the capacitor 4, and then flows to the capacitor 5 through the diode 12 of the second semiconductor switching device 6. When the energy in the leakage transformer 2 is fully transferred to the capacitor 5, the current starts to flow through the switched-on IGBT13,and conversely from the capacitor 5 to the leakage transformer. If the IGBT13 is turned off at this time, the current in the leakage transformer 2 flows to the capacitor 51 through the diode 11 of the first semiconductor switching device 3, and the voltage of the capacitor 51 abruptly rises. Fig. 19A shows a current waveform I of the IGBT10 of the first semiconductor switching device 3 at the time of occurrence of a spark_C1Current waveform I of the sum diode_D1And fig. 19B shows the voltage waveform V of the capacitor 51_C51. In fig. 19A, the positive direction of the coordinates represents I_C1And negative direction represents I_D1. A spark (near short circuit) occurs at the arrow. Obviously, an over-current I_C1Current and voltage V_C51Suddenly dropping. Thus, an overcurrent I_D1Flows to the diode 11 in the next cycle, thereby abruptly increasing V_C51. Fig. 19C shows an output signal of the comparator 54. When V is_C51When it suddenly increases and exceeds a certain level, an output signal is sent. This signal stops the drive circuit 7 from operating.

When the voltage V is_C51It can also be detected when there is a sudden drop. In other words, when the output voltage of the voltage detector 52 is smaller than the voltage of the reference voltage source 53, the comparator transmission signal may be constituted.

As described above, the coil impedance in the filter of the magnetron is reduced by starting the drive circuit 7 with a low frequency. This results in a voltage V of the capacitor 51 at start-up_C51Is greater than during normal operation. As a result, the circuit including the voltage detector 52, the reference voltage source 53, and the comparator 54 can be activated (activated) at the time of start-up and can stop the operation of the drive circuit 7. For this reason, the voltage of the reference voltage source 53 is set to a higher level at the time of startup. The transistor of the switching circuit 55 in fig. 18 is interrupted at startup to set the reference voltage source 53 to a high voltage, and turned on during normal operation to set the reference voltage source 53 to a low voltage.

When the ac power supply is subjected to a surge due to lightning or other reasons, the voltage of the dc power supply 1 obtained by rectifying the ac power supply rises. In this case, the voltage rises several tens of times the normal voltage. In order to protect the circuit under such a voltage surge, the operation of the circuit needs to be stopped. For this purpose, a voltage detector 52, a reference voltage source 53, and a comparator 54 for detecting a voltage change of a capacitor 51 forming a filter together with the coil 50 to prevent a high-frequency component from being transmitted to the direct current power supply 1 are provided. The comparator 54 compares the output of the voltage detector 52 with the voltage of the reference voltage source 53, and if the output of the voltage detector 52 is larger, outputs a stop signal to the drive circuit 7 so that the circuit operation can be stopped. This structure can detect overvoltage caused by rapid voltage rise. Therefore, by using this structure, the operation is immediately stopped when an overvoltage occurs in the ac power supply system. However, during normal operation, it is necessary to prevent interference of the voltage detection level of the capacitor 51 and the voltage detection level of the alternating-current power supply system. An ac supply voltage detector 56 is provided which includes a series circuit of a diode 57 and a constant voltage device, such as a zener diode 58. The ac power supply voltage divided by the resistance is supplied to the voltage detector 52 by the ac power supply voltage detector 56. If the ac power supply voltage divided by the resistance is smaller than the sum of the divided voltage of the capacitor 51 input to the voltage detector 52 and the zener voltage of the zener diode 58, no signal is input to the voltage detector 52. Therefore, during normal operation, the output of the ac supply voltage detector 56 does not affect the voltage detector 52. Further, by providing the diode 57, the voltage detector 52 does not affect the ac power supply voltage detector 56. With this structure, the operation of the circuit can be stopped immediately when an overvoltage caused by a lightning surge or a spark occurs. By sharing circuit components, a circuit can be constructed with fewer components.

Tenth example embodiment

Referring to fig. 20, a determination circuit 59 for stopping the operation of the drive circuit 7 in the tenth exemplary embodiment of the present invention is described. The judgment circuit 59 for judging the stop (stop) shown in fig. 20 includes a capacitor 60 charged by the output of the comparator 54 and a comparator 62 for comparing the voltage of the capacitor 60 with the voltage of the reference voltage source 61. The solid line in the waveform shown in fig. 19D represents that of the capacitor 60Voltage waveform V₆₀And the dotted line represents the voltage V of the reference voltage source 61₆₁. The capacitor 60 is charged by the output signal of the comparator 54 which is a pulse signal as shown in fig. 19C, and the voltage of the capacitor 60 exceeds the voltage V of the reference voltage source 61 at the second pulse signal₆₁. At this point, the output signal of the capacitor 62 is inverted to stop the operation of the driving circuit. The determination circuit 59 determines when to output the stop signal, i.e., after how many times the signal is output from the comparator 54, depending on the capacitance of the capacitor 60, the value of the resistance 63 of the current flowing when charging the capacitor 60, or the value of the resistance 64 when discharging the capacitor 60. Such a determination circuit 59 is provided, for example, for the purpose of preventing the stop of operation with a transient overvoltage which does not have a great influence on the circuit operation or external noise. The judging circuit of the invention utilizes the characteristic that sparks and lightning surge have certain continuity, and the external noise and the transient overvoltage are more transient, without continuously distinguishing the lightning surge and the sparks from the external noise and the transient overvoltage.

Industrial applicability

The high-frequency heating apparatus includes a direct-current power supply, a leakage transformer connected to the direct-current power supply, a first semiconductor switching device connected in series to a primary coil side of the leakage transformer, a first capacitor connected in parallel to the primary coil side of the leakage transformer, a second capacitor, a second semiconductor switching device connected in series to the second capacitor, a drive circuit having an oscillator for driving the first and second semiconductor switching devices, a rectifier connected to a secondary coil side of the leakage transformer for full-wave voltage doubler rectification, and a magnetron connected to the rectifier. By connecting the series circuit of the second capacitor and the second semiconductor switching device in parallel with the primary coil side of the leakage transformer, the OFF time of the first semiconductor switching device 3 or the ON time of the second semiconductor switching device 6 can be freely set. Further, by securing a sufficient capacitance of the second capacitor 5 compared to the first capacitor 4, the voltage of the first semiconductor switching device 3 can be fixed.

With the configuration as shown in fig. 3, by connecting the series circuit of the second capacitor 5 and the semiconductor switching device 6 in series with the primary coil side of the leakage transformer 2, the emitters of the first semiconductor switching device 3 and the second semiconductor switching device can be set to have the same potential. This allows the drive circuit to be designed to have a low withstand voltage because the emitter of the second semiconductor switching device 6 is prevented from having a high potential.

With the configuration shown in fig. 4, by connecting the first capacitor 4 in series with the primary coil side of the leakage transformer 2 and then connecting the series circuit of the second capacitor 5 and the second semiconductor switching device 6 in parallel with the primary coil of the leakage transformer, the capacitance of the first capacitor 4 can be made smaller, and the value thereof is equal to that of the first semiconductor switching device 3.

With the structure shown in fig. 5, two effects can be obtained. In particular, by connecting the first capacitor 4 in series with the primary winding side of the leakage transformer 2 and then connecting the series circuit of the second capacitor 5 and the second semiconductor switching device 6 in series with the primary winding of the leakage transformer, the capacitance of the first capacitor 4 can be made smaller, the value of which is equal to that of the first semiconductor switching device 3, and the drive circuit can be designed to have a lower withstand voltage.

With the structure employing the half-wave voltage doubler rectifier system as shown in fig. 6, the structure employing the full-wave rectifier system as shown in fig. 7 or the structure employing the center-tapped system for dividing the secondary coil of the leakage transformer 2 into two parts as shown in fig. 8, the utilization rate of the leakage transformer can be improved.

Further, even if the voltage of the power supply is largely changed by full-wave rectification of the alternating-current power supply to obtain the direct-current power supply, and a frequency modulator is provided in the drive circuit to modulate the frequency according to a signal emitted based ON the voltage of the direct-current power supply, the ON and OFF operations of the semiconductor switching device can be carried out at zero voltage.

By providing a frequency-modulated switching circuit to switch the frequency modulation at start-up and during normal operation, so that the frequency modulation is deleted at start-up to start at a certain frequency, the current supplied to the cathode of the magnetron can be increased.

The frequency-modulated switching circuit further includes a current detector for detecting a current of a diode or a capacitor forming a rectifier connected to the secondary coil side of the leakage transformer, and a determination circuit for determining magnetron oscillation based on an output signal of the current detector. This allows the state of the magnetron to be accurately judged.

By providing a minimum frequency setting circuit in the fm switch circuit, the circuit can also be prevented from operating in the audio frequency band to avoid irritating sounds.

By performing full-wave rectification of an alternating-current power supply to obtain a direct-current power supply and modulating a pulse width of a signal generated based ON a voltage of the direct-current power supply, ON and OFF operations of a semiconductor switching device can be performed at zero voltage even if the power supply voltage greatly changes.

By changing the ratio of the ON times of the pulses for driving the first semiconductor switching device and driving the second semiconductor switching device, but keeping the period constant during controlling the electric power, the rate of change of the cathode current of the magnetron at the time of controlling the output can be kept low.

By setting a lower operating frequency to start operation, it is possible to reduce the impedance of the coil provided in the magnetron without generating a large voltage in the secondary coil of the leakage transformer to supply a sufficient current to the cathode of the magnetron.

By providing a voltage detector for detecting a voltage applied to the first semiconductor switching device, information on the secondary coil voltage of the leakage transformer at the time of startup can be indirectly detected. By controlling the pulse width of the drive circuit based on this information, the secondary coil voltage of the leakage transformer can be set to a specific value at the time of startup.

By forming the voltage detector with a peak hold circuit including a transistor, a resistor, and a capacitor, a signal conforming to a collector voltage envelope of the first semiconductor switching device having a pulse waveform can be generated. The signal can stably control the pulse width of the driving circuit.

The high-frequency heating apparatus of the invention further comprises a direct-current power supply, a coil connected to the direct-current power supply, a capacitor connected to the coil, and a voltage detector for detecting a voltage variation of the capacitor. The voltage detector includes a reference voltage source and a comparator for comparing a detected level of the capacitor voltage with a voltage level of the reference voltage source. The drive circuit is configured to stop operating when the detected level of the capacitor voltage exceeds or falls below the voltage level of the reference voltage source. This enables detection of a large change in the capacitor voltage due to spark generation at the high-voltage portion in the secondary coil of the leakage transformer, allowing prevention of continuous spark generation by stopping the operation of the drive circuit.

By providing a switching circuit for switching the reference voltage source during start-up and normal operation, a voltage change of the capacitor at start-up and a voltage change of the capacitor at the time of occurrence of a spark are distinguished.

The present invention further includes a direct current power supply obtained by full-wave rectifying an alternating current power supply, and an alternating current power supply voltage detector for detecting a voltage of the alternating current power supply. The ac power supply voltage detector includes a series circuit of a diode and a zener diode, and receives an output signal of the ac power supply voltage. This enables detection of abnormal voltage surges due to lightning that may occur in the ac power supply system, and also provides a simpler circuit configuration by sharing a part of the circuit.

The present invention further includes a determination circuit for determining when to output a stop signal to the driving circuit according to the number of times the detected level of the capacitor voltage exceeds or decreases below the voltage level of the reference voltage source. This enables to distinguish between transient overvoltages due to external noise and lightning surges or sparks.

Reference numerals

1 DC power supply

2 leakage transformer

3 first semiconductor conversion device

4 first capacitor

5 second capacitor

6 second semiconductor switching device

7 drive circuit

8 full wave voltage doubler rectifier

9 magnetron

1 IGBT

11, 12 pole tube

13 IGBT

14, 15 capacitor

16 rectifier

17 capacitor

18 diode

19 rectifier

20 leakage transformer

21 center tap

23 polar tube

24 diode

25 pole tube

26 inverting and amplifying circuit

27 amplifier

28 pulse width modulator

29 frequency modulator

30 Oscillator

31 driving pulse signal generator

35, 36 coil

37, 38, 139 capacitor

39 minimum frequency setting circuit

40 frequency modulation conversion circuit

41 Current detector

42 judging circuit

43, 44 switch

45 V_CEDetector

46 capacitor

47, 48 resistors

50 coil

51 capacitor

52 voltage detector

53 reference voltage source

54 comparator

55 switching circuit

56 ac power supply detector

57 diode

58 Zener diode

59 judging circuit

60 capacitor

61 reference voltage source

62 comparator

63, 64 resistance

65 V_DCDetector

Claims (19)

1. A high-frequency heating apparatus for use with a power supply, wherein the power supply is a direct-current power supply supplied from a rectified alternating-current power supply, characterized by further comprising:

a leakage transformer connected to the DC power supply;

a first switching device connected in series with a primary coil side of the leakage transformer;

a first capacitor connected to the first switching device;

a series circuit of a second capacitor and a second switching device, the latter being a semiconductor switching device;

drive means for alternately driving the first switching device and the second semiconductor switching device;

frequency modulation means for modulating a frequency in accordance with a signal generated from a voltage of the power supply supplied to the drive means;

a rectifying device connected to a secondary coil side of the leakage transformer; and

a magnetron connected to the rectifying means and having a plurality of apertures,

wherein the first capacitor, the second capacitor and the leakage transformer form a resonant circuit.

2. The high-frequency heating apparatus according to claim 1, wherein said rectifying means connected to said secondary coil side of said leakage transformer provides one of the following functions:

1) rectifying by a full-wave voltage doubler;

2) rectifying by a half-wave voltage multiplier;

3) full-wave rectification; and

4) for centrally tapping said secondary winding of said leakage transformer and connected to the central tap of the magnetron via a diode.

3. The high-frequency heating apparatus according to claim 1, further comprising a frequency-modulation switching device for switching the frequency modulation means to start operation at a fixed frequency.

4. The high-frequency heating apparatus according to claim 3, wherein said rectifying means includes a diode and a capacitor connected to said secondary coil side of said leakage transformer, and said frequency-modulation switching device includes:

current detection means for detecting a current of at least one of the diode and the capacitor forming the rectifying means; and

and a judging device for determining the oscillation of the magnetron according to the current detecting device and the output signal of the current detecting device.

5. The high-frequency heating apparatus according to claim 4, wherein said frequency-modulation switching device comprises a minimum frequency setting means.

6. The high-frequency heating apparatus according to claim 1, wherein said power supply is a rectified alternating-current power supply, and said high-frequency heating apparatus further comprises pulse width modulation means for modulating a pulse width in accordance with a signal generated based on said power supply voltage.

7. The high-frequency heating apparatus according to claim 1, wherein power is controlled by changing an ON time ratio of a pulse for driving the first switching device to a pulse for driving the second switching device and keeping a period constant.

8. The high-frequency heating apparatus according to claim 1, wherein said leakage transformer further comprises a three-stage coil for supplying a current to a cathode of said magnetron, said cathode of said magnetron having a filter including a capacitor and a coil; and the driving means further comprises start frequency setting means for setting a frequency to reduce the impedance of the coil to a certain degree.

9. The high-frequency heating apparatus according to claim 1, wherein said driving means further comprises voltage detecting means for detecting a voltage applied to said first switching device, and said voltage detecting means controls a pulse width.

10. The high-frequency heating apparatus according to claim 9, the voltage detecting means has peak hold means comprising a transistor, a resistor and a capacitor.

11. The high-frequency heating apparatus according to claim 1, wherein said driving means includes a coil connected to said power supply, a capacitor connected to said coil, voltage detecting means for detecting a voltage change of said capacitor, reference voltage setting means, and comparing means for comparing a detected level of a voltage of said capacitor with a reference level of said reference voltage setting means.

12. The high-frequency heating apparatus according to claim 11, further comprising a switching device for switching the reference level during startup and normal operation.

13. The high frequency heating apparatus according to claim 11, wherein said power supply is a rectified alternating current power supply, and said high frequency heating apparatus further comprises alternating current power supply voltage detecting means for detecting a voltage of said alternating current power supply, said alternating current power supply voltage detecting means comprising a series circuit of a diode and a constant voltage device, and an output signal of said alternating current power supply voltage detecting means is inputted to said voltage detecting means.

14. The high-frequency heating apparatus according to claim 11, further comprising stop judging means for outputting a stop signal to said driving means in accordance with the number of times said detected level of the voltage of said capacitor exceeds said reference level.

15. The high-frequency heating apparatus according to claim 1, wherein said series circuit of said first capacitor and said second switching device is constituted so that said first capacitor is connected in parallel to said primary coil side of said leakage transformer, and said series circuit of said second capacitor and said second switching device is connected in parallel to said primary coil side of said leakage transformer.

16. The high-frequency heating apparatus according to claim 1, wherein the series circuit of the first capacitor and the second switching device is constituted so that the first capacitor is connected in parallel to the primary coil side of the leakage transformer, and the series circuit of the second capacitor and the second switching device is connected in parallel to the first switching device.

17. The high-frequency heating apparatus according to claim 1, wherein the series circuit of the first capacitor and the second switching device is constituted so that the first capacitor and the first switching device are connected in parallel, and the series circuit of the second capacitor and the second switching device is connected in parallel with the primary coil side of the leakage transformer.

18. The high-frequency heating apparatus according to claim 1, wherein the series circuit of the first capacitor and the second switching device is constituted so as to connect the series circuit of the second capacitor and the second switching device in parallel with the first capacitor and the first switching device.

19. The high-frequency heating apparatus according to claim 1, wherein said first capacitor is connected to said leakage transformer.