HK1018550A - Dual polarized array antenna with central polarization control - Google Patents

Description

Dual-polarization array antenna with central polarization control

The present invention relates generally to antennas for the communication of electromagnetic wave signals and, more particularly, to a planar array antenna having a dual-polarization wave radiator and a substantially rotationally symmetric radiation pattern obtained by alignment with a ground plane of sufficiently large radio-electric dimensions.

Diversity techniques at the receiving end of a wireless communication link can improve signal performance without creating additional interference. Spatial diversity techniques typically use two or more spatially separated receive antennas in the horizontal plane of the local terrain. Methods of using physical separation to improve the performance of communication systems are generally limited by the degree of cross-correlation of the signals received by the two antennas and the height of the antennas from the local ground. Diversity techniques can best improve performance when the cross-correlation coefficient is zero.

For example, in a space diversity system using two receive antennas, the physical spacing between the receive antennas is typically greater than or equal to 8 times the nominal wavelength of the operating frequency for an antenna having an antenna height of 100 feet (30 meters). Furthermore, for an antenna having an antenna height of 150 feet (50 meters), the physical spacing between antennas is typically greater than or equal to 14 times. For the above-mentioned spacing, the cross-correlation coefficient of the two-branch spatial diversity system is set to 0.7. At an operating frequency of 850MHZ, a separation factor of 8 wavelengths between the receive antennas can produce a power difference of ± 2dB, which is sufficient for improving the performance of signal reception for diversity techniques. For a communication system operating at 850MHZ, the physical separation between the receiving antennas is approximately 9 feet (3 meters).

For lower frequency applications where the wavelength is large, the problem of site installation has become increasingly impractical. For example, assuming the same height criteria are used, a system operating at 450MHZ requires approximately 18 feet of antenna spacing for equivalent diversity performance. Although the problem of site installation will be alleviated at higher frequencies because the baseline spacing required for diversity performance is reduced, there is a need to reduce the physical presence of the base station antenna to improve the overall performance of the antenna within its operating environment and to increase the economy of site installation.

Antennas of current wireless communication systems typically use vertical linear polarization as a reference or fundamental polarization characteristic for the transmitting and receiving base station antennas. The polarization of an antenna in a given direction is the polarization of the waves emitted by the antenna. For a field vector of a single frequency at a fixed point in space, its polarization state is a property that describes the shape and direction of the trajectory at the end of the field vector, as well as the tangential direction of that trajectory. The orthogonal polarization is the polarization perpendicular to the reference polarization.

For a receive antenna, the spatially diverse antennas typically have the same polarization state with vertical characteristics. When used with a single polarization state antenna, spatial diversity cannot recover signals with polarization characteristics different from the receiving antenna. In particular, signal power polarized orthogonally to the receive antenna polarization cannot be efficiently coupled to the antenna. Therefore, a space diversity system using a single polarization antenna has only a certain effectiveness for receiving orthogonally polarized signals. The performance of spatial diversity is also limited by the effect of angles, which occurs when the angle of arrival of the signals is not perpendicular to the baseline of the spatially separated array, so that the apparent baseline distance between physically separated antennas is reduced.

Polarization diversity may replace the use of space diversity in base stations of wireless communication systems, particularly those supporting Personal Communication Services (PCS) or cellular mobile radio-telephony (CMR). A prerequisite for the potential effectiveness of polarization diversity is that the transmit polarization state of a typical linearly polarized mobile or portable communication unit does not always coincide with the vertical linear polarization of the antenna at the base station, or is not necessarily a linear polarization state (e.g., an elliptical polarization state). For example, depolarization, which is the conversion of power from a reference polarization state to an orthogonal polarization state, may occur in the propagation path between a mobile user and a base station. Multipath propagation is always accompanied to some extent by depolarization of the signal.

By using an antenna with dual polarization states at the same time, polarization diversity of two branches can be achieved. Dual polarization states may allow the implementation of a base station antenna to be reduced from physically separate antennas to a single antenna with dual characteristic polarization states. Dual polarized antennas are typically used in communications between satellites and ground stations. For satellite communications applications, a typical satellite antenna is a reflector antenna having a relatively narrow field of view, typically ranging between 15 and 20 degrees, to cover a certain range of the earth. Dual polarization antennas for satellites are typically multibeam antennas comprising an array of individual energy supplying elements and either a grating reflective optical element having different focal points for two orthogonal linear polarizations or separate reflective optical elements for each of two orthogonal circular polarizations. The ground station antenna typically comprises a high gain dual polarization antenna with a relatively narrow "pencil" beam having a Half Power Beamwidth (HPBW) of a few degrees or less.

The present invention provides polarization diversity advantages by employing an antenna having dual polarization radiating elements arranged in a planar array and having a substantially rotationally symmetric radiation pattern over a wide field of view. The antenna of the present invention substantially maintains rotational symmetry for HPBW between 45 and 120 degrees compared to previous dual polarization antennas. A high orthogonality is achieved between the polarization states of the pair of antennas, irrespective of the viewing angle (look angle) within the field of view of the antennas. The dual polarization states of the antenna may be determined by a centrally located polarization state control network connected to the array of dual polarization state radiators and capable of receiving the polarization states of the received signal and outputting signals having different predetermined polarization states. The antenna of the present invention can be implemented in a compact structure to achieve a low radio-electric space occupancy and can be easily and relatively inexpensively manufactured.

The present invention relates to a dual polarization state planar array antenna having radiating elements that simultaneously have dual polarization states and have a substantially rotationally symmetric radiation pattern. A substantially rotationally symmetric radiation pattern is a co-polarized pattern response that has "pseudo-circular polarization symmetry" and a difference in the main plane (E-and H-) patterns of no more than about 3 for any value of theta within the field of view of the antenna. 1 dB. Alternatively, a substantially rotationally symmetric radiation pattern may be considered a co-polarized pattern response having "pseudo-circular polarization symmetry" and a ratio of orthogonal polarization states less than about-15 dB within the field of view of the antenna. A Beam Forming Network (BFN), typically a distribution network, is connected to each dual polarization radiator and exchanges electromagnetic signals with each of the radiating elements.

The dual polarization state planar array antenna may include a ground plane and a central polarization state control network. This ground plane is generally parallel to the radiating elements and at a predetermined distance from them. The ground plane typically has sufficient radio-electric (radio-electric) range in the transverse plane of the antenna to image the radiating elements over a wide coverage area, thereby making the radiation pattern in the azimuth plane of the antenna independent of any number of radiators. The PCN, which is connected to the distribution network, can control the polarization state of the received signal, which is transmitted by the radiating elements via the distribution network.

More particularly, the present invention provides a dual polarization planar array antenna having radiating elements that both simultaneously have dual polarizations and have a substantially rotationally symmetric radiation pattern. The array radiation pattern includes a first radiation pattern in an elevation plane (tilt plane) of the antenna and a second radiation pattern in an azimuth plane of the antenna. The first radiation pattern is determined by the geometry of the antenna system and the second radiation pattern is determined by the dual polarization radiating element and the characteristics of the ground plane.

Each dual polarized radiating element may be implemented by an orthogonal pair of electric dipoles, i.e. a first electric dipole element perpendicular to a second electric dipole element. Each orthogonal pair of electric dipoles may be parallel to the conductive plane of the ground plane and lie in the vertical plane of the antenna to form a linear array. The orthogonal pair of electric dipoles, together with the ground plane, may have a rotationally symmetric radiation pattern for a linearly polarized electromagnetic signal of any arbitrary direction.

For example, the polarization states of an orthogonal pair of electric dipoles may be left-tilted and right-tilted. These polarization states are orthogonal, thus minimizing the response of the orthogonal polarization of any electromagnetic signal received by the antenna. The polarization state can be maintained over a wide coverage range of at least 45 degrees (half power beamwidth) in the azimuth plane of the antenna.

The BFN includes a distribution network having a first power splitter coupled to each first radiating element having a first polarization state and a second power splitter coupled to each second radiating element having a second polarization state. The pair of distribution networks is connected between the radiating element and the PCN.

The PCN may have a pair of duplexers, a first duplexer and a second duplexer, and a power combiner. The first duplexer is connected to the first power divider and has a first receiving port and a first transmitting port. The second duplexer is connected to the second power divider and has a second receiving port and a second transmitting port. The first and second receiving ports output a received signal in response to an electromagnetic signal received by the radiating element. First and second transmit ports coupled to the power combiner receive a transmit signal.

The PCN may also comprise a hybrid coupler in the shape of a 0 degree/180 degree "ring" connected to the first and second receiving ports of the duplexer. For example, if the antenna includes an array of orthogonal electric dipole pairs with left-tilted and right-tilted polarizations, the hybrid coupler can receive a received signal from the receive port of the duplexer and can output a received signal with vertical linear polarizations. The hybrid coupler is also capable of receiving the receive signals and outputting a receive signal having a horizontal linear polarization in turn.

Alternatively, the PCN may also include a 0 degree/90 degree quadrature hybrid coupler connected to the first and second receiving ports of the duplexer. For an antenna comprising an array of orthogonal electric dipole pairs having left-handed and right-handed polarization states, the hybrid coupler is capable of receiving a received signal from the receive port of the duplexer and outputting a received signal having a left-handed circular polarization state. The hybrid coupler is also capable of receiving the received signals and outputting a received signal having a right-hand circular polarization state in sequence.

As mentioned above, the flexibility of selecting polarization state pairs is determined only by the variation of relatively few elements in the PCN. It will be appreciated that in the case of a number of array elements greater than 2, the PCN of the present invention contains a much smaller number of elements than the number of array elements. Thus, for a given design, the structure and detailed implementation of the antenna is largely the same, and there is flexibility to select the polarization state by changing a few elements. This feature is important for mass production because polarization state diversity may require different polarization state pairs depending on the particular application of the communication system, the type of diversity combiner, and the type of environment (e.g., rural, urban, indoor, etc.). The PCN also enables the antenna to operate in full duplex mode in both transmit and receive modes when the transmit polarization state is different from the dual receive polarization state.

The ground plane may be a solid conductive plane having a length dimension corresponding to the size of the array. Alternatively, the ground plane may comprise a solid conductive plane and a non-solid conductive plane. The lateral extent of the conductive plane of the solid should be sufficiently large to allow a vertical polarization state element to achieve the desired polarization state. In contrast, the non-solid conductive plane comprises a pair of parallel spaced apart conductive elements aligned in the horizontal plane of the antenna and symmetrically disposed along each lateral extent of the solid conductive plane. The lateral extent of the solid conductive plane is approximately one wavelength of the selected center frequency and the spatial distance (center to center) of each grid element is approximately 1/3 or 1/2 wavelengths of the selected center frequency.

The ground plane may also be a substantially planar sheet containing conductive material. Alternatively, the ground plane may be designed as a substantially non-parallel, continuously curved sheet of conductive material or a piecewise curved sheet of conductive material.

Because the electrical centers of the two polarization states of the antenna of the present invention are preferably coincident, the antenna is generally not referred to as a spatially separated application. However, this coincidence of electrical centers takes up only minimal space in the lateral direction and meets the requirements of the present invention to equalize the time delay between signals coupled to each polarization state. A significant advantage of the antenna polarization diversity provided by the present invention is that the size of the antenna and the complexity of the antenna installation are reduced.

In view of the foregoing, it is an object of the present invention to provide an antenna whose radiating elements are characterized by simultaneous dual polarization states and have a substantially rotationally symmetric radiation pattern.

It is another object of the present invention to provide a radiating element of the type employing orthogonal electric dipole pairs arranged in a planar array structure, wherein the direction of the electric dipole pairs is ± 45 degrees with respect to an axis parallel to the antenna.

It is another object of the present invention to provide an array of radiating elements of the electric dipole pair type in combination with a radio-electric ground plane to produce a radiation pattern that is rotationally symmetric, or nearly rotationally symmetric.

The invention will be better understood from the following detailed description, with reference to the drawings, and from the reading of the appended claims.

FIG. 1 is a block diagram of the basic components of a preferred embodiment of the present invention.

Fig. 2 is an illustration of a perspective view of the structure of a preferred embodiment of the present invention.

Fig. 3 is a front view of a preferred embodiment of the present invention.

Fig. 4 is a top view of a preferred embodiment of the present invention.

Fig. 5 is an illustration of a typical mounting structure of an antenna provided by a preferred embodiment of the present invention.

Fig. 6A,6B and 6C, collectively referred to as fig. 6, are alternative aspects and side views of a dielectric substrate of a radiating element in accordance with a preferred embodiment of the present invention.

Fig. 7A,7B,7C and 7D, collectively referred to as fig. 7, are perspective and side views of a radiating element of a preferred embodiment of the present invention.

Fig. 8 is a dimension of one radiating element of the preferred embodiment of the present invention.

Fig. 9A,9B,9C and 9D, collectively referred to as fig. 9, are side, top and perspective views of a combination of a radiating element and a mounting plate of the preferred embodiment of the present invention.

FIG. 10 is a block diagram of a polarization state control network in accordance with the preferred embodiment of the present invention.

FIG. 11 is a block diagram of a polarization state control network in accordance with an alternative embodiment of the present invention.

FIG. 12 is a block diagram of a polarization state control network in accordance with an alternative embodiment of the present invention.

FIG. 13 is a block diagram of a polarization state control network in accordance with an alternative embodiment of the present invention.

FIG. 14 is a block diagram of a polarization state control network in accordance with an alternative embodiment of the present invention.

Fig. 15 is a schematic diagram of a radio-to-electrical ground plane of an alternative embodiment of the present invention.

Fig. 16 is a schematic diagram of a radio-to-electrical ground plane of an alternative embodiment of the present invention.

Fig. 17 is a schematic diagram of a radio-to-electrical ground plane of an alternative embodiment of the present invention.

Fig. 18 is a schematic diagram of a radio-to-electrical ground plane of an alternative embodiment of the present invention.

The antenna of the present invention is used in wireless communication applications such as Personal Communication Services (PCS) and cellular mobile radio-telephony (CMR) services. The antenna uses polarization state diversity techniques to mitigate the deleterious fading effects and disruptions caused by complex transmission environments. The antenna includes an array of dual polarization state radiating elements and a Beam Forming Network (BFN) having a power distribution network for array excitation. Together with the radiating element, the conductive surface acting as a radio-electric ground plane can produce a substantially rotationally symmetrical pattern over a wide antenna coverage area. A polarization state control network (PCN) connected to the array via the distribution hub provides a mechanism for controlling the polarization state.

Those skilled in the art will readily recognize that poor antenna polarization state characteristics can limit the power transfer of available communication systems. Before discussing the embodiments of the antenna provided by the present invention, it is useful to review the salient features of an antenna with dual polarization state characteristics.

In general, the fourier expansion of the far field of an antenna in a standard spherical coordinate system is:wherein E_ΘAnd E_ΦThe components of the electric field in the theta and phi directions in the standard spherical coordinate system are shown. Unit vector u_x,u_y,u_zAre coincident with the x, y, z axes in the corresponding rectangular coordinate system having the same origin.

Typically, the coefficients are complex to include all polarization states and azimuthal phase distributions. For the purposes herein, the group phase and the spreading factor common to both field components are ignored. If the beam has "pseudo-circular symmetry", the field can be accurately represented by a single expansion term (m = 1). For u on the axis of symmetry_yDirectional electric field (E-field), 'pseudo-circularly polarized symmetric' field can be expressed as:E ₁(Θ,Φ)=f₁(Θ)sin(Φ) u _Θ+f₂(Θ)cos(Φ) u _Φ

wherein f is₁(theta) and f₂(Θ) is the section line (cuts) of the principal axis plane normalized field pattern, the variation of which is described by the first order cosine and sine harmonics. Unit vector u_ΘAnd u_ΦRespectively in the theta and phi directions. The above formula assumes a standard spherical coordinate system where the electric field plane (E-plane) is Φ =90 ° and the magnetic field plane (H-plane) is Φ =0 °. On axis of symmetry u_xThe directional electric field (E-field) can be expressed as:

E₂(Θ,Φ)=f₃(Θ)cos(Φ)u_Θ-f₄(Θ)sin(Φ)u_Φ

the condition that the two polarization components satisfy orthogonality is as follows:

E₁(Θ,Φ)·E^* ₂(Θ, Φ) =0 wherein · represents an inner product, and^*representing a complex conjugate. From which the following can be derived:thus, orthogonality may be true regardless of the viewing angle if: f. of₁(Θ)f₃ ^*(Θ)-f₂(Θ)f₄ ^*(Θ)=0

With Θ =0 °, the normalized field component is 1, which satisfies the orthogonality condition. Off the axis of symmetry, there will be some separate condition for the principal axis planar radiation pattern of the two fundamental polarization states that satisfy the orthogonality condition. In general, at each value of Θ, the product of the E plane plots of the two fundamental polarization states should be equal to the product of the H plane plots. The problem can be further simplified if the pattern is assumed to have an equiphase distribution, the only condition for orthogonality being satisfied being that the pattern must be circularly symmetric. When the symmetry of the pattern is reduced, the degree of orthogonality also deviates from the ideal case.

Phi-phi in field equation₀The replacement of → Φ completes the polarization state on the symmetry axis of the antenna from being consistent with the x-y axis in the rectangular coordinate system to being consistent with Φ = ± Φ₀A transition of the coincident axes. When viewed in the + z direction, defining a tilt to the left (SL) as a rotation on the antenna's axis of symmetryu _yE-field of direction, tilting to the right (SR) u, rotated on the antenna's axis of symmetry_xA directional E field, then the field can be expressed as:

the definition of "orthogonal polarization" herein adopts definition 3 of a.c. ludwig, "definition of orthogonal polarization", ieee trans. antenna spopagat, vol.ap-21, pp.116-119, January 1973. Definition 3 describes the isofield lines of the theoretical elementary radiants of the known huygens source. A huygens source is a combination of an equal strength and orthogonal electric dipole and a magnetic dipole. The Huygens source is unique among all mixtures of electric and magnetic dipoles, when it surrounds its axis of symmetry (u)_z) When rotated 90 degrees, the generated field (from all perspectives) is exactly orthogonal to the field generated by the non-rotating source. Thus, it is possible to provideIf two huygens sources (exactly 90 degrees apart in the phi direction in a standard spherical coordinate system) are chosen as the two radiating elements of a dual polarization antenna, they will produce a pair of fundamental polarization states that are always orthogonal (independent of viewing angle). Thus, when the two orthogonal radiators are excited with a certain amplitude and phase weight, the polarization states they produce vary only in the tilt direction with respect to the polarization state on the resultant axis of symmetry and are related to the polarization state on the resultant axis of symmetry.

The huygens source is one of the desirable features for orthogonal radiators in polarization diversity applications. Of course, it is better if the tilt angle is not changed; however, it is difficult to define a constant tilt angle due to the difficulty of establishing a definition of the polarization state. Because the communication link is associated with only a single polarization state for a single user, a fundamental problem associated with providing optimal polarization state coverage performance is the orthogonality of the polarization states. Besides the condition for optimizing the polarization state performance of the antenna, some required pattern characteristics are attached.

For purposes of describing the principal features of the preferred embodiment of the invention, the array of radiating elements lies along the y-axis in a standard angular coordinate system and lies in the x-y plane. The tilt plane of the array is defined as the plane through the beam peak and along the y-axis. The azimuthal plane is perpendicular to the tilt plane and the sectional line of the main planar pattern passes through the beam peak.

The pattern requirements needed for polarization state optimization can be applied to one radiating element alone if the mutual coupling between elements in the array is sufficiently low. The polarization state of the field generated by one huygens source array is the same as the polarization state of the field generated by a single huygens source. Because it is a pattern of an isotropic array of radiators, the array factor has no polarization properties. This is important in the present invention because the intensity of the radiation pattern in the tilt plane can be substantially controlled by the geometry of the array, while the polarization state of the radiation wave can be achieved exactly by the choice of the elements of the array, as can the pattern features in the azimuth plane.

For a linear array, the preferred orientation of the polarization state of the elements is tilted (+ -45 °) relative to the array (y-axis) to achieve the best balance in element pattern symmetry even in the presence of mutual coupling between array elements. The boundary condition of a finite radio-electric ground plane along the long and short axes of the array in one direction is the same as for two orthogonal polarization state elements when the elements are centered on the ground plane.

Definition of 3, on the axis of symmetry of the antennau _yThe reference (co-planar) for the directional E-field and the unit vector for the orthogonal polarization field are defined as: e.g. of the type_ref(Θ,Φ)=sin(Φ)u_Θ+cos(Φ)u_Φe_cross(Θ,Φ)=cos(Φ)u_Θ-sin(Φ)u_Φ

U on the axis of symmetry of the antenna_xThe reference (co-planar) for the directional E-field and the unit vector for the orthogonal polarization field are defined as: e.g. of the type_ref(Θ,Φ)=cos(Φ)u_Θ-sin(Φ)u_Φe_cross(Θ,Φ)=sin(Φ)u_Θ+cos(Φ)u_Φ

The unit vector definitions of the reference and orthogonal polarization states can be obtained for the SL and SR polarization states by substituting phi rotated by 45 deg. and in the same manner as just.

Some of the features of the antenna provided by the present invention are illustrated by discussing the graphical features of an array of polarization states featuring a double tilting element in the azimuth plane phi =0 deg.. First, the electric field distribution can be written in terms of reference and orthogonal polarization components:

the orthogonal polarization state pattern constitutes half of the difference in the main (E-and H-plane) patterns of the radiating element. The zero orthogonal polarization state implies the condition of complete rotational symmetry of the coplanar patterns. The zero orthogonal polarization state corresponds to the orthogonality of the dual polarization state source.

In addition, the oblique polarization field and the symmetry axis of the antenna are u_yThe inner product of the reference polarization states of the directional E-field produces a pattern that normalizes the product factor of half of the coplanar H-plane pattern for the radiating element. U on the symmetry axis of the oblique polarization field and the antenna_xThe inner product of the reference polarization states of the directional E-field produces a pattern that normalizes the product factor of half of the coplanar E-plane pattern for the radiating element. Except only when the radiating element is patternedThe coverage in the azimuthal plane is the same except for a constant factor of all half for full rotational symmetry. In addition to the constant factor, having the same pattern distribution is an important feature of antennas used in a communication system that uses polarization state diversity techniques. Otherwise, when the collimation deviates up to 45 °, the magnitude difference of the polarization state coupling of a linearly polarized signal to the linearly polarized antenna will be larger than the ideal polarization state mismatch, resulting in non-optimal polarization state diversity performance. When polarization orthogonality exists, the coupling strength of this polarization is weakened, i.e., coupling becomes smaller relative to the ideal case, resulting from the degree of orthogonality.

Another feature of a rotationally symmetric radiation pattern is that the azimuthal pattern characteristics of the array will remain unchanged when two beams corresponding to the polarization states of the dual polarizing element characteristics are combined together with different weights to form polarization state pairs different from the polarization state of the natural element. This capability is a very interesting application area of the present invention. Although the example used to illustrate the principal polarization state feature is the case of linear polarization, it is true for other polarization state pairs as well. It is to be understood that dual circular polarization states (right-handed and left-handed) may also be used in wireless communication systems employing polarization state diversity.

Turning now to the drawings, wherein like reference numerals refer to like elements, FIG. 1 is a block diagram illustrating the primary elements of a preferred embodiment of the present invention. Referring to fig. 1, an antenna 10 exchanges electromagnetic signals using a high frequency spectrum associated with a conventional wireless communication system. The antenna 10 may be a planar array of radiator elements 12, known as wave generators or radiators, wherein the array is oriented parallel to the vertical plane of the antenna when the antenna is viewed from the front. For the preferred linear array embodiment, when not in plan view (mechanical or electrical), the array factor primarily forms coverage in the tilt plane, which coverage in the azimuthal plane is primarily affected by the element pattern features. In general, such a linear array can be classified as a fan beam antenna, which produces a large proportion of the major and minor dimensions of the transverse cross-section of the main lobe.

An antenna 10 capable of transmitting and receiving electromagnetic signals includes a radiating element 12, a ground plane 14, a Beam Forming Network (BFN)16, and a polarization state control network (PCN) 18. The radiating element 12 is a wave generator comprising elements 12a and 12b having dual polarization states, preferably arranged in a linear array and above and at a predetermined distance from the conductive surface of the ground plane 14. The radiating element 12 operates in series with the ground plane 14 to provide the antenna 10 with the desired pattern characteristics. The radiation pattern of the antenna 10 has substantial rotational symmetry, which for ease of illustration is defined as having "pseudo-circular symmetry" and the difference between the main plane (E-and H-plane) patterns does not exceed roughly 3 for any value of theta over all fields of view of the antenna. 1dB of common polarization state pattern response. Alternatively, a highly rotationally symmetric radiation pattern may be defined as a common polarization pattern response having "pseudo-circular symmetry" and a ratio between orthogonal polarization states of less than approximately-15 dB over all fields of view of the antenna. For the preferred embodiment of the antenna 10, a linear array of dual polarization state radiating elements has a rotationally symmetric radiation pattern over a wide field of view, typically in the range of 45 to 120 degrees of Half Power Beamwidth (HPBW).

BFN16, which functions as a distribution network, is connected to the radiating elements 12a and 12b to transmit receive signals from and transmit signals to the radiating elements. The PCN18 coupled to the BFN16 can control the polarization state of the received signal as distributed by the BFN 16. Because the radiating element 12 has dual polarization states, the PCN18 can receive a received signal having either polarization state and output an electromagnetic signal having a polarization state P1 at the first output port 22 and a polarization state P2 at the second output port 24.

Since the antenna 10 is generally used for PCS and CMR applications, those skilled in the art will readily recognize that the best characteristics of the radiating element 12 are generally high efficiency, wide radiation pattern, high polarization purity, and sufficient operating bandwidth. It would be further desirable if the radiating element 12 were lightweight, inexpensive, capable of interfacing directly with BFN16, and capable of being integrated with the antenna package. Dipole antennas meet all of these electrical performance requirements and printed circuit embodiments meet physical requirements. As will be described in detail below with reference to fig. 6, the preferred embodiment of each radiator 12a and 12b is a dipole type antenna having polarization states that are tilted left (SL) and right (SR).

Fig. 2 is a perspective view of the major components of the antenna 10 highlighting the preferred structure of the antenna. Fig. 3 and 4 are front and top views, respectively, of the antenna 10. Referring to fig. 2-4, each radiating element preferably includes two dipole antennas, each having a pair of dipole arms and a dipole base, which are placed together to form an orthogonal dipole pair. The electrical centers of the orthogonal dipole pairs coincide, thereby minimizing any phase delay caused by feeding the dipole antennas. Each orthogonal dipole pair is located above the front conductive surface of the radio-electric ground plane provided by ground plane 14. In particular, the pair of orthogonal dipoles is mounted on a conductive surface of a capacitive plate 20, which in turn is mounted to the ground plane 14. The orthogonal dipole pairs are oriented such that the dipole feeds are located at the base of the dipole, the base of the dipole arms representing the maximum distance that any point on the dipole is spaced from the ground plane. The dipole arms scan in an inverted "V" shape down toward the ground plane 14. The height of the dipole arms above the surface of the ground plane 14 and the angle of the dipole arms can be optimized to obtain a very high rotational symmetry radiation pattern characteristic in the forward direction above the ground plane 14. The preferred dimensions of an antenna design dipole antenna with 90 degree half power azimuth beamwidth and its feed lines are described in detail below with reference to fig. 8.

BFN16 is supported by the leading electrical surface of ground plane 14 and distributes electromagnetic signals to and from the dipole antenna of radiating element 12. The BFN16 uses a pair of distribution networks for a dual polarization state array configuration, one for each polarization state. BFN is preferably a microstrip line design and provides suitable impedance matching between each radiating element 12 and the PCN 18. In addition, BFN16 preferably contains a power divider to distribute the signal to each radiating element 12.

The PCN18 is supported by the leading electrical surface of the ground plane 14 and is located centrally of the antenna assembly and is connected between the distribution network BFN16 and a pair of antenna ports 22 and 24, each of which is connectable to a feeder cable. The PCN18 distributes electromagnetic signals to and from the radiating elements 12 via the BFN16 and provides a complex weight (amplitude and phase) to these signals. In a preferred embodiment, the PCN18 is designed as a polarization state control mechanism with at least four external interfaces connected to the transmission line. Two of the four external interfaces are connected to the distribution network BFN16 and the remaining two external interfaces are connected to the antenna ports 22 and 24, which in turn are connected to feeder cables to connect the source to the antennas.

While it is preferred that the PCN18 be mounted within the antenna module, it is understood that the PCN18 may be mounted outside of the antenna base. If the PCN18 is not installed in the antenna 10 module, the distribution network BFN16 may provide a suitable impedance match between the radiating element 12 and each of the feeder lines connected between the antenna ports 22 and 24. In this embodiment, each antenna port 22 and 24 corresponds to one of two polarization states, thereby suppressing signal reflections along this transmission line. It will be appreciated that the PCN18 can be mounted within the antenna 10 module or outside the antenna base depending on the particular application for the antenna. For example, the PCN18 may be mounted at a mounting point of the base, while the combination of the radiating element 12, ground plane 14, and BFN16 may be mounted within an antenna module at the antenna.

The conductive surface of the ground plane 14 serves as the whole antenna module and as a structural component of the radio-electric ground plane forming the dipole element. The preferred embodiment of the ground plane is a solid, substantially flat sheet of conductive material. The radio-electrical dimension (width) of the transverse plane of the antenna array is approximately 5/3 wavelengths to form radiator elements over a wide field of view (typically greater than 60 degrees) without the limited boundary conditions of the conductive ground plane 14 significantly affecting the radiation characteristics. When the radio-electric dimensions of the ground plane 14 satisfy the above conditions, the orientation of the radiating element 12 can be rotated and aligned with the main axis of the array without seriously degrading the rotational symmetry of the antenna radiation pattern. However, the preferred and optimal orientation is such that the natural antenna axis of symmetry polarization is 45 ° with respect to the main plane of the array.

Empirically derived data demonstrates that while larger lateral dimensions generally result in reduced power of the back radiation pattern, they do not contribute significantly to improved rotational symmetry. For some applications, a low power radiation pattern in the backward direction, called the back lobe (backsobe), is desirable, but the degree of reduction of the back lobe needs to be traded off against the increase in size, weight, cost, and wide load characteristics.

Measurements made on radio-electric ground planes with smaller lateral dimensions show that the lateral dimensions are approximately 1. 5 wavelengths can result in undesirable divergence of the beamwidth of the pattern. And a ground plane with a smaller lateral dimension enables the azimuthal beamwidth to be more sensitive to the number of array elements. This disadvantage is also accompanied by a deviation from the desired rotationally symmetrical radiation pattern.

Measurements also show that the radio-electric dimension of the ground plane 14 in the array transverse plane can be made much smaller than the above criteria for radiators perpendicular to the direction parallel to the array plane, without making the azimuth beamwidth very sensitive to dimensions over a wide range of smaller values. However, horizontally polarized components (physical or synthesized via PCN) do not have this independence. Since dual polarization states are required in this application and preferably with coincident electrical centers, the two polarization states need to adopt the same dimensional criteria, with the condition of the horizontal polarization component being a determining factor.

A protective radome 26 comprising a thermoplastic material may contain the combination of the array of radiating elements 12, BFN16, PCN18, each capacitive plate 20, and the front conductive surface of the ground plane 14. A radome 26 is attached to the edge of the ground plane 14 by screws 28 and surrounds the leading electrical surface of the ground plane 14 and the components mounted thereon. The sealing of the antenna within the enclosed housing formed by the ground plane 14 and the radome 26 may protect the antenna elements from environmental influences such as direct sun exposure, direct water, contaminants, debris, and humidity. The thermoplastic material contained by the radome 26 is preferably a thermoplastic material such as that available from kleerdexcompany of aiken, south carolina under the trademark "KYDEX", such as "KYDEX 100" acrylic PVC alloy sheet.

The antenna may be mounted to a mounting post via a pair of mounts 30 that are attached to the rear conductive surface of the ground plane 14. A staple (not shown) may be used with the mounting bracket 30 to secure the antenna module to a mounting post. Although the preferred mounting structure for the antenna 10 is via a mounting post, it will be appreciated that many other conventional mounting methods can be used to support the antenna 10, including tower supports, sectional supports or other separate elements. One exemplary installation of the antenna 10 is shown in fig. 5, and fig. 5 will be described in detail below.

The antenna ports 22 and 24 are preferably designed as coaxial cable compatible receptacles, such as N-type receptacles, and are connected to the rear surface of the ground plane 14 via capacitive plates 32 and 34. Each capacitive plate 32 and 34 comprises a conductive plate and a dielectric layer adjacent to the conductive plate that is approximately the same size as the conductive plate. When the antenna module is installed, the conductive sheet is installed adjacent to the coaxial cable compatible jack of each port 22 and 24, and the dielectric layer is sandwiched between the ground plane 14 and the conductive sheet to form a sandwich structure. In this manner, the radio-electrical connection of the current path between the antenna ports 22 and 24 and the ground plane 14 is a "capacitive coupling". The area of the conductive sheet is large enough to achieve a low impedance in the operating frequency range. The dielectric layer acts as a Direct Current (DC) barrier by avoiding direct metal-to-metal connections between the antenna ports 22 and 24 and the ground plane 14. This capacitive coupling to reduce the effects of passive cross modulation is U.S. patent application serial NO, filed on 27/2/1995, owned by the same assignee. 08/396,158, which are herein incorporated by reference.

The antennas shown in fig. 2-4 are primarily intended to support communication services in the 1850-1990MHZ Personal Communication Services (PCS) frequency range. However, it will be appreciated by those skilled in the art that the dimensions of the antenna may be varied to support typical cellular telephone communications applications that preferably operate within the frequency range of approximately 805 and 896 MHZ. Similarly, the size of the antenna may be varied to support European communications services, including Global System for Mobile communications (GSM) with a frequency range of 870-960MHz or European PCS with a frequency range of 1710-1880 MHz. These frequency ranges represent examples of the antenna operating frequency ranges, but may extend below or above these frequency ranges associated with PCS applications.

Importantly, the antenna 10 shown in fig. 1-4 is a planar array of radiating elements having dual polarization states and has a radiation pattern with substantial rotational symmetry over a wide field of view. For example, the illustrated antenna design has a 90 degree HPBW in the azimuth plane of the antenna, which is achieved by the combination of a dual polarization state radiating element and a ground plane. In contrast, the half-power beamwidth in the tilt plane is pre-realized by the antenna array size, i.e., the number of radiating elements and the inter-element distance in the planar array. Although the antenna shown in fig. 1-4 has a 90 degree HPBW, other embodiments may have an HPBW beamwidth of from 45 degrees to 120 degrees. It is important that the relevant embodiment of the antenna 10 should have a radiation pattern with a high rotational symmetry within an HPBW of at least 45 degrees.

Fig. 5 is an illustration of a typical installation of an antenna 10 for an antenna system of a PCS system. Fig. 5 emphasizes that the antenna 10 is very useful in a sectored cell structure where the azimuth coverage is divided into K independent cells. In this representative example, there are three antennas, the point of the three sectors (K =3) of antennas 10a,10b and 10c being located at the center of the base station, each antenna having a coverage of 120 degrees (radians) in the azimuth plane, the effective coverage radius of which is determined by the gain of the antenna, the altitude, and the angle of depression of the beam. The antennas 10a,10b and 10c are mounted to a mounting post 40 via top and bottom mounting brackets 42 secured to the rear surface of each antenna. Although the antenna 10 of fig. 5 uses a mounting post, it will be appreciated that the mounting hardware may be used to attach the antenna module flat against the side of the modular support frame, as well as to mount the module to a post or a tower supported cylindrical structure.

The example of fig. 5 illustrates that the large antenna structure that would normally be required to physically separate the antennas is replaced when the station switches from space diversity to polarization diversity techniques. With the polarization diversity feature of the preferred antenna, three antenna modules can be mounted to a single mounting post with mounting hardware to achieve three-sector coverage. It will make the antenna module coverage area on the ground smaller, which is very advantageous, and its impact on the visual environment will be smaller than in current space diversity technology systems.

Fig. 6, which includes fig. 6A,6B, and 6C, illustrates a front view, a side view, and a back view, respectively, of a dielectric slab supporting a preferred embodiment of a radiating element. Referring first to fig. 6C, a dipole antenna 52 of each radiating element 12 is formed on one side of a dielectric plate 50 and metallized to form the necessary conductive strips for a pair of dipole arms 54 and dipole bodies 56. Dipole antenna 52 is lithographically (also referred to as a photomask) formed on the dielectric substrate of dielectric plate 50. The width of the conductive strips forming the dipole arms 54 is selected to provide sufficient bandwidth of operating impedance for the radiating elements. The dipole arms 54 occupy the same surface on which the dipole body 56 is located, the dipole body 56 comprising a pair of parallel conductive strips electrically connecting the dipole arms 54 to the capacitive plate 20 (fig. 2). The capacitive plate 20, which will be described in detail later with reference to fig. 9, serves as a mechanical support and forms a radio-electrical connection between the orthogonal dipole pair and the conductive surface of the ground plane 14. The length of these conductive strips at the orthogonal position forming a feed line 58 (fig. 6A) on the back of the dielectric plate is approximately one quarter of the center wavelength of the selected band and serves as a balun. The width of these conductive strips is increased and close to the base of the dipole element to provide a better radio-to-electrical ground plane for the microstrip feed line 58 (fig. 6A) on the back side of the dielectric slab.

As shown in fig. 6A, on the back side of the dipole antenna 52, there is a feed line 58 in the form of a microstrip line that couples energy to the dipole arms 54 (fig. 6C). This microstrip feed line 58 is lithographically printed on the surface of the dielectric plate 50 as before. The feed line 58 terminates in an open circuit, the open end of the feed line 58 having a length, as measured from the crossover location, of approximately one quarter of the center wavelength of the operating frequency band. The preferred embodiment of feed line 58 is a 50 ohm impedance from the base of dipole antenna 52 (fig. 6C) to the area near the crossover point.

As shown in the side view of fig. 6B, the dielectric plate 50 is a relatively thin piece of dielectric material that may be a wide variety of low loss dielectric materials used in radio-electric circuits. The preferred embodiment material is MC-5, which has a low tangent loss, a relative dielectric constant of 3.26, is relatively non-hygroscopic, and is relatively inexpensive. MC-5 is manufactured by Glasteel Industrial amines, a division of Alpha corporation in Collierville, Tennessee. Lower priced alternatives such as FR-4 (an epoxy glass mixture) are hygroscopic and must generally be sealed to adequately prevent water absorption when used in outdoor environments. Water absorption will reduce the wear properties of the material. Higher priced polytetrafluoroethylene (Teflon) based materials can also be used as a substitute, but there are no obvious advantages.

Although the preferred embodiment of each radiating element 12 is in the form of a printed dipole antenna, it will be appreciated that other dipole antenna embodiments can be used to construct the antenna 10. Other conventional dipole antenna embodiments may also be used to construct the antenna 10. Further, it is also understood that the radiating element 12 can function as an antenna without employing a dipole antenna.

Fig. 7A,7B,7C and 7D, collectively referred to as fig. 7, illustrate different views of orthogonal dipole pairs. Referring first to fig. 7A and 7B, the non-metallized dielectric plate center portion of each dielectric plate 50 has a slot 60 that separates the parallel strips of dipole body 56. A set of crossed slots 60 in a pair of dielectric slabs 50 implements a pair of dipole antennas 52 oriented perpendicular to each other in physically orthogonal directions. As shown in fig. 7C and 7D, the microstrip feed lines 58 are alternately disposed at the crossing regions in an up-and-down configuration to prevent the two feed lines from contacting each other. The pair of orthogonally oriented dipole antennas 52 has substantially the same characteristics except for the differences in detail near the intersection area of the feed lines 58. The width of the dipole bodies 56 is varied to provide an equivalent identical impedance match to a reference location on the base of the radiating element.

Referring now to fig. 8, fig. 8 is a preferred size of a dipole antenna structure for the PCS frequency range, wherein each radiating element 12 contains dipole arms 54 having a swept-down design to form an inverted "V" shape. The height of the dipole arms from the ground plane 14 when mounted is approximately.26 wavelengths. The angle of the dipole arms 54 is approximately 30 degrees. The overall extent of the pair of dipole arms 54 is approximately one-half wavelength and its width is approximately. 38 wavelength. The height of the apex of the lower side of the dipole arm 54 and the height of the body 56 are. 19 wavelengths. The center of gravity of dipole arm 54 is near the apex of dipole antenna 52 and its height is approximately. 22 wavelength. It will be appreciated that the width of the dipole arms 54 is predetermined from the perspective of the frequency band range. For example, a narrow dipole arm has a small operating impedance band. Furthermore, it will be appreciated that details of the geometry of the lower edge of the dipole arm 54 and the apex of the body 56 do not significantly affect antenna performance other than impedance characteristics.

Fig. 9A, B, C and D, collectively referred to as fig. 9, illustrate different views of a preferred structure for mounting orthogonal pairs of radiating elements onto a radio-electric ground plane. Referring to fig. 9, the radio-electrical connection of the current path between each dipole 52 and ground plane 14 is formed by a capacitive coupling connection. In particular, a capacitive plate 20 connects each dipole 52 of an orthogonal dipole pair to the conductive surface of the ground plane 14. The capacitive plates can be nested together to make production easier. Capacitive plate 20 has a conductive plate 70 and a dielectric layer 72. The conductive surface area of the conductive plate 70 is sufficiently large to provide a low impedance path in the operating frequency band. This thin dielectric layer 72 serves two functions: blocks Direct Current (DC) and may be bonded on both sides to mechanically constrain the location of the orthogonal dipole pair module on ground plane 14. The capacitive plate 20 avoids the use of a direct connection to a metal-to-metal junction that produces passive cross-frequency modulation when operating at high radio-power levels, such as several hundred watts.

The conductive plate 70 is preferably a tin plate copper structure to form a mechanically supported orthogonal radiator pair and has a desired shape to enable soldering of the electrical connection of the conductive microstrip connecting the capacitive plate to the microstrip of the dipole body. For the preferred embodiment designed for PCS applications, the conductive plate 70 is approximately 0.010-0.020 inches thick. The dielectric layer 72 preferably has St. Both sides of a dielectric material called ScotchVHB manufactured by 3M company of Paul, Minnesota is adhesive. For the preferred embodiment, the dielectric material is selected to be 0.002 inches thick and at least as wide as the capacitive plates, and is preferably cut to the same width as the capacitive plates.

Fig. 10 is a block diagram of the PCN preferred components of the antenna 10. Referring now to fig. 10, the preferred PCN and a pair of duplexers 80 and 82, and a power combiner 84. Each duplexer 80 and 82 is connected between BFN16 and power combiner 84. In particular, diplexer 80 is connected to the distribution network of radiating elements 12 having a polarization state that is tilted to the left, and diplexer 82 is connected to the distribution network of radiating elements 12 having a polarization state that is tilted to the right. In response to a receive signal having a left-hand tilted polarization state from distribution network BFN16, duplexer 80 outputs the receive signal via an output port. In response to the received signal from distribution network BFN16, diplexer 82 outputs the received signal with the polarization state tilted up and down through an output port. Power combiner 84 receives a transmit signal from a transmitting source and passes the signal to duplexer 80 and duplexer 82. Diplexer 80 and diplexer 82 can receive the transmit signal from power combiner 84 and, in turn, output the transmit signal to BFN 16. Upon application of equal phase excitation of the two fundamental polarization states, the antenna 10 radiates an equivalent vertical polarization state signal.

It will be appreciated that the antenna 10 is not limited to receiving signals having a right-angled polarization and a left-angled polarization and transmitting signals having a vertical polarization. As shown in fig. 11, a PCN18a includes a first polarization state control block 81 that receives a pair of transmit signals from a transmit source and a second polarization state control block 83 that outputs a pair of receive signals. The first polarization state control block 81 and the second polarization state control block 83 are connected to the duplexers 80 and 82. In response to transmission signals TX1 and TX2, first polarization state control module 81 outputs transmission signals to duplexers 80 and 82. In addition, the duplexers 80 and 82 output the receiving signals to the second polarization state control module 83, and the second polarization state control module 83 outputs the receiving signals RX1 and RX2 in turn. In this manner, the four ports of the pair of duplexers 80 and 82 may be combined to provide the desired transmit and receive signal pairs. The polarization state control blocks 81 and 83 may be implemented as a 0/90 hybrid coupler of the type commonly referred to as a quadrature hybrid coupler, or as a 0/180 hybrid coupler of the type commonly referred to as a ring hybrid coupler.

FIG. 12 is a block diagram of another alternative embodiment of a polarization state control network. Referring now to FIG. 12, a PCN18b includes a 0/180 hybrid coupler 85, a duplexer 86, and Low Noise Amplifiers (LNAs) 87a and 87 b. The hybrid coupler 85, which is connected to the BFN16, duplexer 86, LNA87a, carries signals to and from the distribution network BFN 16. Further, the hybrid coupler 85 outputs a reception signal having a horizontal polarization state to the LNA87a and outputs a reception signal having a vertical polarization state to the duplexer 86. Duplexer 86 has a common port connected to hybrid coupler 85, a receive port connected to LNA87b, and a transmit port. The common port of the duplexer 86 receives the reception signal having the vertical polarization state from the hybrid coupler 85 and distributes the transmission signal having the vertical polarization state to the hybrid coupler 85, the reception port of the duplexer 86 outputs the reception signal having the vertical polarization state to the LNA87b, and the transmission port receives the transmission signal having the vertical polarization state. Thus, it will be appreciated that the duplexer 86 may separate the received signal from the transmitted signal based on the spectral characteristics of the signal. The LNAs 87a and 87b connected to the hybrid coupler 85 and the duplexer 86, respectively, amplify the received signal to improve the signal-to-noise performance of the signal. The LNA87a amplifies a reception signal having a horizontal polarization state, and the LNA87b amplifies a reception signal having a vertical polarization state. It is understood that LNA87a and LNA87b may be eliminated from the structure of PCN18b when the PCN is located at the receiver side of the wireless communication system rather than at the antenna.

A PCN implemented with hybrid couplers may perform the mathematical function of transforming the dual linear tilted polarization state (SL/SR) of the preferred embodiment to a vertical/horizontal (V/H) polarization state pair or a right-handed circular polarization/left-handed circular polarization (RCP/LCP) pair, respectively. These polarization state transitions do not change the antenna azimuth pattern beamwidth of the coplanar radiating element when the radiation pattern is rotationally symmetric. A necessary condition for using these hybrid couplers to achieve conversion of the polarization state without changing the beamwidth is that the group electrical path (phase delay) lengths of the paths corresponding to the natural characteristic polarization state of the excitation antenna array have been reasonably matched. This same matching condition is essential for the amplitude characteristic.

FIG. 13 is a block diagram of another alternative embodiment of a polarization state control network. Referring now to fig. 13, a PCN18c includes a 0/180 hybrid coupler 88 of the type and switches 89a-d that provide four polarization states, specifically vertical, horizontal, left-angled and right-angled, for polarization state diversity. The common port of switches 89a and 89b is connected to distribution network BFN 16. In addition, the normally closed ports of switches 89a and 89b are connected to hybrid coupler 88, while the normally open ports thereof are directly connected to switches 89c and 89 d. In a similar manner, the normally closed ports of switches 89c and 89d are connected to hybrid coupler 88, while the normally open ports thereof are connected directly to switches 89a and 89 b. The common port of switches 89c and 89d serves as an output port for providing a received signal having a selected polarization state.

For the normally closed state of switches 89a-d, hybrid coupler 88 is inserted into operation of PCN18c, while the normally open state of switches 89a-d is used to avoid coupler 88. Thus, for a normally open state, the common port of switches 89c and 89d functions to provide a received signal having left-slanted and right-slanted polarization states. In contrast, for the normally closed state, the common port of switches 89c and 89d serves to output a received signal having vertical and horizontal polarization states. This can allow the user to select a desired polarization state for the received signal at the base station receiver.

The switches 89a and 89b may be implemented as single pole double throw switches and the switches 89c and 89d may be implemented as single pole double throw switches or a single pole four throw switch.

FIG. 14 is a block diagram of another alternative embodiment of a polarization state control network. As shown in fig. 14, PCN18d includes more than one element to allow polarization state switching without changing the beamwidth of the pattern in the event of amplitude and/or phase imbalance between two natural polarization state elements. PCN18d may be divided into a variable power distribution network where the relative phase shift of phase shifters 96 and 98 determines the power distribution among the PCN ports. PCN18d includes a pair of hybrid couplers 90 and 92 connected in between by a transmission module 94 for applying an unbalanced phase shift. Hybrid coupler 90, which is a preferred embodiment of an 0/90 degree type hybrid coupler, is functionally connected between input ports 1 and 2 and transmission module 94. The hybrid coupler 92, which is preferably implemented as a-0/180 degree type hybrid coupler, is functionally connected between the outlets 3 and 4 and the transmission module 94. A pair of phase shifters 96 and 98 inserted in the transmission lines of the transmission module 94 may form a phase shift between the hybrid couplers 90 and 92. The preferred embodiment of phase shifters 96 and 98 can be transmission lines of different lengths, i.e., a passive phase shifter, or a variable phase shifter capable of controlling the phase shift between hybrid couplers 90 and 92 as shown in fig. 14. In addition, a pair of phase shifters 100 and 102 may be inserted between the input port and the hybrid coupler 90 to fully control the phase shift of the signal entering the PCN18 d. This configuration of PCN18d allows complete control over the combination of polarization states, for example, any two orthogonal polarization state pairs can be generated as the characteristic polarization state of the antenna. If one or more of the passive phase shifting elements is replaced by a controllable phase shifter, flexibility in polarization state can be achieved without changing the pattern beamwidth.

Referring again to fig. 2-4, for PCS frequencies, the horizontal radio-to-electric dimension of the ground plane is scaled to 10 inches (5 λ)₀/3) to obtain the desired polarization state properties. When this parameter is changed to accommodate lower frequencies, such as the typical cellular mobile radio-telephone band centered at 851MHZ, the lateral physical dimensions of the radio-electric ground plane increase. For this typical cellular frequency, the equivalent lateral dimension of the ground plane 14 is approximately 22.5 inches. The size of the array plane is changed in the same way to get the same antenna directivity and to make the number of array elements constant. It will be appreciated that it is desirable to minimize the lateral physical dimensions to reduce wind load and cost, and to improve overall performance by reducing antenna size.

Fig. 15 is an illustration of an alternative embodiment of the ground plane of the antenna 10 a. Referring to fig. 1 and 15, it will be appreciated that for an array in which the horizontal component lies in a transverse plane, the transverse dimension of a radio-to-electrical ground plane is excited by the pattern and polarization state characteristics of the horizontal polarization state component. The electromagnetic boundary conditions for the horizontal polarization state can be met without seriously affecting the performance of the vertical polarization state components. This can be achieved by using a non-solid conductive surface beyond the minimum lateral dimension required to achieve the required performance characteristics of the vertical polarization state component. The non-solid conductive surface, as shown by the gate posts 110a and 110b of fig. 15, generally comprises a pair of gate posts each having the same size and parallel conductive elements 112. The posts 110a and 110b are located in the horizontal plane of the antenna 10a and symmetrically at both edges of the antenna to form the lateral dimension of the antenna, i.e., the sides of the ground plane 14 a. Typical construction techniques for each of the gate posts 110a and 110b may be an array of metal wires, metal posts, metal tubes, and metal strips. Radome 26a contains slots to receive the top ends of each of the column elements 112 of columns 110a and 110 b.

Test data demonstrates that for most geometries, the energy of the vertical (perpendicular) polarization state is hardly affected by the gate posts 110a and 110 b. The center spacing (S) of the elements 112 of each grid column is approximately S = λ₀3 to lambda₀/2. This element spacing allows the columns 110a and 110b to effectively act as extensions of the ground plane 14a and avoids introducing large transmission losses to parallel (horizontal) polarization state components.

If the grid post elements 112 are implemented as conductive strips facing outwardly along the edges and facing the surface of the antenna 10a, the loss of the transmitted signal of the component with parallel polarization states is greater and the reflectivity of the equivalent conductive surface is increased. Thus, it can be appreciated that a tradeoff can be made between center-to-center spacing and depth to achieve the desired performance.

For PCS frequencies, empirical data indicate that a solid ground plane 14a with a lateral dimension of 4-6 inches will provide good performance for vertical polarization state components. For a physical implementation of this ground plane 14a, the length of the grid post elements 112 of the pair of horizontally oriented grid posts 110a and 110b should be approximately 2-3 inches to provide the desired polarization state and coverage equivalent to a radio-electric ground plane having a solid conductive surface size of 10 inches.

For a cellular frequency of 851MHZ center frequency, a solid conductive surface ground plane 14a of nominal 12 inch lateral dimension is believed to provide good electrical performance and reasonable wind load characteristics along with a pair of horizontal pickets 110a and 110b having a picket element length of 6 inches. Thus, the preferred structure for a 851MHZ radio-electric ground plane uses a hybrid system shown in fig. 15 that includes a solid conductive surface and a pair of posts adjacent to the solid conductive surface.

Another advantage of using grating columns is that the superposition of the equiphase field at each segment edge geometry at the back of the antenna array is partially destroyed, thus effectively increasing the front-to-back ratio pattern envelope performance for most signal polarization states.

At lower frequencies of operation, the use of an array of grid column elements is more important from a practical physical implementation standpoint. For example, for 450MHZ, the effective transverse radio-to-electrical dimension of the ground plane should be approximately 43 inches. Using the principles of the present invention, a radio-electric ground plane can be implemented with a solid conductive surface having a general dimension of 22 inches, together with an array of a pair of pillar elements, each extending along the length of a parallel side of the solid conductive surface by a general dimension of 10.5 inches.

Fig. 16 and 17 are alternative embodiments of a radio-electric ground plane using the antenna of the present invention. Referring now to fig. 1,16 and 17, fig. 16 illustrates an antenna 10b having a curved ground plane 14b, while fig. 17 illustrates an antenna 10c having a piecewise curved ground plane 14 c. The ground plane 14b is a convex shaped conductive surface in which the radiating element 12, BFN16, and PCN18 may be centrally mounted along the outer edge of the convex shape of the structure of the semicircular radio-electric ground plane. In contrast, the ground plane 14c of the antenna 10c is a conductive surface having a piecewise curved shape composed of a central horizontal element and a pair of azimuth elements extending along each side of the central horizontal element. Although the radiating element 12 is preferably supported by the horizontal element of the ground plane 14c, the BFN16 and PCN18 may be supported by the horizontal surface of the central element and the azimuthal plane of the edge element. The nature of the bending of the ground planes 14b and 14c is to reduce the influence of the limited boundaries of the conductive surface of the radio-electric ground plane on the antenna radiation characteristics.

Referring now to fig. 18, a solid ground plane 122 thereof has a depth of approximately one quarter wavelength (λ) of the center frequency of the operating band on both sides₀/4) the antenna 10d with one or more choke grooves 120 can reduce the overall edge diffraction coefficient of the horizontal polarization state components and provide coverage pattern and polarization state performance similar to larger sized radio-to-electrical ground planes. By making the opening of choke groove 120 level with the plane defined by the surface of the conductive plane of ground plane 122, the size of ground plane 122 may be reduced to approximately one wavelength (λ)₀). Choke groove 120 comprises a section of transmission line in the shape of a parallel plate and is short-circuited at approximately one quarter wavelength at the exit. The parallel plate transmission line can be folded at the back of the radio-electric ground plane to reduce the depth of the whole module. As shown in fig. 18, a single choke groove 120 along the edge of the array's major axis is simply constructed perpendicular to the plane without folding.

There will be a greater improvement in performance when there is more than one choke groove along the main axis of the antenna. However, when the width of the parallel plates is typically one tenth of the wavelength (λ)₀10), even if the module is given increased thickness and has two or more grooves on each side, the advantage of reduced size will disappear and tend to be full size (5 λ)₀And/3) a ground plane. The increased module complexity of fabricating two or more choke grooves on each side would be unattractive compared to the simplicity of solid or hybrid solid/non-solid ground plane implementations.

It is intended that only the following claims define the scope of the invention and that the foregoing description be considered as illustrative of various embodiments of the invention. In particular, the scope of the invention may extend beyond any of the detailed embodiments described in this specification.

Claims (32)

1. An antenna system for transmitting and receiving electromagnetic signals with polarization state diversity, comprising:

a plurality of dual-polarization state radiators characterized by radiation patterns having both dual polarization states and substantial rotational symmetry; and

a distribution network connected to each dual polarization state radiator for exchanging electromagnetic signals from and to each dual polarization state radiator.

2. The antenna system of claim 1, further comprising a ground plane generally parallel to the dual polarization radiator and spaced a predetermined distance therefrom.

3. The antenna system of claim 1, wherein said dual polarization radiators each have a rotationally symmetric radiation pattern in response to a fixed linearly polarized electromagnetic signal in any direction within 45 degrees of the coplanar orientation of the antenna symmetry axes.

4. The antenna system of claim 2, wherein the polarization states are orthogonal to each other, thereby minimizing cross-polarization response of any electromagnetic signal received by the antenna system.

5. The antenna system of claim 2, wherein the electrical centers of the dual polarization states are coincident within the antenna system.

6. An antenna system as claimed in claim 2, wherein the ground plane has sufficient radio-electric dimensions in a transverse plane of the antenna system to form dual-polarization radiators within a wide coverage area, thereby enabling a radiation pattern in an azimuth plane (azimuth plane) of the antenna system independent of any number of dual-polarization radiators.

7. The antenna system of claim 2, wherein each dual polarization radiator comprises an orthogonal dipole pair having a first dipole element and a second dipole element disposed orthogonally to each other.

8. The antenna system of claim 7, wherein the polarization state of the dual polarization state radiator can be maintained constant over a wide coverage (half power beamwidth) of at least 45 degrees in the azimuth plane of the antenna system.

9. The antenna system of claim 7, wherein the dual polarization radiators are positioned along the ground plane to form a linear array, and each of the orthogonal dipole pairs is positioned along the ground plane and in a vertical plane of the antenna system.

10. The antenna system of claim 7, further comprising a central polarization control network connected to the distribution network and the at least one antenna port to control the polarization of the dual polarization radiators.

11. The antenna system of claim 10, wherein the distribution network comprises a first power divider connected to each first dipole element and a second power divider connected to each second dipole element, the polarization control network comprising a first duplexer connected to the first power divider and having a first receive port and a first transmit port, and a second duplexer connected to the second power divider and having a second receive port and a second transmit port, the first receive port outputting a receive signal having a left-handed polarization state, the second receive port outputting a receive signal having a right-handed polarization state, the first and second transmit ports connected to a power combiner for receiving a transmit signal having a vertical polarization state.

12. The antenna system of claim 11, further comprising a 0 degree/180 degree hybrid coupler connected to the first receiving port and the second receiving port for receiving the reception signal having the polarization state tilted to the left and the reception signal having the polarization state tilted to the right and outputting the reception signal having the polarization state vertically and for receiving the reception signal having the polarization state tilted to the left and the reception signal having the polarization state tilted to the right and outputting the reception signal having the polarization state horizontally.

13. The antenna system of claim 11, further comprising a 0-degree/90-degree hybrid coupler connected to the first receiving port and the second receiving port for receiving the received signal having a polarization state tilted to the left and the received signal having a polarization state tilted to the right and outputting the received signal having a polarization state circularly polarized to the left, and for receiving the received signal having a polarization state tilted to the left and the received signal having a polarization state tilted to the right and outputting the received signal having a polarization state circularly polarized to the right.

14. The antenna system of claim 7, wherein the level plane of each dipole pair is +/-45 degrees with respect to a vertical axis of the antenna system.

15. The antenna system of claim 7, wherein the polarization states of the orthogonal dipole pairs are a left-tilted polarization state and a right-tilted polarization state.

16. The antenna system of claim 7, wherein the radiation pattern comprises a first radiation pattern in a tilt plane of the antenna system and a second radiation pattern in an azimuth plane of the antenna system, the first radiation pattern being determined by the geometry of the antenna system and the second radiation pattern being determined by the characteristics of the dual polarization radiator and the ground plane.

17. The antenna system of claim 7, wherein the ground plane comprises a solid conductive surface having a lateral dimension large enough to achieve a desired polarization state for the vertical polarization state component, and a non-solid conductive surface comprising an array of parallel and spaced apart conductive elements in a horizontal plane of the antenna system and symmetrically disposed at each lateral edge of the solid conductive surface, the conductive elements having a lateral dimension large enough to achieve a desired polarization state for the horizontal polarization state component.

18. The antenna system of claim 17 wherein the lateral dimension of the solid conductive surface is approximately one wavelength of the selected center frequency and the center spacing of each conductive element of the non-solid conductive surface is approximately 1/3 to 1/2 wavelengths of the selected center frequency.

19. The antenna system of claim 7, wherein the ground plane is a substantially planar sheet comprising a conductive material.

20. The antenna system of claim 7, wherein the ground plane is a substantially non-planar sheet comprising a conductive material.

21. An antenna system for transmitting and receiving electromagnetic signals with polarization state diversity, comprising:

a plurality of dual-polarization state radiators characterized by radiation patterns having both dual polarization states and substantial rotational symmetry;

a distribution network connected to each dual-polarization state radiator for exchanging electromagnetic signals from and to each dual-polarization state radiator;

a ground plane spaced a predetermined distance from the dual polarization state radiator; and

a polarization state control network coupled between the distribution network and the at least one antenna port to control the polarization state of the electromagnetic signal distributed by the distribution network.

22. The antenna system of claim 21, wherein the polarization state control network comprises a first duplexer connected to the first power divider and having a first receive port and a first transmit port, and a second duplexer connected to the second power divider and having a second receive port and a second transmit port, the first receive port outputting the receive signal having a left-handed polarization state, the second receive port outputting the receive signal having a right-handed polarization state, the first and second transmit ports connected to a power combiner for receiving a transmit signal having a vertical polarization state.

23. The antenna system of claim 22, further comprising a 0 degree/180 degree hybrid coupler connected to the first receiving port and the second receiving port, (1) for receiving the received signal having a polarization state tilted to the left and the received signal having a polarization state tilted to the right and outputting the received signal having a polarization state vertically and (2) for receiving the received signal having a polarization state tilted to the left and the received signal having a polarization state tilted to the right and outputting the received signal having a polarization state horizontally.

24. The antenna system of claim 22, further comprising a 0-degree/90-degree hybrid coupler connected to the first receiving port and the second receiving port, (1) for receiving the received signal having a polarization state tilted to the left and the received signal having a polarization state tilted to the right and outputting a received signal having a polarization state circularly polarized to the left, (2) for receiving the received signal having a polarization state tilted to the left and the received signal having a polarization state tilted to the right and outputting a received signal having a polarization state circularly polarized to the right.

25. The antenna system of claim 21, wherein each dual polarization radiator comprises an orthogonal dipole pair having a first dipole element and a second dipole element disposed orthogonally to each other, the polarization of the orthogonal dipole pair being maintained over a wide coverage (half power beamwidth) of at least 45 degrees in the azimuthal plane of the antenna system.

26. The antenna system of claim 25, wherein the ground plane comprises a solid conductive surface having lateral dimensions large enough to achieve a desired polarization state for the vertical polarization state component, and a non-solid conductive surface comprising an array of parallel and spaced apart conductive elements located in a horizontal plane of the antenna system and symmetrically located at each lateral edge of the solid conductive surface.

27. The antenna system of claim 26, wherein the ground plane is a substantially planar sheet comprising a conductive material.

28. The antenna system of claim 26, wherein the ground plane is a substantially non-planar sheet comprising a conductive material.

29. An antenna system for transmitting and receiving electromagnetic signals with polarization state diversity, comprising:

a plurality of dual-polarization state radiators characterized by radiation patterns having both dual polarization states and substantial rotational symmetry;

a distribution network connected to each dual-polarization state radiator for exchanging electromagnetic signals from and to each dual-polarization state radiator;

a ground plane spaced a predetermined distance from the dual polarization state radiator; and

a polarization state control network connected between the distribution network and the at least one antenna port to control the polarization state of the electromagnetic signal distributed by the distribution network,

each dual polarization radiator comprises an orthogonal dipole pair having a first dipole element and a second dipole element disposed orthogonally to each other, the polarization of the orthogonal dipole pair being maintained over a wide coverage (half power beamwidth) of at least 45 degrees in the azimuthal plane of the antenna system.

30. The antenna system of claim 29, wherein the ground plane comprises a solid conductive surface having lateral dimensions large enough to achieve a desired polarization state for the vertical polarization state component, and a non-solid conductive surface comprising an array of parallel and spaced apart conductive elements located in a horizontal plane of the antenna system and symmetrically located at each lateral edge of the solid conductive surface.

31. The antenna system of claim 30, wherein the ground plane is a substantially planar sheet comprising a conductive material.

32. The antenna system of claim 30, wherein the ground plane is a substantially non-planar sheet comprising a conductive material.