HK1001870B - Noise reduction filter system for a coriolis flowmeter - Google Patents

Description

Noise reduction filter system for coriolis flowmeter

The present invention relates to a filter system, and more particularly to a noise reduction filter system for a coriolis flowmeter.

Techniques for measuring mass flow and other information of a material flowing through a conduit using a mass flow meter that uses the coriolis effect are known. Such a meter is disclosed, for example, in us 4109524 at 29.8.1978, in us 4491025 at 1.1.1985, and in us 31450 reissue at 11.2.1982, both of which are issued to j.e. smith et al. These meters each have one or more straight or curved flow tubes. Each flow tube in a coriolis mass flowmeter has a set of natural vibration modes that may be simple bending, simple torsion, or a combination thereof. Each flow tube can be driven to resonate in one of these natural modes. The material flows into the flowmeter through a pipeline connected with the input side of the flowmeter, directly flows through the flow pipe, and then flows out of the flowmeter through the output side of the flow pipe. For natural vibration modes of vibration, the fluid filling the system is defined as a portion of the combined mass of the flow tube and the material within the flow tube.

When no fluid flows through the flowmeter, all points along the flow tube vibrate at the same phase under the applied drive force. When the material begins to flow in, coriolis accelerations will be generated at each point along the flow tube and will therefore have a different phase. The phase at the input side of the flow tube lags the vibrator phase and the phase at the output side of the flow tube leads the vibrator phase. The sensors may be positioned on the flow tube to produce a sinusoidal signal that reflects the motion of the flow tube. The phase difference between the two sensor signals is proportional to the mass flow rate of the material flowing through the flow tube.

One complicating factor in such measurements is that the density of conventional process fluids can vary during the measurement. This change in density will result in a change in the natural frequency of vibration. Since the control system of the flow meter is to be kept at resonance, the resonance frequency will vary with the density. In this case, the mass flow rate will be proportional to the ratio of the phase difference to the vibration frequency.

The coriolis flowmeter disclosed in the aforementioned U.S. reissue patent 31450 to Smith does not require measurement of both phase difference and vibration frequency. It determines the phase difference by measuring the time lag between the horizontal crossing points of two sinusoidal signals of the flow meter. When this method is used, the variation in the vibration frequency can be eliminated and the mass flow rate is proportional to the measured time lag. In the following, this measurement method is referred to as a measurement time lag or Δ t value measurement method.

Information representative of the properties of a material flowing through a coriolis mass flowmeter is typically given by a device that measures the phase or time lag between the output signals of two sensors of the flowmeter. The measurement is of considerable accuracy since the flow rate information obtained is read with an accuracy of at least 0.15% or more. The output signal of the flowmeter is a sinusoidal signal and varies with time or phase as a function of the coriolis force generated by the flowmeter as the material flows through. A signal processing circuit receiving the sensor output signals measures the phase difference with high accuracy and produces a desired characteristic parameter of the process material with a desired reading accuracy of at least 0.15%.

To achieve this accuracy, the signal processing loop must operate at a higher accuracy, measuring the phase drift of the two signals received by the flow meter. Since the phase drift between the two output signals of the flow meter is the information used by the processing loop to obtain the material properties, the processing loop must not introduce any phase drift that could affect the phase drift information provided by the output signals of the flow meter. In practice, the processing loop must have a very low intrinsic phase drift in order to have the phase drift of each input signal be less than.001 °, and in some cases less than a few parts per million. Phase accuracy of this order is required if the accuracy requirement of the information obtained for the process material is below 0.15%.

The frequency of the coriolis flowmeter output signal is within the frequency range of many industrial noises. Moreover, the magnitude of the flow meter output signal is often small, and in many cases, not much larger than the magnitude of the noise signal. This limits the sensitivity of the flow meter and makes the extraction of useful information very difficult.

To date, nothing has been able to shift the output signal frequency of the flow meter outside of the noise band or increase the amplitude of the output signal. The design of a practical coriolis sensor and flowmeter requires a combination of factors, and therefore produces an output signal with poor signal-to-noise ratio and poor dynamic range. These limits determine the characteristics of the flow meter and determine a characteristic curve that includes the maximum and minimum flow rates that can be obtained from the output signal of the flow meter.

In between tests used in both U.S. Pat. No. 5231884 and U.S. Pat. No. 5228327 to Bruck. These patents disclose a signal processing circuit in a coriolis flowmeter that includes three identical channels with precision integrators as filters. A first of the three channels is always connected to a pickup signal, say the left side. The other two channels (second and third) are alternately connected to the right pickup signal at each instant in successive time intervals. When one of the channels, say the second channel, is connected to the right pickup signal, the third channel is connected to the left pickup signal in parallel with the first channel. In a first time interval, the intrinsic phase lag between the first and third channels is measured by comparing the time difference between the outputs of the two channels currently connected to the left signal. Once the characteristic lag is determined, the connection status of the third channel and the second channel to the right pickup signal is switched during a second time interval. In this new configuration, the second channel is in calibration of its hysteresis characteristics, while the third channel is connected to the right pickup signal. The second and third channels may be alternately switched with the control loop approximately once per minute. In this one minute time interval (about 30 to 60 seconds), its aging, temperature and other effects do not have any effect on the phase drift, so their phase relationship is known and can be considered deterministic.

The fine correction integrator used by Zolock has an improved signal-to-noise ratio in the amplitude transfer function of the integrator of up to about 6 db/octave frequency response drop. However, this improvement in the drop in the 6 db/octave frequency response is not satisfactory in many environments where coriolis flowmeters are used. The reason for this is that its single-pole filter, such as the Zolock integrator, etc., has a fairly wide frequency band. Therefore, noise signals generated by unwanted flow tube vibrations, noisy environments, material flow noise, various electromagnetic wave frequency disturbances, and the like cannot be removed, and the high sensitivity of the flowmeter required to ensure accuracy cannot be obtained. Their amplitude may also be reduced as their frequency differs, but when measuring low quality materials such as gases and the like, they still affect the accuracy of the time lag measurement between the two pickup output signals.

There is another source of error in Zolcock and Bruck systems. Integrator time lag measurements are taken at three points defined by the sine wave pickup signal. Only if the two pickoff signals are identical in shape and symmetrical around the peak of the wave, they are in an ideal state. However, when the two magnetic loops (sensors) that produce the pick-up signal are not identical, the resulting non-ideal waveform contains many different harmonic quantities. They have an indeterminate phase state and may change their shape and thus their symmetry properties. This variation may cause the Zolock integrator to be calibrated with one waveform and measured with another waveform during a conventional measurement. Such differences between waveforms, due to the uncertain and varying phase of its harmonic components and its harmonics, can produce uncertain and unpredictable amounts of error.

Techniques such as digital signal processing and filtering are known to overcome the above problems and to improve the signal-to-noise ratio of the processed signal. However, these techniques are relatively complex, expensive, and in many cases require a compensating design for use in non-ideal conditions. It can thus be seen that a need remains for a solution for processing the output signals of a coriolis flowmeter using circuitry that maintains the initial phase shifts of the two output signals without causing any unknown and unwanted phase shifts or other signal changes in the circuitry, thereby not degrading the accuracy of the output signals generated by the processing circuitry that are representative of the characteristics of the material flowing through the flowmeter.

The present invention solves the above-described problems and achieves an improvement in the art. The present invention provides an additional and improved filtering apparatus and method for coriolis flowmeter output signal processing circuits using Zolock and Bruck techniques. The integral filter of Zolock is a fairly broadband filter with 6 db/octave frequency response drop, which can effectively filter out frequencies far above the output frequency of the flowmeter, and at the same time, reduce the influence on the phase drift by the least component variable method, and keep it as an accurate quantity. However, because it is not sharp cutoff, it does not effectively reduce or eliminate noise signals whose frequency is close to the frequency of the meter output signal. Therefore, even if the error due to long-term phase drift is reduced by the Zolock self-calibration feature, a noise signal very close to the frequency of the processed signal will still be output by the Zolock circuit.

In the present invention, the inventor sets a multi-pole filter, for example, a filter having eight or more poles, in front of the precision integrator of the Zolock type. Since this filter is multi-polar, it has a sharp frequency response dip characteristic and can provide a narrow passband to effectively filter out any noise signals whose frequencies are close to the frequencies of the meter pickoff signals being processed. The use of such a filter in the Zolock circuit provides for better filtering of the unwanted signals and thus very effective reduction of the noise component in the processed Coriolis pick-up signal. These signals can therefore be processed to produce highly accurate information representative of various characteristics of the material being processed.

When such a filter is used, it is not necessary to make the stability meet the requirement that its phase drift is less than 0.001 °, so the phase lag between the two pickup signals can be measured with high accuracy. Such filters may have an uncertain and unknown phase drift on the order of hundreds of degrees or more. Although a filter with unknown phase drift, in the order of hundreds of degrees or more, is inserted into a circuit that measures the time difference between two signals with an accuracy of 0.001 ° or more, problems may arise. However, the phase drift of such a multipole filter, although quite large and unknown, is relatively stable over a period of several minutes. The Zolock system implements its conventional channel switching and calibration to effectively eliminate errors caused by phase drift between channels.

The use of such a multipole sharp cutoff filter in a Zolock system allows the system to have a very high db/octave frequency response dip, so that it constitutes a filter that eliminates substantially all of the unwanted noise in the processed pick-up signal. The phase drift of such a multipole filter is not problematic because its variation is very slow over a relatively long period of time, for example, as both excited and non-excited. All phase shifts can be measured once per minute by the Zolock calibration loop and the state of the switchable channel is changed from the active state to the calibration state or vice versa at successive time intervals.

Another advantage of the present invention is that the driver circuit can utilize a filtered pick-up signal from which unwanted components have been removed by the filter of the present invention. This modified sinusoidal drive signal does not only produce self-excitation, but also other undesirable vibration modes that may contribute to a noisy pick-up signal.

These and other advantages and features of the invention will be better understood upon reading the following description in conjunction with the accompanying drawings.

Figure 1 depicts the present invention used in a coriolis material flow rate measurement system.

FIG. 2 depicts the electrical components of FIG. 1 in further detail.

Fig. 1 shows a mechanical assembly 10 and an electrical assembly 20 of a coriolis flowmeter. The electrical assembly 20 is connected to the mechanical assembly 10 by wires 100 and supplies density, mass flow rate, volume flow rate and summed mass flow information to the passageway 26.

The mechanical assembly 10 includes a pair of tube-type elements 110 and 110 ', and manifold elements 150 and 150'. A pair of parallel flow tubes 130 and 130', a drive mechanism 180, and a pair of rate sensors 170L and 170R. The flow tubes 130 and 130 ' have two substantially straight input side branch tubes 131 and 131 ' and output side branch tubes 134 and 134 ' which converge forwardly toward each other at elements 120 and 120 ' having surfaces 121 and 121 '. The diagonal tension tubes 140 and 140 'define axes W and W' of each flow tube as it vibrates.

Side branches 131 and 134 of flow tubes 130 and 130 'are fixedly attached to surfaces 121 and 121' of members 120 and 120 'and the latter are fixedly attached to manifold members 150 and 150'. This results in a continuous, tightly connected material flow path through coriolis flowmeter mechanical assembly 10. When the mechanical assembly 10 with the recessed edges 103 and 103 ' is connected with the fluid ports 101 and 101 ', material will enter the flow meter through the fluid port 101 in the recessed edge 103 of the input side end 104, enter the input side manifold 110 and be directed to the element 120 with the surface 121 through a channel course with a gradually changing cross section, through the input side end 104 and the output side end 104 ' connected with the flow tube system (not shown) for guiding the process material to be tested. Here, the material will flow through the branch tubes 131 and 131 ', the flow tube members 130 and 130', respectively. Upon exiting flow tubes 130 and 130 ' and branch tubes 134 and 134 ', the treatment material will again be stranded in element 120 having surface 121 and then flow through output manifold 150 '. In the output side manifold 150 ', material of another channel route with a stepwise change in its cross section will flow from the input side manifold 150 to the orifice 101 ' at the output side end 104 '. The outlet-side end 104 ' is connected to the pipe system (not shown) via a flange 103 ' with a threaded bore 102 '.

Flow tubes 130 and 130 'may be suitably selected and suitably mounted on members 120 and 120' to enable substantially the same mass distribution, moment of inertia and modulus of elasticity, respectively, to be formed corresponding to bending axes W-W and W '-W'. These bending axes are located near the respective flow tube diagonal draw tubes 140 and 140 'and elements 120 and 120', respectively. The diagonal pull tubes 140 and 140 ' with bending axes W and W ' contain out of phase bending axes of the flow tubes 130 and 130 ' when driven out of phase by the driver 180. Manifold members 120 and 120 'include in-phase bending axes with flow tubes 130 and 130'. These flow tubes may produce in-phase vibrations under the influence of environmental disturbances such as earth vibrations, human activity, and motion generated by nearby machinery. The flow tubes extend outwardly from the mounting member in a substantially parallel manner and have substantially the same mass distribution, moment of inertia and modulus of elasticity corresponding to the respective bending axes.

The two flow tubes 130 are driven in opposite directions by a driver 180 along respective bending axes W and W' at a frequency that is the first anti-phase natural frequency of the meter. This mode of vibration is also referred to as the antiphase bending mode. The two flow tubes 130 and 130' vibrate in anti-phase like the two prongs of a tuning fork. The drive mechanism 180 may be any known drive mechanism, such as one that includes a magnet mounted to the flow tube 130' and an opposing coil mounted to the flow tube 130 and through which an alternating current is passed to vibrate the two flow tubes, and the like. The consumer assembly 20 may also send an appropriate drive signal to the drive mechanism 180 via the lead 185.

The drive mechanism 180 and the resulting coriolis force will cause the flow tube 130 to vibrate in a periodic manner. In the first coriolis vibration half cycle of flow tube 130, the satellite side legs 131 and 131 'are acted upon closer than their mating side legs 134 and 134' and reach the end of the drive earlier than the above-mentioned components with which they mate, where their vibration is zero. During the second half of the coriolis vibration, the flow tube 130 will have an opposite motion, i.e., the satellite side legs 134 and 134 'will be acted upon closer together than the side legs 131 and 131' with which they mate. Thus, the branch pipes 134 will reach the end points of the transmission, where their vibrations are zero, earlier than the branch pipes 131. The time interval (hereinafter referred to as the frequency-specific phase difference, or time lag difference, also referred to as the "Δ t" value), during which the pair of satellite side branches reach their transmission endpoints, is substantially proportional to the mass flow rate of the material flowing through the mechanical assembly 10, prior to the arrival of their respective transmission endpoints by the components with which they are mated (i.e., the components that are forced apart). The drive end points of each flow tube are conventional measurement points for measuring Δ t. The Δ t relationship between the two flow tubes is the same throughout the range of motion of the flow tubes.

To measure the time lag interval Δ t, sensors 170L and 170R may be attached near the free ends of flow tubes 130 and 130'. These sensors may be any known type of sensor. The signals generated by the transducers 170L and 170R give a curve profile of the velocity (or displacement, also referred to as acceleration rate) of the composite drive of the flow tube and can be processed in any known manner to calculate time intervals and hence the mass flow rate of the material flowing through the meter.

Sensors 170L and 170R provide left and right velocity signals to leads 165L and 165R, respectively. The time lag difference, or Δ t, is measured to provide a representation of the phase difference produced between the left and right rate sensor signals.

Electrical component 20 receives left and right rate output signals from conductors 165L and 165R, respectively. The electrical assembly 20 also generates a drive signal that is sent through the lead 185 to the drive mechanism 180, which drives the flow tubes 130 and 130' into vibration. The electrical assembly 20 processes the received left and right velocity signals to calculate the mass flow rate, the volumetric flow rate, and the material density of the material flowing through the mechanical assembly 10.

The properties of the process material flowing through the flow meter 10 are given by the electrical component 20 measuring the phase or time delay Δ t between the output signals of the two transducers 165L and 165R. Such time delay measurements must be very accurate. This requires that the flow rate accuracy of the flow output information of the treatment material obtained when the signal is applied to the output passage 26 of the electrical component 20 should be at least 0.15% of the flow rate.

In order to obtain the accuracy of the output signals, it is necessary for the signal processing circuit in the consumer assembly 20 to determine the phase difference between the sensor output signals with high accuracy. Since the phase difference between the sensor output signals includes input information for the processing circuitry to use in measuring the process material information, the signal processing circuitry in the electrical assembly 20 must not introduce any phase drift to correct or change the phase difference between the sensor output signals provided by the flow meter. This signal processing circuit must have a constant or substantially stable phase difference in order to cause the phase of the sensor signals 165R and 165L to drift by an amount less than 0.001 deg., and in some cases less than a few parts per million. This level of phase accuracy is required if the electrical component 20 is to achieve less than 0.15% accuracy for the signal applied to the output path 26.

Fig. 2 is explained below.

Fig. 2 shows an electrical circuit comprising the electrical assembly 20 as shown in fig. 1. As shown, the electrical assembly 20 includes a drive circuit 27 connected to a drive coil 180 by a path 185. Drive circuit 27 applies an appropriate signal at path 185 to excite drive coil 180 at an appropriate amplitude and frequency to produce the desired opposite phase bending vibrations to flow tubes 130 and 130'. Circuit 27 may be any known driver circuit and will not be discussed in detail herein since no particular form of driver circuit is required in any part of the present invention. It is worth noting that, in this regard, the reader is referred to U.S. patent 5009109 issued to p.kalotay et al, 23/4/1991; united states patent 4934196 to p.romano was granted on 19/6/1990. U.S. patent 4876879 to ruesch, 10/31 1989. All of these patents are owned by the effective assignee of the present invention and disclose various flow tube drive circuits

Embodiments are described.

Electrical assembly 20 also includes a flow measurement circuit 23, which flow measurement circuit 23 is connected on its left side to sensors 170L and 170R by way of passages 165L and 165R to receive sensor output signals. The phase difference contained between the output signals of the two sensors can be received and processed by the input information measurement circuit 23 to obtain accurate information of the process material. Input signals 165L and 165R are input to multiplexer 31, the output of which is input to channels 44, 54 and 64 (channels A, C and B) via paths 45, 55 and 65, respectively. These channels operate in a manner to process the sensor signals and control the operation of counters 74 and 76, the output of which represents the phase difference between the sensor output signals.

The outputs of counters 74 and 76 are applied through channel 87 to microprocessor 80, which has the functions described below, to obtain the desired information for the material being processed. The information obtained is input via path 91 to output circuit 90 which sends the desired treatment material output information to connecting lead 26 via leads 263 and 262. The operation of the counters 74 and 76, and the multiplexer 31, are controlled by control logic 72, described below.

Each channel A, B and C contains a multi-pole filter, an integrating amplifier and at least one potential detector. The output of the multipole filter of each channel is applied to an integrating amplifier of that channel, the output of the latter being correspondingly applied to a potential detector, the output of which is in turn applied to counters 74 and 76, the counters 74 and 76 measuring the time intervals in the form of clock pulses that occur between corresponding changes in the detector output signal. The counter outputs a known value for Δ t, which varies with the mass flow rate of the fluid being treated. The resulting Δ t values and count values are applied through passage 87 to microprocessor 80, which, in response to this information, calculates the mass flow rate and the volume flow rate and generates the desired output information corresponding to the material being processed.

The circuitry contained in channels 44, 54 and 64 will introduce a phase difference into the mass flow information generated by microprocessor 80. Each circuit will not only have an inherent phase lag of different measurements from one another at the input of the multipole filter to the output of the potential detector, but the phase lag is also a function of temperature and their variations with temperature, age, etc. will often be different from one another. This variation may result in an uncontrolled inter-channel phase difference that is an error component in the measured Δ t values of output signals 165L and 165R. Since these signals give rise to a fairly small value of at, the effect of any phase lag introduced by the channel circuit will be significant.

To eliminate such errors in the channel circuit, the circuit shown in fig. 2 further extends the technical teaching given in the aforementioned U.S. patent 5231884 to Zolock. The above patents are herein incorporated by reference as if fully set forth herein. The circuit shown in fig. 2 in the application example of the present invention is more than the technical teaching given by Zolock, especially the technical teaching shown in fig. 3A and fig. 3B in the patent to Zolock. In the technique shown in fig. 2, mass flow and mass flow rate values are generated using meters that are insensitive to phase errors generated by processing circuitry in channels a, B, and C shown in fig. 2. This circuit does not introduce any unwanted phase drift that may have an effect on the phase drift information output by the sensors of paths 165L and 165R. The phase drift information received by flow measurement circuit 23 through paths 165L and 165R will be processed with high accuracy without introducing unwanted phase drift by the channel circuits. Thus, the resulting processed flow meter output signals applied by counters 74 and 76 to microprocessor 80 via path 87 represent the same phase drift information received by the flow meter via paths 165L and 165R. This allows microprocessor 80 to accurately obtain information about the process material in the conduit flowing through the flow meter.

The circuit shown in fig. 2 will produce an intrinsic phase drift in the three channels a, B and C that can be measured and thus effectively cancelled. Multiplexer 31 is connected to the two outputs of the flow meter by paths 165L and 165R so that the input signal can be applied to two of the three channels at a time. If channel C is permanently connected to path 165L, the inputs of the other two channels A and B will be connected to path 165R sequentially at predetermined time intervals. In the state shown in fig. 2, the sensor output 165R on the right side is connected to the channel B through the path 65 via the movable switch contacts f and e. During the next time interval, these switch contacts change their positions in sequence, so that path 165R is connected to the input in channel B on path 65 through contacts f and d. When a switchable channel, say channel B, is connected to path 165L, that channel is referred to as the excitation channel. When the other switchable channel, say channel a, is connected to path 165R, it is also referred to as the excitation channel. It is generally said that when a switchable channel a or B is connected to a via 165L and a channel C is permanently connected to the via 165L, the switchable channel a or B is said to be in its aligned state.

The outputs of the three channels a, B and C are connected by conductors 49, 69 and 59 to the inputs of counters 74 and 76. Through these channels, counter 74 receives input signals from the outputs of channels a and C. Counter 76 receives input signals through the outputs of channels C and B. This enables the counter 74 to measure the phase lag between the output signals of channels a and C. Similarly, counter 76 is capable of measuring the phase lag between the output signals of channels B and C. When channel B is in the calibration state, the inherent phase lag between channels B and C can be measured by connecting the input of channel B to channel 165L through contacts f and d and applying the output signals of channels B and C to counter 76 which measures the phase difference between the output signals of channels B and C. Since both channels are connected to the same input point, the phase difference measured between channels B and C is now determined by the inherent circuit elements of the two channels. When the input of channel a is connected to input channel 165L through switch contacts C and b, the inherent phase lag between channels a and C can be determined in a similar manner. The phase difference between the two channels in paths 49 and 59 can be determined by a counter 74. The phase difference measured by counters 74 and 76 when the corresponding channel a or B is in the calibration state may be input to RAM83 of microprocessor 80 via channel 87. This calibration information is temporarily stored in RAM 83.

In this manner, the phase of each of the non-excited channels can be determined by comparing its output with the output of channel C. Then, when one of the non-drive channels is sequentially switched to the drive channel at the next time interval, its inherent phase lag, now stored in RAM83, may be algebraically coupled to the respective output signals of the respective counters 74 and 76 to determine the true phase lag between the two input signals 165L and 165R.

The output of counter 74 represents the phase difference between signals 165L and 165R, and the error due to the inherent phase drift between channels A and C, during the time interval in which channel A is the excitation channel. However, the intrinsic phase lag of excitation channel A, corresponding to channel B, is a known parameter and is stored in RAM 83. The phase lag given by counter 74 is received by microprocessor 80 via path 87. The measured phase difference from counter 74 is algebraically superimposed on the phase lag correction parameter from RAM83, allowing microprocessor 80 to sum the two quantities to obtain phase lag information representing only the phase difference between signals 165L and 165R. Channel B may act in a similar manner, corresponding to counter 76. In this process, counter 76 stores calibration phase lag information for channel B in the calibration state in RAM83, and does not change the phase difference between signals 165L and 165R during the time interval when channel B is in the excitation state. These two signals are summed by microprocessor 80 to obtain the true phase difference between signals 165L and 165R when channel B is in the energized state.

Channel B may operate in a similar manner with respect to its counter 76, i.e., calibration phase lag information when channel B is in the calibration state is stored in RAM83, and the phase difference between signals 165L and 165R is measured when channel B is in the excitation state.

Since each channel changes state generally once per minute, the current phase lag information given by both counters 74 and 76 is reflected in the phase information given by the sensor output signals 165L and 165R. Since the measured value Δ t given by each excitation channel pair is corrected with the current intrinsic phase lag corresponding to that excitation channel, the Δ t value does not contain any significant channel-induced phase error.

In summary, a flow measurement circuit such as that shown in FIG. 2 utilizes three channels A, B and C, by measuring the phase difference between the channels for each of two pairs of excitation channels, say channels A and C, and channels B and C, respectively. Channels a and B are referred to as switchable channels. Channel C is always connected to via 165L and may be referred to as a reference circuit. Path 165L is connected to the input of channel a at one time interval and to the input of channel B at the next time interval via multiplexer 31. During this time interval, one path, say channel B, is in calibration and not connected to path 165R, and its input is connected in parallel with the input of channel C on path 165L. During the calibration time interval, the phase difference between channel C and channel B is measured. The outputs of channels C and B are connected to the inputs of a counter 76, so that the output of the latter represents the phase lag between the outputs of channels B and C.

All three channels may have different and unstable phase lags. However, since the measurement of the phase lag of channels A and B is made relative to reference channel C, the difference in phase lag between channels A-C and channels B-C may be known. As long as it is connected in a calibrated interval of one channel, say channel B, it is possible to switch the input to channel B in the next interval to make it the excitation channel. At this time, the channel A becomes a calibration channel, and the connection of its input terminal is changed so as to be parallel to the input terminal of the channel C and is simultaneously connected to the path 165L.

When one pair of channels, say channels B and C, is in calibration mode, the other pair of channels, channels a and C, is in measurement mode and is used to measure the phase difference between the signals in paths 165L and 165R. For any pair of channels, the Δ t values measured from input signals 165L and 165R, which may be obtained from a counter, such as counter 74, are corrected algebraically by superimposing the measured Δ t values with the previously measured inherent phase lag of the two channels currently stored in RAM 83.

The two counters 74 and 76 switch the state of the respective channels at predetermined time intervals, for example, once every minute. Thus, for the time interval in which channel A is the excitation channel and is connected to path 165R, counter 74 is an excitation counter and its inputs are connected to the outputs of channels A and C. Channel B is now the calibration channel and is referred to as the calibration counter since the input of counter 76 is connected to the output of channels B and C and the inherent phase difference between the two channels is being measured. Subsequently, in the next time interval in which the states of channels a and B are switched so that channel B is the excitation channel, since the input of counter 76 is connected to the output of channels B and C, the phase difference between the signals of paths 165L and 165R is measured, it is called the excitation counter. The input of counter 74 is connected to the outputs of channels a and C, and the inputs of both are connected to path 165L at this time through multiplexer 31, so it is referred to as a calibration counter. At this point, the output of counter 74 represents the inherent phase difference between channels A and C, while the output of counter 76 represents the phase difference between meter signals 165L and 165R, which contains the phase difference error due to the inherent phase difference between channels B and C when channels B and C are the excitation channels. However, this inter-channel phase difference has been previously measured in a previous time interval and the error information is stored in RAM83 of microprocessor 80. Therefore, when the microprocessor 80 receives the phase difference information from the counter 74, the measured phase difference information can be corrected algebraically using the calibration phase difference stored in the RAM83 to eliminate errors due to the phase difference between the channels. This allows the microprocessor to obtain information about the treatment material only as to the phase difference between the signals 165L and 165R received by the flow meter.

Each channel includes a multi-pole filter, an integrating amplifier and a potential detector. For channel a, these components consist of multipole filter 41, integrating amplifier 46 and potential detector 48. The multipole filter may be any high order filtering type filter having a relatively high sharp cut-off at a corner frequency slightly higher than the frequency of the sensor output signal. However, the pick-up characteristics of the filter are relatively sharp, so that noise signals above or below the sensor signal frequency will be greatly attenuated. The multi-pole filter may contain active or passive components and may also have an indeterminate phase drift on the order of hundreds of degrees. There is no need to try to control the long-term drift of the phase shift of said multi-stage filter. However, its short-term phase characteristics, say in a few minutes, should be fairly stable and not subject to significant changes. The output of the multipole filter is applied to an integrating amplifier 46 which is a monopole filter with a 6 db/octave frequency response dip. Integrating amplifier 46 is used primarily to cancel signal dc offsets that may occur on conductor 42 connecting multipole filter 41 and integrating amplifier 46. Any dc offset in the integrating amplifier output signal must be removed to enable the potential detector element 48 to operate with high accuracy.

The level detector element 48 is an efficient window comparator that gives a change in the level of the output signal even when the output signal produced by the integrating amplifier is greater than a predetermined positive level, or less than a predetermined negative level.

Each of the passages A, B and C has substantially the same function. However, in channel B, the potential detector 58 is a simple potential detector, rather than a window comparator, which is used to detect when the output signal of the integrating amplifier 56 crosses zero potential.

Control logic 72 controls the operation of counters 74 and 76 and the switching operation of multiplexer 31. Element 72 is a finite state machine that defines a period of time intervals and a recurring sequence of corresponding states. These intervals can be labeled as the sequence …, n-1, n, n +1, …, according to scientific labeling methods commonly used in the mathematical and physical disciplines. Control logic 72 operates in coordination with counters 74 and 76 to determine timing measurements for channel pairs A-C and B-C, respectively. The control logic 72 may be formed of a known combination, and other logic may be used. After initiation of the flow tube cycling process for the calibration and switching intervals, control logic circuit 72 generates a signal that is applied to path 34 and causes multiplexer 31 to be actuated by control element 32, thereby causing the output signal of sensor 165R to be input to one of channels A or B at different time intervals, thereby causing the channel pair to repeatedly cycle between its calibration and excitation states. Control logic 72 also generates appropriate control signals to reset counters 74 and 76 at the beginning of each timing interval via paths 79 and 77.

When control logic 72 cycles between the different states, it records the value of the current state in an internal memory (not shown) connected to microprocessor 80 via channel 85. The microprocessor reads this data and uses it to process the count values given by the counters 74 and 76. Microprocessor 80 receives the phase difference measurements between the channels as the state of each channel pair changes during operation, as well as the delta t value for each channel pair. Microprocessor 80 also applies appropriate signals on paths 82 and 84 to control the operation of control logic circuit 72.

Microprocessor 80 is connected by way of path 91 to a known output circuit 90 for outputting a series of standard output signals such as the 4-20MA output signal on path 263 and the scaled frequency information on path 262. The passages 263 and 262 are configured as an output passage 26, and the output passage 26 is connected to a known circuit (not shown) that can utilize the processed material information obtained by the circuit shown in fig. 2.

Microprocessor 80 may be comprised of any known, economically feasible processor having an effective random access memory 83 and ROM 86. Since its program is constructed using an event-driven task architecture, a database is coupled to microprocessor 80 to facilitate data transfer, sharing measured and calculated data among different jobs. For each channel pair, including timing measurements of phase difference and Δ t values between the channels, microprocessor 80 can correct the Δ t values resulting from the measurements given by each channel pair to account for the phase lag between the channel pair currently being energized.

The provision of a multi-pole filter in each channel greatly improves the signal processing capability of the circuit shown in figure 2. Thus, the arrangement of the multipole filter in each channel provides a narrow pass band or sharp cutoff filter, which greatly reduces the amplitude of the noise signal appearing on the conductor paths 165L and 165R. By now the prior art, which did not use the multi-pole filter of the present invention in each channel, the operational sensitivity of the flow meter was limited by the amplitude of the noise signal. In each channel, an integrating amplifier such as that shown at 46 does not contribute to the above problem since it is a single-pole filter with 6 db/octave frequency response droop. This 6 db/octave frequency response drop is not applicable for the cancellation of noise signals whose frequency is very close to the sensor signal frequency. The addition of a multi-pole filter in front of the integrating amplifier solves the above problem because it provides sharp cut-off and sharp attenuation of the received noise signal whose frequency is very close to the frequency of the sensor output signal. Thus, the use of a multi-pole filter can greatly reduce the amplitude of the noise signal at the input conductor path of the integrating amplifier, which in turn can reduce the amplitude of the noise signal at the input of potential detectors 48, 58 and 68. Since the input signals to the potential detectors are substantially noise-free, they can be operated with good accuracy to determine the time lag between the sensor output signals, which can be much less than can be handled in the prior art.

The integrating amplifier also requires a secondary filter and a dc suppressor to remove dc offsets that may be present in the multipole filter output signal.

In contrast to the technique described in U.S. patent 5,231,884 to Zolock, the operation of the circuit shown in FIG. 2 further includes a processing program that controls the operation of microprocessor 80. In contrast to the technique described in U.S. patent 5228327 to Bruck, the present invention further includes the operational control of microprocessor 80 and the components connected thereto.

It will be understood that the invention described above also includes other variations and modifications within the subject and scope of the invention disclosed. For example, the invention is described in connection with a flow meter as shown in fig. 1, but it will be understood that the invention is not limited to this form of flow meter as shown in fig. 1. It is also applicable to any type of coriolis-based flow meter having a single tube, double tube, straight tube, or irregular tube configuration. Furthermore, the flow meter of the present invention may be used without the need for having the particular flange and orifice structure shown in FIG. 1, and may be attached thereto by any suitable means for interfacing with a line to which the flow meter is connected. The multipole filters 41, 51 and 61 shown in fig. 2 are described by way of example as low-pass sharp-cut filters, which are not limitative either. These filters may also be quasi-band-pass filters with the same sharp cut-off, if desired, except that their processing frequency filtering capability is lower than the band-pass center frequency of the sensor output signal. Such a filter can remove various noises above and below the resonance frequency of the flow tube, thereby improving its accuracy, signal-to-noise ratio, and the like. Since this filter is a narrow pass filter, it can follow flow tube frequencies that vary in response to changes in density and mass flow rate. Such quasi-bandpass filters are also well known in the art and need not be described in further detail.

Claims (9)

1. Signal processing apparatus for determining a phase difference between two received input signals (165L, 165R) comprising unwanted signal components, said signal processing apparatus comprising:

a first channel (54) and a second channel (44);

an input (55) of the first channel is connected to receive a first one (165L) of the input signals;

a switching circuit (31) for controllably connecting an input of said second channel to an input of said first channel for receiving said first input signal during an nth time interval and disconnecting an input of said second channel from an input of said first channel for inputting a second one (165R) of said input signals to an input of said second channel during an n +1 th time interval;

phase difference measuring circuitry (74, 76) connected to the outputs of said first and second channels and operative during said (n + 1) th time interval to measure the phase difference between the output signals of said first and second channels;

it is characterized in that the equipment also comprises;

a multipole filter (41, 51) which is phase shifted in each of said first and second channels, each multipole filter being connected between an input and an output of each channel and being operable to filter unwanted signal components from an input signal received at said input so that no unwanted signal components are present in an output signal transmitted from said output; and

a processor (80) operative during said nth time interval to obtain a correction factor indicative of a phase difference between signals output from said first and second channels when said first signal (165L) is received at said inputs of said first and second channels;

the processor (80) is further operative to obtain a corrected phase difference between the first and second input signals by mixing the first correction factor with the phase difference measured from between the filtered output signals of the first and second channels during the n +1 time interval.

2. The signal processing apparatus of claim 1, wherein said signal processing apparatus further comprises:

a third channel (64) having an input (65) and an output; and

a multipole filter (61) having a phase shift in said third channel and connected between said input and said output of said third channel and operative to filter undesired signal components from an input signal received at said input so that an output signal delivered at said output is free of said undesired signal components;

said switching circuit (31) operating during an (n-1) th time interval to connect an input of said third channel with an input of said first channel to receive said first input signal on said first and third channels;

said processor (80) operating in an n-1 th time interval to generate a second correction parameter indicative of a phase difference between output signals of said first and third channels;

when the switch circuit (31) works in the nth time interval, the input end of the third channel is disconnected from the input end of the first channel, and the input end of the third channel is connected to the second input signal;

said phase difference measuring circuit (74, 76) is further connected to an output of said third channel and operates at an nth time interval to measure a phase difference between output signals of said first and third channels; and is

The processor (80) is operative to mix the second correction parameter with the phase difference measured during the nth time interval from between the first and third channel filtered output signals to derive a corrected phase difference between the first and second input signals.

3. The signal processing apparatus of claim 2 wherein each of said channels further comprises:

an operational amplifier (46, 56, 66) having an input coupled to said output of said multipole filter (41, 51, 61), said operational amplifier being effective to cancel any DC offset in the filtered signal applied to its input by the output of said multipole filter;

a potential detector (48, 58, 68) having an input connected to the output of said operational amplifier, said potential detector being responsive to receipt at its input of an ac signal without dc compensation to generate an output signal representative of the signal applied at the input (45, 55, 65) of said channel;

said phase difference measuring circuit having a first counter (74) connected to the outputs of said first and second channels to produce an output signal indicative of the phase difference between the signals from the outputs of said first and third channels; and a second counter (76) connected to the outputs of said first and third channels to generate output signals indicative of the phase difference between the signals at the outputs of said first and third channels, the output signal of each of said counters being indicative of the measured phase difference between said input signals of the respective channel in one of said time intervals and indicative of the phase correction parameter between the respective channels in the other of said time intervals;

the processor (80) receives the output signals of the counters to derive a phase difference between the received signals applied to the inputs of the channels associated with each counter.

4. The apparatus of claim 3, used in combination with:

a coriolis flowmeter (10) including a process material flowing through a vibrating flow tube of said flowmeter;

a sensor (170L, 170R) coupled to said vibrating flow tube for generating an output signal representative of Coriolis motion of said vibrating tube caused by flow of said process material;

leads connecting said sensor to said signal processing device for applying the output signal of said sensor as said received input signal (165L, 165R) to said signal processing device;

wherein said processor (80) of said signal processing apparatus determines information relating to said treatment material based on the derived corrected phase difference between said input signals.

5. The apparatus of claim 4 wherein said coriolis flowmeter includes a driver (180) for vibrating said flow tube;

it is characterized in that the device also comprises:

a driver circuit (27) having an output coupled to said driver for applying a stimulus signal to said driver circuit and an input coupled to an output of said operational amplifier of one of said channels (54) for receiving a filtered signal at said input of said driver circuit.

6. A method for operating a signal processing apparatus for determining a phase difference between two received input signals including an unwanted signal component, said signal processing apparatus comprising: a first channel (54) and a second channel (44);

the method comprises the following steps:

inputting a first one of said input signals to an input (55) of said first channel (54);

connecting the input of said second channel to the input of said first channel to receive said first input signal (165L) during an nth time interval, and disconnecting the input (44) of said second channel from the input (55) of said first channel to apply a second one (165R) of said input signals to the input (45) of said second channel during an n +1 th time interval;

measuring the phase difference between the output signals of said first and second channels during said (n + 1) th time interval;

characterised in that the apparatus further comprises a multi-pole filter (51, 61) having a phase shift in each of said first and second channels (54, 55), each multi-pole filter being connected between an input and an output of each channel and being operable to filter unwanted signal components from an input signal received at said input so that an output signal delivered at said output is free of unwanted signal components;

and the method further comprises the following steps:

obtaining a first correction parameter representative of a phase difference between output signals generated by said first and second channels in response to said first input signal applied to the inputs of said first and second channels during an nth time interval;

in the (n + 1) th time interval, the first correction parameter is superimposed with the measured phase difference between the filtered output signals of the first and second channels to obtain a corrected phase difference between the first input signal (165L) and the second input signal (165R).

7. The method of claim 6, wherein said signal processing apparatus further comprises:

a third channel (64) and a multipole filter (61) having a phase shift in said third channel, said multipole filter (61) being connected between said input and said output of said third channel and being operable to filter unwanted signal components from an input signal received at said first input so that an output signal delivered from said output is free of unwanted signal components;

and the method further comprises the following steps:

connecting the input of said third channel to the input of said first channel in an n-1 time interval to receive said first input signal (165L) on said first and third channels;

generating a second correction parameter representative of a phase difference between the filtered output signals of said first and third channels during an (n-1) th time interval;

disconnecting the input of said third channel from the input of said first channel and connecting the input of said third channel to said second input signal (165L) during an nth time interval in which the input (45) of said second channel is connected to the input (55) of said first channel;

measuring the phase difference between the filtered output signals of said first and third channels during said nth time interval;

in an nth time interval, combining said second correction parameter with said measured phase difference between said first and third channel filtered output signals to obtain a corrected phase difference between said first and second input signals, based on a measurement of the phase difference between said first and third channel filtered output signals.

8. The method of claim 7, wherein said method further comprises the steps of:

an operational amplifier (46, 56, 66) operating in each channel and having an input connected to the output of said multipole filter in each channel, said operational amplifier being operative to remove from its output signal any DC offset in the filtered signal applied to its input by the output of said multipole filter;

applying an AC signal output by said operational amplifier, which does not contain a DC offset, to the input of a potential detector (48, 58, 68) for each channel, said potential detector being responsive to a signal received at its input to produce an output signal representative of the signal applied at the input of said channel;

operating a first counter (74) connected to the outputs of said first and second channels to produce an output signal (87A) representative of the phase difference between the filtered signals at the outputs of said first and second channels;

operating a second counter (76) coupled to the filtered outputs of said first and third channels to produce an output signal (87B) representative of the phase difference between the filtered signals at the outputs of said first and third channels;

the output signal of each of said counters being representative of the measured phase difference between the input signals of the associated channels in one of said time intervals and of the phase correction parameter between the associated channels in the other of said time intervals;

the output signal of the counter is applied to a processor (80) to generate information representative of the phase shift between the received signals applied to the inputs of the channels.

9. The method of claim 7, characterized by operating in conjunction with a coriolis flowmeter (10):

the coriolis flowmeter comprises a process material flowing through a vibrating flow tube of the flowmeter;

the method further comprises the following steps:

coupling a sensor (170L, 170R) to said flow tube (131, 134) to generate an output signal (165L, 165R) representative of coriolis motion of the vibrating flow tube caused by the flow of said process material;

-connecting said sensor to said signal processing means (20) to apply said output signal as a received input signal to said signal processing means;

determining information about said treatment material based on the derived corrected phase shift between said input signals.