HK1099412B - Phased array antenna system with controllable electrical tilt - Google Patents

Description

Phased array antenna system with controllable electrical tilt

The invention relates to a phased array antenna system with controllable electrical tilt. The antenna system is suitable for many telecommunication systems but finds application particularly in cellular mobile radio networks commonly referred to as mobile telephone networks. More particularly, but not by way of limitation, the antenna system of the present invention may be used with second generation (2G) mobile phone networks, such as GSM systems, CDMA (IS95), D-AMPS (IS136) and PCS systems, and third generation (3G) mobile phone networks, such as Universal Mobile Telephone System (UMTS), among other cellular systems.

Operators of cellular mobile radio networks typically employ their own base stations, each having at least one antenna. In cellular mobile radio networks, antennas are a major factor in defining the coverage area in which communications to base stations may occur. The coverage area is typically divided into a number of cells, each cell being associated with a respective antenna and base station.

Each cell contains a base station for radio communication with all mobile radios (mobile stations) in the cell. The base stations are interconnected by other means of communication, typically fixed landline or point-to-point radio links, allowing mobile radios throughout the cell coverage area to communicate with each other and with the public telephone network outside the cellular mobile radio network.

Cellular mobile radio networks using phased array antennas are known: such antennas include arrays having (typically eight or more) individual antenna elements, such as dipoles or patches. The antenna has a radiation pattern including a main lobe and side lobes. In receive mode, the center of the main lobe is the direction of maximum sensitivity of the antenna, while in transmit mode, the center of the main lobe is the direction of its main output radiation beam. A well-known property of phased array antennas is that if the signals received by the antenna elements are delayed due to delays that vary with the distance of the elements from the edge of the array, the main beam of radiation of the antenna is steered in a direction that increases the delay. The angle between the centers of the main radiation beams, i.e. the tilt angle, corresponding to zero and non-zero delay variation depends on the rate at which the delay varies with distance across the array.

The delay can be equivalently implemented by changing the phase of the signal, so there is an expression that: phased array. Thus, by adjusting the phase relationship between the signals fed into the antenna elements, the main beam of the antenna pattern can be changed. This allows steering of the beam to modify the coverage area of the antenna.

Operators of phased array antennas in cellular mobile radio networks require adjustment of the vertical radiation pattern of their antennas, i.e. the cross section of the pattern in the vertical plane. This is necessary to change the vertical angle of the main beam of the antenna, also called "tilt", in order to adjust the coverage area of the antenna. Such adjustments may be needed, for example, to compensate for changes in the cellular network structure or the number of base stations or antennas. It is known to adjust the tilt of the antenna mechanically and electrically, either alone or in combination.

By moving the antenna unit or its housing (radome), the antenna tilt angle can be adjusted mechanically: this is referred to as adjusting the angle of the "mechanical tilt". As described above, the antenna tilt angle can be adjusted electrically by changing the time delay or phase of the signal fed to or received from each antenna array element (or group of elements) without physical movement: this is referred to as adjusting the angle of the "electrical tilt". When used in cellular mobile radio networks, the Vertical Radiation Pattern (VRP) of a phased array antenna has a number of important requirements:

1. high boresight (boresigbt) gain;

2. the first upper side lobe level is sufficiently low to avoid interference with mobile stations using base stations in different cells;

3. the first lower side lobe level is sufficiently high to allow communication in the vicinity of the antenna;

4. the antenna is electrically tilted while maintaining sidelobe levels within predetermined limits.

These requirements are conflicting, for example, increasing boresight gain may increase the level of side lobes. Furthermore, the direction and level of the side lobes may change when the antenna is electrically tilted.

A first upper side lobe maximum level of-18 dB relative to the boresight level is believed to provide a suitable compromise in overall system performance.

The result of adjusting the mechanical or electrical tilt is to reposition the boresight so that for an array in a vertical plane, it points above or below the horizontal plane and thus alters the coverage area of the antenna. It is desirable to be able to change both the mechanical and electrical tilt of a cellular radio base station antenna: this allows maximum flexibility in optimizing cell coverage, since these forms of tilt have different effects on the antenna ground coverage and also on other antennas in the vicinity of the station. Furthermore, if the electrical tilt angle can be adjusted remotely from the antenna device, the operational efficiency is improved. Although the mechanical tilt of an antenna can be adjusted by repositioning its radome, changing its electrical tilt requires additional electronic circuitry, which increases the cost and complexity of the antenna. Furthermore, if a single antenna is shared between multiple operators, it is preferable to provide each operator with a different electrical tilt angle.

The need to obtain individual electrical tilt angles from a shared antenna has so far resulted in an impairment of the antenna performance. Due to the reduction of the antenna effective aperture, the boresight gain will decrease in proportion to the cosine of the tilt angle (which is unavoidable and occurs in all antenna designs). Further reduction in the boresight gain may result as a result of the method used to modify the tilt angle.

The "Antenna Engineers manual" (R.C. Johnson, Antenna Engineers Handbook, 3rd Ed 1993, McGraw Hill, ISBN 0-07-032381-X, Ch 20, Figure 20-2) discloses a known method for adjusting the electrical tilt of a phased array Antenna locally or remotely. In this method, a Radio Frequency (RF) transmitter carrier signal is fed into an antenna and distributed to the radiating elements of the antenna. Each antenna element has a respective phase shifter associated therewith such that the signal phase can be adjusted as the distance across the antenna is varied, thereby changing the electrical tilt angle of the antenna. The power distribution to the antenna elements when the antenna is not tilted is proportional to set the side lobe levels and boresight gain. Optimal control of the tilt angle is obtained when the phase front (phase front) is controlled for all tilt angles such that the side lobe levels do not increase within the tilt range. The electrical tilt angle can be adjusted remotely, if desired, by using a servo mechanism to control the phase shifters.

This prior art approach antenna has several disadvantages. One phase shifter is required for each antenna element. The cost of the antenna is high due to the number of phase shifters required. The cost is reduced by applying the delay means to the respective groups of antenna elements instead of the individual elements, but the side lobe levels are increased. The mechanical coupling of the delay means is used to adjust the delay, but it is difficult to make a correct adjustment; in addition, the need for mechanical links and gears results in delayed non-optimal dispensing. The upper side lobe levels increase as the antenna is tilted downward, creating a potential source of interference for mobile stations using other cells. If the antenna is shared by multiple operators, the operators have a common electrical tilt angle rather than different angles. Finally, if the antenna is used in a communication system (frequency division duplex system) having an uplink and a downlink of different frequencies, which is the usual case, the electrical tilt is different in the transmission and reception modes.

Patent applications PCT/GB2002/004166, PCT/GB2002/004930, GB0307558.7 and GB0311371.9 describe different methods of adjusting the electrical tilt of an antenna locally or remotely by means of a phase difference between two signals fed into the antenna circuit. PCT/GB2004/001297 relates to adjusting the electrical tilt by splitting a carrier signal into two signals, variably phase shifting one signal relative to the other and applying phase-to-power conversion to the resulting signal. The converted signals are separated and power-to-phase converted to be supplied to the antenna elements. The electrical tilt is adjusted by changing the phase shift between the two signals. PCT/GB2004/002016 also involves introducing a variable relative phase shift between two signals, which are then separated into components: a vector combination of these components is formed to provide respective drive signals for the respective antenna elements. Here again the electrical tilt is adjusted by changing the phase shift between the two signals.

However, there is a problem associated with splitting RF signals in that the splitter ratio (splitter ratio) may be too high to be implemented in a single splitting operation: it may require two or more cascaded operations, which increases circuit size, cost, and complexity. The reason for this is the fact that: the splitter is implemented by dividing a microstrip track (track) on the circuit board into narrower strips having different impedances compared to the track before division. With highly complex empirical expressions, the microstrip impedance is related to the track width, but for typical plate substrate thicknesses, a 50 ohm track would be 2.8 millimeters wide. The track narrows with increasing resistance until it is too narrow to be reliably soldered to the substrate. Failure to produce reliable welds occurs when the track width is below about 0.2 mm: this width produces an impedance of about 150 ohms, representing a splitter ratio of 9.5dB, and therefore it is desirable not to exceed this value for a single splitter. PCT/GB2004/001297 requires a splitter ratio of 19dB, which means that cascading at least two splitters operates.

Other potential problems are as follows: a) the required separator output may be too much, exceeding the number that can be implemented in a single separator; b) widely varying splitter ratios reduce the frequency range over which the antenna can be tilted while maintaining the desired low sidelobe levels; and c) a plurality of splitters results in a common signal to the antenna feeding the network, and the lengths of the feed lines to the individual antenna elements are different. Among these problems, c) requires the insertion of additional components such that the signal transit time to each cell is the same, thereby obtaining a phase neutral network and an optimized frequency response. All these problems make it desirable to reduce the number of separators and the separator ratio.

It is an object of the present invention to provide an alternative form of phased array antenna system.

The present invention provides a phased array antenna system with controllable electrical tilt, comprising an antenna with a plurality of antenna elements, characterized in that the system comprises:

a) means for providing two elementary signals with a variable relative delay between them,

b) separation means for splitting these elementary signals into signal components,

c) phase-to-power conversion means for converting the signal components into transformed components having powers that vary as the relative delay varies, an

d) Power-to-phase conversion means for converting the transformed components into antenna element drive signals having phases that vary gradually across the antenna from one antenna element to another as the antenna is electrically tilted and vary individually as the relative delay varies.

The invention provides the advantage that it allows control of the electrical tilt by a single variable relative delay, but multiple delays can be used if desired to increase the achievable range of the electrical tilt, and it requires fewer separate operations.

The phase-to-power conversion component may be a plurality of hybrid radio frequency coupling devices ("hybrids") for providing a sum and a difference of pairs of signal components, each pair having signal components from two fundamental signals. It may be a plurality of 180 degree hybrids for providing a sum and difference of pairs of signal components, each pair having signal components from two elementary signals. Each pair may have signal components of equal magnitude and the component magnitude of each pair is not equal to the component magnitude of the other pair.

The hybrid circuit may be a first hybrid circuit and the power-to-phase conversion means may comprise a plurality of second hybrid circuits for generating the antenna element driving signals. The splitting means may be a first splitting means and the power-to-phase converting means may comprise a second splitting means for splitting the sum and difference into components for input to the second hybrid. The first separating means may be arranged to separate each elementary signal into three signal components. The second separating member may be a plurality of bidirectional separators.

In a preferred embodiment, the invention is arranged such that all paths of the basic signal to the antenna elements contain the same number and type of components.

In another aspect, the present invention provides a method of controlling electrical tilt in a phased array antenna system comprising an antenna having a plurality of antenna elements, characterized in that the method comprises the steps of:

a) two elementary signals are provided with a variable relative delay between them,

b) these elementary signals are split into signal components,

c) converting the signal components into transformed components having powers that vary as the relative delay varies, an

d) The transformed components are converted into antenna element drive signals having phases that vary gradually across the antenna from one antenna element to another as the antenna is electrically tilted and vary individually as the relative delay varies.

The method aspect of the invention may include preferred characteristics commensurate with those of the antenna system aspect, mutatis mutandis.

In order that the invention may be more fully understood, embodiments thereof will now be described, by way of example only, with reference to the accompanying drawings, in which:

figure 1 shows the Vertical Radiation Pattern (VRP) of a phased array antenna with zero and non-zero electrical tilt;

FIG. 2 illustrates a prior art phased array antenna with adjustable electrical tilt;

FIG. 3 is a block diagram of a phased array antenna system of the present invention using a single time delay;

FIG. 4 is a block diagram of a phased array antenna system of the present invention using two time delays;

figure 5 shows a power distribution network for use in the system of figure 3 or figure 4;

FIGS. 6a and 6b illustrate a power distribution network for use in the system of the present invention having twelve element antennas;

FIG. 7 is a schematic diagram of a 180 degree hybrid RF coupling device for use in the networks of FIGS. 5 and 6;

figures 8a and 8b are vector diagrams showing the phases of the antenna element driving signals produced by the network of figure 6;

FIG. 9 shows a 180 degree hybrid circuit 182 that receives two equal A and B inputs having a relative phase shift φ between them, amplitude voltages Va and Vb;

FIG. 10 is a vector diagram of vectors + A, + B, -B, A + B, and A-B;

FIG. 11 shows how the relative amplitudes of A + B and A-B (dotted lines) vary as their relative phase difference φ is adjusted from-180 degrees to 0 degrees to +180 degrees; and

FIG. 12 shows the A + B and A-B phase changes as φ is adjusted from-180 degrees to 0 degrees to +180 degrees.

Referring to fig. 1, Vertical Radiation Patterns (VRPs) 10a and 10b of an antenna 12 are shown, the antenna 12 being a phased array of individual antenna elements (not shown). The antenna 12 is planar, has a center 14 and extends perpendicular to the plane of the figure. VRPs 10a and 10b correspond to zero and non-zero variation, respectively, in the delay or phase of the antenna element signals with array element distance from the array edge across antenna 12. They have respective main lobes 16a, 16b with a central line or "boresight" 18a, 18b, first upper side lobes 20a, 20b and first lower side lobes 22a, 22 b; 18c indicate the boresight direction for a zero delay variation to compare with the boresight direction 18b for a non-zero delay variation. If reference is made to any element in the relevant pair of elements without the suffix a or b, e.g., the side lobe 20, then the reference is to the element without distinction. VRP 10b is tilted (downwardly as shown) relative to VRP 10a, i.e., there is an angle between main beam centerlines 18b and 18c, the magnitude of which depends on the rate at which delay varies with distance across antenna 12.

VRPs must meet a number of criteria: a) high visual axis gain; b) the first upper side lobe 20 should be at a level low enough to avoid causing interference to a mobile station using another cell; c) the first lower sidelobe 22 should be at a sufficient level to enable communication in the vicinity of the antenna 12; and d) the level and direction of the side lobes should remain within predetermined design limits as the antenna is electrically tilted. These requirements conflict with each other, for example, maximizing boresight gain may increase the side lobes 20, 22. The first upper sidelobe maximum level of-18 dB is believed to provide a suitable compromise in overall system performance with respect to the boresight level (length of the primary beam 16). Due to the reduction of the effective aperture of the antenna, the boresight gain decreases in proportion to the cosine of the tilt angle. Depending on the way the tilt angle is altered, a further reduction in the boresight gain may result.

The result of adjusting the mechanical or electrical tilt is to reposition the boresight so that it points above or below the horizontal plane and thus adjusts the coverage area of the antenna. For maximum flexibility of use, the cellular radio base station preferably has available mechanical and electrical tilt, since each tilt has a different effect on the ground coverage and also on other antennas in the vicinity. It is also convenient if the electrical tilt of the antenna can be adjusted away from the antenna. Furthermore, if a single antenna is shared between multiple operators, it is preferable to provide each operator with a different electrical tilt angle, but in the prior art this compromises antenna performance.

Referring now to fig. 2, a prior art phased array antenna system 30 is shown in which the electrical tilt is adjustable. The system 30 includes an input 32 for a Radio Frequency (RF) transmitter carrier signal, which is connected to a power distribution network 34. The network 34 is connected via phase shifters phi.E0, phi.E1L to phi.E [ n ] L and phi.E1U to phi.E [ n ] U to respective radiating antenna elements E0, E1L to E [ n ] L and E1U to E [ n ] U of the phased array antenna system 30: here, the suffix U indicates the upper portion and the suffix L indicates the lower portion, n is any positive integer defining the phased array size, and the dashed lines indicating the relevant elements, such as 36, may be repeated or removed as needed for any desired array size.

The phased array antenna system 30 operates as follows. The RF transmitter carrier signal is fed via an input 32 to a power distribution network 34: network 34 divides (not necessarily averages) this signal among phase shifters phi.e0, phi.e1l to phi.e [ n ] L and phi.e1u to phi.e [ n ] U, which shift and pass their respective divided signals to associated antenna elements E0, E1L to E [ n ] L, E1U to E [ n ] U, respectively, along with the phase shifts. The phase shift is selected to select the appropriate electrical tilt angle. The power split between antenna elements E0, etc. when the tilt angle is zero is selected to set the sidelobe levels and boresight gain appropriately. Optimal control of electrical tilt is obtained when the phase front across the array of elements E0, etc., is controlled for all tilts such that the sidelobe levels do not significantly increase within the tilt range. If desired, the electrical tilt angle can be adjusted remotely by using servos to control the phase shifters phi. E0, phi. E1L to phi.E [ n ] L and phi.E1U to phi.E [ n ] U, which can be actuated mechanically.

The phased array antenna system 30 has several disadvantages:

a) one phase shifter is required for each antenna element or (more disadvantageously) each group of elements;

b) the cost of the antenna is high due to the number of phase shifters required;

c) cost is reduced by applying phase shifters to respective groups of elements instead of individual antenna elements, but side lobe levels are increased;

d) mechanical coupling of the phase shifters to set the delay correctly is difficult and mechanical links and gears are used that result in a non-optimal delay scheme;

e) the upper side lobe levels increase as the antenna is tilted downward, creating a potential source of interference for mobile stations using other base stations.

f) If the antenna is shared by different operators, all operators must use the same electrical tilt angle; and

g) in a system having an uplink and a downlink of different frequencies (frequency division duplex system), an electrical tilt angle at the time of transmission is different from an electrical tilt angle at the time of reception.

Referring now to fig. 3, a phased array antenna system 40 of the present invention with adjustable electrical tilt is shown. The system 40 has an input 42 for an RF transmitter carrier signal: the input 42 is connected to a power splitter 44 which provides two output signals V1a, V1b as input signals to a variable phase shifter 46 and a fixed phase shifter 48, respectively. Since the phase shift and time delay are equivalent at a single frequency, phase shifters 46 and 48 can also be considered time delays. They provide respective output signals V2a and V2b to the power distribution network 50, which will be described in more detail later.

The network 50 provides four drive signals which are passed via fixed phase shifters 58U1, 58U2, 58L1 and 58L2 to four equally spaced antenna elements 60U1, 60U2, 60L1 and 60L2(U upper and L lower) respectively of the phased array antenna 60. The antenna 60 has a center indicated by a dashed line 61. The antenna 60 may have any number of elements, but it has at least two elements.

The phased array antenna system 40 operates as follows. The RF transmitter carrier signal is fed (single feed line) via input 42 to power splitter 44 where it is split into equal power signals V1a and V1 b. Signals V1a and V1b feed variable phase shifter 46 and fixed phase shifter 48, respectively. The variable phase shifter 46 is controlled by the operator to apply a selectable phase shift or time delay, and the degree of phase shift applied here controls the electrical tilt of the phased array antenna 60. Fixed phase shifter 48 (conveniently but not necessarily) applies a fixed phase shift, which for convenience is arranged to be the maximum phase shift φ applicable to variable phase shifter 46_MHalf of that. This allows the phase of V1a to be- φ relative to V1b_M2 to + phi_MThe/2 range is variable and after output from phase shifters 46 and 48, the phase shifted signals become V2a and V2b as described.

Network 50 forms various vector combinations of signals from its input signals V2a and V2b to provide respective drive signals for each antenna element 60U1, etc. The phase of the drive signal varies linearly (or possibly with a shaped (coherent) phase taper) with the change in antenna element distance across the antenna 60 from the antenna element 60U2 or 60L2 at one edge, producing parallel beams that are tilted at some angle to the array boresight from the antenna 60 as desired. As is well known in the phased array art, the angle depends on the rate at which the phase changes with distance across the antenna 60. It can be demonstrated (as will be described later) that the electrical tilt of the array 60 can be changed by using only one variable phase shifter, namely the variable phase shifter 46. This is in contrast to the prior art of figure 2 which requires having multiple variable phase shifters (one corresponding phase shifter per antenna element). The electrical tilt is in one direction when the phase difference introduced by the variable phase shifter 46 is positive, and in the opposite direction when the phase difference is negative.

The fixed phase shifters 58U1, etc. apply fixed phase shifts that vary linearly (ignoring phase taper) between different antenna elements 60U1, etc. depending on the element geometry across the array 60: this will set a zero reference direction (18 a or 18b in fig. 1) for the array 60 boresight when the phase difference between signals V1a and V1b applied by variable phase shifter 46 is zero. Fixed phase shifters 58U1, etc. are not necessary, but are preferred because they can be used to a) properly scale the phase shift introduced by the tilt process, b) optimize the suppression of side lobes within the tilt range, and c) introduce an optional fixed electrical tilt angle.

If there are multiple users, each user may have a corresponding phased array antenna system 40. Alternatively, if the users are required to employ a common antenna 60, each user has a corresponding set of elements 42 to 50 in fig. 3, and the combined network combined signals are required to feed the antenna array 60. Published international patent application WO 02/082581 a2 describes such networks.

Referring now to fig. 4, there is shown yet another phased array antenna system 70 of the present invention that uses two time delays or phase shifts. The system 70 has an RF carrier signal input 72 connected to a first power splitter 74 which provides two output signals V1a, V1b for input to a first variable phase shifter 76 and a first fixed phase shifter 78, respectively. Which provide respective output signals V2a and V2b to a second fixed phase shifter 80 and a second power splitter 82. The first fixed phase shifter 78 and the second fixed phase shifter 80 may be combined in a single device, if desired. The second power splitter 82 splits the signal V2b into two signals V3b1 and V3b2, which are passed to a second variable phase shifter 84 and a third fixed phase shifter 86. The signals V3b1 and V3b2 are then communicated to first and second power distribution networks 88 and 90, respectively, which will be described in more detail later. The signal V2a is passed through a second fixed phase shifter 82 to a third power splitter 92 to be split into two signals V3a1 and V3a2 that feed the first and second power distribution networks 88 and 90, respectively.

Together, networks 88 and 90 provide eight drive signals that are delivered via fixed phase shifters 94U1 through 94L4 to eight equally spaced antenna elements 96U1 through 96L4, respectively, of a phased array antenna 96. The network 90 drives the four innermost antenna elements 96U1, 96U2, 96L1, and 96L2, and the network 88 drives the remaining antenna elements.

The phased array antenna system 70 operates as follows. The RF transmitter carrier signal is fed (single feed) via input 72 to a first power splitter 74 where it is split into equal power signals V1a and V1 b. Signals V1a and V1b are fed into the first variable phase shifter 76 and the fixed phase shifters and 78, respectively. Fixed phase shifter 78 applies a phase shift of one-half of the maximum phase shift applicable by variable phase shifter 76. The first variable phase shifter 76 provides part of the control of the electrical tilt of the phased array antenna 96 and the second variable phase shifter 76 provides the remainder of this control.

The power distribution networks 88 and 90 receive input signals V3a1/V3b1 and V3a2/V3b2, respectively, and they form a vector combination of these signals to provide a corresponding drive signal for each antenna element 96U1, etc. The phase of the drive signal varies linearly with the antenna element distance across the antenna 96. The use of two variable phase shifters 76 and 84 allows a greater range of phase shifts to be applied across the antenna 96 than a single phase variable phase shifter (as in fig. 3) and thus a greater range of electrical tilt can be obtained.

Referring now to fig. 5, a power distribution network 100 of the type typically used at 50, 88 and 90 of fig. 3 and 4 is shown, but which shows more antenna elements than those associated with the power distribution network described above. The network 100 has two inputs 102a and 102b connected to a first splitter component 106a and a second splitter component 106b, respectively. The first separating section 106a divides the input signal or vector a having the amplitude Va into three signals a1.a, a2.a, and a3.a, where a1, a2, and a3 are scalar amplitude separation ratios. Signals a1.a, a2.a and a3.a are fed to first input 1 of first, second and third 180 hybrid RF signal coupling means (hybrid circuits) 110, 112 and 114, respectively. The second splitter component 106B splits an input signal or vector B having an amplitude Vb into three signals B1.B, B2.B, and B3.B, where B1, B2, and B3 are scalar amplitude splitting ratios of the second splitter 106B. These three signals b1.b, b2.b and b3.b are fed to the second input 2 of the hybrids 110, 112 and 114, respectively. The amplitudes of the vectors a and B are equal, i.e., Va equals Vb. The hybrid circuits 110 to 114 are also referred to as sum and difference hybrid circuits.

Each hybrid circuit 110, 112 and 114 has sum and difference outputs S1/D1, S2/D2 and S3/D3, respectively, at which are the vector sum A + B and difference A-B of its input signals A and B. As will be described in more detail later, the property of such a hybrid circuit is that when the input signal amplitudes of such a hybrid circuit are equal, its sum and difference outputs have a fixed phase difference of 90 degrees between them. This is the case even if the phase difference between these input signals is different. The a + B sum signal is in phase with each other, as is the a-B difference signal, and the sum signal is at 90 degrees to the difference signal. When the phase difference between the input signals varies due to the operation of the variable phase shifter 46, the amplitudes of the sum and difference output signals vary: for example, for inphase input signals of equal amplitude, a + B-2A and a-B-0; for inverted equal amplitude input signals, a + B-0 and a-B-2A; for input signals with 90 degrees phase difference and equal amplitude, A + B and A-B are equalThe hybrid circuits 110, 112 and 114 thus act as phase-to-power conversion components, as they convert input signals of constant power but variable phase difference into output signals of variable power but constant phase difference.

The a + B sum signals from the outputs S1, S2, and S3 of the hybrid circuits 110, 112, and 114 are fed to corresponding reference inputs S1, S2, and S3 of the fourth, fifth, and sixth 180 hybrid circuits 116, 118, and 120, respectively. Similarly, the A-B difference signals from the outputs D1, D2, and D3 of the hybrid circuits 110, 112, and 114 are fed to corresponding reference inputs D1, D2, and D3 of the sixth, fifth, and fourth hybrid circuits 120, 118, and 116, respectively.

The fourth, fifth and sixth hybrid circuits 116 to 120 also have sum and difference outputs, in each case indicated by a sum and difference, whose vector sums a + B and differences a-B of the input signals are present at these outputs, respectively. The a + B sum signals are fed through respective fixed phase shifters 122U 1-122U 3 to respective antenna elements 124U 1-124U 3 in the upper half of the six-element phased array antenna 124. Similarly, the a-B difference signals are fed through fixed phase shifters 122L 1-122L 3 to respective antenna elements 124L 1-124L 3 in the lower half of the antenna 124. Strictly speaking, the phase shifters 122U1 through 122L3 and the antenna array 124 are not part of the network 100, as fig. 3 and 4, which include the network, already show these phase shifters and antenna arrays. The fourth, fifth, and sixth hybrid circuits 116 to 120 convert the power difference between their inputs into the phase difference at their outputs, and therefore they function as power-to-phase conversion sections.

Further flexibility exists in setting the required phase and amplitude of each antenna element 124L1 etc. if additional splitters are inserted between the outputs of the first, second and third hybrids 110 to 114 and the inputs of the fourth, fifth and sixth hybrids 116 to 120. To avoid the need to dissipate power other than in the antenna elements, additional hybrids and antenna elements are added to use all RF power as efficiently as possible, as long as the outputs of the first, second or third hybrids 110, 112 or 114 are separated.

The antenna elements 124U1, etc. are associated with respective fixed phase shifters 122U1, etc. for the purpose of a) setting the nominal average tilt of the antenna and b) optimizing the level of the lobe of the antenna 124 over its range of tilt.

In fig. 5, the illustrated hybrids 110 through 120 apply equal weighting to their inputs: that is, for input signals A and B, the sum output is (A + B) and the difference output is (A-B). However, they may also be constructed with unequal weighting of the inputs A and B to produce a sum output (xA + yB) and a difference output (xA-yB). Where x is the weight applied to input a and y is the weight applied to input B. To save energy in an unequal weighted hybrid circuit, the total power input to its input should be equal to the total power output from its output, ignoring heat losses that are inevitable in practical implementations. The use of unequal weighting of the hybrid circuit can yield two advantages: a) adding further flexibility to the design in optimizing antenna element phase and amplitude distribution; b) signal separation can be distributed between two or more splitter assemblies, thereby reducing the maximum separation ratio required for any splitter and improving frequency response.

When used in the system 40, the advantages of the power distribution network 100 are:

a) only one splitting operation is required at splitters 106a and 106b, each splitter splitting into only three signals;

b) the tilting is implemented by a single variable phase shifter or time delay device 46;

c) the signals at the network inputs 102a and 102b and the components into which the signals are converted pass along paths to the antenna elements 124U1 etc. through components of exactly the same number and type, i.e. a splitter and two hybrids (as mentioned, the phase shifter 122U1 etc. is not strictly part of the network 100). These paths should therefore have substantially the same electrical length, ignoring variations due to non-zero manufacturing tolerances. Thus, phase and amplitude errors in the network due to different types of components in different paths are avoided and a good beam shape can be maintained over a tilt range. Furthermore, beam shape is maintained over a larger frequency range, since the phase and amplitude errors for each path to the element vary equally and reduce the error between adjacent antenna elements;

c) ignoring components from ideal properties, the antenna can be implemented without dissipating RF power in any component outside the antenna element;

d) the cost of the phased array antenna is reduced compared to an antenna with similar performance using a plurality of variable time delay devices; and

e) the reliability of the antenna is not compromised by using a large number of variable time delay devices.

A splitter may be inserted between the output of the first three hybrids 110 to 114 and the input of the other hybrids 116 to 120 to introduce even further flexibility in setting the phase and amplitude of the signal fed to the antenna element. This will be described in the next embodiment.

Referring now to fig. 6a and 6b, a further distribution network 140 is shown in two parts 140a and 140 b: network 140 is a power distribution network of the type used with an equally spaced twelve element phased array antenna 148, but otherwise used with 50, 88 and 90 in fig. 3 and 4. Network 140 is comparable to the network described with reference to fig. 5, with the addition of a column of splitters 142c to 142h and sufficient mixing circuitry 144₄To 144₉And phase shifters 146U1 through 144L6 are fixed to provide signals for the added antenna elements 148U1 through 148L6 of the antenna 148. Parts corresponding to the above are labeled in a similar manner and the description will focus on different aspects.

As described above, at the input terminals 102a and 102B, two input signal vectors a and B having respective amplitudes Va and Vb are separated by the separators 106a and 106B into signal portions a1.a, a2.a, a3.a and B1.B, B2.B, B3.B, and fed into the first and second input terminals 1 and 2 of the first, second and third hybrid circuits 110 to 114: that is, signals a [ n +1]. a and B [ n +1]. B are input to the n-th hybrid circuit 110+2n, where n is 0, 1, and 2. The separation ratios are set such that al b1, a 2b 2 and a 3b 3 implement phase-to-power conversion in the hybrid circuits 1l0 to 114.

Referring now also to FIG. 7, this figure schematically illustrates a 180 degree hybrid coupler 132 having inputs A and B and outputs A + B and A-B. The curve 134 indicates the path from input to output, and the associated labels-180 and-360 indicate the phase shift or comparable delay experienced by signals passing along such paths. As shown, the paths 134 from the A input 1 and B input 2 to the A + B "sum" output and A to the A-B "difference" output are all associated with a-180 degree phase shift, while the path 135 from the B input 2 to the A-B "difference" output is associated with a-360 degree phase shift. The 180 degree phase shift inverts the sinusoidal signal or multiplies it by-1, while the-360 degree phase shift keeps it unchanged. Thus, both signals a and B are inverted when passed to the sum output and therefore added together, but only signal B is inverted at the difference output and therefore subtracted from a. As will be described later, if two signal vectors of equal amplitude but different phases are added and subtracted by a 180-degree hybrid circuit, the sum of the results and the difference vector are 90 degrees from each other regardless of the input phase difference. A + B and a-B are therefore 90 degrees out of phase, which is convenient (but not necessary) since it simplifies the calculation of the phase of the antenna element signal, as will be described later. The hybrid circuits 110 to 114 thus function as phase-to-power conversion sections that convert input signals, e.g., (a1.a/b1.b), equal in amplitude but variable in relative phase difference, into sum and difference output signals variable in power but constant in phase difference of 90 degrees. In addition, the A + B outputs of all three hybrids 110-114 are in phase with each other and at 90 degrees to all three A-B outputs of the hybrids.

The a + B outputs of the hybrid circuits 110 to 114 are connected to bidirectional splitters 142c, 142e and 142g, respectively, and the a-B outputs are connected to bidirectional splitters 142d, 142f and 142h, respectively. Splitters 142c to 142h split their input signals into signal portions c1/c2, d1/d2, e1/e2, f1/f2, g1/g2 and h1/h2, respectively: these sections also serve as reference labels for the output of the respective splitter and for input to the fourth to ninth hybrids 144₄To 144₉To the respective corresponding reference inputs c1 to h 2. Fourth to ninth hybrid circuits 144₄To 144₉Has a and B inputs 1 and 2 and a + B and a-B and a and difference outputs "and" difference ", and has the same structure and operation mode as the first, second and third hybrid circuits 110 to 114. Table 1 below shows fourth to ninth hybrid circuits 144₄To 144₉Which input terminals receive which signal portions: here +/-marks indicate addition and/or subtraction, respectively.

TABLE 1

The splitters 142c to 142h split their input signals into signal portions suitable for addition and subtraction to form antenna element drive signals, which may vary gradually with antenna element position across the antenna 148. Table 2 below shows fourth through ninth hybrid circuits 144₄To 144₉Which outputs "and"/"difference" of which drive which antenna elements 148U1 through 148L6 via respective fixed phase shifters 146U1 through 144L 6. The antenna elements 148U1 through 148U6 of the upper half of the antenna 148 are all formed by the fourth through ninth hybrids 144₄To 144₉The sum output of (1) and (b) is driven, but the antenna elements in the lower half are driven by the difference output of these hybrids. Fourth through ninth hybrid circuit outputs 144₄To 144₉Each receives a signal contribution from the sum or difference output of the first to third hybrids 110 to 114, rather than from both types of outputs. The input signals thereof are thus in phase with each other. Fourth to ninth hybrid circuits 144₄To 144₉Thus acting as a power-to-phase conversion component: each converts its two input signals (which are zero in phase difference but not necessarily equal in amplitude) into sum and difference output signals whose phase differences vary between different hybrids but are constant in power (ignoring any provision for amplitude tapering). The illustrated arrangement allows a traveling phase front to be achieved across the antenna 148 and allows all input power to be used efficiently. This ignores the possibility of losses due to power dissipation in non-ideal components. Excluding such losses, the power distribution network 140 does not generate a signal that is not useful for the antenna drive signal, and therefore, does not have to inefficiently process some of the input power.

Fourth hybrid circuit 144₄The outermost antenna element pairs 148U6 and 148L6 are driven. Fifth to ninth hybrid circuits 144₅To 144₉Respectively driving antenna unit pairs 148U5/148L5, 148U4/148L4, 148U3/148L3, 148U2/148L5, and 148U1/148L1, the pairs of elements being progressively closer to the antenna center 150, with each pair centered on it.

Table 2 below shows the slave hybrid circuit 144₄To 144₉The output signal of (1). The separator section c1, etc. must be scalar, but the terms in column 4 brackets of table 2, e.g., (a1.a + b1.b) and (a3.a-b3.b), are vector additions and subtractions. The phase difference is applied between Va and Vb as described previously with reference to fig. 3 or 4, and the vectors are indicated by bold characters. In addition, as described previously, the results of vector addition (a1.a + b1.b) and the like between signals of equal amplitude are all in phase with each other, and are 90-degree phase difference from all vector subtractions (a3.a-b3.b) and the like. The vector subtraction is thus all automatically added to the vector in phase 1/4 cycles.

TABLE 2

Antenna unit	Mixing	Output of	Output signal
				148U6	1444	And	c1.(a1.A+b1.B)+d1.(a3.A-b3.B)
148U5	1446	and	c2.(a1.A+b1.B)+d2.(a3.A-b3.B)
				148U4	1446	and	e1.(a2.A+b2.B)+f1.(a2.A-b2.B)
148U3	1447	and	e2.(a2.A+b2.B)+f2.(a2.A-b2.B)
				148U2	1448	and	g1.(a3.A+b3.B)+h1.(a1.A-b1.B)
148U1	1449	and	g2.(a2.A+b3.B)+h2.(a1.A-b1.B)
				148L1	1449	difference (D)	g2.(a3.A+b3.B)-h2.(a1.A-b1.B)
148L2	1448	Difference (D)	g1.(a3.A+b3.B)-h1.(a1.A-b1.B)
				148L3	1447	Difference (D)	e2.(a2.A+b2.B)-f2.(a2.A-b2.B)
148L4	1446	Difference (D)	e1.(a2.A+b2.B)-f1.(a2.A-b2.B)
				148L5	1445	Difference (D)	c2.(a1.A+b1.B)-d2.(a3.A-b3.B)
148L6	1444	Difference (D)	c1.(a1.A+b1.B)-d1.(a3.A-b3.B)

The expression in column 4 of table 2 has the form P + Q, where Q is a vector that differs in phase from vector P by 1/4 cycles. All P vectors are in phase with each other and all Q vectors are in phase with each other. They can therefore be written as P + jQ, where P and Q are scalar magnitudes of P and Q. For example, for antenna element 148U 6:

P＝c2.(a1.A+b1.B)and Q＝d2.(a3.A-b3.B) (1)

by P_nAnd Q_nDenotes in-phase and quadrature components of a voltage supplied to nth upper and lower antenna elements 148Un and 148Ln (n ═ 1 to 6), the phase phi of which voltage_nIs represented as follows:

wherein Q is_nPositive for antenna element 148Un in the upper half of antenna 148 and negative for antenna element 148Ln in the lower half.

Scalar magnitude V of nth antenna element voltage_nIs represented as follows:

the splitter ratios in this embodiment of network 140 are shown in table 3 below.

TABLE 3

All contributions to the signal arriving at antenna elements 148U1 through 148L6 from inputs 102a and 102b (e.g., c1.(a1.a + b1.b)) are passed through the same number and type of components: that is, each contribution is passed through a path that includes a three-way splitter, a hybrid circuit, a bi-directional splitter, another hybrid circuit, and a fixed phase shifter. No phase padding element, i.e. no additional element to correct for phase shifts in different paths that are not uniform, is needed. The use of two separators in each path moderates the separation ratio: this is useful because, as mentioned above, it is desirable that the splitter ratio not exceed 9.5 dB.

The three-way separators 106a and 106b primarily set the amplitude taper and the two-way separators 142c through 142h primarily set the phase taper: here "taper" means the amplitude or phase distribution across antenna elements 148U1 through 148L 6. By repeating the functional blocks, the design of the network 140 is symmetrical and facilitates relatively easy optimization. It can also be easily adapted to a different number of antenna elements in the antenna by changing the number of splitters and hybrids. Having relatively few splitters, taking into account the number of antenna elements in the array 140.

Fig. 8a is a vector diagram of the drive signals generated by network 140 for antenna elements 148U1 through 148U6 in the upper half of antenna 148: for convenience, the effects of the phase shifters 146U1 through 146L6 are ignored. Horizontal, vertical and oblique arrows such as 160, 162 and 164 indicate the in-phase component, quadrature component and actual antenna element signal vector, respectively. Circled numbers 1 through 6 such as at 166 indicate that adjacent signal vectors are associated with antenna elements 148U1 through 148U6, respectively. By having each vertical arrow 162 extend downward from the horizontal axis 168 rather than upward, i.e., by producing a corresponding mirror image of the signal vector 164 at the reflection (reflection) of the horizontal axis 168, an equivalent vector (not shown) of drive signals for the antenna elements 148L1 through 148L6 in the lower half of the antenna 148 is obtained. Fig. 8a shows that network 140 produces antenna element drive signals with phases that travel correctly across antenna 148. The best performance of the antenna 148 is obtained when the maximum tilt angle corresponding to the maximum allowable sidelobe level at tilt is selected. The splitter ratio is then selected to give a linear phase front for this maximum tilt angle.

Figure 8b is a full vector diagram corresponding to figure 8a but showing the antenna element drive signal vectors indicated by solid line arrows such as 169 for the entire antenna array 140.

Referring now to fig. 9-12, fig. 9 shows an arrangement 180 of a single 180 degree hybrid circuit 182 that receives a and B inputs having two equal amplitude voltages Va and Vb with a relative phase shift Φ therebetween. These voltages are obtained by using a single voltage V at input 184, splitting it into two equal voltages at 186, and passing one of the resulting voltages through a variable phase shifter 188. The mixing circuit 182 generates sum and difference output signals a + B and a-B from the input signals a and B.

FIG. 10 is a vector diagram of vectors + A, + B, -B, A + B, and A-B, the latter two vectors being dotted lines. Since A and B are equal, + A, + B, and-B may be shown as radii of circle 200, circle 200 being a circle circumscribing the triangle formed by vectors + A, + B, and A + B. Vectors + B and-B are equal and opposite, together providing the diameter of circle 200, and geometrically, at other points on the circle, such as origin O, the diameter subtends a right angle. However, the vectors A + B and A-B connect the origin O to the respective ends of the + B/-B diameter, and thus, the vectors A + B and A-B have a right angle (or 90 degree relative phase shift) therebetween, regardless of the phase difference φ between + A and + B.

FIG. 11 shows how the relative amplitudes of A + B and A-B (dotted lines) change as the relative phase difference φ is adjusted from-180 degrees to 0 to +180 degrees: a + B goes from 0 to 1 to 0 in a sinusoidal manner, and A-B goes from 1 to 0 to 1 in a cosine manner. FIG. 12 shows how the phase of A + B and A-B (dotted lines) changes as φ is adjusted from-180 degrees to 0 to +180 degrees: a + B changes from-90 degrees to +90 degrees and a-B initially changes from 0 at-180 degrees to +90 degrees at-0 and then moves rapidly to-90 degrees when passing 0 and then smoothly changes to 0 at +180 degrees.

The present invention provides the following electrical tilt control. As described above, the drive signal for each antenna element 148U1, etc. in fig. 6 is a vector that can be written as P + jQ. When the phase difference between the input vectors a and B (or the voltages Va and Vb) is zero, that is, Φ is 0, the difference output a-B of all the hybrid circuits 110 and the like is also zero as shown in fig. 11. Therefore, when the antenna is not tilted, the drive signals of all the antenna elements 148U1, etc. have the same phase, i.e., the "untilted" phase, and Q is 0 in P + jQ.

As the phase difference between the a and B vectors increases, fig. 11 shows that the difference output of the hybrid circuit increases and the sum output decreases. The value of Q is thus increased and the value of P is decreased. Accordingly, the phase angle of the drive signal of each antenna element 148U1, etc. is altered. Increasing phase wavefronts across the antenna elements are achieved by taking progressively smaller values for P for antenna elements progressively closer to the centerline 150 (e.g., 148U1/148L1) and progressively larger values for P for antenna elements progressively further from the centerline 150 (e.g., 148U6/148L 6). Thus transferring a portion of the antenna drive power from the center of the antenna 148 to its ends. This is achieved by appropriately connecting the outputs of the hybrids 110 to 114.

Thus, in fig. 5, the center hybrid 112 of the first, second and third hybrids feeds signals to the antenna elements 124U2 and 124L2 at an intermediate position between the center of the antenna, shown as a dashed line, and the end elements 124U3/124L3 of the antenna 124, while the other two left-most hybrids 110 and 114, each "interchanging" an a-B difference output, i.e., are connected to the fourth or sixth hybrid 116 or 120 that receives the a + B output of the other (114 or 110) hybrid. This arrangement shifts the in-phase power (P-vector component) from the center to the end of the antenna 124, achieving a traveling phase front.

Fig. 11 shows that the phase of the hybrid circuit difference output is altered 180 degrees depending on whether the phase difference between vectors a and B is positive or negative. This ensures that there is a phase front that travels across the antenna whether the antenna is tilted up or down.

The described embodiment of the invention uses a 180 degree hybrid circuit. They may be replaced with, for example, 90 degree "quadrature" hybrids and 90 degree phase shifters added to achieve the same overall function, but this is less practical.

The examples of the invention described with reference to fig. 3 to 12 are discussed in terms of operation in transmission. However, all components are reversible, and those examples may also operate as receivers. The hybrid circuit and phase shifter are reversible and the splitter becomes the re-combiner in reverse upon reception as needed.

Claims (18)

1.A phased array antenna system with controllable electrical tilt comprising an antenna (124) with a plurality of antenna elements (124U1 to 124L3), characterized in that the system (40) has:

a) means (46) for providing two elementary signals with a variable relative delay between them,

b) separating means (106a, 106b) for splitting the basic signal into signal components,

c) phase-to-power conversion means for converting the signal component into a transformed component having a power that varies as the relative delay varies, an

d) Power-to-phase conversion means for converting the transformed components into antenna element driving signals having phases that vary gradually across the antenna (124) from one antenna element (e.g., 124U1) to another antenna element (e.g., 124U2) as the antenna (124) is electrically tilted, and vary individually as the relative delay varies.

2.A system as claimed in claim 1, characterised in that the phase-to-power conversion means is a plurality of hybrid radio frequency coupling devices ("hybrids") for providing a sum and a difference of pairs of signal components (110 to 114), each pair having signal components from two elementary signals.

3.A system as claimed in claim 1, characterised in that the phase-to-power conversion means is a plurality of 180 degree hybrids (110 to 114) for providing a sum and a difference of pairs of signal components, each pair having signal components from two elementary signals.

4. A system as claimed in claim 3, characterised in that each pair has signal components of equal amplitude, but the component amplitude of each pair is not equal to the component amplitude of the other pair.

5. A system according to claim 3, characterized in that the hybrid circuit is a first hybrid circuit (110 to 114) and the power-to-phase conversion means comprises a plurality of second hybrid circuits (116 to 120) for generating antenna element driving signals.

6. The system of claim 5, characterized in that the splitting component is a first splitting component and the power-to-phase conversion component comprises a second splitting component for splitting the sum and difference into components for input to the second hybrid circuit (1444 to 1449).

7. The system of claim 6, wherein said first separating means is for splitting each of said base signals into three signal components.

8. The system of claim 6, wherein said second separator element is a plurality of bi-directional separators.

9. A system as claimed in claim 1, characterized in that the system is arranged such that all paths extending from the component providing the basic signal to the antenna element contain the same number and type of components.

10. A method of controlling electrical tilt of a phased array antenna system (40) including an antenna (124) having a plurality of antenna elements (124U1 through 124L3), characterized in that the method comprises the steps of:

a) two elementary signals are provided with a variable relative delay between them,

b) separating the base signal into signal components,

c) converting the signal component into a transformed component having a power that varies with the relative delay variation, an

d) Converting the transformed components into antenna element driving signals having phases that vary gradually across the antenna (124) from one antenna element (e.g., 124U1) to another antenna element (e.g., 124U2) as the antenna (124) is electrically tilted and vary individually as the relative delay varies.

11. A method as claimed in claim 10, characterized in that step c) is implemented using a plurality of hybrids (110 to 114) for providing sum and difference pairs of signal components, each pair having signal components from two elementary signals.

12. A method as claimed in claim 10, characterised in that step c) is implemented using a plurality of 180 degree hybrids (110 to 114) for providing sum and difference pairs of signal components, each pair having signal components from two elementary signals.

13. A method as claimed in claim 12, characterised in that each pair has signal components of equal amplitude, but the component amplitude of each pair is not equal to the component amplitude of the other pair.

14. A method according to claim 12, characterized in that the hybrid circuit is a first hybrid circuit and step d) is implemented using a plurality of second hybrid circuits (116 to 120) for generating the antenna element driving signals.

15. A method according to claim 14, characterized in that the separation in step b) is a first separation and in step d) a second separation is carried out to split the sum and the difference into components for input to the second hybrid circuit (116 to 120).

16. The method of claim 15 wherein said first splitting splits each of said base signals into three signal components.

17. The method of claim 15, wherein the second split is a plurality of bi-directional splits.

18. A method according to claim 10, characterized in that all paths extending from the element providing the basic signal to the antenna element (124U1 to 124L3) contain the same number and type of components.