HK1099141A

HK1099141A - Method and apparatus for processing data for transmission in a multi-channel communication system using selective channel inversion

Info

Publication number: HK1099141A
Application number: HK07105762.8A
Authority: HK
Inventors: J．W．凯淳; J.R.沃尔顿
Original assignee: 高通股份有限公司
Priority date: 2001-06-14
Filing date: 2005-04-15
Publication date: 2007-08-03

Description

Method and apparatus for transmitting data using selective channel inversion processing in a multi-channel communication system

The present application is a divisional application of the chinese patent application entitled "method and apparatus for processing reverse transmission data using selective channel in a multi-channel communication system" as filed on application No. 02814932.7 on 6/2002/13.

Background

FIELD

The present invention relates generally to data communications, and more particularly to a novel and improved method and apparatus for transmitting data using selective channel reverse processing within a wireless communication system.

Background

Multi-channel communication systems are commonly used to provide increased transmission capabilities for various types of communications, such as voice, data, and the like. Such a multi-channel system may be a multiple-input multiple-output (MIMO) communication system, an Orthogonal Frequency Division Modulation (OFDM) system, a MIMO system employing OFDM, or some other type of system. MIMO systems use spatial diversity to support multiple spatial subchannels, each of which may be used to transmit data, using multiple transmit antennas and multiple receive antennas. An OFDM system effectively partitions a frequency band of operation into a plurality of frequency subchannels (or bins), each of which is associated with a respective subcarrier upon which data may be modulated. A multi-channel communication system thus supports multiple "transmission" channels, each of which corresponds to a spatial subchannel within a MIMO system, a frequency subchannel within an OFDM system, or a spatial subchannel of a frequency subchannel in a MIMO system employing OFDM.

The transmission channels of a multi-channel communication system typically experience different link conditions (e.g., due to different fading and multipath effects) and may achieve different signal-to-noise-and-interference ratios (SNRs). The transmission capacity (i.e., information bit rate) that a channel may support for a particular performance level may vary from channel to channel. Also, link conditions typically change over time. The bit rate supported by the transmission channel also changes over time.

The different transmission capacities of the transmission channels, coupled with the time-varying nature of these capacities, make it difficult to provide an efficient coding and modulation scheme that can process data prior to transmission over the channels. Furthermore, the coding and modulation schemes should be simple to implement and usable in both transmitter and receiver systems in view of practical circumstances.

There is therefore a need in the art for a technique to efficiently and effectively process data transmitted on multiple transmission channels of different capacities.

SUMMARY

Aspects of the present invention provide a technique for handling transmission of data on a plurality of transport channels selected from all available transport channels. The available transmission channels (e.g., spatial subchannels and frequency subchannels in a MIMO system employing OFDM) are divided into one or more groups, each group including any number of transmission channels. In one aspect, the data processing includes coding and modulation data for each group according to a common coding and modulation scheme selected for each group to provide and weight modulation symbols for each selected transmission channel according to weights assigned to the channels. The weights effectively "invert" the selected transmission channels within each group so that the channels achieve substantially similar received signal-to-noise-plus-interference ratios (SNRs).

In one embodiment, referred to herein as Selective Channel Inversion (SCI), only "good" transmission channels are selected for data transmission, and "bad" transmission channels are not used, if the SNRs (or power gains) within each group are at or above a certain (SNR or power gain) threshold. Using this selective channel inversion, the total available transmit power for each group is distributed (unevenly) among the good transmission channels, resulting in improved efficiency and performance. In another embodiment, all available transmission channels in each group are selected, and channel inversion is performed for all available channels in the group.

Each group of transmission channels may be associated with (1) a respective (SNR or power gain) threshold for selecting a transmission channel for data transmission, (2) a respective coding and modulation scheme for coding and modulating data for the group. For a MIMO system employing OFDM, each group may correspond to a respective transmit antenna, and the transmission channels within each group may be frequency subchannels of the corresponding transmit antenna.

Channel inversion techniques simplify coding/modulation at the transmitter system and decoding/demodulation at the receiver system. Moreover, the selective channel inversion technique also provides improved performance because the combined benefits of (1) using only N within each group selected from all available transmission channels within the group_SThe best transmission channel and (2) match the received SNR for each selected transmission channel to the SNR required by the coding and modulation scheme used by the group to which the channel belongs.

The present invention also provides methods, systems, and apparatus that implement various aspects, embodiments, and features of the present invention, as described in detail below.

Brief description of the drawings

The features, nature, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like parts bear like designations and wherein:

FIG. 1 is a diagram of a multiple-input multiple-output (MIMO) communication system designed to implement various aspects and embodiments of the present invention;

FIG. 2A is a flow diagram of a process for determining an amount of transmit power allocated to each selected transport channel based on a selective channel inversion, according to an embodiment of the present invention;

fig. 2B is a flowchart of a process for determining a threshold α for selecting a transmission channel for data transmission according to an embodiment of the present invention;

FIG. 3 is a diagram of a MIMO communication system in which various aspects and embodiments of the present invention may be implemented;

FIGS. 4A through 4D are block diagrams of four MIMO transmitter systems capable of processing data in accordance with four particular embodiments of the present invention;

FIG. 5 is a block diagram of a MIMO receiver system capable of receiving data in accordance with an embodiment of the present invention;

FIGS. 6A and 6B are respective block diagrams of embodiments of a channel MIMO/data processor and an interference canceller in the MIMO receiver system shown in FIG. 5; and

fig. 7 is a block diagram of a MIMO receiver system capable of receiving data according to another embodiment of the present invention.

Detailed Description

The various aspects, embodiments and features of the present invention may be applied to any multi-channel communication system in which a plurality of transmission channels are available for data transmission. Such multi-channel communication systems include multiple-input multiple-output (MIMO) systems, Orthogonal Frequency Division Modulation (OFDM) systems, MIMO systems employing OFDM, and the like. A multi-channel communication system may also implement Code Division Multiple Access (CDMA), Time Division Multiple Access (TDMA), Frequency Division Multiple Access (FDMA), or some other multiple access technique. Multiple access communication techniques can support concurrent communication for multiple terminals (i.e., users).

Fig. 1 is a diagram of a multiple-input multiple-output (MIMO) communication system 100 designed to implement various aspects and embodiments of the invention. MIMO System 100 uses multiple (N)_T) A plurality of transmitting antennas and a plurality of (N)_R) The receive antennas are used for data transmission. MIMO system 100 is effectively formed as a multiple-access communication system with a Base Station (BS)104 communicating with multiple terminals (T)106 simultaneously. In this case, the base station 104 uses multiple antennas and represents Multiple Input (MI) for uplink transmission and Multiple Output (MO) for downlink transmission. The downlink (i.e., forward link) refers to transmission from the base station to the terminal, and the uplink (i.e., reverse link) refers to transmission from the terminal to the base station.

MIMO systems using multiple (N)_T) Transmitting antenna and a plurality of (N)_R) The receive antennas are used for data transmission. N is a radical of_TA transmitting antenna and N_RMIMO channel formed by multiple receiving antennas can be decomposed into N_CA separate channel of which N_C≤min{N_T，N_R}。N_CEach of the individual channels is also referred to as a spatial subchannel of the MIMO channel and corresponds to a dimension. In a common MIMO system implementation, N_TA plurality of transmit antennas are located at and associated with a single transmitter system, and N_RThe individual receive antennas are similarly located within and associated with a single receiver system. It is also possible to efficiently form MIMO systems for multiple access communication systems each with one base station communicating with multiple terminals simultaneously. In this case, the base station has multiple antennas and each terminal may have one or more antennas.

An OFDM system effectively divides the operating band into a number (N)_F) Frequency subchannels (i.e., frequency bins or sub-bands). At each slot, the modulation symbol may be at N_FEach of the frequency subchannels. Each time slot corresponding to a particular fetchDepending on the time interval of the frequency subchannel bandwidth.

A multi-channel communication system may be used to transmit data over multiple transmission channels. For MIMO systems that do not employ OFDM, there is typically only one frequency subchannel and each spatial subchannel may be referred to as a transmission channel. For a MIMO system employing OFDM, each spatial subchannel for each frequency subchannel may be referred to as a transmission channel. For OFDM systems that do not use MIMO, there is only one spatial subchannel for each frequency subchannel, and each frequency subchannel may be referred to as a transmission channel.

Transmission channels within a multi-channel communication system typically experience different link conditions (e.g., due to different fading or multipath effects) and may acquire different signal-to-noise-and-interference ratios (SNRs). As a result, the capacity of the transmission channel may vary from channel to channel. The capacity may be measured by the information bit rate (i.e., the number of information bits PER modulation symbol) transmitted over the transmission channel for a particular level of performance, such as a particular Bit Error Rate (BER) or Packet Error Rate (PER). Since link conditions typically vary over time, the information bit rate supported by the transmission channel also varies over time.

To fully exploit the capacity of the transmission channel, it is possible to determine (typically at the receiver system) and provide to the transmitter system Channel State Information (CSI) describing the link conditions. The transmitter system may then process (e.g., code modulate and weight) the data such that the information bit rate transmitted by each transmission channel matches the transmission capacity of the channel. CSI may be categorized as "full CSI" or "partial CSI". Full CSI is included in N_T×N_RSufficient characteristics (e.g., amplitude and phase) over the entire system band of the transmission path between each transmit-receive antenna pair within the MIMO matrix (i.e., each transmission channel characteristic). The partial CSI may include, for example, SNRs of the transmission channel.

Various techniques may be used to transmit the pre-processed data over multiple transmission channels. In one technique, the data for each transmission channel may be coded and modulated according to a particular coding and modulation scheme selected for the channel based on the CSI for the channel. By separately coding and modulating each transmission channel, the coding and modulation may be optimized for the SNR of each transmission channel. In one embodiment of this technique, the data is encoded using a fixed base code, and the coded bits for each transmission channel are then punctured (i.e., selectively removed) to obtain the code rate supported by that channel. In this implementation, the modulation scheme for each transmission channel is also selected according to the code rate and SNR of the channel. This CODING and modulation SCHEME is described in further detail in U.S. patent No. 09/776075, entitled "CODING SCHEME FOR A WIRELESS COMMUNICATION SYSTEM", filed on 2/1/2001, assigned to the assignee of the present invention and incorporated herein by reference. For this technique, practical implementation complexity is typically associated with each transmission channel with a different code rate and modulation scheme.

In accordance with an aspect of the present invention, techniques are provided to (1) process data for all selected transmission channels according to a common coding and modulation technique to provide modulation symbols, and (2) weight the modulation symbols for each selected transmission channel according to the CSI for the channel. The weighting effectively "inverts" the selected transmission signal so that the general SNRs are approximately similar at the receiver for all selected transmission channels. In one embodiment, referred to herein as Selective Channel Inversion (SCI), only "good" transmission channels with SNRs (or power gains) at or above a certain SNR (or power gain) threshold are selected for data transmission, and "bad" transmission channels are not used. With selective channel inversion, the total available transmit power is allocated over good transmission channels, resulting in improved efficiency and performance. In another embodiment, all available transport channels are selected for use and channel inversion is achieved for all transport channels.

In another embodiment, all available transport channels are grouped and the selective channel inversion is applied independently for each channel group. For example, the frequency subchannels for each transmit antenna may be grouped together and the selective channel inversion applied independently for each of the transmit antennas. This grouping allows for optimization on a per group (e.g., per transmit antenna) basis.

These channel inversion techniques may be advantageously used when there is full CSI or partial CSI at the transmitter. These techniques improve the complexity associated with the channel-specific coding and modulation techniques described above while also achieving high performance. Moreover, selective channel inversion techniques may also provide improved performance over channel-specific coding and modulation techniques, since the combined benefits of (1) using only N of the available transmission channels_SThe best transmission channel and (2) match the received SNR for each selected transmission channel to the SNR required by the selected coding and modulation scheme.

For MIMO systems employing OFDM and with full CSI available, the transmitter system may have knowledge of the complex-valued gain of the transmission path between each transmit-receive antenna pair for each frequency subchannel. This information may be used to orthogonalize the MIMO channel so that each eigenmode (i.e., spatial subchannel) may be used for an independent data stream.

For MIMO systems employing OFDM with partial CSI available, the transmitter may have limited knowledge of the transmission channel. The independent data streams may be transmitted on corresponding transmission channels on available transmit antennas, and the receiver system may use certain linear (spatial) or non-linear (space-time) processing techniques (i.e., equalization) to separate the data streams. Equalization provides a separate data stream for each transmission channel (e.g., each transmit antenna and/or each frequency subchannel), and each of these data streams has an associated SNR.

If the set of SNRs for the transmission channel is available at the transmitter system, this information may be used to select the appropriate coding and modulation scheme and allocate the total available transmit power for each group (possibly only one group). In one embodiment, the available transmission channels in each group are arranged in order of decreasing received SNR, and the total available transmit power allocation is used for N within the group_SThe best transmission channel. In one embodiment, transmission channels with received SNRs below a particular SNR threshold are not selected for use. Possibly selecting SThe NR threshold is to optimize throughput or some other criteria. The total available transmit power for each group is allocated across all selected used transmission channels within the group such that the transmitted data streams have substantially similar received SNRs at the receiver system. A similar process may be implemented if channel gain is available at the transmitter system. In an embodiment, a common coding scheme (e.g., a particular Turbo code with a particular code rate) and a common modulation scheme (e.g., a particular PSK or QAM constellation) are used for all selected transmission channels within each group.

Transmission channel inversion

If a single (common) coding and modulation scheme can be used at the transmitter system, a single (e.g., convolutional or Turbo) encoder and code rate may be used to encode data for all transmission channels selected for data transmission, and the resulting code bits may be mapped to modulation symbols using a single (e.g., PSK or QAM) modulation scheme. The resulting modulation symbols are then all derived from the same "alphabet" of possible modulation symbols and encoded with the same code and code rate. This may simplify data processing at both the transmitter and receiver ends.

However, transmission channels within a multi-channel communication system typically experience different link conditions and thus obtain different SNRs. In this case, if the same amount of transmit power is used for each selected transmission channel, the transmitted modulation symbol will be received at different SNRs depending on the particular channel transmitted by the modulation symbol. The result may be a symbol error probability that varies widely across the set of selected transmission channels and an associated loss in band efficiency.

According to an aspect of the invention, a power control mechanism is used to set or adjust the transmit power level of each transmission channel selected for data transmission to achieve a particular SNR at the receiver system. By obtaining similar received SNRs for all selected transport channels, it is possible to use a single coding and modulation scheme for all selected transport channels, which may greatly reduce the coding/modulation complexity at the transmitter system and the complexity of the complementary demodulation/decoding process at the receiver system. Power control may be achieved by "inverting" the selected transmission channels and appropriately allocating the total available transmit power on all selected channels, as will be described in detail below.

If the same amount of transmit power is used for all available transmission channels in a MIMO system employing OFDM, the received power for a particular channel can be expressed as:

img id="idf0001" file="A20061014253700101.GIF" wi="302" he="50" img-content="drawing" img-format="GIF"/

wherein

P_rx' (j, k) is the received power of the transmission channel (j, k) (i.e., the jth spatial subchannel of the kth frequency subchannel),

P_lxis the total transmit power available at the transmitter,

N_Tis the number of the transmitting antennas and,

N_Fis the number of frequency subchannels, an

H (j, k) is the complex-valued "effective" channel gain of the transmission channel (j, k) from the transmitter to the receiver.

For simplicity, the channel gain H (j, k) includes the processing effect at the transmitter and receiver. And for simplicity, assume that the number of spatial subchannels is equal to the number of transmit antennas, and N_T·N_FRepresenting the total number of available transport channels. If the same amount of power is transmitted for each available transport channel, the total received power P of all available transport channels_{rx_total}Can be expressed as:

img id="idf0002" file="A20061014253700102.GIF" wi="335" he="48" img-content="drawing" img-format="GIF"/

equation (1) shows that the received power of each transmission channel depends on the power gain of the channel, | H (j, k) & gt². To achieve equal received power on all available transmission channels, the modulation symbols for each channel may be weighted at the transmitter with a weight W (j, k), which may be expressed as:

img id="idf0003" file="A20061014253700103.GIF" wi="313" he="46" img-content="drawing" img-format="GIF"/

where c is a factor selected such that the received power of all transmission channels is approximately equal at the receiver. As shown in equation (3), the weight of each transmission channel is inversely proportional to the gain of that channel. The weighted transmit power of the transmission channel (j, k) may be expressed as:

img id="idf0004" file="A20061014253700104.GIF" wi="309" he="50" img-content="drawing" img-format="GIF"/

where b is a "normalization" factor used to allocate the total transmit power over the available transmission channels. The normalization factor b can be expressed as:

img id="idf0005" file="A20061014253700111.GIF" wi="298" he="70" img-content="drawing" img-format="GIF"/

wherein c is²B. As equation (5) shows, the calculation of the normalization factor b is the sum of the inverse of the power gain of all available transmission channels.

The modulation symbols for each transport channel are effectively "inverted" by weighting them with W (j, k). This channel inversion results in the amount of transmit power per transmission channel being inversely proportional to the channel power gain, as shown in equation (4), which then provides a receiver-side specific received power. The total available transmit power is thus effectively distributed (non-uniformly) over all available transmission channels according to their gains, such that all transmission channels have approximately equal received powers, which can be expressed as:

P_rx(j，k)＝bP_ix (6)

if the noise variance is equal for all transmission channels, equal received power allows the generation of modulation symbols for all channels based on a single common coding and modulation scheme, which greatly simplifies the encoding and decoding process.

If all available transport channels are used for data transmission regardless of their channel gains, the worse transport channel is allocated more power out of the total transmit power. In fact, to obtain similar received power for all transmission channels, the worse the transmission channel, the more transmit power needs to be allocated to that channel. When one or more transmission channels become poorly conditioned, the amount of transmit power required for those channels deprives the good channels of power, which may result in a severe drop in overall system throughput.

Selective channel inversion based on channel gain

In an aspect, the channel inverse is selectively applied and only transmission channels whose received power is at or above some particular threshold α compared to the total received power are selected for data transmission. Transmission channels whose received power is below the threshold are removed (i.e., not used). For each selected transmission channel, the modulation symbols are weighted at the transmitter such that all selected transmission channels are received at substantially similar power levels. The threshold may be selected to maximize throughput or based on some other criteria. The selective channel inversion scheme retains the simplicity of using a common coding and modulation scheme for all transport channels while also providing the high performance typically associated with individual coding per transport channel.

The average power gain L is initially calculated for all available transmission channels_aveAnd can be represented as:

img id="idf0006" file="A20061014253700112.GIF" wi="380" he="74" img-content="drawing" img-format="GIF"/

modulation symbols for each selected transmission channel are weighted at the transmitterWeighting, which can be expressed as:

img id="idf0008" file="A20061014253700122.GIF" wi="339" he="50" img-content="drawing" img-format="GIF"/

the weight of each selected transmission channel is inversely proportional to the gain of the channel and is determined such that all selected transmission channels are received at approximately equal power. The weighted transmit power for each transport channel may be expressed as:

wherein a is a threshold value, wherein a is,is a normalization factor used to allocate the total transmit power in the selected transmission channel. As shown in equation (9), if its power gain is greater than or equal to the power gain threshold (i.e., | H (j, k))²≥αL_ave) The transmission channel is selected. Normalization factorCalculated only on the basis of the selected transmission channel, can be expressed as:

img id="idf0012" file="A20061014253700126.GIF" wi="329" he="63" img-content="drawing" img-format="GIF"/

equations (7) through (10) effectively allocate the total transmit power to the selected transmission channels according to the power gain of the transmission channels such that all selected transmission channels have approximately equal received powers, which can be expressed as follows:

selective channel inversion based on channel SNRs

In many communication systems, the known quantity at the receiver is the received SNRs of the transmission channel rather than the channel gain (i.e., path loss). In such a system, the selective channel inversion technique may be modified to operate according to the received SNRs rather than the channel gain.

If equal transmit power is used for all available transmission channels and the noise variance σ²The same for all channels, then the received SNR γ (j, k) for the transmission channel (j, k) can be expressed as:

img id="idf0014" file="A20061014253700128.GIF" wi="380" he="48" img-content="drawing" img-format="GIF"/

average received SNR γ per available transmission channel_aveCan be expressed as:

img id="idf0015" file="A20061014253700131.GIF" wi="354" he="51" img-content="drawing" img-format="GIF"/

this assumes equal transmit power over the possible transport channels, the total received SNR γ for all available transport channels_totalCan be expressed as:

img id="idf0016" file="A20061014253700132.GIF" wi="353" he="51" img-content="drawing" img-format="GIF"/

total received SNR γ_totalAccordingly, an even distribution over all available transport channels based on the total transmit power.

The normalization factor β, used to distribute the total transmit power among the selected transmission channels, can be expressed as:

img id="idf0017" file="A20061014253700133.GIF" wi="370" he="60" img-content="drawing" img-format="GIF"/

as shown in fig. 15, the calculation of the normalization factor β is based on the sum of the inverse of the SNRs for all selected transmission channels.

To achieve similar received SNRs for all selected transmission channels, the modulation symbols for each selected transmission channel (j, k) may be weighted with a weight related to the SNR for that channel, which may be expressed as:

img id="idf0018" file="A20061014253700134.GIF" wi="371" he="50" img-content="drawing" img-format="GIF"/

whereinimg id="idf0019" file="A20061014253700135.GIF" wi="54" he="19" img-content="drawing" img-format="GIF"/The weighted transmit power for each transport channel may be expressed as:

as shown in equation (17), only the received SNR is greater than or equal to the SNR threshold (i.e., γ (j, k) ≧ α γ_ave) Is selected for use.

If the total transmit power is allocated across all selected transmission channels such that the received SNR is approximately similar across all selected channels, the resulting received SNR for each transmission channel may be expressed as:

by applying γ of equation (13)_aveAnd γ of equation (14)_totalSubstituting equation (18) yields the following equation:

channel inversion of packets of a transmission channel

In the above description, the channel inversion is applied to all available transmission channels or selectively to a subset of the available transmission channels (selected according to a particular threshold). This enables the use of a common coding and modulation scheme for all transport channels used for data transmission.

Selective channel inversion may be applied to groups of transport channels individually and independently. In this case, the transmission channels available in the communication system are initially divided into a certain number of groups. Any number of groups may be formed and each group may include any number of channels (i.e., the number of channels within each group need not be equal).

A specific amount of transmit power is available for each group depending on the limitations and considerations of the respective system. For the full channel inversion technique, the available transmit power for each group is allocated to all transmission channels within the group such that the received signal qualities of these channels are approximately equal (i.e., similar received SNRs). And for selective channel inversion techniques, all or a subset of the available transmission channels within each group may be selected, e.g., based on a particular threshold determined for the group. The available transmit power for each group is then allocated to selected transmission channels within the group such that the received signal qualities for these channels are approximately equal.

Processing data separately for each transport channel group provides a number of additional flexibility. For example, a full or selective channel inversion may be applied independently to each channel group. Also, for those groups to which selective channel inversion is applied, one threshold may be used for all groups, each group may be assigned a separate threshold or some groups may share the same threshold while other groups may be assigned separate thresholds. It is also possible to use different coding and modulation schemes for each group, which may be selected based on the received SNR achieved for the transmission channels within the group.

For MIMO systems employing OFDM, the construction of MIMO establishes multiple (N) in the spatial domain_S) Transmission channels, OFDM construction establishes a plurality (N) of channels in the frequency domain_T) A transmission channel. The total number of transmission channels available for transmitting data is N ═ N_S·N_F. The N transport channels may then be divided into groups in a number of ways.

In an embodiment, the transmission channels are grouped on a per transmit antenna basis. If the data of the spatial subchannel is equal to the number of transmit antennas (i.e., N)_T＝N_S) Then it is possible to pair N independently_TEach of the transmit antennas applies a full or selective channel inversion. In one embodiment, selective channel inversion is applied to each group, and corresponds to N_TN of transmitting antenna_TGroup may be with N_TEach respective threshold is associated with a threshold for each group or transmit antenna. The selective channel inversion then determines a subset of the transmission channels (or frequency channels) associated with each transmit antenna with sufficiently received SNRs, which may be achieved by dividing each frequency into two or more frequency binsThe received SNR for a subchannel is compared to a threshold for the transmitted channel. The total available transmit power available for each transmit antenna is then allocated to the selected frequency subchannels of the transmit antenna such that the received SNRs for these frequency subchannels are approximately similar.

In another embodiment, the available transmission channels are grouped on a per-frequency subchannel basis. In this embodiment, full or selective channel inversion may be applied independently to N_FEach of the frequency subchannels. If selective channel inversion is used, the spatial subchannels in each group may be selected for data transmission based on the threshold of the group corresponding to the frequency subchannel.

Grouping the available transmission channels into groups enables optimizations to be obtained on a per group (e.g., per transmit antenna or per frequency subchannel) basis, which enables the use of a particular coding and modulation scheme for all selected transmission channels within each group. For example, one or more transmit antennas may be assigned to each scheduled terminal for data transmission. The transmission channels associated with the assigned transmit antennas may be placed into groups and selective channel inversion may be implemented on the groups of transmission channels to enable the use of a single coding and modulation scheme for data transmission to the terminal.

If equal transmit power is used for all available transmission channels in group j and the noise variance σ²The same for all channels, the SNR γ received by the transmission channel k in group j_j(k) Can be expressed as:

img id="idf0023" file="A20061014253700151.GIF" wi="353" he="52" img-content="drawing" img-format="GIF"/

wherein

P_rx，j(k) Is the received power of the transmission channel k within group j,

P_tx，jis the total available transmit power for group j,

H_j(k) is the effective channel gain of the transmission channel k from the transmitter to the receiver within the group j, an

N_jIs the number of transmission channels in group j. Group j may correspond to a particular transmit antenna j, where N_j＝N_F。

Average received SNR γ for each available transmission channel in group j_ave，jCan be expressed as:

img id="idf0024" file="A20061014253700152.GIF" wi="402" he="52" img-content="drawing" img-format="GIF"/

equation (20) assumes N within group j_jThe available transmission channels are of equal transmission power. Total received SNR γ of all available transmission channels in group j_total，jCan be expressed as:

img id="idf0025" file="A20061014253700153.GIF" wi="383" he="56" img-content="drawing" img-format="GIF"/

wherein

img id="idf0026" file="A20061014253700154.GIF" wi="390" he="60" img-content="drawing" img-format="GIF"/

Total received SNR, γ, for group j_total，jIs based on the total transmit power P of the group j_tx，jAre equally distributed over all available transport channels within the group.

Normalization factor beta_jFor allocating a total transmission power P on selected transmission channels within the group j_tx，JIt can be expressed as:

img id="idf0027" file="A20061014253700161.GIF" wi="287" he="59" img-content="drawing" img-format="GIF"/

the normalization factor β is shown in equation (23)_jIs calculated based on the SNRs of all selected transmission channels within group j, which are based on a threshold α determined for that group_jγ_ave，jAnd is selected.

To achieve similar received SNRs for all selected transmission channels in the group, the modulation symbols for each selected transmission channel may be weighted with a weight related to the SNR for that channel, which may be expressed as:

img id="idf0028" file="A20061014253700162.GIF" wi="349" he="52" img-content="drawing" img-format="GIF"/

whereinimg id="idf0029" file="A20061014253700163.GIF" wi="59" he="21" img-content="drawing" img-format="GIF"/The weighted transmit power for each transport channel may then be expressed as:

as shown in equation (25), only the received SNR is greater than or equal to the SNR threshold (i.e., γ -_j(k)≥α_jγ_ave，j) Is selected for use.

If the total transmit power is distributed across all selected transmission channels in the group such that the received SNR is approximately similar for all selected channels, then the received SNR produced by each transmission channel may be represented in the form:

the process described above may be repeated for each group of transmission channels. Each group may differ from the derived threshold value alpha_jγ_ave，jCorrelated to provide the desired performance for the group. This ability to allocate transmit power on a per group (e.g., per transmit antenna) basis can provide increased flexibility and possibly improved performance.

Fig. 2A is a flow diagram of a process 200 for determining an amount of transmit power allocated to each selected transmission channel based on a selective channel inversion, according to an embodiment of the invention. Process 200 assumes that all available transport channels (i.e., a group of transport channels of the communication system) are considered. Process 200 is used if channel gain H (j, k), received SNRs γ (j, k), or some other characteristic is available for the transmission channel. For clarity, process 200 is described below for the case where channel gain is available, and the case where received SNRs are available is shown in brackets.

Initially, the channel gains H (j, k) [ or received SNRs γ (j, k) ] for all available transmission channels are obtained at step 212]. A power gain threshold α γ for selecting a transmission channel for data transmission is determined in step 214_ave[ or SNR threshold value α γ [ ]_ave]. The threshold may be calculated as described in detail below.

Each available transport channel is then evaluated for possible use. Available transmission channels for evaluation (but not yet evaluated) are identified at step 216. For the identified transmission channel, the power gain [ or received SNR ] of the channel is determined in step 218]Whether it is greater than or equal to the power gain threshold (i.e., | H (j, k))²≥αL_ave) [ or SNR threshold (i.e. gamma (j, k) ≧ alpha L_ave]. If the identified transport channel meets the criteria, it is selected for use in step 220. Otherwise if the transport channel does not meet the criteria, it is discarded and not used for data transmission.

A determination is made at step 222 whether all available transport channels have been evaluated. If not, processing returns to step 216 to identify another available transport channel to evaluate. Otherwise, processing continues to step 224.

In step 224, a normalization factor for allocating the total transmit power among the selected transmission channels[ or β ]]Will depend on the channel gain [ or received SNRs ] of the selected channel]And is determined. This may be as equation (10) [ or equation (15)]The shown acquisition. Based on the normalization factor and the gain [ or SNR ] of the channel at step 226]Calculating the weight of each selected transmission channelThe weighting may be as in equation (8) [ or equation (16)]The calculations shown. The weighted transmit power for each selected transport channel is as in equation (9) [ or equation (17)]Shown. The process is terminated.

In the above description, the total available transmit power for each group is (non-uniformly) assigned to selected transport channels within the group according to their respective weights such that the received SNRs for these channels are substantially similar (possibly only one group of transport channels). In some embodiments, the total available transmit power may be equally distributed among the selected transport channels, in which case the weights of the selected transport channels are equal. This may be achieved, for example, if the common coding and modulation scheme of a group is selected based on the average SNR of the selected transmission channels within the group. The desired level of performance may be achieved, for example, by interleaving the data on all selected transport channels within the group or by some other processing scheme.

Threshold selection

The threshold value alpha is used to select a transmission channel for data transmission according to various criteria. In one embodiment, the threshold is set to optimize throughput.

Initially, various set points are defined (i.e.img id="idf0034" file="A20061014253700173.GIF" wi="130" he="30" img-content="drawing" img-format="GIF"/) And a code rate vector (i.e.img id="idf0035" file="A20061014253700174.GIF" wi="118" he="28" img-content="drawing" img-format="GIF"/). The code rate includes the impact of the coding and modulation scheme and represents the number of information bits per modulation symbol. Each vector comprising N corresponding to the number of available code rates_ZThese may be the code rates available to the system. Alternatively, N may be defined based on the operating points supported by the system_ZA set point. Each set point corresponds to a particular received SNR needed to achieve a particular performance level. The set point generally depends on the transmission bit rate (i.e., the number of information bits per modulation symbol), which further depends on the code rate and modulation scheme used for data transmission. As described above, a common modulation scheme is used for all selected transport channels. In this case, the transmission bit rate and the setpoint are directly related to the code rate.

Each code rate r_n(wherein 1. ltoreq. N. ltoreq.N_Z) Corresponding to a set point z_nThe latter is the minimum received SNR required for operation at the code rate to achieve the required level of performance. Required setpoint z_nPossibly based on computer simulations, mathematical derivations and/or empirical measurements, as known in the art. The elements within the two vectors R and Z may also be ordered such thatTo obtainimg id="idf0036" file="A20061014253700181.GIF" wi="130" he="26" img-content="drawing" img-format="GIF"/Andimg id="idf0037" file="A20061014253700182.GIF" wi="133" he="25" img-content="drawing" img-format="GIF"/wherein z is₁Is the maximum set point, and r₁Is the highest supported code rate.

The channel gains of all available transmission channels are used to calculate power gains which are sorted and placed in a list H (λ) in descending order of power gain, where λ is 1 ≦ N_TN_FSo that H (1) { | max { | H (j, k) | Y²}., and H (N)_TN_F)＝min{|H(j，k)|²}。

Possible normalization factor sequenceIt can also be defined as follows:

img id="idf0039" file="A20061014253700184.GIF" wi="380" he="67" img-content="drawing" img-format="GIF"/

if the choice is made to use the lambda best transmission channels, the sequenceMay be used as a normalization factor.

For each code rate r_n(wherein 1. ltoreq. N. ltoreq.N_z) Determining the maximum value of λ_maxSuch that the received SNR of each of the lambda best transmission channels is greater than or equal to the code rate r_nAssociated set point z_n. This condition can be expressed as follows:

img id="idf0041" file="A20061014253700186.GIF" wi="358" he="46" img-content="drawing" img-format="GIF"/

wherein σ²Is the received noise power in a single transmission channel. Maximum value of λ_maxCan be identified by evaluating each possible value of λ that begins with 1 ending when equation (28) no longer holds. The achievable SNR for the λ best transmission channels for each value of λ may be determined as shown by the variable to the left of equation (28). The available SNR is then compared with the SNR z required by the code rate rn_nAnd (6) comparing.

Therefore, for each code rate r_nEach value of λ (for λ 1, 2.., λ)_n，max) Evaluated to determine if the received SNR of each of the λ best transmission channels can reach the associated setpoint z if the total transmit power is (unevenly) distributed over all of the λ channels_n. Maximum value λ of λ satisfying this condition_maxIs reaching the desired set point z_nWhile can be the code rate r_nThe maximum number of transmission channels selected.

And each code rate r_nAssociated threshold value alpha_nMay be expressed as:

img id="idf0042" file="A20061014253700191.GIF" wi="319" he="50" img-content="drawing" img-format="GIF"/

the threshold value alpha_nOptimization requirement set point z_nThe throughput of code rate r. Since a common code rate is used for all selected transmission channels, the maximum achievable throughput T_nCan be calculated as the throughput per channel (i.e., r)_n) Multiplied by the number of selected channels lambda_n，max. The pair of set points z_nMaximum achievable throughput T_nCan be expressed as:

T_n＝λ_n，maxr_n (30)

wherein T is_nThe unit of (b) is information bits per modulation symbol.

The optimal throughput of the setpoint vector can be given by:

T_opt＝max{T_n} (31)

as the code rate increases, more information bits may be transmitted per modulation symbol, however, the required SNR also increases, which requires a variance σ for a given noise²Each selected transmission channel of the plurality of transmission channelsAnd (4) rate. Since the total transmit power is limited, there are fewer transmission channels to achieve the higher required SNR. Thus, the maximum achievable throughput for each code rate within vector R may be calculated, and the particular code rate that provides this maximum throughput may be considered the best code rate under the particular channel condition being evaluated. Optimum threshold value alpha_optIs equal to the corresponding generation T_optSpecific code rate r_nThreshold value alpha of_n。

In the above description, the optimum threshold α_optIs determined according to the channel gains of all transmission channels. If received SNRs are available rather than channel gains, the received SNRs may be sorted and placed in a γ (λ) list in decreasing order of SNRs, where 1 ≦ λ ≦ N_TN_FSuch that the first element λ (1) ═ max { γ (j, k) } within the list, and the last element λ (N) within the list_TN_R) Min { γ (j, k) }. The sequence β (λ) can be determined as:

img id="idf0043" file="A20061014253700192.GIF" wi="283" he="66" img-content="drawing" img-format="GIF"/

for each code rate r_n(wherein 1. ltoreq. N. ltoreq.N_Z) Determining the maximum value of λ_maxSuch that the received SNR for each of the lambda selected transmission channels is greater than or equal to the associated setpoint z_n. This condition can be expressed as follows:

β(λ)N_TN_F≥z_n (33)

once to code rate r_nThe maximum value of lambda is determined_maxThen the threshold α associated with the code rate_nIt can be determined that:

img id="idf0044" file="A20061014253700201.GIF" wi="268" he="47" img-content="drawing" img-format="GIF"/

optimum threshold value alpha_optAnd an optimum throughput T_optMay also be determined as described above.

For the above description, the threshold is selected to optimize the throughput of the available transmission channels. The threshold may also be selected to optimize other performance criteria or metrics and is within the scope of the present invention.

Fig. 2B is a flow diagram of a process 240 for determining a threshold a for selecting a transmission channel for data transmission according to an embodiment of the present invention. Process 240 may be used if channel gain, received SNRs, or some other characteristic of the transmission channel is available. For simplicity, process 240 is described in the following case where channel gains are available and where received SNRs are available and shown in square brackets.

Initially, a setpoint vector is defined at step 250img id="idf0045" file="A20061014253700202.GIF" wi="152" he="27" img-content="drawing" img-format="GIF"/And determining a code rate vector supporting the associated setpointimg id="idf0046" file="A20061014253700203.GIF" wi="162" he="27" img-content="drawing" img-format="GIF"/In step 252, the channel gains H (j, k) [ or received SNRs γ (j, k) ] of all available transmission channels are retrieved]And is discharged from best to worst. In step 254, the channel gain [ shown in equation (27) ] or the received SNRs shown in equation (32) ] is then calculated]Determining a sequence of possible normalization factors[ or β (λ)]。

Each available code rate is then evaluated by the loop. In the first step of the loop, the code rate r (without evaluation)_nIdentified for evaluation at step 256. For a first pass loop, the identified code rate may be the first code rate R within the vector R_l. For the identified code rate r_nAt step 258, a maximum value λ of λ is determined_n，maxSuch that the received SNR of each of the lambda best transmission channels is greater than or equal to the estimated code rate r_nAssociated set point z_n. This can be done by calculating and satisfying equation (28) [ or equation (33)]The conditions shown. At step 260, then according to equation (29) [ or equation (34)]Shown channel lambda_n，maxChannel gain of [ or received SNR ]]Determination and setpoint z_nAssociated threshold value alpha_n. The set point may be determined at step 262 as shown by equation (30)z_nMaximum achievable throughput T_n。

It is then determined whether all of N are present at step 264_ZThe code rates are evaluated. If not, the process returns to step 256 and another code rate is identified for evaluation. Otherwise, the optimal throughput T may be determined at step 266 as shown in equation (31)_optAnd an optimum threshold value alpha_opt. The process then terminates.

In the above description, since selective channel inversion is implemented for all channels, a threshold is determined for all available transmission channels in the communication system. In an embodiment in which the transmission channels are divided into a plurality of groups, a threshold for each group may be determined. The threshold for each group may be determined based on various criteria, such as optimizing the throughput of the transmission channels included in the group.

To determine the threshold for each group, the derivation described above may be used. However, list H for each group_j(lambda) [ or gamma ]_j(λ)]Including only the power gain [ or received SNRs ] of the transmission channels included in the group]. And, the sequence[ or β (λ)]Will include channel gains [ or received SNRs ] according to the transmission channels within the group]But rather a possible normalization factor. Code rate r with group j_nAssociated threshold value alpha_j，nThen it may be expressed as:

optimal threshold value alpha of group j_opt，jEqual to the optimal throughput T for generating group j_opt，jSpecific code rate r_nThreshold value alpha of_j，n。

Each transport channel group may be associated with a corresponding threshold. Alternatively, multiple groups may share the same threshold. For example, it may be desirable to use the same coding and modulation scheme for multiple transmit antennas and to share the available transmit power among the transmit antennas.

In the above description, the threshold may be derived from the (uneven) distribution of total available transmit power among the selected transmission channels to obtain similar received SNRs for these channels. In some embodiments, the threshold may be derived based on some other condition and/or metric. For example, the threshold may be derived from an average distribution of the total available transmit power over (i.e., equal weighting for) the selected transmission channels. In this case, the threshold may be selected to maximize the achieved throughput based on the equal transmit power allocation. In another embodiment, the threshold may be only a certain (fixed) target SNR.

Multi-channel communication system

Fig. 3 is an illustration of a MIMO communication system 300 in which various aspects and embodiments of the invention may be implemented. System 300 includes a first system 310 (e.g., base station 104 in fig. 1) in communication with a second system 350 (e.g., terminal 106). System 300 may be used to use a combination of antenna, frequency, and time diversity to increase spectral efficiency, improve performance, and enhance flexibility.

At system 310, a data source 312 provides data (i.e., information bits) to a Transmit (TX) data processor 314, which (1) encodes the data according to a particular coding scheme, (2) interleaves (i.e., reorders) the encoded data according to a particular interleaving scheme, (3) maps the interleaved bits to modulation symbols for one or more transmission channels selected for data transmission, and (4) weights the modulation symbols for each selected transmission channel. The encoding increases the reliability of the data transmission. Interleaving provides time diversity of the coded bits so that data can be transmitted according to the average SNR of the selected transmission channel, countering fading and further removing the correlation between the coded bits used to form each modulation symbol. Interleaving can further provide frequency diversity if the coded bits are transmitted on multiple frequency subchannels. The weights effectively control the transmit power of each selected transmission channel to achieve a desired SNR at the receiver. In an aspect, the encoding, symbol mapping, and weighting may be implemented in accordance with a control signal provided by controller 334.

TX channel processor 320 receives and demultiplexes the weighted modulation symbols from TX data processor 314 and provides weighted modulation symbols for each selected transmission channel, one weighted modulation symbol per time slot. TX channel processor 320 may further pre-adjust the weighted modulation symbols for the selected transmission channel if full CSI is available.

If OFDM is not employed, TX channel processor 320 provides a weighted modulation symbol stream for each antenna used for data transmission. And if OFDM is employed, TX channel processor 320 provides a vector stream of weighted modulation symbols for each antenna used for data transmission. And TX channel processor 320 provides a stream of pre-conditioned modulation symbols or a stream of pre-conditioned modulation symbol vectors to each antenna for data transmission if full CSI processing is implemented. Each stream is then received and modulated by a respective Modulator (MOD)322 and transmitted via an associated antenna 324.

At the receiver 350, a plurality of receive antennas 352 receive the transmitted signals and provide received signals to respective demodulators (DEMODs) 354. Each demodulator 354 performs processing complementary to that performed at modulator 322. The modulation symbols from all demodulators 354 are provided to a Receive (RX) channel/data processor 356 and processed to recover the transmitted data streams. An RX channel/data processor 356 performs processing complementary to that performed by TX data processor 314 and TX channel processor 320 and provides decoded data to a data sink 360. This processing by the receiver system 350 is described in detail below.

MIMO transmitter system

Fig. 4A is a block diagram of a MIMO transmitter system 310a that processes data in accordance with an embodiment of the present invention. Transmitter system 310a is an embodiment of the transmitter portion of system 310 of fig. 3. System 310a includes (1) a TX data processor 314a that receives and processes the information bits to provide weighted modulation symbols and (2) a TX channel processor 320a that demultiplexes the modulation symbols for the selected transmission channel.

In the illustrated embodiment of fig. 4A, TX data processor 314A includes an encoder 412, a channel interleaver 414, a puncturer 416, a symbol mapping element 418, and a symbol weighting element 420. Encoder 412 receives the information bits to be transmitted in aggregate and encodes the received bits according to a particular coding scheme to provide coded bits. Channel interleaver 414 interleaves the coded bits according to a particular interleaving scheme to provide diversity. Truncator 416 truncates (i.e., deletes) zero or more of the interleaved and coded bits to provide the desired number of coded data. The symbol mapping element 418 maps the non-truncated bits to modulation symbols for the selected transmission channel. And symbol weighting element 420 weights the modulation symbols for each selected transmission channel to provide weighted modulation symbols. The weight for each selected transmission channel may be determined based on the SNR achieved for that channel, as described above.

Pilot data (e.g., data of a known pattern) may also be encoded and multiplexed with the processed information bits. The processed pilot data may be transmitted (e.g., in a Time Division Multiplexed (TDM) manner) on a subset or all of the selected transport channels or a subset or all of the available transport channels. The pilot data may also be used at the receiver to perform channel estimation, as described below.

As shown in fig. 4A, data encoding, interleaving, and puncturing may be achieved in accordance with one or more encoding control signals that identify the particular encoding, interleaving, and puncturing scheme to be used. The symbol mapping may be obtained from a modulation control signal that identifies the particular modulation scheme to be used. And the symbol weights may be obtained based on the weights provided to the selected transmission channels.

In one coding and modulation scheme, coding is achieved by using a fixed base code and adjusting puncturing to achieve a desired code rate supported by the SNR of a selected transmission channel. The base code may be a Turbo code, a convolutional code, a rank chain code, or some other code. The base code may also have a particular code rate (e.g., code rate 1/3). For this scheme, puncturing may be performed after channel interleaving to achieve a desired code rate for a selected transmission channel.

Symbol mapping element 416 may be designed to group the unpunctured sets of bits to form non-binary symbols and map each non-binary symbol to a point on the signal constellation corresponding to the modulation scheme selected for the selected transmission channel. The modulation scheme may be QPSK, M-PSK, M-QAM, or some other scheme. Each mapped signal point corresponds to a modulation symbol.

The encoding, interleaving, puncturing, and symbol mapping at the transmitter 310a may be implemented according to a number of schemes. One particular approach is described in the aforementioned U.S. patent application serial No. 09/776075.

The number of information bits that may be transmitted PER modulation symbol for a particular performance level (e.g., one percent packet error rate, PER) depends on the received SNR. Thus, the coding and modulation schemes of the selected transmission channels may be determined based on channel characteristics (e.g., channel gain, received SNRs, or some other information). The channel interleaving may also be adjusted according to the coded control signal.

Table 1 lists various combinations of code rates and modulation schemes that are possible for multiple ranges of received SNRs. The supported bit rate for each transport channel may be achieved by using one of a number of possible combinations of coding rates and modulation schemes. For example, one information bit per modulation symbol may be obtained using a combination of (1)1/2 coding rate and QPSK modulation, (2)1/3 coding rate and 8-PSK modulation, (3)1/4 coding rate and 16-QAM, or some other combination of coding rate and modulation scheme. In Table 1, QPSK, 16-QAM, and 64-QAM are used for the listed SNR ranges. Other modulation schemes, such as 8-PSK, 32-QAM, 128-QAM, etc., may also be used and are within the scope of the present invention.

TABLE 1

Received (a)

Information per symbol

Modulation code element

Per symbol element is coded

Coding rate

SNR Range	Number of bits		Number of bits of
SNR Range	Number of bits		Number of bits of		1.5-4.4	1	QPSK	2	1/2
4.4-6.4	1.5	QPSK	2	3/4	1.5-4.4	1	QPSK	2	1/2
4.4-6.4	1.5	QPSK	2	3/4	6.4-8.35	2	16-QAM	4	1/2
8.35-10.4	2.5	16-QAM	4	5/8	6.4-8.35	2	16-QAM	4	1/2
8.35-10.4	2.5	16-QAM	4	5/8	10.4-12.3	3	16-QAM	4	3/4
12.3-14.15	3.5	64-QAM	6	7/12	10.4-12.3	3	16-QAM	4	3/4
12.3-14.15	3.5	64-QAM	6	7/12	14.15-15.55	4	64-QAM	6	2/3
15.55-17.35	4.5	64-QAM	6	3/4	14.15-15.55	4	64-QAM	6	2/3
15.55-17.35	4.5	64-QAM	6	3/4	＞17.35	5	64-QAM	6	5/6

The weighted modulation symbols from TX data processor 314a are provided to TX channel processor 320a, which is an embodiment of TX channel processor 320 in fig. 3. Within TX channel processor 320a, a demultiplexer 424 receives and demultiplexes the weighted modulation symbols into a plurality of modulation symbol streams, one for each transmission channel used to transmit the modulation symbols. Each modulation symbol stream is provided to a respective modulator 322. If OFDM is employed, the weighted modulation symbols at each slot of all selected frequency subchannels for each transmit antenna are combined into a vector of weighted modulation symbols. Each modulator 322 converts the weighted modulation symbols (for systems that do not employ OFDM) or vector of weighted modulation symbols (for systems that employ OFDM) into an analog signal and further amplifies, filters, quadrature modulates, and upconverts the signal to generate a modulated signal suitable for transmission over a wireless link.

Fig. 4B is a block diagram of a MIMO transmitter system 310B that processes data in accordance with another embodiment of the present invention. Transmitter system 310b is another embodiment of the transmitter portion of system 310 of fig. 3 and includes a TX data processor 314b and a TX channel processor 320 b.

In the illustrated embodiment of fig. 4B, TX data processor 314B includes an encoder 412, a channel interleaver 414, a symbol mapping element 418, and a symbol weighting element 420. Encoder 412 receives the information to be transmitted in its entirety, bits, and encodes them according to a particular coding scheme to provide coded bits. The encoding may be performed according to a particular encoding and code rate selected by the controller 334, which is identified by the encoding control signal. The coded bits are interleaved by a channel interleaver 414 and symbol mapping element 418 maps the interleaved bits to modulation symbols for the selected transmission channel. Symbol weighting element 420 weights the modulation symbols for each selected transmission channel according to the corresponding weights to provide weighted modulation symbols.

In the embodiment shown in fig. 4B, transmitter 310B can pre-adjust the weighted modulation symbols based on full CSI. Within TX channel processor 320b, a channel MIMO processor 422 demultiplexes the weighted modulation symbols into a plurality (up to N)_COne for each spatial subchannel (i.e., eigenmode) used to transmit the modulation symbols. For full-CSI processing, channel MIMO processor 422 pre-adjusts the (up to NC) weighted modulation symbols at each slot to generate NT pre-adjusted modulation symbols, as follows:

img id="idf0050" file="A20061014253700251.GIF" wi="408" he="105" img-content="drawing" img-format="GIF"/

wherein, b₁，b₂，...bN_CN are spatial subchannels 1, 2,. N, respectively_CWeighted modulation symbols of (1);

e_ijis an element of an eigenvector matrix E related to transmission characteristics from the transmit antennas to the receive antennas;

and

img id="idf0051" file="A20061014253700252.GIF" wi="242" he="19" img-content="drawing" img-format="GIF"/

img id="idf0052" file="A20061014253700253.GIF" wi="247" he="19" img-content="drawing" img-format="GIF"/and

img id="idf0053" file="A20061014253700254.GIF" wi="271" he="19" img-content="drawing" img-format="GIF"/

the eigenvector matrix E may be calculated by the transmitter or provided to the transmitter by the receiver. The elements of matrix E are also considered in determining the effective channel gain H (j, k).

For full CSI processing, each pre-adjusted modulation symbol x for a particular transmit antenna_iRepresents up to N_CLinear combinations of weighted modulation symbols for the spatial subchannels. For each time slot, up to N generated by channel MIMO processor 422_TThe individual pre-adjusted modulation symbols are demultiplexed by demultiplexer 424 and provided to (up to) N_TA modulator 322. Each modulator 322 converts a pre-adjusted modulation symbol (for systems that do not employ OFDM) or a vector of pre-adjusted modulation symbols (for systems that employ OFDM) into a modulated signal suitable for transmission over a wireless link.

Fig. 4C is a block diagram of a MIMO transmitter system 310C employing OFDM that processes data in accordance with another embodiment of the present invention. Transmitter system 310c is an embodiment of the transmitter portion of system 310 of fig. 3 and includes a TX data processor 314c and a TX channel processor 320 c. TX data processor 314c may be used to independently code and modulate each transport channel group according to a coding and modulation scheme specifically selected for the group. Each group may correspond to one transmit antenna and the transmission channel of each group may correspond to a frequency subchannel of the transmit antenna.

In the embodiment shown in fig. 4C, TX data processor 314C includes multiple spatial subchannel data processors 410a through 410t, one data processor 410 enabling each transport channel group to be independently coded and modulated. Each data processor 410 includes an encoder 412, a channel interleaver 414, a symbol mapping element 418, and a symbol weighting element 420. These elements of data processor 410 are used to encode the information bits for the groups processed by the data processor, interleave the encoded bits, map the interleaved bits to generate modulation symbols, and weight the modulation symbols for each selected transmission channel within the group. As shown in fig. 4C, it is possible to specifically provide coding and modulation control and weighting for each group.

The weighted modulation symbols from each data processor 410 are provided to a respective combiner 434 within TX channel processor 320c, which combines the weighted modulation symbols for the particular transmit antenna. If each group includes frequency subchannels selected by a particular transmit antenna, combiner 434 combines the weighted modulation symbols for the selected frequency subchannels to form vectors of modulation symbols for each transmission channel, which are then provided to respective modulators 322. The process by which each modulator 322 generates a modulated signal is described below.

Fig. 4D is a block diagram of a MIMO transmitter system 310D employing OFDM that processes data in accordance with another embodiment of the present invention. In this embodiment, it is possible to process the transmission channel of each frequency subchannel independently. Within TX data processor 314c, the information bits to be transmitted are demultiplexed by demultiplexer 428 into multiple (up to NL) frequency subchannel data streams, one for each frequency subchannel used for data transmission. Each frequency subchannel data stream is provided to a respective frequency subchannel data processor 430.

Each data processor 430 processes data for a respective frequency subchannel of the OFDM system. Each data processor 430 may be implemented in a manner similar to TX data processor 314A of fig. 4A, TX data processor 314B of fig. 4B, or some other design. In one embodiment, the data processor 430 demultiplexes the frequency subchannel data stream into a plurality of data substreams, one for each of the spatial subchannels of the frequency subchannel. Each data substream is then encoded, interleaved, symbol mapped, and weighted to generate weighted modulation symbols for the data substream. The coding and modulation of each frequency sub-channel data stream or each data sub-stream may be adjusted according to coding and modulation control signals and the weighting may be implemented according to weights. Each data processor 430 then provides up to N selected for a frequency subchannel_CUp to N of spatial subchannels_CIs weightedModulating the symbol stream.

For a MIMO system employing OFDM, modulation symbols may be transmitted from multiple transmit antennas on multiple frequency subchannels. Within MIMO processor 320d, up to NC streams of modulation symbols from each data processor 430 are provided to a respective subchannel spatial processor 432, which processes the received modulation symbols based on channel control and/or available SI. Each spatial processor 432 may simply be implemented as a demultiplexer (as shown in fig. 4A) if full CSI processing is not performed, or as a channel MIMO processor followed by a demultiplexer (as shown in fig. 4B) if full CSI processing is performed. For MIMO systems employing OFDM, it is possible to achieve full CSI processing (i.e., pre-conditioning) on each frequency subchannel.

Each subchannel spatial processor 432 will have up to N per slot_CDemultiplexing of individual modulation symbols into up to N for the transmit antennas selected for the frequency subchannel_TAnd a modulation symbol. For each transmit antenna, combiner 434 receives up to N selected for that transmit antenna_LThe modulation symbols for each frequency subchannel, the symbols for each slot are combined into a vector of modulation symbols V, and the vector of modulation symbols is provided to the next processing stage (i.e., the corresponding modulator 322).

MIMO processor 320d thus receives and processes the modulation symbols to provide up to N_TVector of modulation symbols V_lTo V_NtA vector of modulation symbols is provided for each transmit antenna selected for data transmission. Each vector of modulation symbols V covers a single time slot and each element of the vector of modulation symbols V is associated with a particular frequency subchannel with a unique subcarrier on which the modulation symbols are transmitted.

Fig. 4D also shows an embodiment of a modulator 322 for OFDM. Vector of modulation symbols V from MIMO processor 320c_lTo V_NtAnd correspondingly to modulators 322a through 322 t. In the embodiment shown in fig. 4D, each modulator 322 includes an Inverse Fast Fourier Transform (IFFT)440, a cyclic prefix generator 442, and an upconverter444。

IFFT440 IFFT converts each received vector of modulation symbols into its time-domain representation (referred to herein as an OFDM symbol). The IFFT440 may be designed to implement an IFFT over any number of frequency subchannels (e.g., 8, 16, 32, etc.). In one embodiment, for each modulation symbol that is transformed into an OFDM symbol, cyclic prefix generator 442 repeats a portion of the OFDM symbol time domain representation to form a "transmission symbol" for the particular transmit antenna. The cyclic prefix ensures that the transmission symbol retains its orthogonal properties in the presence of multipath delay spread, thus improving the effect of degraded path effects. The implementation of the IFFT440 and cyclic prefix generator 442 is known in the art and will not be described in detail herein.

The time domain representation from each cyclic prefix generator 442 (i.e., the transmission symbols for each antenna) is then processed (e.g., converted to analog signals, modulated, amplified, and filtered) by an upconverter 444 to generate a modulated signal, which is then transmitted from the corresponding antenna 324.

OFDM Modulation is described in the title "Multicarrier Modulation for Data Transmission: a paper by An IdeaWhose Time Has Comm "by John A.C. Bingham, IEEE communication Magazine, 1990, 5 months, incorporated herein by reference.

Fig. 4A through 4D illustrate four designs of a MIMO transmitter capable of implementing various aspects and embodiments of the present invention. The invention can also be applied in OFDM systems that do not employ MIMO. In this case, the available transmission channels correspond to frequency subchannels of the OFDM system. A number of other transmitter designs are possible to implement the various inventive techniques described herein and are within the scope of the present invention. Some of these transmitter designs are described in further detail in the following patent applications, assigned to the assignee of the present invention and incorporated herein by reference:

● U.S. patent application No. 09/776075, described above

● U.S. patent application Ser. No. 09/532492, entitled "HIGH EFFICIENCY, HIGHPERFORMANCE COMMUNICATIONS SYSTEM EMPLOYING MULTIPLE-CARRIER MODULATION", filed on 3/22/2000.

● U.S. patent application Ser. No. 09/826481 entitled "METHOD AND APPARATUS FORMULATION CHANNEL STATE INFORMATION IN A WIRELESS COMMUNICATION SYSTEM" filed on 3/23 of 2001.

● U.S. patent application Ser. No. 09854235 entitled "METHOD AND APPARATUS FOR PRECURSOSSING DATA IN A MULTIPLE-INPUT MULTIPLE-OUTPUT (MIMO) COMMUNICATION SYSTEM UTILIZING CHANNEL STATE INFORMATION", filed 3, 11 days 2001.

These patent applications also describe MIMO processing and CSI processing in detail.

In general, transmitter system 310 encodes and modulates all selected transport channels (or all selected transport channels within each group) according to a particular common coding and modulation scheme. The modulation symbols are further weighted by weights assigned to the selected transmission channels such that a desired level of performance is obtained at the receiver. The techniques described herein may be applied to multiple parallel transmission channels supported by MIMO, OFDM, or any other communication scheme (e.g., CDMA scheme) that can support multiple parallel transmission channels.

Fig. 4C illustrates an embodiment in which data for each transmit antenna may be separately encoded and modulated according to the coding and modulation scheme selected for that transmit antenna. Likewise, fig. 4D illustrates an embodiment in which the data for each frequency subchannel may be separately encoded and modulated according to the coding and modulation schemes selected for that frequency subchannel. In general, all available transmission channels (e.g., all spatial subchannels of all frequency subchannels) may be divided into any number of groups of any type, and each group may include any number of transmission channels. For example, each group may include spatial subchannels, frequency subchannels, or subchannels in both domains.

MIMO receiver system

Fig. 5 is a block diagram of a MIMO receiver system 350a capable of receiving data in accordance with an embodiment of the present invention. Receiver system 350a is a particular embodiment of receiver system 350 of fig. 3 and implements a successive cancellation receiver processing technique to receive and recover the transmitted signal. From (up to) N_TThe transmitted signals from the transmitting antennas are composed of N_REach of the antennas 352a through 352r receives and is routed to a respective demodulator (DEMOD)354 (referred to herein as a front-end processor).

Each demodulator 354 conditions (e.g., filters and amplifies) the corresponding received signal, downconverts the conditioned signal to an intermediate frequency or baseband, and digitizes the downconverted signal to provide samples. Each demodulator 354 may also demodulate samples with a received pilot to generate a stream of received modulation symbols, which may be provided to an RX channel/data processor 356 a.

If OFDM is used for data transmission, each demodulator 354 further implements processing complementary to that implemented by modulator 322 shown in fig. 4D. In this case, each demodulator 354 includes an FFT processor (not shown) that generates a translated representation of the samples and provides a vector stream of modulation symbols. Each vector comprising up to N_LUp to N of selected frequency sub-channels_LOne modulation symbol and one vector is provided for each slot. For a transmit processing scheme in which each frequency subchannel is processed independently (e.g., as shown in FIG. 4D), from all N_RThe stream of vectors of modulation symbols from the FFT processors of the demodulators is provided to a demultiplexer (not shown in fig. 5) which "channelizes" the stream of vectors of modulation symbols from each FFT processor to up to N_LA stream of modulation symbols corresponding to the number of frequency subchannels to be used for data transmission. The demultiplexer will then add up to N_LEach of the individual modulation symbol streams is provided to a respective RX MIMO/data processor 356 a.

For MIMO systems that do not employ OFDM, one RX MIMO/data processor 356a may be used to process the secondary N_RN from one receiving antenna_RA stream of modulation symbols. For MIMO systems employing OFDM, one RX MIMO/data processor 356a may be used to process N_RA plurality of modulation symbol streams from up to N for data transmission_LN of each of the frequency sub-channels_RA receiving antenna. Alternatively, a single RX channel/data processor 356a may be used to separately process the set of modulation symbol streams associated with each frequency subchannel.

In the embodiment illustrated in fig. 5, RX channel/data processor 356a, which is an embodiment of RX channel/data processor 356 in fig. 3, includes multiple successive (cascaded) receiver processor stages 510, one for each transmitted data stream recovered by receiver system 350 a. In a transmit processing scheme, selective channel inversion is applied to all available transport channels. In this case, the selected transport channel may be used to transmit one or more data streams, each of which may be independently encoded with a common coding scheme. In another transmit processing scheme, selective channel inversion is applied separately for each transmit antenna. In this case, the selected transmission channel for each transmit antenna may be used to transmit one or more data streams, where each data stream may be independently encoded with the coding scheme selected for that transmit antenna. In general, if one data stream is independently encoded and transmitted on each spatial subchannel, successive cancellation receiver processing techniques may be used to recover the transmitted data stream. For purposes of simplicity, RX channel/data processor 356a is described for the embodiment where one data stream is independently encoded and transmitted on each spatial subchannel for a given frequency subchannel processed by data processor 356 a.

Each receiver processing stage 510 (except for the last stage 510n) includes a channel MIMO/data processor 520 coupled to an interference canceller 530, while the last stage 510n includes only a channel MIMO/data processor 520 n. For the first receiver processing stage 510a, a channel MIMO/data processor 520a receives and processes N from demodulators 354a through 354r_RA stream of modulation symbols to provide a decoded data stream (or first transmitted signal) for a first transport channel and for each of second through last stages 510b through 510nA channel MIMO/data processor 520 receives and processes N from interference canceller 520 in a preceding stage_RA modified symbol stream to derive a decoded data stream for the transport channel processed by the stage. Each channel MIMO/data processor 520 further provides CSI (e.g., the received SNR) for the associated transmission channel.

For the first receiver processing stage 510a, interference canceller 530a operates from all N_RA demodulator 354 receives N_RA stream of modulation symbols. And for each of the second through penultimate stages, interference canceller 530 receives N from the interference canceller in the previous stage_RA modified symbol stream. Each interference canceller 530 also receives a decoded data stream from a channel MIMO/data processor 520 in the same stage and performs processing (e.g., encoding, interleaving, modulation, channel response, etc.) to derive N_RA remodulated symbol stream that is an estimate of an interference component of the received modulated symbol stream due to the decoded data stream. The remodulated symbol stream is then subtracted from the received modulated symbol stream to derive N_RA modified symbol stream that includes all but the subtracted (i.e., cancelled) interference component. N is a radical of_RThe modified symbol stream is then provided to the next stage.

In fig. 5, a controller 540 is shown coupled to RX channel/data processor 356a and may be used to direct various steps in the successive cancellation receiver processing performed by processor 356 a.

Fig. 5 shows a receiver structure that may be used directly when each data stream is transmitted on a respective transmit antenna (i.e., one data stream for each transmitted signal). In this case, each receiver processing stage 510 may be configured to recover one of the transmitted signals sent to receiver system 350a and provide a decoded data stream corresponding to the recovered transmitted signal.

For some other transmit processing schemes, the data streams may be transmitted over multiple transmit antennas, frequency subchannels, and/or time intervals to provide spatial, frequency, and time diversity, respectively. For these schemes, receiver processing begins by deriving a received stream of modulation symbols for the channel transmitted on each transmit antenna for each frequency subchannel. Modulation symbols for multiple transmit antennas, frequency subchannels, and/or time intervals may be combined in a complementary manner, such as demultiplexing implemented at a transmitter system. The combined modulation symbol streams are then processed to provide corresponding decoded data streams.

Fig. 6A is a block diagram of an embodiment of a channel MIMO/data processor 520x, which is an embodiment of channel MIMO/data processor 520 shown in fig. 5. In this embodiment, channel MIMO/data processor 520x includes a space/space-time processor 610, a CSI processor 612, a selector 614, a demodulation element 618, a deinterleaver 618, and a decoder 620.

The space/space-time processor 610 is at N_RPerforming linear spatial processing on received signals for non-dispersive MIMO channels (i.e., with flat fading), at N_RSpace-time processing is performed on the received signal for dispersive MIMO channels (i.e., with frequency selective fading). Spatial processing may be obtained using linear spatial processing techniques, such as Channel Correlation Matrix Inversion (CCMI) techniques, Minimum Mean Square Error (MMSE) techniques, and others. These techniques may be used to remove undesired signals or to maximize the received SNR of each constituent signal in the presence of noise and interference from other signals. The space-time processing may be implemented using linear space-time processing techniques, such as an MMSE linear equalizer (MMSE-LE), a Decision Feedback Equalizer (DFE), a Maximum Likelihood Sequence Estimator (MLSE), and so forth. The CCMI, MMSE-LE, and DFE techniques are further described in the aforementioned U.S. patent application Ser. No. 09854235. DFE and MLSE techniques are described by s.l. ariyavistakul et al in the title "optimal Space-time processors with dispersion Interference: the Unified Analysis and RequiredFilter Span "paper is described in detail in IEEE trans. on communication, Vol.7, month 7 1999, and is incorporated herein by reference.

The CSI processor 612 determines CSI for each transmission channel used for data transmission. For example, the CSI processor 612 may estimate a noise covariance matrix from the received pilot signals and then calculate the SNR for the kth transmission channel of the data stream to be decoded. The SNR may be estimated similar to conventional pilot-assisted single-carrier and multi-carrier systems, as is known in the art. The SNR for all transmission channels used for data transmission may include CSI reported back to the transmitter system. CSI processor 612 may further provide a control signal to selector 614 that identifies the particular data stream to be recovered by the receiver processing stage.

Selector 614 receives the multiple symbol streams from space/space-time processor 610 and extracts the symbol streams corresponding to the data streams to be decoded, as indicated by the control signals from CSI processor 612. The decimated stream of modulation symbols is then provided to a demodulation element 614.

For the embodiment shown in fig. 6A, where the data streams for each transmission channel are independently encoded and modulated based on a common coding and modulation scheme, the recovered modulation symbols for the selected transmission channel are demodulated (e.g., M-PSK, M-QAM) according to a demodulation scheme that is complementary to the common modulation scheme used for the transmission channel. The demodulated data from demodulation element 616 is then deinterleaved by deinterleaver 618 in a manner complementary to that performed by channel interleaver 614, and the deinterleaved data is further decoded by decoder 620 in a manner complementary to that implemented by encoder 612. For example, if Turbo or convolutional coding is implemented at the transmitter system, a Turbo decoder or a Viterbi decoder may be used for decoder 620. The decoded data stream from decoder 620 represents an estimate of the recovered transmitted data stream.

Fig. 6B is a block diagram of an interference canceller 530x, which is an embodiment of interference canceller 530 of fig. 5. Within interference canceller 530x, the decoded data stream from channel MIMO/data processor 520 in the same stage is re-encoded, interleaved, and re-modulated by a channel data processor 628 to provide re-modulated symbols, which are estimates of the modulation symbols at the transmitter system prior to MIMO processing and channel distortion. Channel data processor 628 executes at the transmitter systemThe same processing (e.g., encoding, interleaving, and modulation) is performed on the data stream. The remodulated symbols are then provided to a channel simulator 630, which processes the symbols with an estimated channel response to provide an estimate of the interference due to the decoded data streamThe channel response estimate may be derived from data transmitted by the pilot and/or transmitter system and according to techniques described in the aforementioned U.S. patent application serial No. 09/854235.

Interference vectorInner N_RThe elements correspond to symbol streams transmitted at the kth transmit antenna at N_RA component of the received signal on each of the receive antennas. Each element of the vector represents an estimated component of a decoded data stream within a corresponding received modulation symbol stream. These components are for_RWithin each received modulation symbol stream (i.e., vector)r ^k) The interference of the transmitted signal (not yet detected) remains and is derived by summer 632 from the received signal vectorr ^kIs subtracted (i.e., eliminated) to provide a modified vector with components from the removed decoded data streamr ^k+1. Modified vectorr ^k+1As input vectors to the next receiver processing stage, as shown in fig. 5.

Various aspects of successive cancellation receiver processing are described in further detail in the aforementioned U.S. patent application serial No. 09/854235.

Fig. 7 is a block diagram of a MIMO receiver system 350b capable of receiving data in accordance with another embodiment of the present invention. From (up to) N_TThe transmitted signals from the transmitting antennas are composed of N_REach of the antennas 352a through 352r receives and is routed to a corresponding demodulator 354. Each demodulator 354 conditions, processes, and digitizes a respective received signal to provide samples, which are then provided toAn RX MIMO/data processor 356 b.

Within RX MIMO/data processor 356b, the samples for each receive antenna are provided to a respective FFT processor 710, which generates a transformed version of the received samples and provides a corresponding vector stream of modulation symbols. The vector streams of modulation symbols from FFT processors 710a through 710r are then provided to a processor 720. Processor 720 channelizes the vector stream of modulation symbols from each FFT processor 710 into a plurality of up to N_LA stream of subchannel symbols. Processor 720 may also perform spatial processing or space-time processing on the subchannel symbol streams to provide processed modulation symbols.

For each data stream transmitted on multiple frequency subchannels and/or multiple spatial subchannels, processor 720 may also combine all of the modulation symbols for the frequency and spatial subchannels used to transmit the data stream into one stream of processed modulation symbols and provide it to data stream processor 730. Each data stream processor 730 performs demodulation, deinterleaving and decoding complementary to those performed on the data streams at the transmitter unit and provides corresponding decoded data streams.

Receiver systems that use successive cancellation receiver processing computations as well as those that do not use successive cancellation receiver processing computations may be used to receive, process, and recover the transmitted data streams. Some receiver systems capable of processing signals received on multiple transmission channels are described in the above U.S. patent application serial nos. 09/776075, 09/826481, and 09/532492, entitled "HIGH EFFICIENCY, HIGH radio MODULATION transmission multiple-CARRIER MODULATION", filed 3.2000 and assigned to the assignee of the present invention and incorporated herein by reference.

Obtaining CSI for a transmitter system

For the sake of brevity, various aspects and embodiments of the present invention have been described in the context of CSI being comprised of SNRs. In general, the CSI may comprise any type of information indicative of the characteristics of the communication link. Various types of information may be provided as CSI, some examples of which are described below.

In one embodiment, the CSI consists of SNR, which is derived as the ratio of the signal power to the power of the noise plus interference. The SNR is typically estimated and provided for each transmission channel (e.g., each transmitted data stream) used for data transmission, although an aggregate SNR may also be provided for multiple transmission channels. The SNR estimate may be quantized to a value with a certain number of bits. In an embodiment, the SNR estimate is mapped to an SNR index, e.g., using a look-up table.

In another embodiment, the CSI consists of power control information for each spatial subchannel for each frequency subchannel. The power control information may include a single bit per transmission channel to indicate a request for more or less power, or it may include multiple bits to indicate the magnitude of the requested power level change. In this embodiment, the transmitter system may use power control information fed back from the receiver system to determine which transmission channels to select and which power to use for each transmission channel.

In another embodiment, the CSI consists of signal gain and interference plus noise power. These two components may be separately derived and provided to each transmission channel for data transmission.

But in another embodiment the CSI consists of signal power, interference power and noise power. These three components are derived and provided to each transmission channel for data transmission.

In another embodiment, the CSI is formed by the signal-to-noise ratio plus a list of interference powers for each observable interference term. This information may be derived and provided to each transmission channel for data transmission.

In another embodiment, the CSI is formed by signal components in the form of a matrix (e.g., N for all transmit-receive antenna pairs)_T×N_RComplex term) and a matrix-form noise plus interference component (e.g., N)_T×N_RPlural terms). Hair-like deviceThe transmitter system may then combine the signal components and the noise-plus-interference components, as appropriate, for the appropriate transmit-receive antenna pairs to derive a quality for each transmission channel used for the data transmission (e.g., a post-processing SNR for each transmitted data stream received at the receiver system).

In another embodiment, the CSI consists of a data rate indicator for each transmitted data stream. The quality of a transmission channel used for data transmission may be initially determined (e.g., based on an estimated SNR for the transmission channel) and a data rate corresponding to the determined channel quality may then be identified (e.g., based on a look-up table). The identified data rate indicates a maximum data rate that may be transmitted at the transmission channel for the required level of performance. The data rate is then mapped and represented as a Data Rate Indicator (DRI), which can be efficiently encoded. For example, if the transmitter system of each transmit antenna supports (up to) seven possible data rates, a 3-bit value may be used to represent the DRI, where, for example, zero may indicate a zero data rate (i.e., no transmit antenna is used) and 1 to 7 may be used to indicate seven different data rates. In a typical implementation, the quality measurements (e.g., SNR estimates) are mapped directly to the DRI according to, for example, a look-up table.

In another embodiment, the CSI consists of an indication of the particular processing scheme used at the transmitter system for each transmitted data stream. In this embodiment, the indication may identify a particular coding scheme and a particular modulation scheme to use for transmitting the data stream to achieve the desired performance.

In another embodiment, the CSI consists of a differential indicator of a particular quality measurement of the transmission channel. Initially, the SNR or DRI or some other quality measure of the transmission channel is determined and reported as a reference measurement. Thereafter, quality monitoring of the transmission channel is continued and the difference between the last reported measurement and the current measurement is determined. The difference may be quantized to one or more bits and the quantized difference mapped to and represented by a difference indicator, which is then reported. The differential indicator may indicate that the most recently reported measurement was increased or decreased (or maintained) by a particular step size. For example, the differential indicator may indicate that (1) the observed SNR for a particular transmission channel has increased or decreased by a particular step size, or (2) the data rate should be adjusted by a particular amount, or some other change. The reference measurements may be transmitted periodically to ensure that errors in the differential indicators and/or erroneous reception of these indicators do not accumulate.

In another embodiment, the CSI consists of the channel gain for each available transmission channel, as estimated at the receiver system from the signal transmitted by the transmitter system.

Other forms of CSI may also be used and are within the scope of the invention. In general, the CSI includes any type and form of information for (1) selecting a set of transmission channels that will yield optimal or near optimal throughput, (2) weighting factors for each selected transmission channel that will yield transmission channels at or near SNRs, and (3) deriving optimal or near optimal code rates for the selected transmission channels.

The CSI may be derived based on signals transmitted from the transmitter system and received at the receiver system. In an embodiment, the CSI is derived based on a pilot reference included in the transmitted signal. Alternatively or additionally, the CSI may be derived based on data included within the transmitted signal. While data may be transmitted on only selected transmission channels, pilot data may be transmitted on unselected transmission channels to enable the receiver system to estimate channel characteristics.

In another embodiment, the CSI is comprised of one or more signals transmitted from the receiver system to the transmitter system. In some systems, there may be some correlation between the uplink and downlink (e.g., in a Time Division Duplex (TDD) system where the uplink and downlink share the same frequency band in a time division multiplexed manner). In these systems, the quality of the uplink may be estimated (to the necessary accuracy) based on the quality of the downlink, and vice versa based on the transmitted signal (e.g., pilot signal) from the receiver system. The pilot signal is a method by which the transmitter system can estimate the CSI observed at the receiver system. For this type of CSI, there is no need to report channel characteristics.

The signal quality may be estimated at the transmitter system according to various techniques. Some of these techniques are described in the following patents, which are assigned to the assignee of the present invention and are incorporated herein by reference:

● U.S. Pat. No. 5799005 entitled "SYSTEM AND METHOD FOR DETERMINING RECEIVED PILOT POWER AND PATH LOSS IN A CDMA COMMUNICATION SYSTEM" filed on 25/8 of 1998,

● U.S. Pat. No. 5903554 entitled "METHOD AND APPATUS FOR MEASURING LINKQUALITY IN A SPREAD SPECTRUM COMMUNICATION SYSTEM", filed on 11/5 of 1999,

● U.S. Pat. Nos. 5056109 and 5265119, both entitled "SYSTEM AND METHOD FOR ROTATING TRANSMISSION POWER IN A CDMA CELLULAR MOBILE TELEPHONESYSTEM", filed 1991, 8.10 and 23.11.1993,

● U.S. Pat. No. 6097972 entitled "METHOD AND DAPPARTUS FOR PROCESSOR PROCEDURE CONTROL SIGNALS IN CDMA MOBILE TELEPHONE SYSTEM" filed on 8/1/2000.

Schemes for estimating a single transmission channel from pilot signals or data transmissions can also be found in various papers in the art. One such channel estimation method is described by f.ling in its paper entitled "optical Reception, Performance Bound, and current-Rate Analysis of referred Coherent code CDMA Communications with applications", IEEE transmission On Communication, 10 months 1999.

Information on various types of CSI, as well as various CSI reporting mechanisms, are also described in U.S. patent application Ser. No. 08963386 entitled "METHOD AND APPARATUS FOR HIGH RATE PACKET DATATRANSMISSION", filed on 3.11.1997, assigned to the assignee of the present invention, AND also described in "TIE/EIA/IS-856 cdma2000 High Rate Packet Data Air interface Specification", both of which are incorporated herein by reference.

The CSI may be reported back to the transmitter using various CSI transmission schemes. For example, the CSI may be transmitted in full or differential form or in a combination thereof. In an embodiment, the CSI is reported periodically and differential updates are sent based on previously sent CSI. In another embodiment, the CSI is only sent when there is a change (e.g., if the change exceeds a certain threshold), which may reduce the effective rate of the feedback channel. As an example, SNRs may be sent back (e.g., in differential form) when there is a change in them. For OFDM systems (sampled or without MIMO), correlation in the frequency domain may be used to reduce the amount of CSI fed back. As an example of an OFDM system, if the SNR of a particular spatial subchannel corresponding to M frequency subchannels is the same, the SNR and the first and last frequency subchannels for which this condition holds may be reported. Other compression or feedback channel error recovery techniques to reduce the amount of data for the fed back CSI may also be used and are within the scope of the invention.

Referring to fig. 3, the CSI determined by RX channel/data processor 356 (e.g., the received SNR) is provided to a TX data processor 362, which processes the CSI and provides processed data to one or more modulators 354. Modulator 354 further conditions the processed data and transmits CSI back to transmitter system 310 via the reverse channel.

At system 310, the transmitted feedback signal is received by an antenna 324, demodulated by a demodulator 322, and provided to a RX data processor 332. RX data processor 332 performs processing complementary to that performed by TX data processor 362 and recovers the reported CSI prior to providing it to controller 334.

Controller 334 uses the reported CSI to perform a variety of functions, including (1) selecting N for data transmission_SA set of best available transmission channels, (2) determining the coding and modulation to be used for data transmission on the selected transmission channelA scheme, and (3) determining weights for the selected transport channels. Controller 334 may select a transmission channel for high throughput based on some other performance criteria or metric and may further determine a threshold for selecting a transmission channel, as described above.

Characteristics of the transmission channels available for data transmission (e.g., channel gain or received SNRs) may be determined in accordance with the various techniques described above and provided to the transmitter system. The transmitter system may then use this information to select N_SThe set of best transmission channels, properly encode and modulate the data, further weighting the modulation symbols.

The techniques described herein may be used for data transmission from a base station to one or more terminals on the downlink and may also be used for data transmission from each of one or more terminals to a base station on the uplink. For the downlink, the transmitter system 310 of fig. 3 and 4A through 4D may represent a portion of a base station and the receiver system 350 in fig. 3, 5, and 6 may represent a portion of a terminal. For the uplink, the transmitter system 310 of fig. 3 and 4A through 4D may represent a portion of a terminal, and the receiver system 350 in fig. 3, 5, and 6 may represent a portion of a base station.

Elements of the transmitter and receiver systems may be implemented using one or more Digital Signal Processors (DSPs), Application Specific Integrated Circuits (ASICs), processors, microprocessors, controllers, microcontrollers, Field Programmable Gate Arrays (FPGAs), programmable logic devices, other electronic components, or a combination of any of the foregoing. Some of the functions and processes described herein may also be implemented in software executing on a processor. Aspects of the invention may also be implemented using a combination of software and hardware. The calculations to determine the threshold a and select the transmission channel may be implemented, for example, in accordance with program code executing on a processor (controller 334 in fig. 3).

Headings are included herein for reference only and to aid in locating certain sections. These headings are not intended to limit the scope of the inventions described below, and these concepts may have applicability in other sections throughout the entire specification.

The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims

1. In a multi-channel communication system, a method of determining a threshold for selecting a transmission channel for data transmission, comprising:

defining a set of code rates, wherein each code rate is selected to encode data prior to transmission;

defining a set of set points, wherein each set point corresponds to a respective code rate and indicates a target signal-to-noise-plus-interference ratio required for a particular performance level at the corresponding code rate;

determining a specific number of transmission channels supported by each code rate, and obtaining a set point of the corresponding code rate;

determining a performance metric for each code rate based in part on the number of supported transmission channels; and

a threshold is derived based on a performance metric for the code rate in the set, and wherein a transmission channel for data transmission is selected based on the threshold.

2. The method of claim 1, wherein the number of supported transport channels per code rate is determined by allocating the total available transmit power among the supported transport channels such that a setpoint corresponding to the code rate is obtained for each supported transport channel.

3. The method of claim 1, wherein the performance metric for each code rate is an achievable overall throughput of a supported transport channel.

4. The method of claim 1, wherein the setpoint is a minimum received signal-to-noise-plus-interference ratio required for a particular level of performance at a corresponding code rate.

5. The method of claim 1, further comprising the step of ordering all available transmission channels according to their channel gains.

6. The method of claim 1 wherein the threshold is an optimization threshold corresponding to a code rate that results in an optimized throughput.

7. The method of claim 1, wherein the particular number of transport channels is a maximum number of transport channels that can be selected for each of the code rates while reaching a set point corresponding to that code rate

8. The method of claim 1, further comprising the step of ordering all available transmission channels according to their signal-to-noise ratios.

9. The method of claim 3, wherein the maximum achievable throughput is the throughput per the transmission channel multiplied by a particular number of the transmission channels.

10. The method of claim 5, further comprising the step of determining at least one normalization factor.