HK1097051B - Inverter controller with feed-forward compensation and its control method - Google Patents

Description

Converter controller with feedforward compensation, conversion system and control method thereof

Technical Field

The present invention relates to a display device, and more particularly, to a converter controller for controlling a backlight power of a display device and a control method thereof.

Background

Discharge lamps, such as Cold Cathode Fluorescent Lamps (CCFLs), are widely used as background light sources in Liquid Crystal Displays (LCDs) of devices such as portable computers and televisions. The CCFL is activated by a DC/AC converter that converts a direct current signal to an alternating current signal and converts the input voltage to the higher voltage required by the CCFL. Typically, a controller is used to adjust the variation of the input voltage to control the CCFL current. One method of regulating the input voltage is to use a Pulse Width Modulation (PWM) technique, in which the CCFL current is modulated by a PWM control signal in a feedback loop.

Us patent No. 6,876,157 discloses a CCFL converter controller that employs PWM techniques to regulate CCFL input voltage variations. A feedback signal representing CCFL current is input to one input of an error amplifier, the other input of the error amplifier receiving a signal representing CCFL reference brightness. A compensation capacitor is connected between the output terminal of the error amplifier and the ground terminal, and generates a PWM control voltage signal based on the output of the error amplifier to control the PWM circuit. When the input voltage increases, the value of the error signal decreases; when the input voltage decreases, the value of the error signal increases. Therefore, the error amplifier needs to charge and discharge the compensation capacitor, that is, the controller needs to respond to the input voltage with a transient variation for a while, so the controller adjusts the input voltage with a slow variation speed, and cannot control the CCFL current at the transient variation of the input voltage. In addition, since the error signal varies with the variation of the input voltage, the time for charging and discharging the compensation capacitor by the error signal also varies. Therefore, when the CCFL brightness is adjusted by using the low-frequency PWM dimming signal, the duty ratio of the PWM signal outputted by the PWM circuit is also changed, thereby affecting the CCFL brightness.

Another CCFL converter controller employs feed-forward compensation of the CCFL input voltage to regulate the CCFL current. The controller includes an error amplifier for receiving a current offset signal inversely proportional to the input voltage to effect feed forward compensation of the input voltage. However, it is difficult to control the current deviation to be exactly proportional to the input voltage in the integrated circuit, that is, the input voltage cannot be accurately compensated, and thus, the feedforward compensation performance of the controller is not good. In addition, the circuitry will also become complex, thereby increasing cost.

Disclosure of Invention

It is an object of the present invention to provide a converter controller with feedforward compensation that accurately feed forward compensates the power supply supplying the load, controlling the load current at the instant of power supply change.

It is another object of the present invention to provide a method for accurate feed forward compensation of a power supply supplying a load, thereby avoiding variations in load current with variations in the power supply.

To achieve the above object, the converter controller with feedforward compensation to control the load current of the present invention includes an error amplifier, a pulse signal generator, and a driver connected in series; and an oscillator connected between a power supply supplying power to the load and a ground terminal. The pulse signal generator is connected between the error amplifier and the driver. The oscillator outputs an oscillating signal having an amplitude related to the magnitude of the power supply. The error amplifier receives a reference signal and a feedback signal indicative of the load current and outputs an error signal. The error signal is input to a first input terminal of the pulse signal generator, and the oscillation signal of the oscillator is input to a second input terminal of the pulse signal generator. The pulse signal generator outputs a pulse signal to the driver according to the error signal and the oscillation signal, and the amplitude of the oscillation signal controls the duty ratio of the pulse signal, so that the feed-forward compensation of the power supply is realized.

According to the converter controller with feed-forward compensation for controlling the load current, the amplitude of the oscillating signal is proportional to the power supply size.

The converter controller with feed forward compensation to control load current according to the present invention further comprises a time component connected between the power supply and ground.

The converter controller with feedforward compensation to control the load current of the invention, the time component comprises a resistor and a capacitor connected in series, and the joint between the resistor and the capacitor is connected to the output end of the oscillator.

According to the converter controller with feedforward compensation for controlling the load current, the oscillating signal is a sawtooth wave with a fixed frequency.

The converter controller with feedforward compensation for controlling load current of the present invention further includes a compensation capacitor connected between the output terminal of the error amplifier and ground.

The converter controller with feedforward compensation to control the load current is characterized in that the power supply is a direct current voltage source.

The present invention also provides a switching system with feed forward compensation to control load current, comprising: a power conversion circuit connected to a power supply that supplies power to the load and converting the power supply into a signal required by the load; and a controller for controlling the power conversion circuit, including an error amplifier, a pulse signal generator and a driver connected in series, an oscillator connected between the power supply and a ground terminal, the pulse signal generator being connected between the error amplifier and the driver, the error amplifier receiving a reference signal and a feedback signal representing a load current and outputting an error signal based on the reference signal and the feedback signal, the error signal being input to a first input terminal of the pulse signal generator, an oscillation signal output from the oscillator being input to a second input terminal of the pulse signal generator, the pulse signal generator outputting a pulse signal based on the error signal and the oscillation signal, the pulse signal controlling an output of the power conversion circuit via the driver, an amplitude of the oscillation signal varying with a magnitude of the power supply, thereby adjusting the duty cycle of the pulse signal to achieve feed forward compensation of the power supply.

According to the conversion system with feed-forward compensation for controlling the load current, the amplitude of the oscillating signal is proportional to the power supply size.

The switching system with feed forward compensation for controlling load current according to the present invention, the controller includes a time component connected between the power supply and ground.

The switching system with feedforward compensation to control the load current comprises a resistor and a capacitor which are connected in series, and the junction between the resistor and the capacitor is connected to the output end of the oscillator.

According to the conversion system with the feedforward compensation function for controlling the load current, the oscillating signal is a sawtooth wave with a fixed frequency.

The conversion system with feed-forward compensation to control the load current is characterized in that the power supply is a direct-current voltage source.

According to the conversion system with the feedforward compensation function for controlling the load current, the power supply conversion circuit converts the direct-current voltage source into the alternating-current voltage required by the load based on the pulse signal.

The switching system with feedforward compensation for controlling load current according to the present invention includes a compensation capacitor connected between the output terminal of the error amplifier and ground.

The invention discloses a method for realizing feedforward compensation of a power supply for supplying power to a load, which comprises the following steps: generating an oscillating signal having an amplitude proportional to the magnitude of said power supply; generating a feedback signal indicative of the load current; comparing the feedback signal with a reference signal to generate an error signal; and comparing the error signal with the oscillation signal to generate a pulse signal with a duty ratio varying with the amplitude of the oscillation signal so as to control the current of the load.

The method for realizing feedforward compensation of the power supply for supplying power to the lamp source generates a low-frequency pulse width modulation dimming signal for controlling the error signal so as to adjust the current of the lamp source.

Compared with the prior art, when the power supply voltage changes, the amplitude of the oscillation signal generated by the oscillator is changed to realize feedforward control, which is equivalent to changing the duty ratio of the pulse signal along with the power supply voltage, so that the error signal is kept unchanged. Therefore, the converter controller with feedforward compensation can immediately and accurately adjust the duty ratio of the pulse signal when the power supply voltage changes, thereby avoiding the influence caused by the change of the power supply voltage, and the power supply voltage can have a wider range.

Drawings

FIG. 1 is a circuit block diagram of a converter system including a converter controller of the present invention;

FIG. 2 is a circuit diagram of a high frequency oscillator in the converter controller of the present invention;

FIG. 3 is a simplified diagram of the circuit of FIG. 2 when the high frequency oscillator is in a normal operating mode;

FIG. 4 is a timing diagram of various signal waveforms in the converter controller of FIG. 1;

fig. 5 is another timing diagram of various signal waveforms in the converter controller of fig. 1.

Detailed Description

Fig. 1 is a block circuit diagram of a conversion system 100 of the present invention. The system 100 includes a power conversion circuit 160 and a converter controller 110 that controls the output of the power conversion circuit 160. The power conversion circuit 160 is a DC/DC or DC/AC circuit that provides power to the lamp source. In the present embodiment, the lamp source is at least one Cold Cathode Fluorescent Lamp (CCFL)170, and the inverter circuit 160 is a DC/AC circuit for converting the DC voltage source VIN to an AC voltage required by the CCFL 170, such as a full bridge, half bridge, push-pull and/or Class D inverter circuit. In one embodiment, the conversion circuit 160 converts the dc voltage source VIN into a sinusoidal voltage for transmission to the cold cathode fluorescent lamp 170.

The inverter controller 110 of the present invention includes an error amplifier 120, a high frequency oscillator 130 with feed forward compensation, a comparator 140 that generates a PWM drive signal, and a driver 150 that drives the CCFL inverter circuit 160. Error amplifier 120, comparator 140, driver 150 and switching circuit 160 are connected in series, the non-inverting input of error amplifier 120 receives a reference signal, such as a voltage reference signal VREF indicative of the desired brightness of the CCFL, and the inverting input thereof receives a voltage feedback signal VFB via a feedback circuit 180 that is proportional to the current actually flowing through the CCFL. The error amplifier 120 compares the reference signal with the feedback signal and generates an error signal CMP at its output. The error signal CMP is input to the non-inverting input of the comparator 140. A compensation capacitor 11 is connected between the output terminal of the error amplifier 120 and the ground terminal GND. The high frequency oscillator 130 is connected between a dc voltage source VIN for supplying power to the cold cathode fluorescent lamp 170 and a ground terminal, and an output terminal thereof is connected to a junction 123 between an external resistor 12 and an external capacitor 13 connected in series between the dc voltage source VIN and the ground terminal, the resistor 12 and the capacitor 13 being time components for determining a frequency of an oscillation signal RTCT outputted from the high frequency oscillator 130 in a normal operation mode. The oscillation signal RTCT output from the high-frequency oscillator 130 at its output terminal is input to the inverting input terminal of the comparator 140. The comparator 140 outputs a PWM driving signal according to the error signal CMP and the oscillation signal RTCT. The driver 150 receives the PWM driving signal and controls the output of the conversion circuit 160. The error amplifier 120, the high frequency oscillator 130, the comparator 140 and the driver 150 may be integrated on the same chip.

Fig. 2 shows a circuit diagram of the high-frequency oscillator 130 in the converter controller 110 according to the present invention. The high-frequency oscillator 130 includes a resistor divider connected between the dc voltage source VIN and the ground, and the resistor divider includes a first resistor 21 and a second resistor 22 connected in series. A resistor 32 and switches 34 and 36 connected in series are provided between the dc voltage source VIN and the contact 123. In one embodiment switches 34, 36 are N-channel enhancement mode MOS transistors. The high-frequency oscillator 130 further includes a first comparator 23 and a second comparator 24. The junction CTIN between switches 34 and 36 is connected to the non-inverting input of the first comparator 23 and the inverting input of the second comparator 24. The first comparator 23 compares the signal at the junction CTIN with a first reference voltage VRH obtained at the junction 212 after the dc voltage source VIN is divided by the resistor divider, and VRH is obtained by the following formula: VRH ═ VIN × (R2/(R1+ R2)); where R1 and R2 represent the resistance values of the first resistor 21 and the second resistor 22, respectively. It can be seen that the first reference voltage VRH is proportional to the dc voltage source VIN. In one embodiment, the dc voltage source VIN is typically in the range of 6 to 30 volts, and the first reference voltage VRH is set to 1/8 to 1/10 of the dc voltage source VIN by selecting the first resistor 21 and the second resistor 22 with appropriate sizes. At the same time, the second comparator 24 compares the signal at the junction CTIN with the second reference voltage VRL. The second reference voltage VRL is a voltage signal that is invariant with the dc voltage source VIN and has a fixed value, preferably a value close to zero voltage, such as 0.1 v. The high frequency oscillator 130 further comprises a flip-flop 26 for receiving the outputs of the first comparator 23 and the second comparator 24. In the present embodiment, the flip-flop 26 is an RS flip-flop, and obviously other types of flip-flops can be used, the high frequency oscillator 130 further includes a switch 25 connected across the capacitor 13, that is, between the junction 123 and the ground terminal, the on state of the switch 25 is controlled by the output of the flip-flop 26. In one embodiment, switch 25 is an N-channel enhancement MOS transistor, although other types of transistors may be used. The operation of the high-frequency oscillator 130 according to this embodiment will be described in detail below.

The high frequency oscillator 130 can be operated in the following three different modes by the lamp lighting detection signal LOB indicating whether the cold cathode fluorescent lamp 170 is lit and the enable signal POFF from the external circuit of the inverter system 100 to control whether the cold cathode fluorescent lamp 170 is operated or not to control the on state of the switches 34, 36: CCFL ignition mode, CCFL normal operation mode and standby mode. The enable signal POFF controls the on state of the switch 36 through the inverter 38. When the cold cathode fluorescent lamp 170 is in the glow starting mode, the enable signal POFF is at a low level, and the detection signal LOB is at a high level, so as to control both the switches 34 and 36 to be turned on. When the cold cathode fluorescent lamp 170 is in a normal operating mode after being turned on, the enable signal POFF is at a low level to control the switch 36 to be turned on; at the same time, the detection signal LOB is low, thereby turning off the switch 34. In the standby mode, the enable signal POFF is at a high level, the switch 36 is turned off, and the high frequency oscillator 130 does not operate, thereby turning off the current of the whole circuit, so that the current leakage can be reduced, the power consumption can be reduced, and the power saving can be achieved in the standby mode. The operation of the high frequency oscillator 130 in the normal operation mode and the ignition mode will be described in detail below.

When the high-frequency oscillator 130 is in the normal operation mode, the enable signal POFF and the lamp ignition detection signal LOB are both low, so that the switch 34 is controlled to be off and the switch 36 is controlled to be on, since the switch 36 is equivalent to a short circuit in the on state, i.e., the contact CTIN coincides with the contact 123, the circuit of the high-frequency oscillator 130 in the normal operation can be represented by the circuit shown in fig. 3. The operation of the high frequency oscillator 130 in the normal operation mode is described below with reference to fig. 3. Dc voltage source VIN charges capacitor 13 via resistor 12 to generate a voltage signal RTCT at junction 123. The voltage signal RTCT is compared with the first reference voltage VRH by the first comparator 23. The output of the first comparator 23 remains low until the voltage signal RTCT equals the first reference voltage VRH. At the same time, the second comparator 24 compares the voltage signal RTCT with the second reference voltage VRL. The output of the second comparator 24 remains low until the voltage signal RTCT is less than the second reference voltage VRL. When the voltage signal RTCT is equal to the first reference voltage VRH, the output of the first comparator 23 changes from low to high, and the output of the second comparator 24 remains low, and then the output of the RS flip-flop 26 changes to high to control the switch 25 to turn on, and the switch 25 turns on to release the charge on the capacitor 13, when the capacitor 13 discharges to a level where the voltage signal RTCT is less than the second reference voltage VRL, the output of the first comparator 23 changes to low again, and the output of the second comparator 24 changes to high, and then the output of the RS flip-flop 26 changes to low to control the switch 25 to turn off, so that the power source VIN continues to charge the capacitor 13. It can be seen that the voltage signal RTCT at the junction 123 is an oscillating signal. The oscillation frequency f in the normal operation mode is obtained by the following formula:

where R denotes the resistance value of the external resistor 12 and C denotes the capacitance value of the external capacitor 13.

Referring to fig. 2, when the high-frequency oscillator 130 is in the ignition mode, the enable signal POFF is at a low level, and the lamp ignition detection signal LOB is at a high level, so that both the switches 34 and 36 are turned on. At this time, in addition to the first current branch from the dc voltage source VIN to the junction 123 via the external resistor 12 as shown in fig. 3, there is an additional current branch from the dc voltage source VIN to the junction 123 via the resistor 32. The dc voltage source VIN charges the capacitor 13 via the external resistor 12 and the resistor 32, thereby generating a voltage signal RTCT at the junction 123. Since the switch 36 is in the on state, it is equivalent to a short circuit, i.e. the contact CTIN coincides with the contact 123. Therefore, the signal of the contact CTIN coincides with the voltage signal RTCT of the contact 123. Compared with fig. 3, the operation of the high-frequency oscillator 130 is the same as that described in fig. 3 except that an additional current branch is added to charge the capacitor 13, and is omitted here for avoiding redundancy. The oscillation frequency f' in the glow mode is obtained by the following equation:

where R' represents a resistance value of the external resistor 12 and the resistor 32 connected in parallel, and C represents a capacitance value of the external capacitor 13. It follows that the oscillation frequency in the ignition mode is higher than the frequency in the normal operation mode. By selecting the resistor 32 with a proper size, the starting frequency of the cold cathode fluorescent lamp 170 is increased by thirty percent compared with the normal operating frequency, so that the normal starting is realized.

As shown in fig. 4 and 5, the oscillation signal RTCT43 of the oscillator 130 is a sawtooth signal with a fixed frequency, and its maximum value/peak value is equal to the first reference voltage VRH42, and its minimum value/valley value is zero. In the preferred embodiment, the switch 25 has a larger size, so that the conducting capability is strong, and the current flowing at the moment when the switch 25 is conducted is large, so that the capacitor 13 is discharged quickly. Therefore, the discharge time of the capacitor 13 is much shorter than the period of the oscillating signal RTCT, so that the discharge time of the capacitor 13 can be ignored, thereby avoiding the frequency of the oscillating signal RTCT varying with the dc voltage source VIN. In practical applications, the first reference voltage VRH and the second reference voltage VRL can be connected to the ground terminal through a voltage buffer or a decoupling capacitor to reduce noise interference of the circuit.

As can be seen from the above description, the amplitude of the oscillation signal RTCT output by the high-frequency oscillator 130 is proportional to the dc voltage source VIN, and when the dc voltage source VIN changes, the amplitude of the oscillation signal RTCT changes with the dc voltage source VIN, while the error signal CMP remains unchanged. The comparator 140 adjusts the duty ratio of the PWM driving signal according to the error signal CMP and the oscillation signal RTCT, so as to realize the feed-forward compensation of the dc voltage source VIN.

In the exemplary switching system 100 of FIG. 1, CCFL current is controlled on and off using a low frequency pulse width modulation (LPWM) dimming signal that is proportional to the duty cycle of the LPWM dimming signal, which may be generated by digitizing an analog signal input by a user, for the purpose of controlling CCFL brightness. The frequency of the LPWM dimming signal is much lower than the frequency of the PWM driving signal outputted from the comparator 140, for example, the frequency range of the PWM driving signal is 35kHz to 80kHz, the frequency range of the LPWM dimming signal is 50 to 200hz, the LPWM dimming signal controls the error signal CMP outputted from the error amplifier 120, when the LPWM dimming signal is at a low level, an enabling circuit (not shown) absorbs the charge on the compensation capacitor 11, so that the error signal CMP is reduced to a low level. At this time, the PWM drive signal is obtained by comparing the lowest value of the oscillation signal RTCT with the error signal CMP. Therefore, when the LPWM dimming signal is at a low level, the PWM driving signal is at a low level and the CCFL current is substantially zero. When the LPWM dimming signal is at a high level, the enable circuit does not act on the compensation capacitor 11, the error amplifier 120 recharges the compensation capacitor 11 to an initial value, and the error signal CMP goes high, indicating the output of the CCFL at maximum brightness. With the feedforward compensated converter controller of the invention, the value of the error signal CMP remains unchanged. Therefore, under the control of the LPWM dimming signal, the error signal CMP keeps the charging time TRISE (shown in fig. 4) and the discharging time TFALL (shown in fig. 4) of the compensation capacitor 11 unchanged, and the luminance of the CCFL is not affected by the change of the voltage source VIN.

Fig. 4 is a timing diagram illustrating various signal waveforms in the converter controller 110 according to the present invention. Curve 41 shows a waveform of the variation of the dc voltage source VIN that powers the CCFL. In one embodiment, dc voltage source VIN varies between 6 and 30V. Curve 42 is a waveform of the first reference voltage VRH, which is proportional to the dc voltage source VIN. In one embodiment, the ratio between the first reference voltage VRH and the dc voltage source VIN is 0.1, i.e., VRH equals 0.6V when VIN equals 6V; when VIN equals 30V, VRH equals 3V. Curve 43 is a waveform diagram of the oscillation signal RTCT output from the high-frequency oscillator 130. The amplitude of the oscillating signal RTCT is proportional to the dc voltage source VIN. Curve 44 is a waveform diagram of the LPWM dimming signal. Curve 45 is a waveform diagram showing the error signal CMP output by the error amplifier 120 under control of the LPWM dimming signal. When the LPWM dimming signal is at a high level, the error signal CMP is at a high level; when the LPWM dimming signal is low, the error signal CMP is low. Curve 46 is a waveform diagram of the PWM drive signal output by comparator 140. When the dc voltage source VIN is increased, the error signal CMP remains unchanged, and the amplitude of the oscillation signal RTCT is increased and proportional to the dc voltage source VIN, so that the duty ratio of the PWM driving signal is decreased to adjust the voltage transmitted from the power conversion circuit 160 to the CCFL, thereby controlling the variation of the CCFL current.

Fig. 5 is a timing chart showing different signal waveforms when the dc voltage source VIN changes from the first voltage value to the second voltage value. Curve 43A represents the waveform of the oscillating signal RTCT outputted by the high frequency oscillator 130 when the dc voltage source VIN has the first voltage value, and the amplitude thereof is proportional to the first voltage value. Curve 43B represents the waveform of the oscillating signal RTCT outputted by the high frequency oscillator 130 when the dc voltage source VIN decreases from the first voltage value to the second voltage value, and the amplitude thereof is proportional to the second voltage value. Curve 45 shows the waveform of the error signal CMP output by the error amplifier 120 when the LPWM dimming signal is at a high level. Curve 46A represents a waveform of the PWM driving signal outputted by the comparator 140 after comparing the error signal CMP with the oscillation signal RTCT represented by curve 43A. Curve 46B represents a waveform of the PWM driving signal outputted by the comparator 140 after comparing the error signal CMP with the oscillation signal RTCT represented by curve 43B. When the oscillation signal RTCT has a higher amplitude as shown by the curve 43A, the PWM drive signal has a smaller duty ratio. When the oscillation signal RTCT has a lower amplitude as shown by the curve 43B, the PWM drive signal has a larger duty ratio. Therefore, the amplitude of the oscillation signal RTCT adjusts the duty ratio of the PWM driving signal, and the feedforward compensation of the dc voltage source VIN is realized.

The converter controller with feed forward compensation of the present invention can realize instantaneous compensation when the power supply voltage changes, and maintain tight control of the lamp current. The feed-forward control also improves the input voltage regulation rate and makes the correlation between the start-up transient process and the input voltage lower. When the power supply voltage changes, the amplitude of a sawtooth wave signal generated by an internal oscillator is changed to realize feedforward control, which is equivalent to changing the duty ratio of a PWM driving signal along with the power supply voltage, so that an error signal CMP is kept unchanged, a compensation capacitor does not need to be charged or discharged, and the response of a controller to the change of the power supply voltage is basically instantaneous.

It will be apparent that other embodiments, which are obvious to those skilled in the art, may be made without departing from the spirit and scope of the invention, as defined in the appended claims.

Claims (17)

1. A converter controller with feed forward compensation for controlling load current, comprising: comprises an error amplifier, a pulse signal generator and a driver which are connected in series, an oscillator connected between a power supply for supplying power to a load and a grounding terminal, said pulse signal generator being coupled between said error amplifier and said driver, the oscillator outputting an oscillating signal having an amplitude related to the magnitude of said power supply, the error amplifier receiving a reference signal and a feedback signal indicative of the load current and outputting an error signal, the error signal is input to a first input terminal of a pulse signal generator, the oscillation signal of the oscillator is input to a second input terminal of the pulse signal generator, the pulse signal generator outputs a pulse signal to the driver according to the error signal and the oscillation signal, the amplitude of the oscillation signal controls the duty ratio of the pulse signal, so that the feed-forward compensation of the power supply is realized.

2. A converter controller with feedforward compensation to control load current according to claim 1, wherein: the amplitude of the oscillating signal is proportional to the magnitude of the power supply.

3. A converter controller with feedforward compensation to control load current according to claim 1, wherein: the converter controller also includes a time component connected between the power supply and ground.

4. A converter controller with feed forward compensation to control load current according to claim 3, wherein: the time component comprises a resistor and a capacitor connected in series, the junction between the resistor and the capacitor being connected to the output of the oscillator.

5. A converter controller with feedforward compensation to control load current according to claim 1, wherein: the oscillating signal is a sawtooth wave having a fixed frequency.

6. A converter controller with feedforward compensation to control load current according to claim 1, wherein: the converter controller further includes a compensation capacitor connected between the output terminal of the error amplifier and ground.

7. A converter controller with feedforward compensation to control load current according to claim 1, wherein: the power supply is a direct current voltage source.

8. A switching system with feed forward compensation for controlling load current, comprising:

a power conversion circuit connected to a power supply that supplies power to the load and converting the power supply into a signal required by the load; and

a controller for controlling the power conversion circuit, including an error amplifier, a pulse signal generator and a driver connected in series, an oscillator connected between the power supply and a ground terminal, the pulse signal generator being connected between the error amplifier and the driver, the error amplifier receiving a reference signal and a feedback signal representing a load current and outputting an error signal based on the reference signal and the feedback signal, the error signal being input to a first input terminal of the pulse signal generator, an oscillation signal output by the oscillator being input to a second input terminal of the pulse signal generator, the pulse signal generator outputting a pulse signal based on the error signal and the oscillation signal, the pulse signal controlling an output of the power conversion circuit via the driver, an amplitude of the oscillation signal varying with a magnitude of the power supply, thereby adjusting the duty cycle of the pulse signal to achieve feed forward compensation of the power supply.

9. The switching system with feed forward compensation for controlling load current of claim 8, wherein: the amplitude of the oscillating signal is proportional to the magnitude of the power supply.

10. The switching system with feed forward compensation for controlling load current of claim 8, wherein: the controller includes a time component connected between the power supply and ground.

11. The switching system with feed forward compensation for controlling load current of claim 10, wherein: the time component comprises a resistor and a capacitor connected in series, the junction between the resistor and the capacitor being connected to the output of the oscillator.

12. The switching system with feed forward compensation for controlling load current of claim 8, wherein: the oscillation signal is a sawtooth wave with a fixed frequency.

13. The switching system with feed forward compensation for controlling load current of claim 8, wherein: the power supply is a direct current voltage source.

14. The switching system with feed forward compensation for controlling load current of claim 13, wherein: the power conversion circuit converts the direct-current voltage source into alternating-current voltage required by the load based on the pulse signal.

15. The switching system with feed forward compensation for controlling load current of claim 8, wherein: the controller includes a compensation capacitor connected between the output terminal of the error amplifier and a ground terminal.

16. A method of implementing feed forward compensation for a power supply supplying a lamp source, comprising:

generating an oscillating signal having an amplitude proportional to the magnitude of said power supply;

generating a feedback signal indicative of the lamp source current;

comparing the feedback signal with a reference signal to generate an error signal; and

comparing the error signal with the oscillation signal to generate a pulse signal with a duty ratio varying with the amplitude of the oscillation signal so as to control the current of the lamp source.

17. A method of implementing feed forward compensation of a power supply supplying a lamp source as set forth in claim 16, wherein: generating a low frequency pulse width modulated dimming signal that controls the error signal to adjust the current of the lamp source.