HK1086125B - Mimo wlan system - Google Patents

Description

MIMO WLAN system

Claiming priority in accordance with 35U.S.C. § 119

This application claims priority from U.S. provisional patent application No. 60/421,309, entitled "MIMOWLAN system," filed on 25/10/2002.

Background

Technical Field

The present invention relates generally to data communications, and more particularly to a multiple-input multiple-output (MIMO) Wireless Local Area Network (WLAN) communication system.

Background

Wireless communication systems are widely deployed to provide various types of communication such as voice, packet data, and so on. These systems may be multiple-access systems capable of supporting communication with multiple users sequentially or simultaneously by sharing the available system resources. Examples of multiple-access systems include Code Division Multiple Access (CDMA) systems, Time Division Multiple Access (TDMA) systems, and Frequency Division Multiple Access (FDMA) systems.

Wireless Local Area Networks (WLANs) are also widely used to allow communication between wireless electronic devices, such as computers, via wireless links. WLANs may employ access points (or base stations) that operate like hubs and provide connectivity for wireless devices. The access point may also connect (or "bridge") the WLAN to a wired LAN, thereby enabling wireless devices to access LAN resources.

In a wireless communication system, a Radio Frequency (RF) modulated signal from a transmitter unit may reach a receiver unit via multiple propagation paths. The characteristics of the propagation path may change over time due to a number of factors, such as fading and multipath. To provide diversity against the effects of bad paths and improve performance, multiple transmit and receive antennas may be used. If the propagation paths between the transmit and receive antennas are linearly independent (i.e., the transmission on one path is not a combination of the transmissions on the other paths), which is at least somewhat true, then as the number of antennas increases, the probability of correctly receiving a data transmission also increases. In general, as the number of transmit and receive antennas increases, diversity also increases and performance also improves.

MIMO systemBy a plurality of (N)_T) Transmitting antenna and multiple antennas (N)_R) And the receiving antenna is used for data transmission. From N_TRoot transmitting antenna and N_RThe MIMO channel formed by the root receiving antenna can be decomposed into N_SA spatial channel, N_S≤min{N_T，N_R}。N_SEach of the spatial channels corresponds to a dimension. MIMO systems can provide improved performance (e.g., increased transmission capacity and/or greater reliability) if the additional dimensionalities created by the multiple transmit and receive antennas are utilized.

The resources of a given communication system are generally limited by various regulatory constraints and other practical considerations. However, the system may be required to support multiple terminals, provide various services, achieve specific performance goals, and so on.

Therefore, there is a need in the art for a MIMO WLAN system that can support multiple users and provide high system performance.

Disclosure of Invention

A multiple access MIMO WLAN system having various capabilities and capable of high performance is described herein. In an embodiment, the system employs MIMO and Orthogonal Frequency Division Multiplexing (OFDM) to maintain high throughput, combat deleterious path effects, and provide other benefits. Each access point in the system can support multiple user terminals. The resource allocation for the downlink and uplink depends on the requirements of the user terminals, channel conditions and other factors.

Channel structures that support efficient downlink and uplink transmissions are also provided herein. The channel structure comprises a plurality of transport channels that can be used for a plurality of functions, such as signalling of system parameters and resource allocation, downlink and uplink data transmission, random access of the system, etc. Various attributes of these transport channels are configurable, which enables the system to easily adapt to changing channel and load conditions.

MIMO WLAN systems support multiple rates and transmission modes in order to maintain high throughput when supported by channel conditions and user terminal capabilities. The rate may be configured based on an estimate of the channel condition and may be selected independently for the downlink and uplink. Different transmission modes may also be used depending on the number of antennas at the user terminal and the channel conditions. Each transmission mode is associated with different spatial processing at the transmitter and receiver and may be selected for use under different operating conditions. Spatial processing facilitates data transmission from multiple transmit antennas and/or data reception with multiple receive antennas for higher throughput and/or diversity.

In one embodiment, a MIMO WLAN system uses a single frequency band for the downlink and uplink, which share the same operating frequency band using Time Division Duplexing (TDD). For TDD systems, the downlink and uplink channel responses are reciprocal. Calibration techniques are provided herein to determine and compensate for differences in frequency response of transmit/receive chains at an access point and a user terminal. Techniques for utilizing reciprocal characteristics of the downlink and uplink, as well as calibration, to simplify spatial processing at the access point and user terminal are also described herein.

A pilot structure with several types of pilots for different functions is also provided. For example, beacon pilots may be used for frequency and system acquisition, MIMO pilots may be used for channel estimation, steered indices (i.e., steered pilots) may be used for improved channel estimation, and carrier pilots may be used for phase tracking.

Various control loops for proper system operation are also provided. Rate control may be carried out independently on the downlink and uplink. Power control may be exercised for particular transmissions (e.g., fixed rate services). Timing control may be used for uplink transmissions to compensate for different propagation delays of the user terminals located in the system.

Random access techniques are also provided to enable user terminals to access the system. These techniques support access to the system by multiple user terminals, fast acknowledgement of system access attempts, and fast allocation of downlink/uplink resources.

According to the present invention, there is provided a method of exchanging data in a wireless Time Division Duplex (TDD) Multiple Input Multiple Output (MIMO) communication system, comprising: receiving a pilot from a user terminal on an uplink; deriving at least one control vector for a downlink of a user terminal based on the received pilots; and spatially processing a first data transmission sent to the user terminal on the downlink with the at least one control vector.

According to the present invention, there is also provided an apparatus in a wireless Time Division Duplex (TDD) multiple-input multiple-output (MIMO) communication system, comprising: means for receiving a pilot on the uplink from the user terminal; means for deriving at least one control vector for a downlink of a user terminal based on the received pilots; and means for spatially processing a first data transmission sent to the user terminal on the downlink with the at least one control vector.

According to the present invention, there is also provided a method of transmitting and receiving a pilot in a wireless multiple-input multiple-output (MIMO) communication system, comprising: transmitting a MIMO pilot from a plurality of antennas and over a first communication link, wherein the MIMO pilot comprises a plurality of pilot transmissions from the plurality of antennas, wherein the pilot transmission from each antenna is identifiable by a communication entity receiving the MIMO pilot; and receiving a steered pilot from the communication entity via at least one eigenmode of a second communication link, wherein the steered pilot is generated based on the MIMO pilot.

According to the present invention, there is also provided an apparatus in a wireless multiple-input multiple-output (MIMO) communication system, comprising: means for transmitting a MIMO pilot from a plurality of antennas and over a first communication link, wherein the MIMO pilot comprises a plurality of pilot transmissions transmitted from the plurality of antennas, wherein the pilot transmission from each antenna is identifiable by a communication entity receiving the MIMO pilot; and means for receiving a steered pilot from the communication entity via at least one eigenmode of the second communication link, the steered pilot generated based on the MIMO pilot.

According to the present invention, there is also provided a method of channel estimation in a wireless multiple-input multiple-output (MIMO) communication system, comprising: receiving a steered pilot from a user terminal via at least one eigenmode of an uplink; and estimating a channel response for at least one eigenmode of an uplink of the user terminal based on the received steered pilot.

According to the present invention, there is also provided an apparatus in a wireless multiple-input multiple-output (MIMO) communication system, comprising: means for receiving a steered pilot from a user terminal via at least one eigenmode of an uplink; and means for estimating a channel response for at least one eigenmode of an uplink of the user terminal based on the received steered pilot.

Various aspects and embodiments of the invention are described in further detail below.

Drawings

The features and nature of the present invention will become more apparent in the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein:

fig. 1 illustrates a MIMO WLAN system;

fig. 2 shows a layer structure of a MIMO WLAN system;

FIGS. 3A, 3B, and 3C illustrate a TDD-TDM frame structure, an FDD-TDM frame structure, and an FDD-CDM frame structure, respectively;

FIG. 4 shows a TDD-TDM frame structure with five transport channels, BCH, FCCH, FCH, RCH and RACH;

FIGS. 5A through 5G illustrate various Protocol Data Unit (PDU) formats for five transport channels;

FIG. 6 illustrates one structure of an FCH/RCH packet;

fig. 7 shows an access point and two user terminals;

FIGS. 8A, 9A and 10A show three transmitter units for diversity mode, spatial multiplexing mode and beam steering mode, respectively;

FIGS. 8B, 9B and 10B illustrate three transmit diversity processors for diversity mode, spatial multiplexing mode and beam steering mode, respectively;

FIG. 8C illustrates an OFDM modulator;

FIG. 8D shows an OFDM symbol;

FIG. 11A shows a framing unit and scrambler within a transmit data processor;

FIG. 11B shows an encoder and repetition/puncturing unit within a transmit data processor;

FIG. 11C shows another transmit data processor that may be used in a spatial multiplexing mode;

fig. 12A and 12B show state diagrams for user terminal operation;

fig. 13 shows a timeline of RACH;

fig. 14A and 14B illustrate procedures for controlling transmission rates of a downlink and an uplink, respectively;

FIG. 15 illustrates the operation of the power control loop; and

fig. 16 shows a process for adjusting uplink timing for a user terminal.

Detailed Description

The word "exemplary" is used herein to mean "serving as an example, instance, or illustration. Any embodiment described herein as "exemplary" is not necessarily to be construed as preferred or advantageous over other embodiments or designs.

1. General system

I. General system

Fig. 1 illustrates a MIMO WLAN system 100 that supports multiple users and is capable of implementing embodiments of various aspects of the present invention. MIMO WLAN system 100 includes a plurality of Access Points (APs) 110 that support communication for a plurality of user terminals. For simplicity, only two access points 110 are shown in fig. 1. An access point is generally a fixed station used for communicating with user terminals. An access point may also be referred to as a base station or some other terminology.

User terminals 120 may be distributed throughout the system. Each user terminal may be a fixed or mobile terminal capable of communicating with the access point. A user terminal may also be called a mobile station, a remote station, an access terminal, a User Equipment (UE), a wireless device, or some other terminology. Each user terminal may communicate with one, possibly multiple, access points on the downlink and/or uplink at any given moment. The downlink (i.e., forward link) refers to transmission from the access point to the user terminal, and the uplink (i.e., reverse link) refers to transmission from the user terminal to the access point.

In fig. 1, access point 110a communicates with user terminal 120a via 120f, and access point 110b communicates with user terminal 120f via 120 k. Depending on the particular design of system 100, an access point may communicate with multiple user terminals simultaneously (e.g., over multiple code channels or sub-channels) or sequentially (e.g., via multiple time slots). At any given moment, the user terminal may receive downlink transmissions from one or more access points. The downlink transmission from each access point may include overhead data to be received by multiple user terminals, user-specific data to be received by a particular user terminal, other types of data, or any combination thereof. Overhead data may include pilots, paging and broadcast messages, system parameters, and so on.

The MIMO WLAN system is based on a centralized controller network architecture. In this manner, system controller 130 is coupled to access point 110 and further to other systems and networks. For example, system controller 130 may be coupled to a Packet Data Network (PDN), a wired Local Area Network (LAN), a Wide Area Network (WAN), the internet, a Public Switched Telephone Network (PSTN), a cellular communications network, and so forth. System controller 130 may be designed for a number of functions such as (1) coordination and control of the access points coupled thereto, (2) routing of data among the access points, (3) access and control of communications with the user terminals served by the access points, and so forth.

MIMO WLAN systems may be able to provide much higher throughput with much greater coverage capability than conventional WLAN systems. MIMO WLAN systems may support synchronous, asynchronous, and isochronous data/voice services. MIMO WLAN systems may be designed to provide the following features:

high service reliability

Guaranteed quality of service (QoS)

High instantaneous data rate

High spectral efficiency

Extended coverage.

MIMO WLAN systems may operate in various frequency bands (e.g., 2.4GHz and 5.x GHz U-NII bands), subject to bandwidth and radiation limitations that are specific to the selected operating band. The system is designed to support indoor and outdoor use, with a typical maximum cell size of 1km or less. The system supports fixed terminal applications, however some modes of operation also support portable and limited mobile operation.

MIMO, MISO and SIMO

In certain embodiments, and as described in this specification, each access point is equipped with four transmit and receive antennas for data transmission and reception, where the same four antennas are used for transmission and reception. The system also supports the case where the transmit and receive antennas of a device (e.g., access point, user terminal) are not shared, even though the configuration typically provides lower performance than when the antennas are shared. MIMO WLAN systems can also be designed such that: so that each access point is equipped with some other number of transmit/receive antennas. Each user terminal may be equipped with a single transmit/receive antenna or multiple transmit/receive antennas for data transmission and reception. The number of antennas employed per user terminal type depends on various factors such as the services (e.g., voice, data, or both) supported by the user terminal, cost considerations, regulatory constraints, security issues, and so forth.

For a given one-to-many antenna access point and multi-antenna user terminal, the MIMO channel consists of N available for data transmission_TRoot transmitting antenna and N_RA root receive antenna. Different MIMO channels are formed between the access point and different multi-antenna user terminals. Each MIMO channel may be decomposed into N_SA spatial channel, N_S≤min{N_T，N_R}。N_SThe data stream may be in N_SIs transmitted on a spatial channel. Spatial processing is required at the receiver and may or may not be performed at the transmitter in order to perform at N_SMultiple data streams are transmitted over multiple spatial channels.

N_SThe spatial channels may or may not be orthogonal to each other. This depends on various factors such as (1) whether spatial processing is performed at the transmitter to obtain the orthogonal spatial channels, and (2) whether spatial processing is performed at both the transmitter and the receiver when orthogonalizing the spatial channels. N if no spatial processing is performed at the transmitter_SN can be used for one spatial channel_SThe root transmit antennas perform and are not likely to be orthogonal to each other.

By decomposing the channel response matrix for the MIMO channel, N, as described below_SThe spatial channels may be orthogonal. If N is present_SThe spatial channels are orthogonal using decomposition, each of which is referred to as an eigenmode of the MIMO channel, the decomposition requiring spatial processing at the transmitter and receiver. In this case, N_SThe data stream may be in N_SThe eigenmodes are transmitted orthogonally. However, eigenmodes generally refer to theoretical structures. For various reasons, N_SThe spatial channels are generally not perfectly orthogonal to each other. For example, if (1) the transmitter knows the MIMO channel, or (2) the transmitter and/or receiver has the MIMO channelThe spatial channels are not orthogonal if not fully estimated. For simplicity, in the following description, the term "eigenmode" is used to denote the case where an attempt to orthogonalize a spatial channel with decomposition is attempted, even if the attempt is not completely successful due to incomplete channel estimation or the like.

For a given number (e.g., four) of antennas at the access point, the number of spatial channels available to each user terminal depends on the number of antennas employed by the user terminal and the characteristics of the wireless MIMO channel coupling the access point antenna and the user terminal antenna. If a user terminal is equipped with one antenna, four antennas at the access point and a single antenna at the user terminal form a multiple-input single-output channel (MISO) for the downlink and a single-input multiple-output channel (SIMO) for the uplink.

MIMO WLAN systems may be designed to support multiple transmission modes. Table 1 lists the transmission modes supported by the exemplary design of the mimo wlan system.

TABLE 1

Transmission mode	Description of the invention
		SIMO	For receive diversity, data is transmitted from a single antenna, but may be received by multiple antennas.
Diversity	Data is sent redundantly from multiple transmit antennas and/or multiple subbands to provide diversity.
		Beam steering	Data is transmitted at full power on a single (optimal) spatial channel using the phase control information of the dominant eigenmodes of the MIMO channel.
Spatial multiplexing	Data is transmitted over multiple spatial channels to achieve higher spectral efficiency.

For the sake of brevity, the term "diversity" refers to transmit diversity in the following description, unless otherwise specified.

The transmission modes available for the downlink and uplink for each user terminal depend on the number of antennas employed at the user terminal. Table 2 lists the transmission modes available for different terminal types for the downlink and uplink, assuming multiple (e.g., four) antennas at the access point.

TABLE 2

MISO (on downlink)/SIMO (on uplink)	X	X	X	X
					Diversity	X	X		X
Beam steering	X	X		X
					Spatial multiplexing		X		X

For the downlink, all transmission modes except the spatial multiplexing mode may be used for single-antenna user terminals and all transmission modes may be used for multi-antenna user terminals. For the uplink, all transmission modes may be used by a multi-antenna user terminal, while a single-antenna user terminal transmits data from one available antenna using a MIMO mode. Receive diversity (i.e., receiving data transmissions with multiple receive antennas) may be used for SIMO, diversity, and beam-steering modes.

MIMO WLAN systems may also be designed to support various other transmission modes, which are within the scope of the present invention. For example, a beamforming mode may be used to transmit data on a single eigenmode, using the amplitude and phase information of that eigenmode (rather than using only phase information, the latter being used entirely by the beam steering mode). As another example, an "uncontrolled" spatial multiplexing mode may be defined in which the transmitter transmits only multiple data streams from multiple transmit antennas (without any spatial processing), and the receiver performs the necessary spatial processing to isolate and recover the data streams transmitted from the multiple transmit antennas. As yet another example, a "multi-user" spatial multiplexing mode may be defined in which an access point transmits multiple data streams (using spatial processing) in parallel on the uplink from multiple transmit antennas to multiple user terminals. As yet another example, a spatial multiplexing mode may be defined in which the transmitter performs spatial processing to attempt to orthogonalize the multiple data streams transmitted over the multiple transmit antennas (but may not be fully successful due to incomplete channel estimation), and the receiver performs the necessary spatial processing to isolate and recover the data streams transmitted from the multiple transmit antennas. In this way, spatial processing to transmit multiple data streams via multiple spatial channels may be performed at the following locations: (1) at both the transmitter and the receiver, (2) only at the receiver, or (3) only at the transmitter. Different spatial multiplexing may be used depending on factors such as the capabilities of the access point and the user terminal, available channel state information, system requirements, etc.

In general, the access point and the user terminal may be designed with any number of transmit and receive antennas. For simplicity, specific embodiments and designs are described below in which each access point is equipped with four transmit/receive antennas and each user terminal is equipped with four or fewer transmit/receive antennas.

2.OFDM

In one embodiment, a MIMO WLAN system employs OFDM to effectively divide the overall system bandwidth into multiple (N)_F) Orthogonal subbands. These subbands may also be referred to as tones, frequency bins, or frequency channels. According to OFDM, each subband is associated with a respective subcarrier, which may be modulated with data. For MIMO systems using OFDM, each spatial signal for each subbandThe lanes may be viewed as an independent transmission channel whereby the complex gain associated with each sub-band is constant over the sub-band bandwidth.

In one embodiment, the system bandwidth is divided into 64 orthogonal subbands (i.e., N)_F64) to indices-32 to + 31. Of these 64 subbands, 48 subbands are used for data (e.g., with indices of ± {1, …,6, 8, …,20, 22, …, 26}), 4 subbands are used for pilot and possibly signaling (e.g., with indices of ± {7, 21}), the DC subband (index of 0) is unused, the odds subband is also unused and serves as a guard subband. This OFDM subband structure is further detailed in the document of IEEE standard 802.11a, entitled "Part 11: wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) specificities: High-Speed Physical Layer in the 5GHz Band, ", filed 9.1999, is publicly available and is incorporated by reference. It is within the scope of the invention for a MIMO WLAN system to use a different number of subbands and various other OFDM subband structures. For example, a total of 53 indices from-26 to +26 may be used for data transmission. As another example, a 128 subband structure, a 256 subband structure, and a subband structure having some other number of subbands may be used. For clarity, a MIMO WLAN system having the above-described 64-subband structure is described below.

For OFDM, data to be transmitted on each subband is first modulated (i.e., symbol mapped) using a particular modulation scheme selected for that subband. Zero values are provided for unused subbands. For each symbol period, all N_FThe modulation symbols and zero values of the subbands are transformed to the time domain using an Inverse Fast Fourier Transform (IFFT) to obtain a signal containing N_FA transformed symbol of time domain samples. The duration of each transformed symbol is inversely related to the bandwidth of each subband. In a particular design of a MIMO WLAN system, the system bandwidth is 20MHz, N_FThe bandwidth of each subband is 312.5KHz and the duration of each transformed symbol is 3.2 microseconds, 64.

OFDM can provide certain advantages, such as the ability to combat frequency selective fading, which is characterized by different channel gains at different frequencies of the overall system bandwidth. It is well known that frequency selective fading causes inter-symbol interference (ISI), a phenomenon in which each symbol in a received signal acts as interference to subsequent symbols in the received signal. ISI distortion degrades performance by affecting the ability to correctly decode received symbols. Frequency selective fading can be easily addressed with OFDM by repeating a portion of each transformed symbol (or attaching a cyclic prefix thereto) to form a corresponding OFDM symbol, which is then transmitted.

The length of the cyclic prefix (i.e., the amount to be repeated) for each OFDM symbol depends on the delay spread of the wireless channel. In particular, to effectively combat ISI, the cyclic prefix should be longer than the maximum expected delay spread of the system.

In an embodiment, cyclic prefixes of different lengths may be used for OFDM symbols, depending on the expected delay spread. For the particular MIMO WLAN system described above, a cyclic prefix of 400 nanoseconds (8 samples) or 800 nanoseconds (16 samples) may be selected for the OFDM symbol. The "short" OFDM symbol uses a cyclic prefix of 400 nanoseconds with a duration of 3.6 microseconds. The "long" OFDM symbol uses a cyclic prefix of 800 nanoseconds and has a duration of 4.0 microseconds. A short OFDM symbol may be used if the maximum expected delay spread is equal to or less than 400 nanoseconds, and a long OFDM symbol may be used if the delay spread is greater than 400 nanoseconds. Different cyclic prefixes may be selected for different transport channels, and the cyclic prefixes may also be dynamically selected, as described below. Higher system throughput may be achieved by using shorter cyclic prefixes when possible, since more OFDM symbols of shorter duration may be transmitted in a given fixed time interval.

It is within the scope of the invention that MIMO WLAN systems may be designed not to use OFDM.

3. Layer structure

Fig. 2 illustrates a layer structure 200 that may be used for a MIMO WLAN system. The layer structure 200 includes: (1) application and higher layer protocols corresponding approximately to layer 3 and above (higher layers) of the ISO/OSI reference model, (2) protocols and services corresponding to layer 2 (link layer), and (3) protocols and services corresponding to layer 1 (physical layer).

Higher layers include various applications and protocols such as signaling services 212, data services 214, voice services 216, circuit data applications, and so forth. The signaling is typically provided as messages and the data is typically provided as packets. Services and applications in higher layers start and stop messages and packets according to the semantics and timing of the communication protocol between the access point and the user terminal. The higher layers use the services provided by layer 2.

Layer 2 supports the delivery of messages and packets generated by higher layers. In the embodiment shown in fig. 2, layer 2 includes a Link Access Control (LAC) sublayer 220 and a Medium Access Control (MAC) sublayer 230. The LAC sublayer implements a data link protocol that enables correct transmission and delivery of messages generated by higher layers. The LAC sublayer uses services provided by the MAC sublayer and layer 1. The MAC sublayer is responsible for transmitting messages and packets using services provided by layer 1. The MAC sublayer controls access to layer 1 resources by applications and services in the higher layers. The MAC sublayer may include a radio link protocol (232), which is a retransmission mechanism for providing higher reliability for packet data. Layer 2 provides a Protocol Data Unit (PDU) to layer 1.

Layer 1 includes a physical layer 240 and supports the transmission and reception of wireless signals between the access point and the user terminal. The physical layer performs coding, interleaving, modulation, and spatial processing for each transport channel used to transmit messages and packets generated by higher layers. In this embodiment, the physical layer includes a multiplexing sublayer 242, and the multiplexing sublayer 242 multiplexes the PDUs processed for the respective transport channels into the correct frame format. The layer 1 provides data in units of frames.

Fig. 2 illustrates a particular embodiment of a layer structure that may be used for a MIMO WLAN system. Various other suitable layer structures may also be designed and used for the MIMOWLAN system and are within the scope of the present invention. The functions performed by each layer are described in further detail below.

4. Transmission channel

Multiple services and applications may be supported by a MIMO WLAN system. In addition, other data required for proper system operation may need to be sent by the access point and exchanged between the access point and the user terminal. Multiple transmission channels may be defined for a MIMO WLAN system to transmit various types of data. Table 3 lists an exemplary set of transport channels and also provides a brief description of each transport channel.

TABLE 3

As shown in table 3, the downlink transport channels used by the access point include BCH, FCCH and FCH. The uplink transport channels used by the user terminal include RACH and RCH. Each of these transport channels is described in further detail below.

The transmission channels listed in table 3 represent one particular embodiment of a channel structure that may be used for a MIMO WLAN system. Fewer, additional, and/or different transmission channels may also be defined for use in a MIMO WLAN system. For example, specific functions may be supported by function-specific transport channels (e.g., pilot, paging, power control, and synchronization channels). In this way, other channel structures with different sets of transmission channels may be defined and used for MIMO WLAN systems, which is within the scope of the present invention.

5. Frame structure

A plurality of frame structures may be defined for the transport channel. The particular frame structure to be used for a MIMO WLAN system depends on various factors such as (1) whether the same or different frequency bands are used for the downlink and uplink, and (2) the multiplexing scheme used to multiplex the transmission channels together.

If only one frequency band is available, the downlink and uplink may be transmitted on different phases of a frame using Time Division Duplexing (TDD), as described below. If two frequency bands are available, the downlink and uplink are transmitted on different frequency bands using Frequency Division Duplexing (FDD).

For TDD and FDD, the transport channels may be multiplexed together in Time Division Multiplexing (TDM), Code Division Multiplexing (CDM), Frequency Division Multiplexing (FDM), and so on. For TDM, each transport channel is assigned to a different portion of a frame. For CDM, the transport channels are sent in parallel, but each transport channel is channelized by a different channelization code, similar to the channelization performed in a Code Division Multiple Access (CDMA) system. For FDM, each transmission channel is assigned to a different portion of the link band.

Table 4 lists various frame structures that may be used to transmit transport channels. Each of these frame structures is described in further detail below. For clarity, the frame structure is described for the set of transport channels listed in table 3.

TABLE 4

	Shared frequency bands for downlink and uplink	Separate frequency bands for downlink and uplink
			Time diversity	TDD-TDM frame structure	FDD-TDM frame structure
Code diversity	TDD-CDM frame structure	FDD-CDM frame structure

Fig. 3A illustrates an embodiment of a TDD-TDM frame structure 300a that may be used when a single frequency band is used for both the downlink and uplink. Data transmission occurs in units of TDD frames. Each TDD frame may be defined to span a particular time duration. The frame duration may be selected based on various factors such as (1) the bandwidth of the operating band, (2) the expected size of the PDUs of the transmission channel, and so on. In general, shorter frame durations can provide reduced delay. However, a longer frame duration may be more efficient because the header and overhead may represent a smaller portion of a frame. In a particular embodiment, each TDD frame is 2 milliseconds in duration.

Each TDD frame is divided into a downlink phase and an uplink phase. For three downlink transport channels-BCH, FCCH and FCH, the downlink phase is further divided into three segments. For two uplink transport channels, RCH and RACH, the uplink phase is further divided into two segments.

The segments of each transport channel may be defined as a fixed duration or a variable duration that varies from frame to frame. In one embodiment, the BCH segment is defined to have a fixed duration and the FCCH, FCH, RCH, and RACH segments are defined to have variable durations.

The segments of each transport channel may be used to transmit one or more Protocol Data Units (PDUs) for that transport channel. In the particular embodiment shown in FIG. 3A, in the downlink phase, BCH PDUs are transmitted in a first segment 310, FCCH PDUs are transmitted in a second segment 320, and one or more FCH PDUs are transmitted in a third segment 330. On the uplink phase, one or more RCH PDUs are transmitted in a fourth segment 340 and one or more RACH PDUs are transmitted in a fifth segment 350 of the TDD frame.

Frame structure 300a represents a particular layout of various transport channels within a TDD frame. This arrangement may provide certain benefits, such as reduced delay, for data transmission on the downlink and uplink. The BCH is first sent within the TDD frame because it conveys system parameters for PDUs available for other transport channels within the same TDD frame. The FCCH is then sent because it conveys channel allocation information indicating which user terminals are designated to receive downlink data on the FCH and which user terminals are designated to receive uplink data on the RCH within the current TDD frame. Other TDD-TDM frame structures may also be defined and used for MIMO WLAN systems and are within the scope of the present invention.

Fig. 3B illustrates an embodiment of an FDD-TDM frame structure 300B that may be used when two separate frequency bands are used to transmit the downlink and uplink. Downlink data is transmitted in downlink frames 302a and uplink data is transmitted in uplink frames 302 b. Each downlink and uplink frame may be defined across a particular time duration (e.g., 2 milliseconds). For simplicity, downlink and uplink frames may be defined to have the same duration and further defined to be aligned on frame boundaries. However, different frame durations and/or non-aligned (i.e., offset) frame boundaries may also be used for the downlink and uplink.

As shown in fig. 3B, for three downlink transport channels, the downlink frame is divided into three segments. For two uplink transport channels, the uplink frame is divided into two segments. The segments of each transport channel may be defined to have a fixed or variable duration and may be used to transmit one or more PDUs for that transport channel.

In the particular embodiment shown in FIG. 3B, the downlink frame transmits one BCH PDU, one FCCH PDU, and one or more FCH PDUs in segments 310, 320, and 330, respectively. The uplink frame transmits one or more RCH PDUs and one or more RACH PDUs in segments 340 and 350, respectively. This particular arrangement may provide the benefits described above (e.g., reduced latency for data transfer). As described below, transport channels may have different PDU formats. Other FDD-TDM frame structures may also be defined and used for MIMO WLAN systems and are within the scope of the present invention.

Fig. 3C illustrates an embodiment of an FDD-CDM/FDM frame structure 300C that may also be used when the downlink and uplink are sent using separate frequency bands. Downlink data may be sent in downlink frames 304a and uplink data may be sent in uplink frames 304 b. Downlink and uplink frames may be defined to be of the same duration (e.g., 2 milliseconds) and aligned at frame boundaries.

As shown in fig. 3C, three downlink transport channels are transmitted in parallel within a downlink frame and two uplink transport channels are transmitted in parallel within an uplink frame. For CDM, the transport channels for each link are "channelized" with a different channelization code, which may be a Walsh code, an Orthogonal Variable Spreading Factor (OVSF) code, a quasi-orthogonal function (QOF), and so on. For FDM, the transmission channel of each link is allocated to a different portion of the link frequency band. Different amounts of transmit power may also be used for different transport channels in each link.

Other frame structures may also be used for the downlink and uplink transport channels and are within the scope of the present invention. Furthermore, different types of frame structures may be used for the downlink and uplink. For example, a TDM-based frame structure may be used for the downlink and a CDM-based frame structure may be used for the uplink.

In the following description, it is assumed that the MIMO WLAN system uses one frequency band for downlink and uplink transmission. For clarity, the TDD-TDM frame structure shown in fig. 3A is used for a MIMO WLAN system. For clarity, a specific implementation of the TDD-TDM frame structure is described in this specification. For this implementation, the duration of each TDD frame is fixed to 2 milliseconds, and the number of OFDM symbols per TDD frame is a function of the length of the paging prefix used for the OFDM symbols. The fixed duration of the BCH is 80 microseconds and an 800 nanosecond paging prefix is used for the transmitted OFDM symbols. The remainder of the TDD frame contains 480 symbols if an 800 nanosecond paging prefix is used, and 533 OFDM symbols plus 1.2 microseconds of excess time if a 400 nanosecond cyclic prefix is used. The excess time may be added to the guard interval at the end of the RACH segment. Other frame structures and other implementations may also be used and are within the scope of the invention.

II. Transmission channel

Transport channels are used to transmit various types of data and can be classified into two groups: common transport channels and dedicated transport channels. Since common and dedicated transport channels are used for different purposes, different processing may be used for the two sets of transport channels, as described in further detail below.

A common transport channel. The common transport channels include BCH, FCCH and RACH. These transport channels are used to transmit data to or receive data from a plurality of user terminals. For improved reliability, the BCH and FCCH are sent out by the access point in diversity mode. On the uplink, the RACH is transmitted by the user terminal in a beam-steering mode (if supported by the user terminal). The BCH operates at a known fixed rate so that the user terminal can receive and process the BCH without any additional information. The FCCH and RACH support multiple rates to allow for higher efficiency. As used herein, each "rate" or "rate set" is associated with a particular code rate (or coding scheme) and a particular modulation scheme.

A dedicated transport channel. The dedicated transport channels include the FCH and RCH. These transport channels are typically used to transmit user-specific data to a particular user terminal. The FCH and RCH may be dynamically allocated to user terminals as needed and as available. The FCH may also be used in broadcast mode to send overhead, paging and broadcast messages to user terminals. Overhead, paging and broadcast messages are typically sent before any user-specific data on the FCH.

Fig. 4 illustrates an exemplary transmission on BCH, FCCH, FCH, RCH, and RACH based on a TDD-TDM frame structure 300 a. In this embodiment, one BCH PDU 410 and one FCCH PDU 420 are transmitted in BCH segment 310 and FCCH segment 320, respectively. The FCH segment 330 may be used to transmit one or more FCH PDUs 430, each FCH PDU 430 may be directed to a particular user terminal or user terminals. Similarly, one or more RCH PDUs 440 may be transmitted by one or more user terminals in the RCH segment 340. The start of each FCH/RCH PDU is represented by the FCH/RCH offset from the end of the previous segment. RACH PDU 450 may be transmitted by multiple user terminals in RACH segment 350 for accessing the system and/or for transmitting short messages, as described below.

For clarity, the transport channels are described for the particular TDD-TDM frame structures shown in fig. 3A and 4.

1. Broadcast Channel (BCH) -Downlink

The access point transmits the beacon pilot, the MIMO pilot, and the system parameters to the user terminal using the BCH. The user terminal uses the beacon pilot to acquire system timing and frequency. The user terminals use the MIMO pilots to estimate the MIMO channel formed by the access point antennas and their own antennas. The beacon pilots and MIMO pilots are described in further detail below. The system parameters specify various attributes of the downlink and uplink transmissions. For example, since the durations of the FCCH, FCH, RACH, and RCH segments are variable, system parameters specifying the length of each of these segments for the current TDD frame are transmitted in the BCH.

FIG. 5A illustrates an embodiment of a BCH PDU 410. In this embodiment, BCH PDU 410 includes a preamble portion 510 and a message portion 516. Preamble portion 512 also includes a beacon pilot portion 512 and a MIMO pilot portion 514. Portion 512 transmits the beacon pilot with a fixed duration TCP-8 microseconds. Portion 514 transmits the MIMO pilot with a fixed duration TMP of 32 microseconds. Part 516 transmits a BCH message with a fixed duration TBM of 40 microseconds. The duration of BCH PDU is fixed at TCP + TMP + TBM 80 microseconds.

The preamble may be used to transmit one or more types of pilots and/or other information. The beacon pilot includes a particular set of modulation symbols transmitted from all transmit antennas. The MIMO pilot comprises a specific set of modulation symbols transmitted from all transmit antennas with different orthogonal codes to enable the receiver to recover the pilot transmitted from each antenna. Different sets of modulation symbols may be used for the beacon and the MIMO pilot. The generation of beacons and MIMO pilots is described in further detail below.

The BCH message conveys system configuration information. Table 5 lists various fields of an exemplary BCH message format.

TABLE 5-BCH message

Field/parameter name	Length (bit)	Description of the invention
			Frame counter	4	TDD frame counter
Network ID	10	Network Identifier (ID)
			AP ID	6	Access point ID
AP Tx Lvl	4	Access point transmit level
			AP Rx Lvl	3	Reception level of access point
FCCH Length	6	Duration of FCCH (Unit is OFDM code element)
			FCCH Rate	2	Physical layer rate of FCCH
Length of FCH	9	Duration of FCH (Unit is OFDM code element)
			Length of RCH	9	Duration of RCH (unit is OFDM code element)
RACH length	5	Duration of RACH (unit is RACH time slot)
			RACH slot size	2	Duration of each RACH time slot (unit is OFDM code element)
RACH guard interval	2	Guard interval at the end of RACH
			Duration of cyclic prefix	1	Duration of cyclic prefix
Paging bit	1	"0" — broadcast message sent on FCH "1" — no paging message to send
			Broadcast bit	1	"0" for broadcast message sent on FCH "1" for no broadcast message to send
RACH acknowledgement bit	1	"0" for RACH acknowledgement sent on FCH "1" for no RACH acknowledgement to send
			CRC	16	CRC value of BCH message
Tail bits	6	Tail bits for convolutional encoder
			Retention	32	Is reserved for future use

The frame counter may be used to synchronize various processes (e.g., pilot, scrambling, cover, etc.) at the access point and the user terminal. The frame counter may be implemented with a wrap around 4-bit counter. The counter is incremented by one at the beginning of each TDD frame, and the counter value is included in the frame counter field. The network ID field indicates an Identifier (ID) of a network to which the access point belongs. The AP ID field indicates the ID of the access point within the network ID. The AP Tx Lvl and AP Rx Lvl fields represent the maximum transmit power level and the expected receive power level at the access point, respectively. The user terminal may use the desired received power level to determine the initial uplink transmit power.

The FCCH length, FCH length, and RCH length fields indicate the length of the FCCH, FCH, and RCH fields, respectively, of the current TDD frame. The length of these fields is given in units of OFDM symbols. The OFDM symbol duration of BCH is fixed at 4.0 microseconds. The OFDM symbol duration of all other transport channels (i.e., FCCH, FCH, RACH, and RCH) is variable, and depends on the cyclic prefix selected, which is specified by the cyclic prefix duration field. The FCCH rate field indicates the rate used by the FCCH for the current TDD frame.

The RACH length field indicates the length of the RACH field, which is given in units of RACH slots. The duration of each RACH slot is given by the RACH slot size field in units of OFDM symbols. The RACH guard interval field indicates the amount of time between the last RACH slot and the start of the BCH segment of the next TDD frame. These various fields of the RACH are detailed further below.

The paging bit and the broadcast bit indicate whether a paging message and a broadcast message are transmitted on the FCH in the current TDD frame, respectively. These two bits may be set independently for the TDD frame. The RACH acknowledgment bit indicates whether an acknowledgment for the PDU is sent on the RACH prior to sending the TDD frame on the FCCH in the current TDD frame.

The CRC field includes a CRC value for the entire BCH message. This CRC value can be used by the user terminal to determine whether the received BCH message was decoded correctly (i.e., good) or in error (i.e., erased). The tail bits field includes a set of zero values that are used to reset the convolutional encoder to a known state at the end of the BCH message.

As shown in table 5, the BCH message includes a total of 120 bits. These 120 bits may be transmitted with 10 OFDM symbols using the processing detailed below.

Table 5 shows one particular embodiment of the format of the BCH message. Other BCH message formats with fewer, additional, and/or different fields may also be defined and used, and are within the scope of the present invention.

2. Forward Control Channel (FCCH) -Downlink

In an embodiment, the access point can allocate resources for the FCH and RCH on a frame-by-frame basis. The access point uses the FCCH to communicate resource allocations (i.e., channel allocations) for the FCH and RCH.

Fig. 5B illustrates one embodiment of an FCCH PDU 420. In this embodiment, the FCCH PDU includes only portion 520 of the FCCH message. The FCCH message has a variable duration that may vary from frame to frame depending on the amount of scheduling information transmitted on the FCCH of the frame. The FCCH message duration is an even number of OFDM symbols and is given by the FCCH length field on the BCH message. The duration of the messages transmitted using diversity mode (e.g., BCH and FCCH messages) is given in an even number of OFDM symbols because the diversity mode transmits OFDM symbols in pairs, as described below.

In one embodiment, the FCCH can be transmitted at four possible rates. The specific rate used for the FCCH PDUs in each TDD frame is represented by the FCCH physical Mode (Phy Mode) field in the BCH message. Each FCCH rate corresponds to a particular code rate and a particular modulation scheme and is further associated with a particular transmission mode, as shown in table 26.

The FCCH message may include zero, one, or multiple Information Elements (IEs). Each information element may be associated with a particular user terminal and used to provide information indicative of FCH/RCH resource allocation for that user terminal. Table 6 lists various fields of an exemplary FCCH message format.

TABLE 6 FCCH message

Field/parameter name	Length (bit)	Description of the invention
			N_IE	6	Number of IEs included in FCCH message

N _ IE information elements, each comprising:

IE type	4	IE type
			MAC ID	10	ID assigned to user terminal
Control field	48 or 72	Control field for channel allocation
			Stuffing bits	Variable	Implementing padding bits for an even number of OFDM symbols in an FCCH message
CRC	16	CRC value for FCCH messages
			Tail bits	6	Tail bits for convolutional encoder

The N _ IE field indicates the number of information elements included in the FCCH message transmitted within the current TDD frame. For each Information Element (IE) included in the FCCH message, the IE type field indicates the specific type of the IE. Multiple IE types are defined for allocating resources for different types of transmissions, as described below.

The MAC IE field indicates a specific user terminal to which the information element is directed. Each user terminal registers with the access point at the start of a communication session and is assigned a unique MAC ID by the access point. The MAC ID is used to identify the user terminal during the session.

The control field is used to convey channel allocation information for the user terminal and is described in detail below. The padding bits field includes a sufficient number of padding bits such that the total length of the FCCH message is an even number of OFDM symbols. The FCCH CRC field includes a CRC value that the user terminal can use to determine whether the received FCCH message was decoded correctly or in error. The tail bit field includes a zero value for resetting the convolutional encoder to a known state at the end of the FCCH message. Some of these fields are detailed further below.

As shown in table 1, the MIMO WLAN system supports multiple transmission modes for the FCH and RCH. Furthermore, the user terminal may be active or idle during the connection. Multiple classes of IEs are defined to allocate FCH/RCH resources for different types of transmissions. Table 7 lists an exemplary set of IE types.

TABLE 7 FCCH IE types

IE type	IE size (bit)	IE type	Description of the invention
				0	48	Diversity mode	Diversity only mode
1	72	Spatial multiplexing mode	Spatial multiplexing mode-variable rate service
				2	48	Idle mode	Idle state-variable rate service
3	48	RACH acknowledgement	RACH acknowledgement-diversity mode
				4		Beam steering mode	Beam steering mode
5-15	-	Retention	Reserved for future use

For IE types 0, 1, and 4, resources are allocated to particular user terminals for the FCH and RCH (i.e., in channel pairs). For IE type 2, the user terminal is allocated the least resources on the FCH and RCH to maintain the latest estimate of the link. Exemplary formats of the respective IE types are described below. In general, the rates and durations of the FCH and RCH may be independently allocated to user terminals.

Ie type 0, 4-diversity/beam-steering mode

IE types 0 and 4 are used to allocate FCH/RCH resources for diversity mode and beam-steering mode, respectively. For fixed low rate services (e.g., voice), the rate remains fixed for the duration of the call. For variable rate services, the rate may be independently selected for FCH and RCH. The FCCH IE indicates the location of FCH and RCH PDUs allocated to the user terminal. Table 8 lists various fields of exemplary IE type 0 and 4 information elements.

TABLE 8-FCCH IE types 0 and 4

Field/parameter name	Length (bit)	Description of the invention
			IE type	4	IE type

MAC ID	10	Temporary ID assigned to user terminal
			FCH shift	9	FCH offset from TDD frame start (expressed in OFDM symbols)
FCH preamble type	2	FCH preamble size (expressed in OFDM symbols)
			FCH rate	4	Rate of FCH
RCH offset	9	RCH offset from TDD frame start (in OFDM symbol)
			RCH preamble type	2	RCH preamble size (expressed in OFDM code element)
Rate of RCH	4	Rate of RCH
			RCH timing adjustment	2	Timing adjustment parameters for RCH
RCH power control	2	Power control bits for RCH

The FCH and RCH offset fields represent the time offset from the start of the current TDD frame to the start of the FCH and RCH PDUs, respectively, allocated by the information element. The FCH and RCH rate fields represent the rates of the FCH and RCH, respectively.

The FCH and RCH preamble type fields indicate the size of the preamble in the FCH and RCH PDUs, respectively. Table 9 lists the values of the FCH and RCH preamble type fields and the associated preamble sizes.

TABLE 9 leader types

Type (B)	Bits	Preamble size
			0	00	0 OFDM code element
1	01	1 OFDM code element
			2	10	4 OFDM code elements
3	11	8 OFDM code elements

The RCH timing adjustment field includes two bits used to adjust the timing of uplink transmissions from the user terminal identified by the MAC ID field. The timing adjustment is used to reduce interference in a TDD based frame structure, such as the frame structure shown in fig. 3A, where downlink and uplink transmissions are time division duplexed. Table 10 lists the values of the RCH timing adjustment field and the associated actions.

TABLE 10 RCH timing adjustment

Bits	Description of the invention
		00	Maintaining current timing
01	Advancing uplink transmit timing by 1 sample

10	Delaying uplink transmit timing by 1 sample
		11	Is not used

The RCH power control field includes two bits used to adjust the transmit power of the uplink transmission from the identified user terminal. The power control field is used to reduce interference on the uplink. Table 11 lists the values of the RCH power control field and the associated actions.

TABLE 11 RCH constant Power control

Bits	Description of the invention
		00	Maintaining current transmit power
01	The uplink transmit power is increased by δ dB, where δ is a system parameter.
		10	The uplink transmit power is reduced by δ dB, where δ is a system parameter.
11	Is not used

The channel allocation for the identified user terminals may be provided in various ways. In an embodiment, the user terminal is allocated FCH/RCH resources only for the current TDD frame. In another embodiment, the FCH/RCH resources are allocated to the terminal for each TDD frame before cancellation. In yet another embodiment, FCH/RCH resources are allocated to user terminals for every n TDD frames, which is referred to as "decimated" scheduling of TDD frames. The different types of allocations may be indicated by an allocation type field in the FCCH information element.

Ie type 1-spatial multiplexing mode

IE type 1 uses spatial multiplexing mode to allocate FCH/RCH resources to user terminals. The rates of these user terminals are variable and can be selected independently for the FCH and RCH. Table 12 lists various fields of an exemplary IE type 1 information element.

TABLE 12 FCCH IE type 1

Field/parameter name	Length (bit)	Description of the invention
			IE type	4	IE type

MAC ID	10	Temporary ID assigned to user terminal
			FCH shift	9	FCH offset from the end of FCCH (in OFDM symbols)
FCH preamble type	2	FCH preamble size (expressed in OFDM symbols)
			FCH spatial channel 1 Rate	4	FCH rate for spatial channel 1
FCH spatial channel 2 Rate	4	FCH rate for spatial channel 2
			FCH spatial channel 3 Rate	4	FCH rate for spatial channel 3
FCH spatial channel 4 Rate	4	FCH rate of spatial channel 4
			RCH offset	9	RCH offset from the end of FCH (in OFDM symbols)
RCH preamble type	2	RCH preamble size (expressed in OFDM code element)
			Rate 1 of RCH spatial channel	4	RCH rate for spatial channel 1
RCH spatial channel 2 Rate	4	RCH rate for spatial channel 2
			Rate 3 of RCH spatial channel	4	RCH rate for spatial channel 3
RCH spatial channel 4 Rate	4	RCH rate for spatial channel 4
			RCH timing adjustment	2	Timing adjustment parameters for RCH
Retention	2	Reserved for future use

For IE type 1, the rate of each spatial channel may be independently selected on the FCH and RCH. The rate interpretation of the spatial multiplexing mode is generally because it can specify the rate of each spatial channel (up to four spatial channels for the embodiment shown in table 12). If the transmitter performs spatial processing to transmit data on the eigenmodes, a rate is given per eigenmode. If the transmitter simply transmits data from the transmit antennas and the receiver performs spatial processing to isolate and recover the data (for an uncontrolled spatial multiplexing mode), then a rate is given per antenna.

The information element includes the rate of all enabled spatial channels, with a value of zero for the channels that are not enabled. A user terminal with fewer than four transmit antennas sets the unused FCH/RCH spatial channel rate field to zero. Since the access point is equipped with four transmit/receive antennas, user terminals with more than four transmit antennas can use them to transmit up to four independent data streams.

C.ie type 2-idle mode

IE type 2 is used to provide control information for user terminals operating in the idle state (as described below). In one embodiment, the control vectors used by the access point and the user terminals for spatial processing are continuously updated while the user terminals are in an idle state so that data transmission can begin quickly while continuing. Table 13 lists various fields of an exemplary IE type 2 information element.

TABLE 13 FCCH IE type 2

Field/parameter name	Length (bit)	Description of the invention
			IE type	4	IE type
MAC ID	10	Temporary ID assigned to user terminal
			FCH shift	9	FCH offset from the end of FCCH (in OFDM symbols)
FCH preamble type	2	FCH preamble size (expressed in OFDM symbols)
			RCH offset	9	RCH offset from the end of FCH (in OFDM symbols)
RCH preamble type	2	RCH preamble size (expressed in OFDM code element)
			Retention	12	Reserved for future use

Ie type 3-RACH fast acknowledgement

IE type 3 is used to provide a fast acknowledgement for user terminals attempting to access the system over the RACH. In order to gain access to the system or to send a short message to the access point, the user terminal may send a RACHPDU on the uplink. After the user terminal has transmitted the RACH PDU, it monitors the BCH to determine if the RACH acknowledgement bit is set. This bit is set by the access point if any user terminal successfully accesses the system and an acknowledgement is sent on the FCCH for at least one user terminal. If the bit is set, the user terminal processes the FCCH for the acknowledgement sent on the FCCH. If the access point wishes to acknowledge a RACH PDU it correctly decodes from the user terminal without allocating resources, an IE type 3 information element is sent. Table 14 lists various fields of an exemplary IE type 3 information element.

TABLE 14-FCCH ID type 3

Field/parameter name	Length (bit)	Description of the invention
			IE type	4	IE type
MAC ID	10	Temporary ID assigned to user terminal
			Retention	34	Reserved for future use

Single or multiple types of acknowledgements may be defined and transmitted on the FCCH. For example, a quick acknowledgment and an assignment-based acknowledgment may be defined. The fast acknowledgement may be used to simply acknowledge that a RACH PDU has been received by the access point and that no FCH/RCH resources have been allocated to the user terminal. The assignment-based acknowledgement includes an assignment of FCH and/or RCH for the current TDD frame.

The FCCH may be implemented in other ways and may be transmitted in various ways. In an embodiment, the FCCH is sent at a single rate communicated in the BCH message. The rate may be selected based on, for example, the lowest signal-to-noise-and-interference ratio (SNR) of all users to which the FCCH is transmitted in the current TDD frame. Different rates may be used for different TDD frames depending on the channel conditions of the recipient user terminal within each TDD frame.

In another embodiment, the FCCH is implemented with multiple (e.g., four) FCCH subchannels. Each FCCH subchannel is transmitted at a different rate and is associated with a different required SNR to recover the subchannel. The FCCH subchannels are transmitted in order from lowest rate to highest rate. Each FCCH subchannel may or may not be transmitted within a given TDD frame. The first FCCH subchannel (with the lowest rate) is transmitted first and can be received by all user terminals. This FCCH channel will indicate whether each of the remaining FCCH subchannels will be transmitted in the current TDD frame. Each user terminal will process the transmitted FCCH subchannel to obtain its FCCH information element. Each user terminal may terminate processing of the FCCH when any of the following occurs: (1) fails to decode the current FCCH subchannel, (2) receives its FCCH information element in the current FCCH channel, or (3) has all transmitted FCCH subchannels processed. The user terminal can terminate processing of the FCCH whenever it encounters an FCCH decoding failure because the FCCH subchannel is transmitted at an increased rate and the user terminal cannot decode a subsequent FCCH subchannel transmitted at a higher rate.

3. Random Access Channel (RACH) -uplink

The user terminal uses the RACH to gain access to the system and sends a short message to the access point. The operation of the RACH is based on a slotted Aloha random access protocol, which is described below.

Fig. 5C illustrates one embodiment of RACH PDU 450. In this embodiment, the RACH PDU includes a preamble portion 552 and a message portion 554. If the user terminal has multiple antennas, preamble portion 552 can be used to transmit a controlled reference. The steered reference is a pilot composed of a special set of modulation symbols that are spatially processed before being transmitted on the uplink. Spatial processing causes a pilot to be transmitted on one particular eigenmode of the MIMO channel. The processing of the controlled reference is described in further detail below. Preamble portion 552 has a fixed duration of at least 2 OFDM symbols. Message portion 554 conveys a RACH message and has a variable duration. The duration of the RACH PDU is thus variable.

In one embodiment, four different rates are supported for the RACH. The particular rate used for each RACH message is represented by a 2-bit RACH Data Rate Indicator (DRI). In an embodiment, four different message sizes are also supported for the RACH. The size of each RACH message is represented by a message part field included in the RACH message. Each RACH rate supports 1, 2, 3 or all 4 message sizes. Table 15 lists the four RACH rates, their associated coding and modulation parameters, and the message sizes supported by these RACH rates.

Watch 15

The RACH message conveys a short message and an access request from the user terminal. Table 16 lists the various fields of an exemplary RACH message and each field size for each of four different message sizes.

TABLE 16

The message duration field indicates the size of the RACH message. The MAC PDU type field indicates the RACH message type. The MAC ID field contains a MAC ID that uniquely identifies the user terminal that sent the RACH message. During initial system access, a unique MAC ID is not assigned to the user terminal. In this case, a registration MAC ID (e.g., a specific value reserved for registration purposes) may be included in the MAC ID field. The slot ID field indicates the starting RACH slot on which RACH PDUs are transmitted (RACH timing and transmission are described below). The payload field includes information bits of the RACH message. The CRC field contains the CRC value of the RACH message and the tail bit field is used to reset the convolutional encoder of the RACH. The operation of the RACH and the BCH and FCCH for system access are detailed further below.

RACH may also be implemented with "fast" RACH (F-RACH) and "slow" RACH (S-RACH). The F-RACH and S-RACH may be designed to efficiently support user terminals in different operating states. For example, the F-RACH may be used by a user terminal: (1) registered with the system, (2) compensate for their Round Trip Delay (RTD) by correctly advancing their transmit timing, and (3) achieve the required SNR for operation on the F-RACH. The S-RACH may be used by user terminals that do not use the F-RACH in any case.

Different designs may be used for the F-RACH and S-RACH to facilitate fast access to the system whenever possible and to minimize the amount of system resources required to implement random access. For example, the F-RACH may use shorter PDUs, employ a weaker coding scheme, require the F-RACH PDUs to arrive at the access point approximately time-aligned, and use a slotted Aloha random access scheme. The S-RACH may use longer PDUs, employ stronger coding schemes, allow the S-RACH PDUs to arrive at the access point non-aligned in time, and use a non-slotted Aloha random access scheme.

For simplicity, the following description assumes that a single RACH is used for a MIMO WLAN system.

4. Forward Channel (FCH) -Downlink

The access point uses the FCH to transmit user-specific data to a particular user terminal and to transmit paging/broadcast messages to multiple user terminals. The FCH may be allocated on a frame-by-frame basis. Multiple FCH PDU types are provided to accommodate different uses of the FCH. Table 17 lists an exemplary set of FCH PDU types.

TABLE 17 FCH PDU type

Encoding	FCH PDU type	Description of the invention
			0	Can only take a message	FCH broadcast/paging service/user message
1	Message and preamble sequence	FCH user messages
			2	Preamble only sequence	FCH Idle State

FCH PDU type 0 is used to send paging/broadcast messages and user messages/packets on the FCH and includes only messages/packets. (data for a particular user terminal may be sent as a message or a packet, the two terms being used interchangeably herein.) FCH PDU type 1 is used to send user packets and includes a preamble. FCHPDU type 2 includes only the preamble and does not include any messages/packets and is associated with idle state FCH traffic.

Fig. 5D illustrates one embodiment of an FCH PDU 430a of FCH PDU type 0. In this embodiment, FCH PDU 430a includes only one message portion 534a of a paging/broadcast message or user packet. The message/packet may have a variable length, which is given by the FCH message length field in the FCH PDU. The message length is given in integer number of PHY frames (described below). The rate and transmission mode of the paging/broadcast messages are specified and described below. The rate and transmission mode of the user packets are specified in the associated FCCH information element.

Fig. 5E illustrates one embodiment of an FCH PDU 430b of FCH PDU type 1. In this embodiment, FCH PDU 430b includes a preamble portion 532b and a message/packet portion 534 b. The preamble portion 532b is used to transmit the MIMO pilot or the steered reference and has a variable length given by the FCH preamble type field in the associated FCCH information element. Portion 534b is used to transmit FCH packets and also has a variable length (represented in integer numbers of PHY frames) given by the FCH message length field in the FCH PDU. The FCH packet is sent with the rate and transmission mode specified by the associated FCCH information element.

Fig. 5F illustrates one embodiment of an FCH PDU 430c of FCH PDU type 2. In this embodiment, FCH PDU 430c includes only preamble portion 532c and no message portion. The length of the preamble portion is indicated by the FCCH IE. FCH PDU type 2 may be used to enable the user terminal to update its channel estimate while in idle state.

Multiple FCH message types are provided to accommodate different uses of the FCH. Table 18 lists an exemplary set of FCH message types.

TABLE 18 FCH message types

Encoding	FCH message type	Description of the invention
			0	Paging message	Paging message-diversity mode, rate 0.25bps/Hz
1	Broadcast messages	Broadcast message-diversity mode, rate 0.25bps/Hz
			2	User grouping	Dedicated channel operation-user terminal specific PDU, rate specified in FCCH
3-15	Retention	Reserved for future use

One paging message may be used to page multiple user terminals and sent with FCH PDU type 0. If the paging bit in the BCH message is set, one or more FCH PDUs with pilot messages (i.e., "paging PDUs") are first transmitted on the FCH. Multiple paging PDUs may be transmitted within the same frame. The transmission of paging PDUs uses diversity mode and a minimum rate of 0.25bps/Hz in order to increase the probability of correct reception by the user terminal.

A broadcast message may be used to transmit information to multiple user terminals and is transmitted using FCH PDU type 0. If the broadcast bit in the BCH message is set, one or more FCH PDUs (i.e., "broadcast PDUs") with the broadcast message are transmitted on the FCH immediately after any paging PDU transmitted on the FCH. The transmission of the broadcast PDU also uses diversity mode and a minimum rate of 0.25bps/Hz in order to increase the probability of correct reception.

A user packet may be used to transmit user-specific data and may be transmitted with FCH PDU type 1 or 2. After any paging and broadcast PDUs are sent on the FCH, user PDUs of types 1 and 2 are sent on the FCH. Each user PDU may be transmitted in a diversity, beam-steering, or spatial multiplexing mode. The FCCH information element specifies the rate and transmission mode used for each user PDU sent on the FCH.

A message or packet sent on the FCH includes an integer number of PHY frames. In one embodiment, as described below, each PHY frame may include a CRC value that enables the individual PHY frames in the FCH PDU to be checked and retransmitted as necessary. For asynchronous services, RLP may be employed to segment, retransmit, and reassemble PHY frames within a given FCH PDU. In another embodiment, one CRC value is provided for each message or packet rather than for each PHY frame.

Fig. 6 illustrates one embodiment of the structure of an FCH packet 534. The FCH packet includes an integer number of PHY frames 610. Each PHY frame 610 includes a payload field 622, a CRC field 624, and a tail bits field 626. The first PHY frame of the FCH packet also includes a header field 620 that represents the message type and duration. The last PHY frame in the FCH packet also includes a pad bit field 628, which field 628 contains zero value pad bits at the end of the payload to pad the last PHY frame. In an embodiment, each PHY frame includes 6 OFDM symbols. The number of bits included in each PHY frame depends on the rate used for that PHY frame.

Table 19 lists various fields of an exemplary FCH PDU format for FCH PDU types 0 and 1.

TABLE 19-FCH PDU Format

Tail bits

6

Tail bits for convolutional encoder

The FCH message type and FCH message length fields are sent in the header of the first PHY frame of the FCH PDU. The payload, CRC and tail bit fields are included in the respective PHY frame. The payload portion of each FCH PDU carries information bits for paging/broadcast messages or user-specific packets. The padding bits are used to pad the last PHY frame of the FCH PDU as needed.

A PHY frame may also be defined to include some other number of OFDM symbols (e.g., 1, 2, 4, 8, etc.). Since the OFDM symbols are transmitted in pairs for diversity mode, the PHY frame may be defined with an even number of OFDM symbols, and diversity mode may be used for FCH and RCH. The PHY frame size may be selected based on the expected traffic to minimize inefficiencies. In particular, if the frame size is too large, ineffectiveness is created by using one large PHY frame to transmit a small amount of data. Alternatively, if the frame size is too small, the overhead represents a large portion of the frame.

5. Reverse Channel (RCH) -uplink

The user terminal transmits uplink data and pilot to the access point using the RCH. The RCH may be allocated per TDD frame. One or more user terminals may be designated to transmit on the RCH within any given TDD frame. Multiple RCH PDU types are provided to accommodate different modes of operation on the RCH. Table 20 lists an exemplary set of RCH PDU types.

TABLE 20 RCH PDU type

Encoding	RCH PDU type	Description of the invention
			0	Message only	RCH user message, without preamble sequence
1	Message and preamble sequence, not idle	RCH user message, with preamble sequence
			2	Message and preamble, idle	RCH Idle State message with preamble

RCH PDU type 0 is used to send messages/packets on the RCH and does not include a preamble. The RCHPDU type 1 is used to send messages/packets and includes a preamble. RCH PDU type 2 includes a preamble and a short message and is associated with RCH traffic in an idle state.

FIG. 5D illustrates one embodiment of an RCH PDU of RCH PDU type 0. In this embodiment, the RCH PDU includes only the message portion 534a of a variable-length RCH packet, which is given in integer number of PHY frames by the RCH message length field in the RCH PDU. The rate and transmission mode of the RCH packets are specified in the associated FCCH information element.

Fig. 5E illustrates one embodiment of an RCH PDU of RCH PDUY type 1. In this embodiment, the RCH PDU includes a preamble portion 532b and a packet portion 534 b. The preamble portion 532b is used to transmit a reference (e.g., a MIMO pilot or a steered reference) and has a variable length given by the RCH preamble type field in the associated FCCH information element. Part 534b is used to send an RCH packet and has a variable length, which is given by the RCH message length field in the RCH PDU. The RCH packet is sent using the rate and transmission mode specified in the associated FCCH information element.

FIG. 5G illustrates one embodiment of an RCH PDU 350d of RCH PDU type 2. In this embodiment, the RCH PDU includes a preamble portion 532d and a message portion 535 d. The preamble portion 532d is used to transmit a reference and is 1, 4 or 8 OFDM symbols in length. Part 536d is used to transmit a short RCH message and has a fixed length of one OFDM symbol. Short RCH messages are sent with a particular rate and transmission mode (e.g., rate 1/2 or rate 1/4 and BPSK modulation).

The packet sent on RCH (for PDU types 0 and 1) consists of an integer number of PHY frames. Fig. 6 shows the structure of an RCH packet (the same for FCH packets for PDU types 0 and 1. the RCH packet includes an integer number of PHY frames 610. each PHY frame includes a payload field 622, an optional CRC field 624, and a tail bit field 626. the first PHY frame in the RCH packet also includes a header portion 620, and the last PHY frame in the packet also includes a pad bit field 628.

Table 21 lists various fields of an exemplary RCH PDU format for RCH PDU types 0 and 1.

TABLE 21 RCH PDU Format (PDU type 0 and 1)

The RCH message type, RCH message length, and FCH rate indicator field are sent in the header of the first PHY frame of the RCH PDU. The FCH rate indicator field is used to convey FCH rate information (e.g., the maximum rate supported by each spatial channel) to the access point.

Table 22 lists various fields of an exemplary RCH PDU format for RCH PDU type 2.

TABLE 22 RCH PDU type 2 RCH message

Field/parameter name	Length (bit)	Description of the invention
			FCH Rate indicator	16	Representing the maximum rate of each spatial channel on the FCH
RCH request	1	User terminal request for sending additional data
			Retention	1	Reserved for future use
Tail bits	6	Tail bits for convolutional encoder

The user terminal uses the RCH request field to request additional capacity on the uplink. The short RCH message does not include a CRC and is transmitted in a single OFDM symbol.

6. Dedicated channel activity

The RCH and data transmission on the RCH may occur independently. One or more spatial channels (for beam steering and diversity modes) may be active and used for data transmission for respective dedicated transport channels, depending on the transmission mode selected for RCH and RCH usage. Each spatial channel may be associated with a particular rate.

When all four rates of either FCH only or RCH only are set to zero, the user terminal is idle on that link. The idle terminal will still send an idle PDU on the RCH. When all four rates of FCH and RCH are set to zero, both the access point and the user terminal are powered off and do not transmit. User terminals with fewer than four transmit antennas set the unused rate field to zero. User terminals with more than four transmit antennas use no more than four spatial channels to transmit data. Table 23 shows the transmission rate and channel activity when the rate on all four spatial channels of one (or both) of the FCH or RCH is set to zero.

TABLE 23

FCH rate

Rate of RCH

Channel activity

Status of transmission

There may be situations where both the RCH and FCH are idle (i.e., no data is sent) but still send a preamble. This is called the idle state. As shown in table 13, a control field for supporting the user terminal in the idle state is provided in the FCCH IE type 2 information element.

7. Other designs

For simplicity, specific PDU types, PDU structures, message formats, etc., have been described for the exemplary design. It is also contemplated that fewer, additional, and/or different types, structures, and formats may be used and are within the scope of the present invention.

OFDM subband Structure

In the above description, the same OFDM subband structure is used for all transmission channels. Improved efficiency may be achieved by using different OFDM subband structures for different transmission channels. For example, a 64 subband structure may be used for some transmission channels, a 256 subband structure may be used for some other transmission channels, and so on. In addition, multiple OFDM subband structures may be used for a given transmission channel.

For a given system bandwidth W, the duration of an OFDM symbol depends on the total number of subbands. If the total number of subbands is N, then each transformed symbol (without cyclic prefix) is N/W microseconds in duration (if W is in WHz units). A cyclic prefix is added to each transformed symbol to form a corresponding OFDM symbol. The length of the cyclic prefix is determined by the expected delay spread of the system. The cyclic prefix represents overhead, i.e., the overhead required for each OFDM symbol to combat the frequency selective channel. This overhead represents a larger percentage of OFDM symbols if the symbols are short and a smaller percentage of OFDM symbols if the symbols are long.

Since different transmission channels may be associated with different types of traffic data, an appropriate OFDM subband structure may be selected for each transmission channel to match the expected traffic data type. If a large amount of data is expected to be transmitted on a given transmission channel, a larger subband structure may be defined for that transmission channel. In this case, the cyclic prefix would represent a smaller percentage of OFDM symbols and achieve greater efficiency. Conversely, if a small amount of data is expected to be transmitted on a given transmission channel, a smaller subband structure may be defined for that transmission channel. In this case, even if the cyclic prefix represents a larger percentage of the OFDM symbol, higher efficiency can be achieved by reducing the amount of excess capacity using a smaller OFDM symbol size. Thus, an OFDM symbol may be viewed as a "boxcar", and a correctly sized "boxcar" may be selected for each transport channel based on the amount of data expected to be transmitted.

For example, for the above-described embodiments, data on the FCH and RCH are transmitted in PHY frames, each consisting of 6 OFDM symbols. In this case, another OFDM structure may be defined for the FCH and RCH. For example, a 256 subband structure may be defined for FCH and RCH. A256 subband structure of "large" OFDM symbols may be approximately four times as long in duration as a 64 subband structure of "small" OFDM symbols, but may also be four times as large in data transfer capacity. However, only one cyclic prefix is required for one large OFDM symbol, while four cyclic prefixes are required for the equivalent four small OFDM symbols. Thus, the cyclic redundancy overhead can be reduced by 75% by using a larger 256 subband structure.

This concept can be extended to enable the use of different OFDM subband structures for the same transmission channel. For example, the RCH supports different PDU types, each associated with a particular size. In this case, a larger subband structure may be used for the larger size RCH PDU type, and a smaller size RCHPDU type may be usedUsing a smaller sub-band structure. Combinations of different subband structures may also be used for a given PDU. For example, if one long OFDM symbol is equivalent to four short (OFDM) symbols, N may be used_largeA large OFDM symbol and N_smallTransmitting PDU with small OFDM symbols, where N_largeNot less than 0 and not less than 3 and not less than N_small≥0。

Different OFDM subband structures are associated with different length OFDM symbols. Thus, if different OFDM subband structures are used for different transmission channels (and/or for the same transmission channel), the FCH and RCH offsets for the FCH and RCHPDU may need to be specified with the correct time resolution, which is less than one OFDM symbol period. In particular, the time increment for FCH and RCH PDUs may be given by an integer number of cyclic prefix lengths, rather than an OFDM symbol period.

Rate and transmission mode

The above-mentioned transmission channels are used to transmit various types of data for various services and functions. Each transport channel may be designed to support one or more rates and one or more transmission modes.

1. Transmission mode

Multiple transmission modes are supported for the transport channel. As described below, each transmission mode is associated with specific spatial processing at the transmitter and receiver. Table 24 lists the supported transmission modes for each transport channel.

Watch 24

Transmission of

Transmission mode

Channel with a plurality of channels	SIMO	Transmit diversity	Beam steering	Spatial multiplexing
					BCH	-	X		-
FCCH	-	X	-	-
					RACH	X	-	X	-
FCH	-	X	X	X
					RCH	X	X	X	X

For diversity mode, to achieve spatial, frequency, and/or time diversity, each data symbol is redundantly transmitted over multiple transmit antennas, multiple subbands, multiple symbol periods, or a combination thereof. For the beam steering mode, a single spatial channel is used for data transmission (typically the best spatial channel), and each data symbol is transmitted on the single spatial channel with the full transmit power available to the transmit antenna. For spatial multiplexing mode, multiple spatial channels are used for data transmission, with each data symbol being transmitted on a spatial channel, where a spatial channel corresponds to an eigenmode, a transmit antenna, and so on. The beam-steering mode can be seen as a special case of the spatial multiplexing mode, where only one spatial channel is used for data transmission.

Diversity mode may be used for the common transport channels (BCH and FCCH) of the downlink from the access point to the user terminal. Diversity mode may also be used for dedicated transport channels (FCH and RCH). The use of diversity mode on the FCH and RCH may be negotiated at call setup. Diversity mode uses a pair of antennas for each subband to transmit data on one "spatial mode".

The beam steering pattern may be employed on the RACH by a user terminal having multiple transmit antennas. The user terminal can estimate a MIMO channel based on the MIMO pilot transmitted on the BCH. The channel estimate is then used to perform beam steering on the RACH for system access. The beam steering mode may also be used for dedicated transport channels (FCH and RCH). By utilizing the gain of the antenna array at the transmitter, the beam steering mode may be able to achieve a higher signal-to-noise-and-interference ratio (SNR) at the receiver than the diversity mode. In addition, since the controlled reference includes only symbols for a single "controlled" antenna, the preamble portion of the PDU is reduced. Diversity mode may also be used for RACH.

When channel conditions support, spatial multiplexing mode may be used for both FCH and RCH to achieve higher throughput. The spatial multiplexing mode and the beam steering mode are reference driven and require closed loop control for proper operation. In this way, the user terminal is allocated resources on both the FCH and RCH to support the spatial multiplexing mode. Up to four spatial channels (limited by the number of antennas at the access point) may be supported on the FCH and RCH.

2. Coding and modulation

A plurality of different rates are supported for the transport channel. Each rate is associated with a particular code rate and a particular modulation scheme, which in combination result in a particular spectral efficiency (or data rate). Table 25 lists the various rates supported by the system.

TABLE 25

Rate word	Spectral efficiency (bps/Hz)	Coding rate	Modulation scheme	Information bit/OFDM code element	Coded bit/OFDM code element
						0000	0.0	-	Is free of	-	-
0001	0.25	1/4	BPSK	12	48
						0010	0.5	1/2	BPSK	24	48
0011	1.0	1/2	QPSK	48	96
						0100	1.5	3/4	QPSK	72	96
0101	2.0	1/2	16QAM	96	192
						0110	2.5	5/8	16QAM	120	192
0111	3.0	3/4	16QAM	144	192
						1000	3.5	7/12	64QAM	168	288
1001	4.0	2/3	64QAM	192	288
						1010	4.5	3/4	64QAM	216	288
1011	5.0	5/6	64QAM	240	288
						1100	5.5	11/16	256QAM	264	384
1101	6.0	3/4	256QAM	288	384
						1110	6.5	13/16	256QAM	312	384
1111	7.0	7/8	256QAM	336	384

Each common transport channel supports one or more rates and one transport mode (or possibly more, such as the case with RACH). The BCH is transmitted at a fixed rate using diversity mode. Using diversity mode, the FCCH may be transmitted at one of four possible rates, such as indicated by the FCCH physical mode field in the BCH message. In an embodiment, the RACH may be transmitted at one of four possible rates, each RACH message being one of four possible sizes, as indicated by the RACH DRI embedded in the preamble of the RACH PDU. In another embodiment, the RACH is transmitted at a single rate. Table 26 lists the coding, modulation, and transmission parameters and message sizes supported by each common transport channel.

Table 26 parameters of common transport channel

The size of the FCCH message is variable and is given in an even number of OFDM symbols.

FCH and RCH support all rates listed in table 25. Table 27 lists the coding, modulation, and transmission parameters and message sizes supported by the FCH and RCH.

TABLE 27 parameters of FCH and RCH

5.0	5/6	64QAM	1440	288	1728	288	6
								5.5	11/16	256QAM	1584	720	2304	288	6
6.0	3/4	256QAM	1728	576	2304	288	6
								6.5	13/16	256QAM	1872	432	2304	288	6
7.0	7/8	256QAM	2016	288	2304	288	6

And B, note A: at each rate 1/2, is repeated over two sub-bands to obtain an effective code rate 1/4. The parity bits represent redundant bits introduced by the encoding and are used for error correction of the receiver.

The PHY frame size in table 27 indicates the number of coded bits, modulation symbols, and OFDM symbols of each PHY frame. If 48 data subbands are used for data transmission, each OFDM symbol includes 48 modulation symbols. For diversity and beam-steering modes, one symbol stream is transmitted, and the PHY frame size corresponds to a single rate employed for that symbol stream. For a spatial multiplexing mode, multiple symbol streams may be transmitted on multiple spatial channels, and the total PHY frame size may be determined by the sum of the PHY frame sizes of the individual spatial channels. The PHY frame size of each spatial channel is determined by the rate at which the spatial channel is employed.

For example, assume that a MIMO channel can support four spatial subchannels operating at spectral efficiencies of 0.5, 1.5, 4.5, and 5.5 bps/Hz. The four rates selected for the four spatial channels are then shown in table 28.

TABLE 28-example spatial multiplexing Transmission

Thus, the total PHY frame size is 144+432+1296+1584 information bits or 288+576+1728+2304 coded bits. Even though each of the four spatial channels supports a different number of payload bits, the total PHY frame may be transmitted within 6 OFDM symbols (e.g., 24 microseconds, assuming 4 microseconds/OFDM symbol).

V. physical layer processing

Fig. 7 shows a block diagram of an embodiment of an access point 110x and two user terminals 120x and 120y in a MIMO WLAN system.

On the downlink, at access point 110x, a Transmit (TX) data processor 710 receives traffic data (i.e., information bits) from a data source 708 and signaling and other information from a controller 730 and possibly a scheduler 734. The various types of data may be sent on different transport channels. Tx data processor 710 "frames" (if necessary), scrambles the framed/unframed data, encodes the scrambled data, interleaves (i.e., reorders) the encoded data, and maps the interleaved data to modulation symbols. For simplicity, "data symbols" refer to modulation symbols for traffic data, and "pilot symbols" refer to modulation symbols for pilot. Scrambling is data bit randomization. The encoding improves the reliability of the data transmission. Interleaving provides time, frequency, and/or spatial diversity for the coded bits. Scrambling, encoding, and modulation may be performed based on control signals provided by controller 730, as described in further detail below. TX data processor 710 provides a stream of modulation symbols for each spatial channel used for data transmission.

TX spatial processor 720 receives one or more modulation symbol streams from TX data processor 710 and performs spatial processing on the modulation symbols to provide four transmit symbol streams, one for each transmit antenna. Spatial processing is detailed further below.

Each Modulator (MOD)722 receives and processes a respective transmit symbol stream to provide a respective OFDM symbol stream. Each OFDM symbol stream is further processed to provide a corresponding downlink modulated signal. The four downlink modulated signals from modulators 722a through 722d are then transmitted from four antennas 724a through 724d, respectively.

At each user terminal 120, the transmitted downlink modulated signals are received by one or more antennas 752, each of which provides a received signal to a respective demodulator (DEMOD) 754. Each demodulator 754 performs inverse processing to that performed at modulator 722 and provides received symbols. A Receive (RX) spatial processor 720 then performs spatial processing on the received symbols from all demodulators 754 to provide recovered symbols, which are estimates of the modulation symbols transmitted by the access point.

A rx data processor 770 receives and demultiplexes the recovered symbols into their respective transport channels. The recovered symbols for each transmission channel may be symbol demapped, deinterleaved, decoded, and descrambled to provide decoded data for that transmission channel. The decoded data for each transport channel may comprise recovered packet data, messages, signaling, and so on, which may be provided to a data sink 722 for storage and/or a controller 780 for further processing.

The processing of the access point 110 and the terminal 120 for the downlink is further detailed below. The processing for the uplink may be the same as or different from the processing for the downlink.

For the downlink, at each active user terminal 120, the receive spatial processor 760 further estimates the downlink to obtain Channel State Information (CSI). The CSI may include channel response estimates, received SNR, and the like. Rx data processor 770 may also provide the status of individual packets/frames received on the downlink. The controller 780 receives the channel state information and packet/frame status and determines feedback information to be sent back to the access point. The feedback information is processed by a tx data processor 790 and a tx spatial processor 792 (if present), conditioned by one or more modulators 754, and transmitted back to the access point via one or more antennas 752.

At access point 110, the transmitted uplink signals are received by antennas 724, demodulated by demodulators 722, and processed by a rx spatial processor 740 and a rx data processor 742 in a manner complementary to that performed at the user terminals. The recovered feedback information is then provided to the controller 730 and the scheduler 734.

Scheduler 734 uses the feedback information to perform a variety of functions, such as (1) selecting a set of user terminals for data transmission on the downlink and uplink, (2) selecting a transmission rate and transmission mode for each selected user terminal, and (3) allocating available FCH/RCH resources to the selected terminals. Scheduler 734 and/or controller 730 further processes the downlink transmissions using information (e.g., control vectors) obtained from the uplink transmissions, as described in further detail below.

Multiple transmission modes are supported for data transmission on the downlink and uplink. The processing of each of these transmission modes is described in further detail below.

1. Diversity mode-transmit processing

Fig. 8A shows a block diagram of an embodiment of a transmitter unit 800 capable of performing transmit processing for diversity mode. Transmitter 800 may be used for the transmitter portion of an access point and a user terminal.

Within tx data processor 710a, a framing unit 808 frames the data of each packet to be transmitted on the FCH or RCH. Framing need not be performed for other transport channels. Framing may be performed as shown in fig. 6 to generate one or more PHY frames for each user packet. Scrambler 810 then scrambles the framed/unframed data for each transmission channel to randomize the data.

Encoder 812 receives the scrambled data and encodes the data in accordance with the selected coding scheme to provide coded bits. A repetition/puncturing unit 814 then repeats or punctures (i.e., deletes) some of the code bits to achieve the desired coding rate. In one embodiment, encoder 812 is a rate 1/2, constrained length 7 binary convolutional encoder. The code rate 1/4 may be obtained by repeating each coded bit once. A coding rate greater than 1/2 may be obtained by removing some of the coded bits from the encoder 812. The specific design of framing unit 808, scrambler 810, encoder 812, and repetition/puncturing unit 814 is described below.

Interleaver 818 then interleaves (i.e., reorders) the coded bits from unit 814 based on the selected interleaving scheme. In one embodiment, each set of 48 consecutive coded bits to be transmitted on a given spatial channel is spread over 48 data transmission subbands (or simply data subbands) to provide frequency diversity. The interleaving process is described in further detail below.

Symbol mapping unit 820 then maps the interleaved data in accordance with a particular modulation scheme to provide modulation symbols. As shown in table 26, the diversity mode may use BPSK, 4QAM, or 16QAM depending on the selected rate. In diversity mode, the same modulation scheme is used for all data subbands. Symbol mapping may be achieved by: (1) organizing groups of B bits to form B bit values, wherein B ≧ 1, and (2) mapping each B bit value to a point in the signal constellation corresponding to the selected modulation scheme. Each mapped signal point is a complex value and corresponds to a modulation symbol. Symbol mapping unit 820 provides a stream of modulation symbols to transmit diversity processor 720 a.

In one embodiment, the diversity mode uses space-time transmit diversity (STTD) per subband for dual transmit diversity. STTD supports simultaneous transmission of independent symbol streams on two transmit antennas while maintaining orthogonality at the receiver.

The STTD scheme operates as follows. Suppose two modulation symbols, labeled s, are to be transmitted on a given subband₁And s₂. The transmitter generates two vectors:x ₁＝[s₁ s₂]^Tandwherein "^*"denotes complex conjugation"^T"denotes transposition. Each vector includes two elements to be transmitted from two transmit antennas in one symbol period (i.e., a vector)x ₁Transmitted from two antennas in a first symbol period, vectorx ₂Transmitted from both antennas in the next symbol period).

If the receiver is equipped with a single receive antenna, the received symbol can be expressed as:

r₁＝h₁s₁+h₂s₂+n₁， (1)

wherein r is₁And r₂Is two symbols received by the receiver in two consecutive symbol periods;

h₁and h₂Is the path gain from the two transmit antennas to the receive antenna for the subband under consideration, assuming that the path gain is constant over the subband and remains stationary over a 2-symbol period; and

n₁and n₂Are respectively associated with two received symbols r₁And r₂The associated noise.

The receiver may then derive two transmitted symbols s as follows₁And s₂Estimation of (2):

alternatively, the transmitter may generate two vectorsAndand the two vectors are sequentially transmitted from the two transmit antennas in two symbol periods. The received symbols may then be expressed as:

the receiver then derives estimates of the two transmitted symbols as follows:

the above description can be extended for two or more transmit antennas, N_RA MIMO-OFDM system that receives antennas and a plurality of subbands. Two transmit antennas are used for any given subband. Suppose that two modulation symbols, labeled s, are to be transmitted on a given subband k₁(k) And s₂(k) In that respect The transmitter generates two vectorsx ₁＝[s₁(k) s₂(k)]^TAndor two sets of equivalent symbolsAndeach symbol set includes two elements sequentially transmitted from a corresponding transmit antenna in two symbol periods on subband k (i.e., symbol set { x)_i(k) Is sent from antenna i in two symbol periods on subband k, the set of symbols { x }_j(k) It is transmitted from antenna j on subband k in the same 2-symbol period).

The vector of received symbols at the receive antennas over two symbol periods can be represented as:

r ₁(k)＝h _i(k)s₁(k)+h _j(k)s₂(k)+n ₁(k)，

whereinr ₁(k) Andr ₂(k) is a vector of symbols received at the receiver over subband k in two consecutive symbol periods, each vector comprising N_RN of root receiving antenna_RA plurality of received symbols;

h _i(k) andh _j(k) is from two transmit antennas i and j to N for subband k_RVectors of path gains for the root receive antennas, each vector comprising N from the associated transmit antenna_RChannel gain for each of the receive antennas, where path gain is assumed to be constant over the subband and remains static over a 2 symbol period; and

n ₁(k) andn ₂(k) are respectively associated with two received vectorsr ₁(k) Andr ₂(k) an associated noise vector.

The receiver may then derive two transmitted symbols s as follows₁(k) And s₂(k) Estimation of (2):

alternatively, the transmitter may generate two symbol sets { x }_i(k)}＝{s₁(k) s₂(k) Andand the two symbol sets are transmitted from two transmit antennas i and j. The vector of received symbols can then be expressed as:

the receiver may then derive estimates of the two transmitted symbols as follows:

the STTD scheme is described by S.M. Alamouti in a paper entitled "A Simple Transmit Diversity Technique for Wireless Communications" published in the IEEE journal on selected areas of Communications, volume 8, 16, 1998, page 1451 and 1458. The STTD scheme is also described in the following commonly assigned U.S. patent applications: application No. 09/737,602, entitled "Method and System for incorporated Bandwidth Efficiency in Multiple Input-Multiple Output Channels", filed on 5.1.2001; and application No. 10/179,439 filed 24.6.2002 entitled "Diversity Transmission Modes for MIMO OFDM Communication Systems".

The STTD scheme transmits one modulation symbol in each subband through two transmit antennas in each symbol period. However, the STTD scheme distributes information within each modulation symbol over two consecutive OFDM symbols. Thus, symbol recovery at the receiver is performed based on two consecutive received OFDM symbols.

The STTD scheme uses a pair of transmit antennas for each data subband. Since the access point includes four transmit antennas, each antenna may be selected for half of the 48 data subbands. Table 29 lists exemplary subband-antenna assignment schemes for the STTD scheme.

Watch 29

Sub-band index	Transmitting antenna	Bit index	Sub-band index	Transmitting antenna	Bit index	Sub-band index	Transmitting antenna	Bit index	Sub-band index	Transmitting antenna	Bit index
												-	-	-	-13	1，2	26	1	3，4	1	15	1，2	33
-26	1，2	0	-12	3，4	32	2	1，2	7	16	2，4	39
												-25	3，4	6	-11	1，3	38	3	2，4	13	17	1，3	45
-24	1，3	12	-10	2，4	44	4	1，3	19	18	2，3	5
												-23	2，4	18	-9	1，4	4	5	2，3	25	19	1，4	11

-22	1，4	24	-8	2，3	10	6	1，4	31	20	3，4	17
												-21	1	P0	-7	2，	P1	7	3	P2	21	4	P3
-20	2，3	30	-6	1，2	16	8	3，4	37	22	1，2	23
												-19	1，2	36	-5	3，4	22	9	1，2	43	23	2，4	29
-18	3，4	42	-4	1，3	28	10	2，4	3	24	1，3	35
												-17	1，3	2	-3	2，4	34	11	1，3	9	25	2，3	41
-16	2，4	8	-2	1，4	40	12	2，3	15	26	1，4	47
												-15	1，4	14	-1	2，3	46	13	1，4	21	-	-	-
-14	2，3	20	0	-	-	14	3，4	27	-	-	-

As shown in Table 29, transmit antennas 1 and 2 are for subbands indexed-26, -19, -13, etc., transmit antennas 2 and 4 are for subbands indexed-25, -18, -12, etc., transmit antennas 1 and 3 are for subbands indexed-24, -17, -11, etc., and so on. There are six different antenna pairs for the four transmit antennas. Each of the two antenna pairs is for 8 data subbands, with the 8 data subbands being approximately evenly spaced across the 48 data subbands. The allocation of antennas to subbands is such that different antennas are used for adjacent subbands, which provides greater frequency and spatial diversity. For example, antennas 1 and 2 are used for subband-26, and antennas 3 and 4 are used for subband-25.

The antenna-subband assignments in table 29 also result in the use of all four transmit antennas for each coded bit of lowest rate 1/4, which maximizes antenna diversity. For rate 1/4, each coded bit is repeated and transmitted on two subbands (also referred to as bi-subband repetition coding). The two subbands used for each code bit are mapped to different antenna pairs so that the code bit is transmitted using all four antennas. For example, bit indices 0 and 1 in table 29 correspond to the same coded bit in diversity mode, where the bit with index 0 is transmitted from antennas 1 and 2 on subband-26 and the bit with index 1 is transmitted from antennas 3 and 4 on subband 1. As another example, bit indices 2 and 3 in Table 29 correspond to the same coded bit, with the bit index 2 being transmitted from antennas 1 and 3 on subband-17 and the bit index 3 being transmitted from antennas 2 and 4 on subband 10.

The system may support other transmit diversity schemes and this is within the scope of the invention. For example, the system may support a space-frequency transmit diversity (SFTD), which can achieve space and frequency diversity from each subband pair. An exemplary SFTD scheme operates as follows. Assume that two modulation symbols, s (k) and s (k +1), are generated and mapped to two adjacent subbands of an OFDM symbol. For SFTD, the transmitter may transmit symbols s (k) and s (k +1) from two antennas on subband k, and may transmit symbol s from the same two antennas on subband k +1^*(k +1) and-s^*(k) In that respect Since the channel response is assumed to remain constant for the transmission of two symbol pairs, the modulation symbol pairs use adjacent subbands. The processing at the receiver to recover the modulation symbols is the same as in the STTD scheme, except that the received symbols for the two subbands are processed instead of the received symbols for the two OFDM symbol periods.

Fig. 8B shows a block diagram of an embodiment of the transmit diversity processor 720a that can implement the STTD scheme in diversity mode.

Within Transmit diversity processor 720a, a demultiplexer 832 processes data from the transmitA modulator 710a receives and demultiplexes the modulated symbol stream s (n) into 48 substreams, labeled s, for the 48 data subbands₁(n) to s₁(n) of (a). Each modulation symbol substream includes one modulation symbol for one symbol period, corresponding to a symbol rate (T)_OFDM)^-1Wherein T is_OFDMIs the duration of one OFDM symbol. Each modulation symbol stream is provided to a respective transmit subband diversity processor 840.

Within each transmit subband diversity processor 840, a demultiplexer 842 demultiplexes the modulation symbols for the subband into two sequences of symbols, each sequence having a symbol rate of (2T)_OFDM)^-1. Space-time encoder 850 receives the two sequences of modulation symbols and uses two symbols s in the two sequences for each 2-symbol period₁And s₂Forming two symbol sets for two transmit antennasAndeach symbol set includes two symbols, each symbol from one of the two sequences. By first providing a symbol s₁Second provides a code element s₂ ^*To generate a symbol set { x_iWherein s is obtained through a switch 856a₁By fetching s with unit 852a₂And delays the conjugated symbol by one symbol period by delay unit 854a to obtain s₂ ^*. As shown in Table 29, two symbol sets { x_iAnd { x }_jAre to be transmitted from two antennas i and j allocated to a subband. Space-time encoder 850 assigns a first set of symbols to a first transmit antenna iIs provided to a buffer/multiplexer 870 for a second set of symbols for a second transmit antenna jTo another buffer/multiplexer 870. The two symbols provided by control encoder 850 for each symbol period are referred to as STTD symbols.

Buffers/multiplexers 870a through 870d are used to buffer and multiplex the STTD symbols from all diversity processors 840. Each buffer/multiplexer 870 receives pilot symbols and STTD symbols from the appropriate transmit subband diversity processor 840, as determined by table 29. For example, buffer/multiplexer 870a receives modulation symbols for subbands-26, -24, -22, -19, etc. (i.e., all subbands mapped to antenna 1), buffer/multiplexer 870b receives modulation symbols for subbands-26, -23, -20, -19, etc. (i.e., all subbands mapped to antenna 2), buffer/multiplexer 870c receives modulation symbols for subbands-25, -24, -20, -18, etc. (i.e., all subbands mapped to antenna 3), and buffer/multiplexer 870d receives modulation symbols for subbands-25, -23, -22, -18, etc. (i.e., all subbands mapped to antenna 4).

Each buffer/multiplexer 870 then multiplexes the four pilot, 24 STTD symbols, and 36 zeros for the four pilot subbands, 24 data subbands, and 36 unused subbands, respectively, for each symbol period to form a sequence of 64 transmit symbols for the 64 total subbands. Although there are a total of 48 data subbands, for diversity mode only 24 subbands are used for each transmit antenna, so the actual total number of unused subbands for each antenna is 36 instead of 12. Each transmit symbol is a complex value (which may be zero for unused subbands) that is sent on a subband in one symbol period. Each buffer/multiplexer 870 provides one transmit symbol stream x for one transmit antenna_i(n) of (a). Each transmit symbol stream comprises a concatenated sequence of 64 transmit symbols, one sequence for each symbol period. Referring back to fig. 8A, tx diversity processor 720a provides four transmit symbol streams x to four OFDM modulators 722a through 722d₁(n) to x₄(n)。

Fig. 8C shows a block diagram of an embodiment of an OFDM modulator 722x that may be used for each of OFDM modulators 722a through 722d in fig. 8A. Within OFDM modulator 722x, an Inverse Fast Fourier Transform (IFFT) unit 852 receives a stream of transmit symbols x_i(n) and using a 64 point fast fourier transformThe inverse leaf transform converts each sequence of 64 transmit symbols into its time-domain representation (referred to as a transformed symbol). Each transformed symbol includes 64 time-domain samples corresponding to a total of 64 subbands.

For each transformed symbol, cyclic prefix generator 854 repeats a portion of the transformed symbol to form a corresponding OFDM symbol. As described above, one of two different cyclic prefix lengths may be used. The cyclic prefix of BCH is fixed and is 800 nsec. The cyclic prefix of all other transport channels is optional (either 400nsec or 800nsec) and is represented by the cyclic prefix duration field of the BCH message. For a system with a bandwidth of 20MHz, a sampling period of 50nsec, and 64 fields, each transformed symbol is 3.2 milliseconds in duration (i.e., 64 x 50nsec), and each OFDM symbol is either 3.6 milliseconds in duration or 4.0 milliseconds in duration, depending on whether the OFDM symbol uses a cyclic prefix of 400nsec or 800 nsec.

Fig. 8D illustrates an OFDM symbol. The OFDM symbol consists of two parts: a cyclic prefix (8 or 16 samples) of duration 400 or 800nsec, and a transformed symbol (64 samples) of duration 3.2 microseconds. The cyclic prefix is a copy of the last 8 or 16 samples of the transformed symbol (i.e., cyclic persistence) and is inserted in front of the transformed symbol. The cyclic prefix quebaoOFDM1 symbol retains its orthogonality in the presence of multipath delay spread, thereby improving performance against deleterious path effects such as multipath and channel dispersion caused by frequency selective fading.

Cyclic prefix generator 854 provides a stream of OFDM symbols to transmitter (TMTR) 856. Transmitter 856 converts the stream of OFDM symbols into one or more analog signals, and further amplifies, filters, and upconverts the analog signals to generate a modulated signal, which is transmitted from the associated antenna.

The baseband waveform of the OFDM symbol can be expressed as:

where n represents the symbol period (i.e., the OFDM symbol index);

k represents a subband index;

N_STis the number of pilot and data subbands;

c_n(k) represents the symbols transmitted on subband k of symbol period n; and

wherein T is_CPIs the cyclic prefix duration;

T_Sis the OFDM symbol duration; and

Δ f is the bandwidth of each subband.

2. Spatial multiplexing mode-transmit processing

Fig. 9A shows a block diagram of a transmitter unit 900 capable of performing transmission processing for a spatial multiplexing mode. Transmitter unit 90 is another embodiment of the transmitter portion of the access point and user terminal. For the spatial multiplexing mode, data may be sent on up to four spatial channels, again assuming four transmit antennas and four receive antennas are available. Different rates are used for each spatial channel depending on its transmission capacity. Each rate field is associated with a particular code rate and modulation scheme as shown in table 25. In the following description, it is assumed that N is selected_EA spatial channel for data transmission, wherein N_E≤N_S≤min{N_T，N_R}。

Within tx data processor 710b, a framing unit 808 frames the data of each FCH/RCH packet to generate one or more PHY frames for the packet. Each PHY frame is included within 6 OFDM symbols and may be within all N_EThe number of data bits transmitted in each spatial channel. Scrambler 810 scrambles the data for each transport channel. Encoder 812 receives the scrambled data and encodes it in accordance with the selected coding scheme to provide coded bits. In one embodiment, a common coding scheme is used for all N_EThe data for each spatial channel is encoded by puncturing the code bits with a different puncturing pattern to achieve different code rates for the different spatial channels. Thus, puncturing unit 814 punctures the coded bits to obtain the desired code rate for each spatial channel. The puncturing of the spatial multiplexing mode is described in further detail below.

Demultiplexer 816 receives the coded bits from puncturing unit 814 and demultiplexes the coded bits to be the selected N_EProviding N for each spatial channel_EAnd an encoded bit stream. Each coded bit stream is provided to a respective interleaver 818, which interleaves the coded bits in the stream over 48 data subbands. The coding and interleaving of the spatial multiplexing mode is described in further detail below. FromThe interleaved data for each interleaver 818 is provided to a corresponding symbol mapping unit 820.

In the spatial multiplexing mode, up to four different rates may be used for the four spatial channels depending on the received SNRs achieved for these spatial channels. Each rate is associated with a particular modulation scheme as shown in table 25. Each symbol mapping unit 820 maps the interleaved data in accordance with a particular modulation scheme selected for the associated spatial channel to provide modulation symbols. Of all four spatial channels selected for use, symbol mapping units 820a through 820d provide four streams of modulation symbols for the four spatial channels to TX spatial processor 720 b.

The tx spatial processor 720b performs spatial processing for the spatial multiplexing mode. For simplicity, the following description assumes that four transmit antennas, four receive antennas, and 48 data subbands are used for data transmission. The data subband index is given by the set K, where K ± {1, …,6, 8 …,20, 22, … 26}, for the OFDM subband structure described above.

The model for a MIMO-OFDM system can be expressed as:

r(k)＝H(k)x(k)+n(k)，k∈K， (5)

whereinr(k) Is a "receive" vector having four terms for symbols received through the four receive antennas of subband k (i.e., a symbol vector having four terms for each of the four receive antennas of subband kr(k)＝[r₁(k) r₂(k) r₃(k) r₄(k)]^T)；

x(k) Is a "transmit" vector having four terms for symbols transmitted over four transmit antennas for subband k (i.e., a symbol vector having four terms for each transmit antennax(k)＝[x₁(k) x₂(k) x₃(k) x₄(k)]^T)；

H(k) Is of sub-band k (N)_R×N_T) A channel response matrix; and

n(k) is addition of sub-band kVector of white gaussian noise (AWGN).

Hypothesis noise vectorn(k) Has a zero mean and a covariance matrix ofΛ _n＝σ² IWhereinIIs an identity matrix, σ²Is the noise variance.

Channel response matrix for subband kH(k) Can be expressed as:

wherein h is_ij(k) The term is the connection term (i.e., complex gain) between transmit antenna i and receive antenna j for subband k (for i e {1, 2, 3, 4} and j e {1, 2, 3, 4 }). For simplicity, a channel response matrix is assumedH(k) Is known (for K e K) or can be determined by both the transmitter and the receiver.

Channel response matrix for each sub-bandH(k) May be "diagonal" to obtain N for that subband_SAn eigenmode. This may be by way of a correlation matrixH(k) The eigenvalue decomposition is performed to achieve,R(k)＝H ^H(k)H(k) whereinH ^H(k) To representH(k) The conjugate transpose of (c). Correlation matrixR(k) The eigenvalue decomposition of (a) may be expressed as:

R(k)＝V(k)D(k)V ^H(k)，k∈K， (7)

whereinV(k) Is one (N)_T×N_T) Unitary matrix of which the array is_R(k) Eigenvectors (i.e. ofV(k)＝[v ₁(k) v ₂(k) v ₃(k) v ₄(k)]Each of whichv _i(k) Eigenvectors that are one eigenmode); and

D(k) is thatR(k) Of the eigenvalues (N)_T×N_T) A diagonal matrix.

The unitary matrix is characterized in thatM ^H M＝I. Eigenvectorsv _i(k) Also called (for i e {1, 2, 3, 4}) is the transmit spatial vector for each spatial channel.

Channel response matrixH(k) Singular value decomposition may also be used to diagonalize, as shown below:

H(k)＝U(k)∑(k)V ^H(k)，k∈K， (8)

whereinV(k) Is listed asH(k) A matrix of right eigenvectors of (a);

∑(k) is composed ofH(k) Are diagonal matrices of singular values ofD(k) Diagonal element of (R(k) Eigenvalues of) of the square root; and

U(k) is listed asH(k) Of the left eigenvector of (a).

Singular value decomposition is described by Gilbert Strang in a book entitled "Linear Algebra and Its Applications," second edition academy, 1980. As shown in equations (7) and (8), matrixV(k) Is thatR(k) Eigenvectors of andH(k) the right eigenvector of (a). Matrix arrayU(k) Is thatH(k)H ^H(k) Eigenvectors of andH(k) the left eigenvector of (a).

Diagonal matrix for each sub-bandD(k) Including non-negative real values on the diagonal and zero values at other locations.R(k) Is marked as { { lambda }₁(k)，λ₂(k)，λ₃(k)，λ₄(k) Lambda or { lambda }_i(k) For i e {1, 2, 3, 4 }.

For each of the 48 data subbands, a channel response matrix may be formedH(k) The eigenvalue decomposition is performed independently to determine four eigenmodes for that subband (assuming each matrix isH(k) All in full alignment). Each diagonal matrixD(k) May be ordered such that λ₁(k)≥λ₂(k)≥λ₃(k)≥λ₄(k) Where for subband k, λ₁(k) Is the maximum eigenvalue, λ₄(k) Is the smallest eigenvalue. When each diagonal matrixD(k) When the eigenvalues of (a) are sorted, the correlation matrixV(k) The eigenvectors (or columns) of (a) are also ordered accordingly.

A "wideband" eigenmode may be defined as a set of eigenmodes of the same order in all subbands after ordering (i.e., wideband eigenmode m includes eigenmode m in all subbands). The "dominant" broadband eigenmodes are ordered with each matrixThe eigenmode associated with the largest singular value of (a).

Then form the vectord ^mIncluding the mth row eigenvalues for all 48 data subbands. The vectord ^mCan be expressed as:

d ^m＝[λ_m(-26)…λ_m(-22)…λ_m(22)…λ_m(26)]，m＝{1，2，3，4} (9)

(Vector)d ¹including the eigenvalues of the best or dominant wideband eigenmode. For a MIMO-OFDM system with four transmit antennas and four receive antennas (i.e., a 4 x 4 system), there are up to four wideband eigenmodes.

By making the eigenvalues lambda if the noise variance at the receiver is constant over the operating band and known to the transmitter_m(k) Divided by the noise variance σ²The received SNR for each subband of each wideband eigenmode may be determined. For simplicity, assume that the noise variance is equal to 1(i.e.,. sigma.)²＝1)。

For spatial multiplexing mode, the total transmit power P available to the transmitter_totalMay be assigned to wideband eigenmodes based on various power allocation schemes. In one scheme, the total transmit power P_totalAre uniformly distributed to all four broadband eigenmodes such that P_m＝P_total/4, wherein P_mIs the transmit power allocated to wideband eigenmode m. In another scheme, the total transmit power P is divided using a water-filling (water-filling) process_totalFour wideband eigenmodes are assigned.

The water-filling process allocates power so that the broadband eigenmode with higher power receives a larger portion of the total transmit power. The amount of transmit power allocated to a given wideband eigenmode depends on its received SNR, which in turn depends on the power gain (or eigenvalue) of all subbands of the wideband eigenmode. The water-filling process may allocate zero transmit power to wideband eigenmodes with sufficiently poor received SNR. The water filling process comprises four broadband eigenmode receptionsβ＝{β₁，β₂，β₃，β₄In which is beta_mIs a normalization factor for wideband eigenmode m and can be expressed as:

normalization factor β, as described below_mThe transmit power allocated to wideband eigenmode m is kept constant after applying channel inversion. As shown in equation (10), the normalization factor β_mCan be based on vectorsd ^mAnd assuming that the noise variance is equal to 1 (i.e., σ)²1) is derived.

Then, the water flooding process is based on the setβThe total transmit power to be allocated to each wideband eigenmode is determined so that spectral efficiency or some other criterion can be optimized. The transmit power allocated to wideband eigenmode m by the water-filling process can be expressed as:

P_m＝α_mP_total，m＝{1，2，3，4} (11)

the power distribution of the four broadband eigenmodes can be controlled byα＝{α₁，α₂，α₃，α₄Is given, whereinAnd isIf setαWhere more than one value is non-zero, thenTo select the spatial multiplexing mode.

The process of performing water injection is well known in the art and will not be described herein. A bibliography describing water flooding is "Information Theory and replaceable Communication" by Robert G.Gallager, John Wiley and Sons Press, 1968, which is incorporated herein by reference.

For spatial multiplexing mode, the rate selection for each spatial channel or wideband eigenmode may be based on: the spatial channel/wideband eigenmode is assigned to a transmit power P_mThe achieved received SNR. For simplicity, the following description assumes data transmission on wideband eigenmodes. The received SNR for each wideband eigenmode can be expressed as:

in one embodiment, the rate for each wideband eigenmode is determined based on a table that includes the rates supported by the system and the SNR range for each rate. The table may be obtained by computer simulation, experimental measurements, and the like. The particular rate to be used for each wideband eigenmode is the rate in the table, with a range of SNRs including the received SNR for the wideband eigenmode. In another embodiment, the rate of each wideband eigenmode is selected based on: (1) the received SNR for the wideband eigenmode, (2) SNR offsets to account for estimation errors, variability in the MIMO channel, and other factors, and (3) tables of supported rates and their required SNRs. For this embodiment, the average received SNR for each wideband eigenmode is first calculated as described above, or calculated as the average (in dB) of the received SNR for all subbands of the wideband eigenmode. In either case, an operating SNR is then calculated, equal to the sum of the received SNR and the SNR offset (both in dB). The operating SNR is then compared to the required SNR for each rate supported by the system. The highest rate in the table is then selected for the wideband eigenmode, which requires an SNR less than or equal to the operating SNR. The rates of the transmit diversity mode and the beam steering mode may also be determined in a similar manner.

Transmission power P allocated for each wideband eigenmode_mMay be distributed among the 48 data subbands for the wideband eigenmode so that the received SNRs for all subbands are approximately equal. This non-uniform distribution of power among subbands is called channel inversion. Transmission power P allocated to each sub-band_m(k) Can be expressed as:

wherein beta is_mGiven in equation (10).

As shown in equation (13), the transmission power P_mBased on which the channel power gain is non-uniformly distributed among the data subbands, the channel power gain being defined by an eigenvalue λ_m(k) Given, for K ∈ K. The power distribution is such that approximately equal received SNRs are achieved at the receiver for all data subbands for each wideband eigenmode. This channel inversion is performed independently for each of the four wideband eigenmodes. Channel inversion in wideband eigenmode is further detailed in the following commonly assigned U.S. patent applications: U.S. patent application No. 10/229,209, entitled "Coded MIMO Systems with Selective Channel Inversion applied Per Eigenmode", filed on 27.8.2002.

Channel inversion may be performed in various ways. For full channel inversion, if a wideband eigenmode is selected, all data subbands are used for data transmission. For selective channel inversion, all or a subset of the available data subbands may be selected for use for each wideband eigenmode. Selective channel inversion discards bad subbands whose received SNR is below a certain threshold, and performs channel inversion only on selected subbands. Selective Channel inversion for each wideband Eigenmode is also described in commonly assigned U.S. patent application No. 10/229,209 filed on 27.8.2002, entitled "Coded MIMO Systems with Selective Channel inversion applied Per Eigenmode". For simplicity, the following description assumes that full channel inversion is performed for each wideband eigenmode selected for use.

The gain used for each subband of each wideband eigenmode may be based on the transmit power P assigned to that subband_m(k) To be determined. Gain g per data subband_m(k) Can be expressed as:

a diagonal gain matrix may be defined for each subbandG(k) In that respect The matrixG(k) The gain of the four eigenmodes of subband k is included along the diagonal and can be expressed as:G(k)＝diag[g₁(k)，g₂(k)，g₃(k)，g₄(k)]

for spatial multiplexing mode, transmit vector for each data subbandx(k) Can be expressed as:

x(k)＝V(k)G(k)s(k)，k∈K， (15)

wherein

s(k)＝[s₁(k) s₂(k) s₃(k) s₄(k)]^T，

x(k)＝[x₁(k) x₂(k) x₃(k) x₄(k)]^T

(Vector)s(k) Comprising four modulation symbols, vectors, to be transmitted on the four eigenmodes of subband kx(k) Including four transmit symbols to be transmitted from the four antennas for subband k. For simplicity, equation (15) does not include corrections used to compensate for the difference between the transmit/receive chains at the access point and the user terminalFactors, described in detail below.

Fig. 9B shows a block diagram of an embodiment of tx spatial processor 720B capable of performing spatial processing for spatial multiplexing mode. For simplicity, the following description assumes that all four broadband eigenmodes are selected. However, fewer than four broadband eigenmodes may be selected.

Within processor 720b, a demultiplexer 932 receives four modulation symbol streams (labeled s) to be transmitted on four wideband eigenmodes₁(n) to s₄(n)), each stream is demultiplexed for 48 data subbands, for 48 substreams, and the four modulation symbol streams for each data subband are provided to respective transmit subband spatial processors 940. Each processor 940 performs the process shown in equation (15) for one subband.

At each transmit subband spatial processor 940, four modulation symbol substreams (labeled s)₁(k) To s₄(k) Are provided to four multipliers 942a through 942d which also receive the gains g for the four eigenmodes of the associated subband₁(k)、g₂(k)、g₃(k) And g₄(k) In that respect Each gain g_m(k) May be based on the transmit power P allocated to that subband/eigenmode_m(k) As shown in equation (14). Each multiplier 942 uses its gain g_m(k) To scale its modulation symbols to provide scaled modulation symbols. Multipliers 942a through 942d provide the four scaled modulation symbol substreams to four beamformers 950a through 950d, respectively.

Each beamformer 950 performs beamforming by transmitting one symbol substream over one eigenmode of one subband. Each beamformer 950 receives one symbol substream s of an associated eigenmode_m(k) And an eigenvectorv _m(k) In that respect In particular, beamformer 950a receives eigenvectors of a first eigenmodev ₁(k) Beamformer 950d receives eigenvectors of the second eigenmodev ₂(k) And so on. Beamforming is performed using eigenvectors of the relevant eigenmodes.

Within each beamformer 950, the scaled modulation symbols are provided to four multipliers 952a through 952d, which also receive the associated eigenmodesv _m(k) Four elements v of the eigenvector of (1)_m，1(k)、v_m，2(k)、v_m，3(k) And v_m，4(k) In that respect Each multiplier 952 then uses its eigenvector value v_m，j(k) To multiply the scaled modulation symbols to provide "beamformed" symbols. Multipliers 952a through 952d provide four beamformed symbol substreams (which are to be transmitted from four antennas) to summers 960a through 960d, respectively.

Each summer 960 receives the four beamformed symbols for the four eigenmodes for each symbol period and adds them to provide preconditioned symbols for the associated transmit antenna. Summers 960a through 960d provide the four preconditioned symbol substreams for the four transmit antennas to buffers/multiplexers 970a through 970d, respectively.

Each buffer/multiplexer 970 receives pilot symbols and preconditioned symbols from transmit subband spatial processors 940a through 940k for 48 data subbands. Next, for each symbol period, each buffer/multiplexer 970 multiplexes 4 pilot symbols, 48 preconditioned symbols, and 12 zeros for 4 pilot subbands, 48 data subbands, and 12 unused subbands, respectively, to form a sequence of 64 transmit symbols for the symbol period. Each buffer/multiplexer 970 provides a stream of transmit symbols x for one transmit antenna_i(n), wherein the transmit symbol stream comprises a concatenated sequence of 64 transmit symbols. The transmit symbols may be scaled with a correction factor to account for differences in transmit/receive chains at the access point and the user terminal, as described below. The following OFDM modulation of each transmit symbol stream is described above.

The parallel symbol streams may also be transmitted from four transmit antennas without spatial processing at the access point using an uncontrolled spatial multiplexing mode. For this mode, the beamformer 950 may be omitted from the channel inversion process and beamforming. Each modulation symbol stream is further OFDM processed and transmitted from a respective transmit antenna.

The uncontrolled spatial multiplexing mode may be used in various scenarios, such as where the transmitter is unable to perform the spatial processing necessary to support beam steering based on eigenmode decomposition. This may be because the transmitter has not performed a calibration procedure and cannot generate a good enough estimate of the channel, or does not perform calibration and eigenmode processing at all. For the uncontrolled spatial multiplexing mode, spatial multiplexing is still used to increase transmission capability, but the receiver performs spatial processing to separate the individual symbol streams.

For the uncontrolled spatial multiplexing mode, the receiver performs spatial processing to recover the transmitted symbol stream. In particular, the user terminal may implement a Channel Correlation Matrix Inversion (CCMI) technique, a Minimum Mean Square Error (MMSE) technique, a successive interference cancellation receiver processing technique, or some other receiver spatial processing technique. These techniques are described in detail in commonly assigned U.S. patent application No. 09/993,087, filed on 11/6 of 2001, entitled "Multiple-Access Multiple-Input Multiple-output (MIMO) Communication System". The uncontrolled spatial multiplexing mode may be used for both downlink and uplink transmissions.

The multi-user spatial multiplexing mode supports data transmission on the downlink to multiple user terminals simultaneously based on the "spatial signature" of the user terminals. The spatial signature of the user terminal is given by the channel response vector (for each sub-band) between the access point antenna and the respective user terminal antenna. The access point may obtain the spatial signature based on the controlled reference sent by the user terminal. The access point may process spatial signatures of user terminals desiring data transmission to: (1) selecting a group of user terminals for simultaneous data transmission on the downlink, and (2) deriving a control vector for each independent data stream to be sent to the selected user terminals.

The control vectors for the multi-user spatial multiplexing mode may be derived in various ways. Two exemplary schemes are described below. For simplicity, the following description is for one subband, assuming that each user terminal is equipped with a single antenna.

In a first scheme, the access point obtains the control vector using channel inversion. The access point can select N_apA single antenna user terminal is used for simultaneous transmission on the downlink. The access point obtains a 1 xN for each selected user terminal_apThe channel response row vector and forms an N_ap×N_apChannel response matrixH _muThe matrix has N_apN of individual user terminals_apA row vector. The access point is then N_apObtaining N by each selected user terminal_apMatrix of individual control vectorsH _steer，The access point may also send a controlled reference to each selected user terminal. Each user terminal processes its controlled reference to estimate channel gain and phase, and demodulates the received symbols for its single antenna with the channel gain and phase estimates to obtain recovered symbols.

In a second scheme, the access point pre-decodes a signal to be sent to N_apN of individual user terminals_apSymbol streams such that they experience little crosstalk at the user terminal. The access point may be N_apForming a channel response matrix for each selected user terminalH _muAnd is toH _muPerforming OR factorization such thatH _mu＝F _tri Q _muWhereinT _triAndQ _muis a unitary matrix. Access point then uses matrixT _triTo predecode N_apA stream of data symbols to obtain N_apIs decodedOf the symbol streamaAnd using unitary matricesQ _muFurther processing the pre-decoded symbol stream to obtain a stream for transmission to N_apN of individual user terminals_apAnd a transmission symbol stream. Likewise, the access point may also send a controlled reference to each user terminal. Each user terminal coherently demodulates its received symbols using a controlled reference to obtain recovered symbols.

For the uplink in the multiuser spatial multiplexing mode, the access point may recover the N symbols by MMSE receiver processing, successive interference cancellation, or some other receiver processing technique_apN transmitted by user terminals simultaneously_apA symbol stream. The access point may estimate the uplink channel response for each user terminal and use the channel response estimates for receiver spatial processing and to schedule uplink transmissions. Each single-antenna user terminal may transmit an orthogonal pilot on the uplink. From N_apThe uplink pilots for the individual user terminals may be orthogonal in time and/or frequency. The time orthogonality is achieved by: by having each terminal cover its uplink pilot with an orthogonal sequence assigned to the user terminal. The frequency orthogonality is achieved by: each user terminal is caused to transmit its uplink pilot on a different set of subbands. Uplink transmissions from the user terminals should be approximately time aligned at the access point (e.g., time aligned within the paging prefix).

3. Beam steering mode-transmit processing

Fig. 10A shows a block diagram of a transmitter unit 1000 capable of performing transmission processing for a beam steering mode. Transmitter unit 1000 is yet another embodiment of the transmitter portion of the access point and user terminal.

Within tx data processor 710c, a framing unit 808 frames the data of each FCH/RCH packet to generate one or more PHY frames for the packet. Scrambler 810 then scrambles the data for each transmission channel. Encoder 812 then encodes the framed data in accordance with the selected coding scheme to provide coded bits. Puncturing unit 814 then punctures the coded bits to achieve a desired code rate for the wideband eigenmode used for data transmission. The coded bits from puncturing unit 818 are interleaved across all data subbands. Symbol mapping unit 820 then maps the interleaved data in accordance with the selected modulation scheme to provide modulation symbols. Next, for the beam steering mode, the transmit spatial processor 720c performs transmit processing on the modulation symbols.

The beam steering mode may be used to transmit data on a spatial channel or wideband eigenmode — typically the spatial channel associated with the largest eigenvalue for all data subbands. If the transmit power allocation for the wideband eigenmode is only in the setaA non-zero term is generated, then the beam steering mode may be selected. While the spatial multiplexing mode performs beamforming for each selected eigenmode of each subband based on its eigenvectors, the beam steering mode performs beam steering based on the "normalized" eigenvectors, which principle is to have the eigenmode of each subband transmit data on that single eigenmode.

For the dominant eigenmodes, each eigenvectorv ₁(k) The four elements (for K e K) may have different sizes. Thus, the four per-antenna transmit vectors may have different sizes, each including the preconditioned symbols for all data subbands for a given transmit antenna. If the transmit power of each transmit antenna is limited (e.g., due to limitations of power amplifiers), then the beamforming technique may not fully use the total power available for each antenna.

The beam steering mode uses only eigenvectors from the dominant eigenmodesv ₁(k) Phase information (for K ∈ K) and normalized for each eigenvector so that all four elements in the eigenvector are of equal size. Normalized eigenvectors for subband kCan be expressed as:

where a is a constant (e.g., a ═ 1); and

θ_i(k) is the phase of subband k for transmit antenna i, expressed as:

vector as shown in equation (17)The phase of each element in the array is derived from the eigenvectorv ₁(k) Is obtained (i.e., theta)_i(k) From v_1，i(k) Is obtained in whichv ₁(k)＝[v_1，1(k) v_1，2(k) v_1，3(k) v_1，4(k)]^T)。

Channel inversion may also be performed for the beam steering mode so that a common rate may be used for all data channels. For the beam steering mode, the transmit power allocated to each data subbandCan be expressed as:

whereinIs a normalization factor that keeps the total transmit power unchanged after applying channel inversion;

is the transmit power allocated to each of the four antennas; and

is the power gain for subband k of the dominant eigenmode of the beam steering mode.

Normalization factorCan be expressed as:

transmission powerCan be given as P₁＝P_totalAnd/4 (i.e., the uniform distribution of total transmit power among the four transmit antennas).

Power gainCan be expressed as:

channel inversion results for 48 data subbandsFor K ∈ K. The gain per data subband may then be given as

For beam steering mode, transmit vector per subbandx(k) Can be expressed as:

also for simplicity, equation (21) does not include correction factors to compensate for the difference between the transmit/receive chains at the access point and the user terminal.

As shown in equation (16), the normalized control vector for each sub-bandMay be of equal size but may have different phases. Thus, beam steering generates one transmit vector for each subbandx(k)，x(k) Have the same size but may have different phases.

Fig. 10B shows a block diagram of an embodiment of a tx spatial processor 720c capable of performing spatial processing for the beam steering mode.

Within processor 720c, a demultiplexer 1032 receives and demultiplexes the stream of modulation symbols s (n) into 48 sub-streams of 48 data subbands (labeled s (1) through s (k)). Each symbol substream is provided to a respective transmit subband beam steering processor 1040. Each processor 1040 performs the processing shown in equation (14) for one subband.

Within each transmit subband beam steering processor 1040, the modulation symbol substreams are provided to multipliers 1042, multipliers 1042 also providing associated subband receive gainsThen, the multiplier 1042 uses the gainThe modulation symbols are scaled to obtain scaled modulation symbols, which are then provided to beam steering unit 1050.

Beam steering unit 1050 also receives normalized eigenvectors for the relevant subbandsWithin beam steering unit 1050, is scaledThe amplified modulation symbols are provided to four multipliers 1052a through 1052d, which also receive the normalized eigenvectors, respectivelyFour elements of (2)Each multiplier 1052 normalizes the eigenvector values with itMultiplied by its scaled modulation symbols to provide preconditioned symbols. Multipliers 1052a through 1052d provide the four preconditioned symbol substreams to buffers/multiplexers 1070a through 1070d, respectively.

Each buffer/multiplexer 1070 receives pilot symbols and preconditioned symbols from transmit subband beam-steering processors 1040a through 1040k for 48 data subbands, multiplexes the pilot and preconditioned symbols and zeros for each symbol period, and provides a stream of transmit symbols x for one transmit antenna_i(n) of (a). The subsequent OFDM modulation for each transmit symbol stream is as described above.

The processing of the Beam Steering mode is further detailed in commonly assigned U.S. patent application Ser. No. 10/228,393, entitled "Beam-Steering and Beam-Forming for Wireless MIMO Systems", filed on 27.8.2002. The system may also be designed to support a beamforming mode whereby the data stream is transmitted on the dominant eigenmodes using eigenvectors instead of normalized vectors.

Framing of PHY frames

Fig. 11A illustrates one embodiment of a framing unit 808, the framing unit 808 being used to frame the data of each FCH/RCH packet before subsequent processing by the sending data processor. The framing function bypasses messages sent on BCH, FCCH and RACH. The framing unit generates an integer number of PHY frames for each FCH/RCH packet, where each PHY frame spans 6 OFDM symbols for the embodiments described herein.

For diversity and beam steering modes, only one spatial channel or wideband eigenmode is used for data transmission. The rate of this mode is known and the number of information bits that can be transmitted in the payload of each PHY frame can be calculated. For the spatial multiplexing mode, multiple spatial channels may be used for data transmission. Since the rate of each spatial channel is known, the number of information bits transmitted within the payload of each PHY frame can be calculated for all spatial channels.

As shown in FIG. 11A, the information bits (labeled i) of each FCH/RCH packet are grouped₁i₂i₃i₄…) to CRC generator 1102 and multiplexer 1104 within framing unit 808. CRC generator 1102 generates a CRC value for the bits in the header (if any) and payload fields of each PHY frame and provides the CRC bits to multiplexer 1104. Multiplexer 1104 receives the information bits, CRC bits, header bits, and padding bits (e.g., zero values) and provides these bits in the correct order based on the PHY frame control signal, as shown in fig. 6. The framing function may be bypassed by providing information bits directly through multiplexer 1104. Framed and unframed bits (labeled d)₁d₂d₃d₄…) is provided to scrambler 810.

5. Scrambling

In one embodiment, the data bits for each transport channel are scrambled at the encoder. Scrambling randomizes the data so that long sequences of all ones or all zeros are not transmitted. This may reduce the peak-to-average power variation of the OFDM waveform. Scrambling may be omitted for one or more transmission channels and may also be selectively enabled and disabled.

Fig. 11A also shows one embodiment of scrambler 810. In this embodiment, scrambler 810 implements a generator polynomial:

G(x)＝x⁷+x⁴+x (22)

other generator polynomials may also be used and are within the scope of the present invention.

As shown in fig. 11A, scrambler 810 includes seven delay elements 1112a through 1112g coupled in series. Adder 1114 performs a modulo-2 addition on the two bits held in delay elements 1112d and 1112g for each clock cycle and provides a scrambling bit to delay element 1112 a.

Framed/unframed bits (d)₁d₂d₃d₄) Is provided to an adder 1116, the adder 1116 applying a corresponding scrambling bit to each bit d_nPerforming modulo-2 addition to provide scrambled bits q_n. Scrambler 810 provides a sequence of scrambled bits, labeled q₁q₂q₃q₄…。

At the beginning of each TDD frame, the initial state of the scrambler (i.e., the contents of delay elements 1112a through 1112 g) is set to a non-zero number of 7 bits. As shown in the BCH message, the three Most Significant Bits (MSBs), i.e., delay elements 1112e through 1112f, are always set to one ("1") and the four Least Significant Bits (LSBs) are set to the TDD frame counter.

6. Coding/puncturing

In one embodiment, a single base code is used to encode the data prior to transmission. The base code generates code bits for a code rate. All other code rates supported by the system (as shown in table 25) may be obtained by either repeating the code bits or puncturing the code bits.

Fig. 11B illustrates one embodiment of an encoder 812 that can implement the base code of the system. In this embodiment, the base code is a rate 1/2 convolutional code with a constraint length of 7 (K-7), and the generators are 133 and 171 (octal).

Within the encoder 812, a multiplexer 1120 receives and multiplexes the scrambled bits and tail bits (e.g., zero values). Encoder 812 also includes six delay elements 1122a through 1122f coupled in series. Four adders 1124a to 1124d are also coupled in sequence and are used to implement the first generator (133). Similarly, four adders 1126a through 1126d are also coupled in sequence and are used to implement the second generator (171). As shown in fig. 11B, the adder is further coupled to a delay element in a manner that implements two generators 133 and 171.

The scrambled bits are provided to first delay element 1122a and adders 1124a and 1126 a. For each clock cycle, adders 1124a through 1124d perform modulo-2 addition on the incoming bit and the four previous bits held in delay elements 1122b, 1122c, 1122e, and 1122f to provide the first encoded bit for that clock cycle. Similarly, adders 1126a through 1126d perform modulo-2 addition on the incoming bit and the four previous bits held in delay elements 1122a, 1122b, 1122c, and 1122f to provide the second encoded bit for the clock cycle. The coded bits generated by the first generator are marked a₁a₂a₃a₄…, the coded bits generated by the second generator are marked b₁b₂b₃b₄… are provided. The multiplexer 1128 then receives the two coded bit streams from the two generators and multiplexes them into a single coded bit stream, labeled a₁b₁a₂b₂a₃b₃a₄b₄… are provided. For each scrambled bit q_nGenerating two coded bits a_nAnd b_nThis results in code rate 1/2.

Fig. 11B also illustrates one embodiment of a repetition/puncturing unit 814 that may be used to generate other code rates based on the base code rate of 1/2. Within element 814, the coded bits from rate 1/2 of encoder 812 are provided to repetition element 1132 and truncation 1134. The repetition element 1132 repeats each rate 1/2 code bit once to obtain an effective code rate 1/4. Puncturing unit 1134 may remove some of the code bits at rate 1/2 based on a particular puncturing pattern to provide a desired code rate.

Table 30 lists exemplary puncturing patterns that may be used for various code rates supported by the system. Other truncation patterns may also be used and are within the scope of the present invention.

Watch 30

Coding rate	Truncated pattern
		1/2	11
7/12	11111110111110
		5/8	1110111011
2/3	1110
		11/16	1111101111111010011100
3/4	111001
		13/16	01111011111101110000101100
5/6	1110011001
		7/8	11101010011001

To obtain the code rate k/n, puncturing unit 1134 provides n code bits for each set of 2k rate 1/2 code bits received from encoder 812. Thus, 2k-n code bits are removed from each set of 2k code bits. The bits to be deleted from each group are marked by zeros in the puncturing pattern. For example, to obtain code rate 7/12, two bits are removed from each set of 14 code bits from encoder 812, the removed bits being the 8 th and 14 th code bits in the set, as marked by puncturing pattern "11111110111110". If the desired coding rate is 1/2, no puncturing is performed.

A multiplexer 1136 receives the coded bit stream from the repetition element 1132 and from the puncturing element 1134. Then, if the desired code rate is 1/4, multiplexer 1136 provides the code bits from repetition unit 1132, and if the desired code rate is 1/2 or higher, multiplexer 1136 provides the code bit stream from puncturing unit 1134.

In addition to the coding and puncturing patterns described above, other coding and puncturing patterns may be used and are within the scope of the present invention. For example, the data may be encoded using a Turbo code, a block code, some other code, or any combination thereof. Also, different coding schemes may be used for different transport channels. For example, conventional coding may be used for common transport channels and Turbo coding may be used for dedicated transport channels.

7. Interlacing

In one embodiment, the code bits to be transmitted are interleaved among 48 data subbands. For diversity and beam steering modes, one coded bit stream is transmitted and interleaved across all data subbands. For spatial multiplexing mode, up to four coded bit streams may be transmitted on up to four spatial channels. Interleaving may be performed independently for each spatial channel such that each coded bit stream is interleaved among all data subbands of the spatial channel used to transmit the bit stream. Table 29 shows exemplary coded bit-subband assignments for interleaving that may be used for all transmission modes.

In one embodiment, interleaving is performed between all 48 data subbands in each interleaving interval. For this embodiment, each set of 48 coded bits in a stream is spread over 48 data subbands to provide frequency diversity. The 48 code bits in each group may be assigned indices 0 through 47. Each code bit index is associated with a respective subband. All coded bits with a particular index are transmitted on the associated subband. For example, the first code bit (index 0) in each group is transmitted on subband-26, the second code bit (index 14) is transmitted on subband 1, the third code bit (index 2) is transmitted on subband-17, and so on. This interleaving scheme can be used for diversity mode, beam steering mode, and spatial multiplexing mode. Other interleaving schemes for spatial multiplexing are described below.

Interleaving may alternatively or additionally be performed over time. For example, after interleaving between data subbands, the coded bits for each subband may be further interleaved (e.g., on one PHY frame or one PDU) to provide time diversity. For the spatial multiplexing mode, interleaving may also be performed over multiple spatial channels.

Furthermore, interleaving may be employed across the dimensions of the QAM symbols such that the coded bits forming a QAM symbol are mapped to different bit positions of the QAM symbol.

8. Symbol mapping

Table 31 shows symbol mapping for various modulation schemes supported by the system. For each modulation scheme (other than BPSK), half of the bits are mapped as in-phase (I) components and the other half of the bits are mapped as quadrature (Q) components.

In an embodiment, the signal constellation for each supported modulation scheme may be defined based on Gray (Gray) mapping. According to gray mapping, neighboring points in a signal constellation (in both the I and Q components) differ by only one bit position. Gray mapping reduces the number of bit errors for the more likely erroneous case, which corresponds to the received symbol being mapped to a position near the correct position, in which case only one coded bit would be received in error.

Watch 31

The I and Q values for each modulation scheme shown in Table 31 are normalized by a normalization factor k_normScaling so that the average power of all signal points in the constellation of relevant signals is equal to one. Quantized values of normalization factors for the supported modulation schemes may also be used. Thus, the modulation symbols s in a particular signal constellation will have the following form:

s＝(I+jQ)·K_norm，

where I and Q are the values of the signal constellation in Table 31.

For a given PDU, the modulation may differ between PDUs, and the multiple spatial channels used for data transmission may also differ. For example, for the BCH PDU, different modulation schemes can be used for beacon pilots, MIMO pilots, and BCH messages.

9. Processing of spatial multiplexing modes

For spatial multiplexing mode, a PDU may be transmitted on multiple spatial channels. Various schemes may be used to process data for transmission on multiple spatial channels. Two specific processing schemes for the spatial multiplexing mode are described below.

In a first processing scheme, coding and puncturing are performed per spatial channel to achieve a desired coding rate for each spatial channel. N to be used for data transmission_EThe spatial channels are arranged from highest to lowest received SNR. The data of the entire PDU is first encoded to obtain a rate 1/2 coded bit stream. The coded bits are then punctured to obtain the desired code rate for each spatial channel.

For N_EPuncturing may be performed in order, from the best (i.e., highest SNR) to the worst (i.e., lowest SNR) spatial channel. In particular, the puncturing unit first performs puncturing for the best spatial channel with the highest received SNR. When the correct number of coded bits has been generated for the best spatial channel, the puncturing unit performs puncturing for the next best spatial channel with the next highest received SNR. This process continues until all N_EThe coded bits for each spatial channel have been generated. The order of puncturing is from maximum received SNR to minimum received SNR regardless of the particular coding rate used for each spatial channel.

For the example shown in table 28, 3456 information bits to be transmitted within the total PHY frame are first encoded with a base code of rate 1/2 to obtain 6912 encoded bits. The first 3168 coded bits are punctured with a puncturing pattern at code rate 11/16 to obtain 2304 coded bits, which are provided within a PHY frame of the first spatial channel. The next 2592 code bits are then punctured with a puncturing pattern at code rate 3/4 to obtain 1728 code bits, which are provided within the PHY frame of the second spatial channel. The next 864 coded bits are then punctured with a puncturing pattern at coding rate 3/4 to obtain 576 coded bits that are provided within the PHY frame of the third spatial channel. The last 288 coded bits of the PHY frame are then punctured with a puncturing pattern at coding rate 1/2 to obtain 288 coded bits, which are provided within the PHY frame for the last spatial channel. The four separate PHY frames are further processed and transmitted on four spatial channels. Puncturing of the next total PHY frame is then performed in a similar manner. The first processing scheme may be implemented by transmit data processor 710b in fig. 9A.

In a second processing scheme, coding and puncturing is performed for pairs of subbands. In addition, the coding and puncturing cycles through all of the selected spatial channels for each pair of subbands.

Fig. 11C shows a block diagram illustrating a transmit data processor 710d implementing the second processing scheme. Encoder 812 performs convolutional encoding of rate 1/2 on the scrambled bits from scrambler 810. Each spatial channel is assigned a particular rate that is associated with a particular combination of code rate and modulation scheme, as shown in table 25. Let b_mRepresenting the number of coded bits per modulation symbol for spatial channel m (or equivalently, the number of coded bits sent on each data subband of spatial channel m), r_mRepresenting the coding rate used for spatial channel m. b_mThe value of (d) depends on the constellation size of the modulation scheme used for spatial channel m. In particular, for BPSK, QPSK, 16-QAM, 64-QAM and 256-QAM, b_mEqual to 1, 2, 4, 6 and 8, respectively.

The encoder 812 provides a rate 1/2 encoded bit stream to a demultiplexer 816, and the demultiplexer 816 demultiplexes the received encoded bit stream into four substreams for the four spatial channels. Demultiplexing so that the first 4b₁r₁The coded bits are sent to buffer 813a, followed by 4b, of spatial channel 1₂r₂The coded bits are sent to buffer 813b for spatial channel 2 and so on. Each buffer 813 receives 4b whenever demultiplexer 816 cycles once on all four spatial channels_mr_mAnd (4) encoding bits. For each cycle, there is a totalThe coded bits of rate 1/2 are provided to four buffers 813a through 813 d. Thus, for each b_totalFor each coded bit, the demultiplexer 816 cycles through all four positions, b, of the four spatial channels_totalIs the number of coded bits that can be transmitted on a pair of subbands using all four spatial channels.

Once each buffer 813 uses 4b of the associated spatial channel_mr_mThe code bits in the buffer may be punctured to obtain the code rate for the spatial channel, filled with code chips. Due to 4b_mr_mThe coded bits of rate 1/2 span an integer number of puncturing periods for each puncturing pattern, thus providing, in effect, 2b after puncturing for each spatial channel m_mAnd (4) encoding bits. Then, 2b of each spatial channel_mThe coded bits are distributed (or interleaved) across the data subbands.

In an embodiment, interleaving is performed for each spatial channel in a set of 6 subbands at a time. The punctured coded bits for each spatial channel may be arranged sequentially as c_iFor i ═ 0, 1, 2, …. Maintaining a counter C for each spatial channel_mSo that each group 6b provided for the spatial channel for the puncturing element_mOne coded bit is counted. For example, for b_mQPSK of 2, coded bits c provided for the puncturing unit₀To c₁₁The counter will be set to C_mFor the following coded bit c ═ 0₁₂To c₂₃Will be set as C_mAnd so on for 1. Counter value C for spatial channel m_mCan be expressed as:

to determine the code bit c_iTo which sub-band is assigned, the coding is first determined as followsCode index of code bit:

bit index ═ i mod 6) +6 · C_m (24)

The bit indices are then mapped to the corresponding subbands using table 29.

For the above example, the first set of 6 coded bits c₀To c₅A second group of 6 coded bits c, associated with bit indices 0 to 5, respectively₆To c₁₁And are also associated with bit indices 0 through 5, respectively. As shown in table 29, code bit c₀And c₆Will be mapped to subband-26, encoding bit c₁And c₇Will be mapped to subband 1 and so on. Spatial processing then begins for this first set of 6 subbands. Third group of 6 coded bits c₁₂To c₁₇(C_m1) are associated with bit indices 6 to 11, respectively, and a fourth set of 6 coded bits c₁₈To c₂₃And are also associated with bit indices 6 through 11, respectively. Coded bit c₁₂And c₁₈Will be mapped to subband-25, coded bit c₁₃And c₁₉Will be mapped to subband 2 and so on. Spatial processing then begins for this next set of 6 subbands.

The number 6 in equation (24) results from performing interleaving in groups of 6 subbands. The (mod8) operation in equation (23) stems from having 8 interleaved sets for 48 data subbands. Since each cycle of demultiplexer 816 shown in fig. 11C produces enough code bits to fill both subbands of each wideband eigenmode, a total of 24 cycles are required to provide 48b for one OFDM symbol for each spatial channel_mAnd (4) encoding bits.

Interleaving in groups of 6 subbands at a time may reduce processing delay. In particular, once each set of 6 subbands is available, spatial processing may begin.

In other embodiments, one may be at N_BInterleaving is performed for each spatial channel in groups of subbands, where N is_BMay be any integer (e.g. for all 48)Interleaving over data subbands, N_BMay be equal to 48).

VI calibration

For a TDD system, the downlink and uplink share the same frequency band in a time division duplex manner. In this case, half of the correlation exists between the channel responses of the downlink and uplink. The correlation may be used to simplify channel estimation and spatial processing. For a TDD system, each sub-band of the wireless link is assumed to be reciprocal. That is, ifH(k) Representing the channel response matrix from antenna array A to antenna array B for subband k, then the reciprocal channel means the junction from antenna array B to antenna array AH(k) Is given by the transpose ofH ^T(k)。

However, the response (gain and phase) of the transmit and receive chains at the access point is generally different from the response of the transmit and receive chains at the user terminal. Calibration may be performed to determine the difference between the frequency responses of the transmit/receive chains at the access point and the user terminal and to compensate for the difference so that the calibrated downlink and uplink responses can be represented in terms of each other. Once the transmit/receive chains have been calibrated and complemented, the metrics for one link (e.g., downlink) can be used to derive control vectors for the other link (e.g., uplink).

"effective" downlink and uplink channel responsesH _dn(k) AndH _up(k) including responses of the transmit and receive chains available at the access point and the user terminal, and is represented as:

H _dn(k)＝R _ut(k)H(k)T _ap(k)，k∈K， (25)

H _up(k)＝R _ap(k)H ^T(k)T _ut(k)，k∈K，

whereinT _ap(k) AndR _ap(k) is N_ap×N_apDiagonal moment ofArray whose entries are for subband k, N at the access point respectively_apA term of complex gain associated with transmit and receive chains of a root antenna;

T _ut(k) andR _ut(k) is N_ut×N_utWith entries for subband k and N at the user terminal, respectively_utA term of complex gain associated with transmit and receive chains of a root antenna; and

H(k) is N of the downlink_ut×N_apA channel response matrix.

Combining the two formulas in formula set (25) to obtain the following relationship:

H _up(k)K _ut(k)＝(H _dn(k)K _ap(k))^T，k∈K， (26)

whereinAnd is

The left side of equation (26) represents the "actual" calibrated channel response on the uplink, and the right side represents the transpose of the "actual" calibrated channel response on the downlink. Applying diagonal matrices to the effective downlink and uplink channel responses, respectively, as shown in equation (26)K _ap(k) AndK _ut(k) the aligned channel responses of the downlink and uplink can be represented by transposes of each other. Of access point (N)_ap×N_ap) The diagonal matrix is the receive chain responseR _ap(k) And transmit chain responseT _ap(k) Ratio of (i.e. the) Wherein the ratio is derived element by element. Similarly, of the user terminal (N)_ut×N_ut) Diagonal matrixIs the receive chain responseR _ut(k) And transmit chain responseT _ut(k) The ratio of.

Matrix arrayK _ap(k) AndK _ut(k) including values that can make up for differences between the transmit/receive chains at the access point and the user terminal. This then enables the channel response of one link to be represented by the channel response of the other link, as shown in equation (26).

Calibration may be performed to determine the matrixK _ap(k) AndK _ut(k) in that respect In general, the actual channel responseH(k) And transmit/receive chain responses are unknown, nor can they be determined accurately or easily. But rather can estimate the effective downlink and uplink channel responses based on the pilots transmitted on the downlink and uplink, respectivelyH _dn(k) AndH _up(k) as described below. The channel response estimate may then be based on the downlink and uplink channels, as described belowAndto derive a matrixK _ap(k) AndK _ut(k) is known as the correction matrixAndmatrix arrayAndincluding correction factors that can compensate for differences between the transmit/receive chains at the access point and the user terminal.

The "calibrated" downlink and uplink channel responses observed by the user terminal and access point are expressed as:

whereinH _cdn ^T(k) AndH _cup(k) is an estimate of the "actual" calibrated channel response expression in equation (26). Using the expression of equation (26) to join two equations in equation set (27), one may obtainRelation formulaDepends on the matrixAndthe latter in turn typically depending on the downlink and uplink channel response estimatesAndthe quality of (c).

Calibration may be performed using various schemes. For clarity, a specific calibration scheme is described below. To perform calibration, the user terminal first obtains the timing and frequency of the access point based on the beacon pilot transmitted on the BCH. The user terminal then sends a message on the RACH to start the calibration procedure with the access point. Calibration may be performed in parallel with enrollment/verification.

Since the frequency response of the transmit/receive chains at the access point and the user terminal is generally smooth over most of the noted frequency bands, the phase/gain difference of the transmit/receive chains can be characterized by a small number of subbands. Calibration may be performed for 4, 8, 16, 48, or some other number of subbands specified within the message sent to start calibration. Calibration may also be performed for the pilot subbands. The calibration constants for subbands on which calibration is not explicitly performed may be calculated by interpolating the calibrated subbands. For clarity, it is assumed below that calibration is performed for all data subbands.

For calibration, the access point allocates a sufficient amount of time on the RACH to the user terminal to transmit an uplink MIMO pilot of sufficient duration plus a message. The duration of the uplink MIMO pilot may depend on the number of subbands on which calibration is performed. For example, if calibration is performed for four subbands, 8 OFDM symbols may be sufficient, and more subbands may require more (e.g., 20) OFDM symbols. The total transmit power is typically fixed, so if the MIMO pilot is transmitted on a small number of subbands, a larger number of transmit powers may be used for each of these subbands, with a high SNR for each subband. Conversely, if the MIMO pilot is transmitted on a large number of subbands, a smaller amount of transmit power may be used for each subband, with poor SNR for each subband. If the SNR for each subband is not high enough, more OFDM symbols are transmitted for the MIMO pilot and the OFDM symbols are integrated at the receiver to obtain a higher overall SNR for that subband.

The user terminal then transmits a MIMO pilot on RCH, which is used by the access point to derive an estimate of the effective uplink channel response for each data subbandThe uplink channel response estimate is quantized (e.g., to a 12-bit complex value, having in-phase (I) and quadrature (Q) components) and transmitted to the user terminal.

The user terminal also derives estimates of the effective downlink channel response for the various data subbands based on the downlink MIMO pilot sent on the BCHObtaining efficient uplink and downlink channel response estimates for all data subbandsAndthereafter, the user terminal determines a correction factor for each data subbandAndwhich are used by the access point and the user terminal, respectively. Can correct the vectorIs defined as only comprisingThe diagonal elements of (1), and correcting the vectorsIs defined as only comprisingThe diagonal elements of (a).

The correction factors may be derived in various ways, including by matrix ratio calculations and MMSE calculations. Both of these calculation methods are described in further detail below. Other calculation methods may also be used and are within the scope of the invention.

1. Matrix ratio calculation

To estimate based on the effective downlink and uplink channel responsesAndto determine a correction vectorAndfirst, a one (N) is calculated for each data subband_ut×N_ap) Matrix arrayC(k) The following are:

where the ratio is derived element by element. Thus, it is possible to provideC(k) Can be calculated as follows:

whereinIs thatThe (i, j) th (row, column) element of (a),is thatThe (i, j) th element of (c)_i，j(k) Is thatC(k) The (i, j) th element of (a).

Thus, the correction vector of the access pointIs equal toC(k) Is measured in the mean of the normalized lines of (a). First, the first element in a row is used to pair N in the row_apEach of the elements is scaled to thereby pairC(k) Is normalized. Thus, ifIs thatC(k) Line i of (1), then normalized lineCan be expressed as:

thus, the mean value of the normalization rows is N_utThe sum of the normalized lines divided by N_utExpressed as follows:

due to the standardization, thereforeIs one.

Correction vector for user terminalIs equal toC(k) Is calculated as the mean of the reciprocal of the normalized column. First using the vectorThe jth element of (notation K)_ap，j，j(k) To) scale each element in a column, thereby toC(k) Is normalized. Thus, ifIs thatC(k) Line j of (1), then normalized lineCan be expressed as:

thus, the mean of the reciprocal of the normalized column is N_apThe sum of the reciprocals of the normalized columns divided by N_apExpressed as follows:

wherein the normalized columnThe inverse of (c) is performed element by element.

MMSE calculation

For MMSE calculation, correction factorAndestimating from effective downlink and uplink channel responsesAndsuch that the Mean Square Error (MSE) between the calibrated downlink channel response and the calibrated uplink channel response is minimized. This condition can be expressed as follows:

it can also be written as:

wherein because ofIs a diagonal matrix and is thus

Equation (34) is constrained:is set to one (i.e., the first element of (c))). Without this constraint, a common solution, the matrix, would be obtainedAndis set to zero. In equation (34), a matrix is first obtainedY(k)：Followed by a matrixY(k) N of (A)_ap·N_utEach term of the terms results in the square of the absolute value.

Performing MMSE computation for each specified subband to obtain a correction factor for that subbandAndthe MMSE calculation for one subband is described below. For brevity, the subband index k is omitted in the following description. Also for the sake of brevityClean, downlink channel response estimationIs marked as a_ijEstimation of uplink channel responseIs marked as b_ij}, matrixIs marked as u_i}, matrixIs marked as v_iWhere i ═ 1 … N_apAnd j ═ 1 … N_ut}。

The mean square error can be rewritten from equation (34) as follows:

with the same constraint of u₁1. The minimum mean square error is obtained by taking the partial derivative of equation (35) with reference to u and v and setting the partial derivative to zero. The result of these operations is the following set of equations:

in the formula (36a), u₁1, so there is no partial derivative in this case, the index i is taken from 2 to N_ap。

(N) in the formula sets (36a) and (36b)_ap+N_ut-1) the set of equations can be more conveniently represented in matrix form, as follows:

Ay＝z， (37)

wherein

Matrix arrayAComprising (N)_ap+N_ut-1) lines, first N_ap-1 line corresponds to N in the formula set (36a)_ap1 formula, last N_utThe rows correspond to N in the formula set (36b)_utAnd (4) a formula. In particular, a matrixAThe first row of (a) is generated from formula set (36a) according to i-2, the second row is generated according to i-3, and so on. Matrix arrayAN of (2)_apLines are generated from the formula set (36b) according to j-1, and so on, and the last line is N according to j_utAnd (4) generating. As shown above, the matrixASum vector ofzMay be based on a matrixAndto be obtained by the following steps.

The correction factor being included in the vectoryIn (b), the following is obtained:

y＝A ^-1 z (38)

the result of the MMSE computation is a correction matrix that minimizes the mean square error in the calibrated downlink and uplink channel responsesAndas shown in equation (34). Due to the matrixAndestimating channel response based on downlink and uplinkAndobtaining, thus correcting, a matrixAndis dependent on the channel estimateAndthe quality of (c). MIMO pilot frequency is averaged at receiver to obtainAnda more accurate estimate of.

Correction matrix obtained based on MMSE calculationAndgenerally better than a correction matrix obtained based on a matrix ratio calculation. Some channels have small gain and the noise energy is measured to make the channel gain largeThis is especially true in the case of large degradation.

3. Post-computation

A pair of correction vectors may be determined for each data subbandAndthe computation is simplified because adjacent subbands may be correlated. For example, the calculation may be performed for every n subbands, rather than for each subband, where n may be determined by the expected response of the transmit/receive chain. If calibration is performed for less than all of the data and pilot subbands, then correction factors for "uncalibrated" subbands may be obtained by interpolating correction factors for "calibrated" subbands.

Various other calibration schemes may also be used to derive correction vectors for the access point and user terminal, respectivelyAndhowever, the above scheme enables deriving "compatible" correction vectors for the access point when the calibration is performed by different user terminals.

After derivation, the user terminal corrects the vectors for all data subbandsAnd sent back to the access point. If the access point has been calibrated (e.g., by other user terminals), the current correction vector is updated with the newly received correction vector. Thus, if the access point uses the correction vectorTo transmit a MIMO pilot from which the user terminal derivesTo determine a new correction vectorThe updated correction vector is the product of the current and new correction vectors, i.e.Where the multiplication is performed on an element-by-element basis. Then, the updated correction vectorMay be used by the access point until they are updated again.

Correction vectorAndmay be derived by the same user terminal or by different user terminals. In one embodiment, the updated correction vector is defined asWhere the multiplication is performed on an element-by-element basis. In another embodiment, the updated correction vector may be redefined asWhere α is the factor used to provide the weighted average (e.g., 0 < α < 1). If the calibration updates are infrequent, then α is closeAs best as possible at 1. A smaller value of alpha may be better if the calibration updates are frequent but noisy. Then, the updated correction vectorAre used by the access point until they are updated again.

Access point and user terminal use their respective correction vectorsAndor a corresponding correction matrixAndthe modulation symbols are scaled (for K ∈ K) prior to transmission, as described below. Equation (27) shows the calibrated downlink and uplink channels observed by the user terminal and access point.

VII spatial processing

Spatial processing at the access point and user terminal can be simplified for a TDD system after calibration has been performed to account for differences in the transmit/receive chains. As described above, the calibrated downlink channel response isCalibrated uplink channel response of

1. Uplink spatial processing

Calibrated uplink channel response matrixH _cup(k) The singular value decomposition of (a) may be expressed as:

whereinU _ap(k) Is thatH _cup(k) Of the left eigenvector of (N)_ap×N_ap) A unitary matrix;

∑(k) is thatH _cup(k) Of singular values of (N)_ap×N_ut) A diagonal matrix; and

V _ut(k) is thatH _cup(k) Of the right eigenvector of (N)_ut×N_ut) A unitary matrix.

Accordingly, the calibrated downlink channel response matrixH _cdn(k) The singular value decomposition of (a) may be expressed as:

matrix arrayV _ut ^*(k) AndU _ap ^*(k) are respectivelyH _cdn(k) A matrix of left and right eigenvectors. As shown in equations (39) and (40) and based on the above description, the matrices of the left and right eigenvectors of one link are the complex conjugates of the matrices of the right and left eigenvectors, respectively, of the other link. Matrix arrayV _ut(k)、V _ut ^*(k)、V _ut ^T(k) AndV _ut ^H(k) is a matrixV _ut(k) Of different forms, matricesU _ap(k)、U _ap ^*(k)、U _ap ^T(k) And U_ap ^H(k) Is also a matrixU _ap(k) Different forms of (1). For the sake of brevity, the matrices referred to in the following descriptionU _ap(k) AndV _ut(k) but also to various other forms thereof. Matrix arrayU _ap(k) AndV _ut(k) used by the access point and the user terminal, respectively, for spatial processing and identified by their subscripts. Eigenvectors are also commonly referred to as "control" vectors.

User terminals may estimate calibration based on MIMO pilots transmitted by access pointsThe downlink channel response of. The user terminal may then estimate the calibrated downlink channel responsePerforming a singular value decomposition (for K ∈ K) to obtainIs diagonal matrix ofAnd the matrix of left eigenvectorsV _ut ^*(k) In that respect The singular value decomposition can be given by:where the cap on each matrix "^" indicates that it is an estimate of the actual matrix.

Similarly, the access point may estimate the calibrated uplink channel response based on the MIMO pilot sent by the user terminal. The access point may then estimate the calibrated uplink channel responsePerforming a singular value decomposition (for K ∈ K) to obtainIs diagonal matrix ofAnd the matrix of left eigenvectorsU _ap ^*(k) In that respect The singular value decomposition can be given by:

one (N)_ut×N_ut) Matrix arrayF _ut(k) Can be defined as:

while active, the user terminal continuously estimates the calibrated downlink channelAndmatrix of left eigenvectorsThe latter for updating the matrixF _ut(k)。

User terminal usage matrixF _ut(k) Spatial processing is performed for beam steering and spatial multiplexing modes. For spatial multiplexing mode, transmit vector for each subbandx _up(k) Can be expressed as:

x _up(k)＝F _ut(k)s _up(k)，k∈K， (42)

whereins _up(k) Is a data vector with N to be at subband k_SN transmitted on eigenmodes_SA code element;

F _ut(k) substitution of one in equation (15)V(k) For simplicity, the channel inversion scheme for the channel inversion is omitted from equation (42)G(k) Signal scaling is performed;

x _up(k) is the uplink transmit vector for subband k.

At an access point, a received vector for an uplink transmissionr _up(k) Can be expressed as:

whereinr _up(k) Is the received vector for uplink subband k; and

n _up(k) is the Additive White Gaussian Noise (AWGN) for subband k. Equation (43) uses the following relationship:andat the access point, the received uplink transmission is transmitted as shown in equation (43)Is transformed, the latter beingOf the left eigenvector ofDiagonal matrix composed of singular valuesAnd (4) zooming.

User terminal usage matrixF _ut(k) A controlled reference is transmitted on the uplink. The controlled reference is a pilot transmission on one of the wideband eigenmodes using beam steering or beam forming, as described in more detail below. At the access point, the received uplink controlled reference (in the absence of noise) is approximated asIn this way, the access point can derive a unitary matrix based on the controlled reference transmitted by the user terminalAnd diagonal matrixIs estimatedAnd (6) counting. Various estimation techniques can be used to derive estimates of the unitary and diagonal matrices.

In one embodiment, to deriveFor subband k of wideband eigenmode m, the received vector of the controlled referencer _m(k) First of all with the complex conjugate p of the pilot OFDM symbol transmitted for the steered reference^*(k) Multiplication. The generation of the steered reference and pilot OFDM symbols is described in detail below. For each wideband eigenmode, the result is integrated over multiple received controlled reference symbols to yieldIs estimated by the estimation of (a) a,being a broadband eigenmode mScaled left eigenvector of (a). Since the eigenvectors have unit power, they can be estimated based on the received power of the controlled referenceSingular value (or σ) of_m(k) Received power of a controlled reference may be measured for each subband of each wideband eigenmode.

In another embodiment, a MMSE technique is used to receive vectors based on a controlled referencer _m(k) To obtainIs estimated.

The controlled reference may be transmitted for a wideband eigenmode in any given symbol period and used to derive an estimate of an eigenvector for each subband of the wideband eigenmode. Thus, the receiver can derive an estimate of an eigenvector in a unitary matrix in any given symbol period. Since estimates of multiple eigenvectors of the unitary matrix are derived in different symbol periods, and due to noise and other sources of degradation in the transmission path, the eigenvectors estimated for the unitary matrix may not be orthogonal. If the estimated eigenvectors are then used for spatial processing of data transmissions on other links, any error in the orthogonality of these estimated eigenvectors will result in cross-talk between the eigenmodes, which will degrade performance.

In one embodiment, the eigenvectors estimated for each unitary matrix are forced to be orthogonal to each other. The orthogonality of the eigenvectors may be achieved using various techniques such as QR factorization, minimum mean square error calculation, polarization decomposition, and the like. QR factorization of a matrixM ^T(with non-orthogonal columns) into an orthogonal matrixQ _FAnd an upper triangular matrixR _F. Matrix arrayQ _FIs composed ofM ^TThe columns of (a) form an orthogonal basis.R _FIs at the diagonal element ofQ _FGiven in the direction of the respective columnM ^TLength of the components of each column. Matrix arrayQ _FMay be used for spatial processing on the downlink. Matrix arrayQ _FAndR _Fmay be used to derive an enhanced matched filter matrix for the uplink. QR factorization may be performed by various methods, including Gram-Schmidt procedures, assisted transformations, and so on.

Other techniques for estimating the unitary and diagonal matrices based on a controlled reference may also be used and are within the scope of the present invention.

Thus, the access point may estimate based on the controlled reference transmitted by the user terminalAndboth without the need forSingular value decomposition is performed.

Normalized matched filter matrix for uplink transmission from user terminalsM _ap(k) Can be expressed as:

the matched filtering for uplink transmission at the access point can be expressed as:

whereinIs a vector of modulation symbols transmitted by a user terminal for a spatial multiplexing modes _up(k) Is estimated. For the beam steering mode, only the matrix is usedM _ap(k) To provide a symbol estimate for the eigenmode used for data transmission

2. Downlink spatial processing

For the downlink, the access point uses one (N)_ap×N_ap) Matrix arrayF _ap(k) To perform spatial processing. The matrix can be expressed as:

correction matrixDerived by the user terminal and sent back to the access point during calibration. Matrix arrayMay be derived based on the controlled reference sent by the user terminal on the uplink.

For spatial multiplexing mode, a transmit vector for the downlink for each data subbandx _dn(k) Can be expressed as:

x _dn(k)＝F _ap(k)s _dn(k)，k∈K， (47)

whereinx _dn(k) Is the transmission of the vector or vectors,s _dn(k) is a data vector of the downlink, again omitted for simplicityG(k) Signal scaling to achieve channel inversion.

Receiving vector of downlink transmission at user terminalr _dn(k) Can be expressed as:

as shown in equation (48), at the user terminal,the received downlink transmission isThe transformation is carried out by changing the parameters of the image,is thatMatrix of left eigenvectorsDiagonal matrix composed of singular valuesTo scale.

As described above, by estimating the downlink channel response for calibrationBy performing singular value decomposition, the user terminal can derive a diagonal matrixAnd the matrix of left eigenvectors

The matched filtering at the user terminal for the downlink transmission can then be expressed as:

3. spatial processing of access points and user terminals

Due to the reciprocal channel and calibration of the TDD system, spatial processing at both the access point and the user terminal is simplified. Table 32 summarizes the spatial processing at the access point and the user terminal for data transmission and reception.

Watch 32

Spatial processing of data reception is also referred to as matched filtering.

Due to the existence of reciprocal channels, thereforeIs a user terminal(for transmission) right eigenvectors anda matrix of both left eigenvectors (for reception). In a similar manner to that described above,is an access point(for transmission) right eigenvectors anda matrix of both left eigenvectors (for reception). Singular value decomposition requires only downlink channel response estimation by the user terminal for calibrationIs performed to obtainAndthe access point may derive the controlled reference based on the control sent by the user terminalAndand without the need for uplink channel response estimationSingular value decomposition is performed. Access point and user terminal may be derived from the user terminalAnd different means are used so as to have different forms of matricesFurther, the access point derives a matrix based on the controlled referenceMatrices derived by singular value decomposition, typically with user terminalsDifferent. For the sake of brevity, these differences are not shown in the above derivation.

4. Beam steering

For certain channel conditions, it is preferable to transmit data on only one wideband eigenmode, which is generally the best or dominant wideband eigenmode. This situation may be: the received SNR of all other wideband eigenmodes is sufficiently poor that improved performance can be achieved by using all available transmit power on the dominant wideband eigenmode.

Data transmission on one wideband eigenmode may be achieved with beamforming or beam steering. For beamforming, eigenvectors of the dominant wideband eigenmodes are typically usedOr(i.e. after the sorting,orThe first column) performs spatial processing on the modulation symbols, where K ∈ K. For beam steering, a set of "normalized" (or saturated) eigenvectors of the dominant wideband eigenmodes is typically usedOrThe modulation symbols are spatially processed, where K ∈ K. For clarity, beam steering for the uplink is described below.

For the uplink, each eigenvector of the dominant wideband eigenmodeMay have different sizes, where K ∈ K. Thus, the pre-conditioned symbols for each subband may also have different sizes by combining the modulation symbols for subband k with the eigenvectors for subband kIs obtained by multiplying the elements of (1). Thus, the transmit vectors for each antenna, which each include the preconditioned symbols for all of the data subbands for a given transmit antenna, may have different sizes. If the transmit power of each transmit antenna is limited (e.g., due to limitations of the power amplifier), then the total power available for each antenna is not fully used for beamforming.

Beam steering uses only the eigenvectors of the dominant wideband eigenmodesK ∈ K, and each eigenvector is normalized so that all elements in the eigenvector have equal size. Normalized eigenvectors for subband kCan be expressed as:

where a is a constant (e.g., a ═ 1); and

θ_i(k) is the phase of subband k for antenna i, given as:

vector as shown in equation (52)The phase of each element in the array is derived from the eigenvectorIs derived from the corresponding element of (i.e. theta)_i(k) FromTo obtain itIn)。

5. Uplink beam control

The spatial processing performed by the user terminal on the uplink for beam steering may be represented as:

whereins _up(k) Is the modulation symbol to be transmitted on subband k; and

is the transmit vector for subband k for beam steering.

The normalized control vector for each sub-band, as shown in equation (53)N of (A)_utThe elements may be of equal size but may be of different phase.

The uplink transmissions received at the access point for beam steering may be represented as:

whereinIs the uplink reception direction of subband k for beam steeringAmount of the compound (A).

Matched filter row vectors for uplink transmission using beam steeringCan be expressed as:

matched filter vectorCan be derived as follows. The spatial processing (i.e., matched filtering) at the access point for receiving uplink transmissions using beam steering may be represented as:

wherein(i.e. theIs thatAnd the inner product of its conjugate transpose),

is a modulation symbol s transmitted by a user terminal on the uplink_up(k) Is estimated, and

is post-processed noise.

6. Downlink beam control

The spatial processing performed by the access point for beam steering on the downlink can be expressed as:

whereinIs a normalized eigenvector for subband k, which is based on the eigenvectors of the dominant wideband eigenmodesBut generated as described above for the uplink.

Matched filter row vectors for downlink transmission using beam steeringCan be expressed as:

the spatial processing (i.e., matched filtering) of the received downlink transmission at the user terminal may be represented as:

wherein(i.e. theIs thatAnd the inner product of its conjugate transpose).

7. Spatial processing with channel inversion

For the uplink, the transmission vector of the spatial multiplexing modex _up(k) Can be derived by the user terminal as:

whereinG(k) Is a diagonal matrix of the gain of the channel inversion described above. Equation (60) is similar to equation (15) except thatInstead of the formerV(k) And (c) other than.Is provided to a multiplier 952 within the beamformer 950 of fig. 9B.

Transmit vectors for beam steering mode for uplinkCan be derived by the user terminal as:

whereinIs a vector having four elements of equal magnitude but phase based on the eigenvectors of the principal eigenmodesAnd then the result is obtained. Vector quantityMay be derived similarly to that described above in equations (16) and (17). Gain ofChannel inversion is achieved and may be derived similar to that described above in equations (18) through (20), except that equation (20) is usedAnd (c) other than.Is provided to a multiplier 1052 within the beam steering unit 1050 of fig. 10B.

For downlink, transmission vector of spatial multiplexing modex _dn(k) Can be derived by the access point as:

equation (62) is similar to equation (15) except that instead of replacingV(k) To useAnd (c) other than.May be provided to a multiplier 952 within the beamformer 950 of fig. 9B.

For downlink, transmission vectors of beam steering modeCan be derived by the access point as:

whereinIs a vector having four elements of equal size but whose phase is based on the dominant eigenmodesTo obtain. Gain ofChannel inversion is achieved and may be derived as described above in equations (18) through (20) except for the use of equation (20)And (c) other than.Is provided to a multiplier 1052 within the beam control unit 1050 in fig. 10B.

Pilot structure

A pilot structure is provided for MIMO WLAN systems to enable access points and user terminals to perform timing and frequency acquisition, channel estimation, and other functions needed for proper system operation. Table 33 lists four classes of pilots for an exemplary pilot structure and their short descriptions.

TABLE 33 Pilot types

Pilot frequency type	Description of the invention
		Beacon pilot	Pilots are transmitted from all transmit antennas and used for timing and frequency acquisition.
MIMO pilot	Transmitted from all transmit antennas with different orthogonal codesAnd is used as a pilot for channel estimation.
		Steered reference or steered pilot	Pilots are transmitted on specific eigenmodes of the MIMO channel for a particular user terminal and are used for channel estimation and possibly rate control.
Carrier pilot	A pilot for phase tracking the carrier signal.

Steered reference and steered pilot are synonyms.

In one embodiment, a pilot structure includes: (1) for downlink-beacon pilots, MIMO pilots, steered reference and carrier pilots transmitted by the access point, and (2) for uplink-MIMO pilots, steered reference and carrier signals transmitted by the user terminals.

The downlink beacon pilot and the MIMO pilot are transmitted on the BCH within each TDD frame (as shown in fig. 5A). The user terminal may use the beacon pilot for timing and frequency acquisition and doppler estimation. The user terminal may use the MIMO pilot to: (1) deriving an estimate of the downlink MIMO channel, (2) deriving a steering vector for the uplink transmission (if beam-steering or spatial multiplexing mode is supported), and (3) deriving a matched filter for the downlink transmission. The downlink controlled reference may be used by a particular user terminal for channel estimation.

The uplink controlled reference is transmitted by each active user terminal that supports beam-steering or spatial multiplexing modes and is usable by the access point to: (1) derive a control vector for downlink transmissions, and (2) derive a matched filter for uplink transmissions. Typically, the controlled reference is only transmitted by user terminals that support beam steering and/or spatial multiplexing modes. The reference sent item, whether or not it was correctly controlled (e.g., due to poor channel estimation). That is, since the control matrix is diagonal, the reference also becomes orthogonal per transmit antenna.

If the user terminal is calibrated, it can use the vectorTransmitting a controlled reference (for K ∈ K) on the primary eigenmode on RACH, whereFor the dominant eigenmodesThe column (c). If the user terminal is not aligned, it may use the vectorA pilot is sent on the RACH (for K e K). Vector per subbandv _ut.p(k) Comprising N_utA random control coefficient, their phase theta_i(k) Possibly according to a pseudo-random process, where i e {1, 2, … N_ut}. Since only N is present_utThe relative phase between the control coefficients is related, so that the phase of the first control coefficient can be set to zero (i.e., θ)₁(k) 0). Other N_utThe phase of 1 control coefficient may change at each access attempt, so that each control coefficient is equal toCovers a full 360 degrees, whereinIs N_utAs a function of (c). When using RACH in beam mode before calibration, control vectors are paired at each RACH attemptv _ut，p(k) N of (A)_utThe phase perturbation of the individual elements causes the user terminal not to use bad control vectors for all access attempts. MIMO may be transmitted for or by user terminals that do not support beam steering and/or spatial multiplexing modes.

The access point does not know the channel of any user terminal until the user terminal communicates directly with the access point. When a user wishes to transmit data, it first estimates the channel based on the MIMO pilot transmitted by the access point. ()

Control vectorIs a calibrated uplink channel response estimateMatrix of right eigenvectorsIn the first column, whereinIs thatColumn i. The above assumptionsSingular value sum ofThe columns of (a) are arranged in the above order.

The second symbol of the controlled reference transmitted by the user terminal in the preamble of the RACH comprises the Data Rate Indicator (DRI) of the RACH PDU. By mapping DRI to a particular QPSK symbol s, as shown in Table 15_driEmbedding the DRI in a second controlled reference symbol, then s_driThe symbols are multiplied by pilot symbols p (k) before spatial processing. The second symbol of the controlled reference for the RACH may be represented as:

only the eigenvectors of the dominant eigenmodes, as shown in equations (64) and (65)Is used for the controlled reference for the RACH.

The symbols of the controlled reference transmitted by the user terminal in the preamble of the RCH can be represented as:

whereinx _up，sr，m(m) is the transmit vector for subband k for wideband eigenmode m; and

is the steering vector for subband k of wideband eigenmode m (i.e., theColumn m).

The symbols of the controlled reference sent by the access point in the preamble of the RCH can be represented as:

whereinx _dn，sr，m(m) is the transmit vector for subband k for wideband eigenmode m; and

is a correction matrix for subband k of the access point; and

is the steering vector for subband k for wideband eigenmode m.

Control vectorIs a calibrated downlink channel response estimateRight eigenvector matrix ofColumn m of (1), wherein

The controlled reference may be transmitted in various ways. In an embodiment, one or more eigenvectors are used for the controlled reference for each TDD frame, and the latter is represented by the FCH/RCH preamble type field in the FCCH information element depending on the duration of the controlled reference. Table 36 lists the eigenmodes used for RCH and preamble of RCH for various preamble sizes for an exemplary design.

Watch 36

Type (B)	Preamble size	Eigenmodes used
			0	0 OFDM code element	Without preamble sequence
1	1 OFDM code element	Eigenmode m, where m is frame counter mode 4
			2	4 OFDM code elements	Cycling through all 4 eigenmodes in the preamble
3	8 OFDM code elements	Cycling twice on all 4 eigenmodes in the preamble sequence

As shown in table 36, the controlled reference is transmitted for all four eigenmodes within a single TDD frame when the preamble sequence size is 4 or 8 OFDM symbols. The controlled reference transmitted by the user terminal for the nth OFDM symbol in the preamble of the RCH may be represented as:

where L is the preamble size, i.e. L-4 for type 2 and L-8 for type 3.

Similarly, the controlled reference sent by the access point for the nth OFDM symbol in the preamble of the FCH can be expressed as:

as shown in equations (68) and (69), four eigenmodes are cycled through every 4-symbol period by the (n mod 4) operation of the steering vector. This scheme can be used when the channel changes more quickly and/or early on in a connection when good channel estimates need to be obtained for proper system operation.

In another embodiment, the controlled reference is transmitted for one wideband eigenmode of each TDD frame. For example, a controlled reference of four wideband eigenmodes may be cycled through within four TDD frames. For example, the user terminal may use control vectors for the first, second, third, and fourth TDD frames, respectivelyAndthe particular control vector to be used may be specified by the 2 LSBs of the frame counter value in the BCH message. This scheme can use a shorter preamble portion in the PDU, but may require a longer period of time to obtain a good estimate of the channel.

For both embodiments described above, the controlled reference may be sent on all four eigenmodes used for data transmission, even though less than four eigenmodes are currently used (e.g., because unused eigenmodes are poor and discarded by water-filling). The transmission of the controlled reference over the currently unused eigenmode enables the receiver to determine when the eigenmode has improved to be selected for use.

B. Controlled reference for beam steering

For the beam steering mode, spatial processing at the transmit end is performed with a set of normalized eigenvectors for the dominant wideband eigenmode. The total transfer function with normalized eigenvectors is different from the total transfer function with unnormalized eigenvectors (i.e., with unnormalized eigenvectors)). The controlled reference generated with a set of normalized eigenvectors for all subbands may then be transmitted by the transmitter and used by the receiver to derive matched filter vectors for these subbands of the beam steering mode.

For the uplink, the controlled reference for the beam steering mode can be expressed as:

at the access point, the receive uplink steered reference for the beam steering mode can be expressed as:

to obtain matched filter row vectors for uplink transmission using beam steeringReceived vector of controlled referenceFirst with p^*(k) Multiplication. The result is then integrated over a plurality of received controlled reference symbols to formIs estimated. Thus a vectorIs the conjugate transpose of the estimate.

While operating in beam-steering mode, a user terminal may transmit multiple symbols of a controlled reference, e.g., using normalized eigenvectorsUsing eigenvectors of a dominant wideband eigenmodeAnd possibly one or more symbols using eigenvectors of other wideband eigenmodes. By usingThe generated controlled reference symbols may be used by the access point to derive a matched filter vectorBy usingThe generated controlled reference symbols may be used to obtainAnd then used to derive normalized eigenvectors used for beam steering on the downlinkEigenvectors using other eigenmodesToThe generated controlled reference symbols may be used by the access point to deriveToAnd the singular values of these other eigenmodes. This information is then used by the access point to determine whether to use a spatial multiplexing mode or a beam steering mode for data transmission.

For the downlink, the user terminal may estimate based on the calibrated downlink channel responseDeriving matched filter vectors for beam steering patternsIn particular, the user terminal is selected fromOf (2)Value decomposition to obtainAnd normalized eigenvectors can be derivedThe user terminal may then handleAndmultiply to obtainThen based onExportingAlternatively, the controlled reference may be used by the access point using the normalized eigenvectorTransmitted, the controlled reference being processable by the user terminal in the manner described above to derive

4. Carrier pilot-uplink

The OFDM subband structure depicted herein includes four pilot subbands with indices of-21, -7, and 21. In one embodiment, a carrier pilot is transmitted on four pilot subbands in all OFDM symbols that are not part of the preamble sequence. The carrier pilot may be used by the receiver to track phase changes due to drift of the oscillator at the transmitter and receiver. This may provide improved data demodulation performance.

The carrier pilot comprises four pilot sequences P_c1(n)、P_c2(n)、P_c3(n) and P_c4(n) which are transmitted on the four pilot subbands. The pilot sequence may be defined as:

where n is the index of the OFDM symbol period.

The pilot sequence may be defined based on various data sequences. In one embodiment, the pilot sequence P_c1(n) radicalIn a polynomial G (x) x⁷+x⁴+ x generation, where the initial state is set to all ones, the output bits are mapped to signal values as follows:andthus, for n ═ {1, 2, … 127}, pilot sequence P_c1(n) may be expressed as:

P_c1(n)＝{1，1，1，1，-1，-1，-1，1，-1，-1，-1，-1，1，1，-1，1，-1，-1，1，1，-1，1，1，-1，1，1，1，1，1，1，-1，1，1，1，-1，1，1，-1，-1，1，1，1，-1，1，-1，-1，-1，1，-1，1，-1，-1，1，-1，-1，1，1，1，1，1，-1，-1，1，1，-1，-1，1，-1，1，-1，1，1，-1，-1，-1，1，1，-1，-1，-1，-1，1，-1，-1，1，-1，1，1，1，1，-1，1，-1，1，-1，1，-1，-1，-1，-1，-1，1，-1，1，1，-1，1，-1，1，1，1，-1，-1，1，-1，-1，-1，1，1，1，-1，-1，-1，-1，-1，-1，-1}。

pilot sequence P_c1The values "1" and "-1" in (n) may be mapped to pilot symbols with a particular modulation scheme. For example, by using BPSK, "1" is mapped to "1 + j" and "-1" is mapped to "- (1+ j)". If there are more than 127 OFDM symbols, the pilot sequence is repeated so that P for n > 127_c1(n)＝P_c1(n mod127)

In one embodiment, four pilot sequences are reset for each transmission channel. Thus, on the downlink, the pilot sequence is reset for the first OFDM symbol of the BCH message, again for the first OFDM symbol of the FCCH message, and for the first OFDM symbol sent on the FCH. In another embodiment, the pilot sequence is reset at the beginning of each TDD frame and repeated as needed. For this embodiment, the pilot sequence may be stopped during the pilot sequence portion of the BCH and RCH.

In diversity mode, four pilot sequences are mapped into four subband/antenna pairs as shown in table 29. In particular, P_c1(n) subband-21, P for antenna 1_c2(n) subband-7, P for antenna 2_c3(n) subbands 7, P for antenna 3_c4(n) subbands 21 for antenna 4. Each pilot sequence is then transmitted on an associated subband and antenna.

In spatial multiplexing mode, four pilot sequences are transmitted on the dominant eigenmodes of their respective subbands. The spatial processing of the carrier pilot symbols is similar to the processing performed for the modulation symbols, as described above. In the beam steering mode, four pilot sequences are transmitted on their respective subbands using beam steering. Beam steering of the carrier pilot symbols is also similar to the processing performed for the modulation symbols.

A particular pilot structure is described above for a MIMO WLAN system. Other pilot structures may also be used for the system and are within the scope of the invention.

IX. System operation

Fig. 12A illustrates one particular embodiment of a state diagram 1200 of the operation of a user terminal. The state diagram includes four states — an initial (Init) state 1210, a sleep (Dormant) state 1220, an Access (Access) state 1230, and a Connected (Connected) state 1240. Each state 1210, 1220, 1230 and 1240 is associated with a plurality of sub-states (not shown in FIG. 12A for simplicity).

In an initial state, the user terminal acquires system frequency and timing, and obtains system parameters transmitted on the BCH. In the initial state, the user terminal may perform the following functions:

● system determination-the user terminal determines which carrier frequency to acquire the system.

● frequency/timing acquisition-the user terminal acquires the beacon pilot and adjusts its frequency and timing accordingly.

● parameter acquisition-the user terminal processes the BCH to obtain system parameters associated with the access point from which the downlink signal is received.

After completing the functions required in the initial state, the user terminal transitions to the dormant state.

In the sleep state, the user terminal periodically monitors the BCH for updated system parameters, indications of paging and broadcast messages transmitted on the downlink, and so on. In this state, no radio resource is allocated to the user terminal. In the sleep state, the user terminal may perform the following functions:

● if registration is guaranteed, the user terminal enters an access state according to the registration request.

● if transmitter/receiver calibration is warranted, the user terminal enters the access terminal in accordance with the calibration request.

● the user terminal monitors the BCH for indications of paging and broadcast messages sent on the FCH.

● if the user terminal has data to send on the uplink, it enters the access state according to the resource request.

● the user terminal performs maintenance procedures such as updating system parameters and tracking channels.

● the user terminal may enter a slotted mode of operation to conserve power if that mode is supported by the user terminal.

If the user terminal desires radio resources from the access point for any task, it transitions to an access terminal. For example, the user terminal may transition to an access state in response to a paging or DST indicator sent in a BCH message for registering or requesting calibration, or requesting dedicated resources.

In the access state, the user terminal is in the process of accessing the system. The user terminal may send a short message and/or a request for FCH/RCH resources with the RAHC. Operation on the RACH is described in further detail below. If the subscriber terminal is released by the access point, it transitions back to the dormant state. A user terminal transitions to a connected state if it is allocated resources for the downlink and/or uplink.

In the connected state, the user terminal is allocated FCH/RCH resources, although not necessary for every TDD frame. The user terminal may be actively using the allocated resources or may be idle (while still maintaining the connection) in the connected state. The subscriber terminal remains in the connected state until it is released by the access point or it times out after having not been active for a certain timeout period, in which case it transitions back to the dormant state.

While in the dormant, access or connected state, the subscriber terminal transitions back to the initial state if it is powered off or if the connection is lost.

Fig. 12B illustrates one particular embodiment of a state diagram for the connection state 1240. In this embodiment, the connection state includes three sub-states — a setup sub-state 1260, an open sub-state 1270, and an idle sub-state 1280. The user terminal enters the setup substate after receiving the assignment on the FCCH.

In the setup substate, the user terminal is in the process of setting up a connection, and no data has been exchanged yet. Connection setup may include channel estimation for access points, rate determination, service negotiation, and so on. After entering the setup substate, the ue sets a timer for a specified amount of time. If the timer expires before the subscriber terminal leaves the substate, it transitions to the dormant state. The user terminal transitions to the on sub-state after connection setup is completed.

In the on sub-state, the user terminal and the access point exchange data on the downlink and/or uplink. While in the on sub-state, the user terminal monitors the BCH for system parameters and an indication of paging/broadcast messages. If the BCH message is decoded correctly within a certain number of TDD frames, the UE transitions back to the initial state.

The user terminal also monitors the FCCH for channel allocation, rate control, RCH timing control and power control information. The user terminal uses the BCH beacon pilot and FCH preamble to estimate the received SNR and determine the maximum rate that can be reliably maintained on the FCH.

The FCH and RCH allocation for the user terminal for each TDD frame is given by an information element in the FCCH PDU sent in the current (or possibly previous) TDD frame. For any given TDD frame, no user terminal may be allocated for data transmission on the FCH and/or RCH. For each TDD frame in which a user terminal is not scheduled for data transmission, it does not receive FCH PDUs on the downlink and does not transmit on the uplink.

For each TDD frame in which a user terminal is scheduled, data transmission on the downlink and/or uplink is performed using the rate, transmission mode, and RCH timing offset (for the uplink) indicated in the FCCH allocation (i.e., the FCCH information element addressed to the user terminal). The user terminal receives, demodulates and decodes the FCH PDUs sent to it. The user terminal also transmits an RCH PDU including a preamble sequence and an RCH data rate indicator. The user terminal adjusts the rate used on the RCH in accordance with the rate control information contained in the FCCH allocation. If power control is applied for uplink transmission, the user adjusts its transmit power based on the power control commands included in the FCCH. The data exchange may be bursty, in which case the user terminal enters an idle substate whenever no data is exchangeable. And the user terminal enters an idle sub-state according to the indication of the access point. If the access point does not assign an FCH or RCH to the user terminal within a certain number of TDD frames, the user terminal transitions back to the sleep state and retains its MAC ID.

In the idle substate, both the uplink and downlink are idle. No data is sent in either direction. However, the link is maintained with controlled reference and control messages. In this sub-state, the access point periodically allocates idle PDUs to the user terminal on the RCH and possibly the FCH (not necessarily simultaneously). The user terminal may be able to remain in the connected state indefinitely as long as the access point periodically allocates idle PDUs on the FCH and RCH to maintain the link.

While in the idle substate, the user terminal monitors the BCH. If the BCH message is not decoded correctly within a certain number of TDD frames, the user terminal transitions back to the initial state. The user terminal also monitors the FCCH for channel allocation, rate control, RCH timing control and power control information. The user terminal may also estimate the received SNR and determine the maximum rate supported by the FCH. The user terminal transmits an idle PDU on RCH (when allocated) and sets the RCH request bit in the idle PDU if it has data to transmit. If the access point does not assign an FCH or RCH to the user terminal for a certain number of TDD frames, the user terminal transitions back to the sleep state and retains its MAC ID.

The timeout timer may be set to a particular value after entering any of the three sub-states. If there is no activity while in the sub-state, the timer counts down. While in the set-up, active or idle substate, the terminal may transition back to the dormant state if a timeout timer expires and may transition back to the initial state if the connection is lost. In the active or idle substate, the terminal may also transition back to the dormant state if the connection is released.

Fig. 12A and 12B illustrate one particular embodiment of a state diagram that may be used for a user terminal. Various other state diagrams having fewer, additional, and/or different states and sub-states may also be defined for the system and are within the scope of the invention.

X. random access

In one embodiment, a random access scheme is employed to enable a user terminal to access a MIMO WLAN system. In one embodiment, the random access scheme is a slotted Aloha scheme, whereby the user terminals transmit in randomly selected RACH slots to enable access to the system. The user terminal may send multiple transmissions on the RACH until access is granted or a maximum number of access attempts has been reached. Individual parameters for each RACH transmission may be varied to improve the probability of success, as described below.

Fig. 13 illustrates a RACH timeline, which is divided into RACH slots. The number of RACH slots available within each TDD frame and during the RACH slot duration is a configurable parameter. A maximum of 32 RACH slots may be used per TDD frame. The guard interval between the end of the last RACH slot and the beginning of the BCH PDU of the next TDD frame is also a configurable parameter. These three parameters of RACH may vary from frame to frame and are indicated by the RACH length field, RACH slot size field and RACH guard interval field of the BCH message.

When a user terminal wishes to access the system, it first processes the BCH to obtain the relevant system parameters. The user terminal then transmits a RACH PDU on the RACH. The RACH PDU includes a RACH message containing information needed by the access point to process the access request from the user terminal. For example, the RACH message includes a MAC ID assigned to the user terminal that enables the access point to identify the user terminal. The registered MAC ID (i.e., a specific MAC ID value) may be reserved for unregistered user terminals. In this case, the long ID of the user terminal may be included in the payload field of the RACH message together with the registration MAC ID.

As described below, the RCH PDUs can be transmitted at one of four rates, as listed in table 15. The selected rate is embedded in the preamble of the RACH PDU (as shown in fig. 5C). The RACH PDU also has a variable length of 1, 2, 4 or 8 OFDM symbols (also listed in table 15), which is indicated in the message duration field of the RACH message.

To transmit a RACH PDU, the user terminal first determines the number of RACH slots available for transmission (i.e., the "available" number of RACH slots). This determination is made based on: (1) the number of RACH slots available in the current TDD frame, (2) the duration of each RACH slot, (3) the guard interval, and (4) the length of the RACH PDU to be transmitted. The RACH PDU cannot extend beyond the end of the RACH segment of the TDD frame. Thus, if a RACH PDU is longer than one RACH slot plus a guard interval, the PDU may not be transmitted on one or more of the later available RACH slots. Based on the factors listed above, the number of RACH slots available for transmitting RACH PDUs may be less than the number of available RACH slots. The RACH segment includes a guard interval that is used to prevent uplink transmissions from the user terminals from interfering with the next BCH segment, which is possible for user terminals that do not compensate for their round trip delay.

Next, the user terminal randomly selects one of the available RACH slots to transmit a RACH PDU. Then, the user terminal transmits RACH PDU starting from the selected RACH slot. If the user terminal knows the round trip delay to the access point, it can compensate for this delay by adjusting its timing accordingly.

When the access point receives a RACH PDU, it checks the message using the CRC included in the received RACH message. If the CRC fails, the access point discards the RACH message. If the CRC passes, the access point sets the RACH acknowledgement bit on the BCH in the subsequent TDD frame and sends a RACH acknowledgement on the FCCH in 2 TDD frames. There may be a delay between setting the acknowledgement bit on the BCH and sending the acknowledgement on the FCCH, which is used to compensate for scheduling delays and the like. For example, if the access point receives a message on the RACH, it may set an acknowledgement bit on the BCH and have a delayed response on the FCCH. The acknowledgment bit prevents the user terminal from retrying and enables unsuccessful user terminals to retry quickly except during busy RACH cycles.

If the user terminal is performing registration, it uses the registration MAC ID (e.g., 0x 0001). The access point responds by sending a MAC ID assignment message on the FCH. All other RACH transmission types include the user terminal MAC ID assigned by the system. The access point explicitly acknowledges all correctly received RACH messages by sending an acknowledgement on the FCCH using the MAC ID assigned to the user terminal.

After the user terminal transmits the RACH PDU, it monitors the BCH and FCCH to determine whether its RACH PDU has been received and processed by the access point. The user terminal monitors the BCH to determine if the RACH acknowledgment bit in the BCH message has been set. If the bit is set, this indicates that an acknowledgement for this and/or other user terminals has been sent on the FCCH, and the user terminal further processes the FCCH to obtain the IE type 3 information element containing the acknowledgement. Otherwise, if the RACH acknowledgment bit is not set, the user terminal continues to monitor the BCH or continues its access procedure on the RACH.

FCCH IE type 3 is used to transmit a quick acknowledgement of a successful access attempt. Each acknowledgement information element contains a MAC ID associated with the user terminal for which the acknowledgement is sent. The fast acknowledgement serves to inform the user terminal that its access request has been received but is not associated with the allocation of FCH/RCH resources. Instead, an assignment-based acknowledgment is associated with an FCH/RCH assignment. If the user terminal receives a fast acknowledgement on the FCCH, it transitions to the dormant state. If the user terminal receives an assignment-based acknowledgement, it obtains the scheduling information sent with the acknowledgement and begins using the FCH/RCH assigned in the current TDD frame. (

If the user terminal receives an acknowledgement on the FCCH within a certain number of TDD frames after transmitting the RACH PDU, it continues the access procedure on the RACH. In this case, the user terminal may assume that the access point did not correctly receive the RACH PDU. The user terminal maintains a counter that counts the number of access attempts. The counter is initialized to zero at the first access attempt and then incremented by one for each subsequent access request. If the counter value reaches the maximum number of attempts, the user terminal terminates the access procedure.

For each subsequent access attempt, the user terminal first determines various parameters of this access attempt, including (1) the amount of time to wait before sending a RACH PDU, (2) the RACH slot used for RACH PDU transmission, and (3) the rate of the RACH PDU. To determine the amount of time to wait, the user terminal first determines the maximum amount of time to wait for the next access attempt, which is referred to as a Contention Window (CW). In an embodiment, the contention window (given in units of TDD frames) may grow exponentially for each access attempt (i.e., CW-2^{access_attempt}). Contention window may also be based on accessSome other function of the number of attempts, such as a linear function. The amount of time to wait for the next access attempt is then randomly chosen between zero and CW. The user terminal may wait this amount of time before sending a RACH PDU for the next access attempt.

For the next access attempt, the user terminal decreases the rate of RACH PDUs if the lowest rate is not used for the previous access attempt. The initial rate of the first access attempt may be selected based on the received SNR of the pilot transmitted on the BCH. Failure of the access point to correctly receive the RACH PDU may result in failure to receive the acknowledgement. In this way, the rate of RACH PDUs in the next access attempt is reduced to increase the probability of correct reception by the access point.

After waiting the randomly selected waiting time, the user terminal randomly selects a RACH slot again for transmission of RACH PDUs. The selection of the RACH slot of this access attempt may be performed in a similar manner as the first access attempt described above, except that the RACH parameters (i.e., RACH slot number, slot duration and guard interval) of the current TDD frame (transmitted in the BCH message) are used together with the current RACH PDU length. The RACHPDU is then transmitted in a randomly selected RACH time slot.

The above access procedure continues until any of the following occurs: (1) the user terminal receives an acknowledgement from the access point, or (2) the maximum allowed number of attempts has been reached. For each access attempt, the amount of time to wait before sending a RACH PDU, the RACH slot to use for RACH PDU transmission, and the rate of RACH PDUs may be selected as described above. If an acknowledgement is received, the user terminal operates as indicated by the acknowledgement (i.e., it waits in a sleep state when a fast acknowledgement is received or starts using the FCH/RCH when an allocation-based acknowledgement is received). If the maximum allowed number of access attempts has been reached, the user terminal transitions back to the initial state.

XI Rate, Power and timing control

The access point schedules downlink and uplink transmissions on the FCH and RCH and further controls the rate of all active user terminals. In addition, the access point adjusts the transmit power of a particular active user terminal on the uplink. Various control loops may be maintained to adjust the rate, transmit power, and timing for each active user terminal.

1. Fixed and variable rate services

The access point may support fixed and variable rate services on the FCH and RCH. Fixed rate services may be used for voice, video, and so on. Variable rate services may be used for packet data (e.g., Web browsing).

For fixed rate services on FCH/RCH, a fixed rate is used for the entire connection. The best effort delivery is for FCH and RCH (i.e., no retransmissions). The access point schedules a constant number of FCH/RCH PDUs in each specified time interval to meet the Qos requirements of the service. Depending on the delay requirements, the access point may not need to schedule one FCH/RCH PDU per TDD frame. For fixed rate services, power control is implemented on the RCH instead of the FCH.

For variable rate services over FCH/RCH, the rate used by FCH/RCH can vary with channel conditions. For some synchronous services (e.g., video, audio), the QoS requirements may utilize a minimum rate constraint. For these services, the scheduler at the access point adjusts the FCH/RCH allocation so that a constant rate can be provided. For asynchronous data services (e.g., web browsing, file transfer, etc.), the best effort delivery has a retransmission option. For these services, the rate is the maximum that channel conditions can reliably withstand. The scheduling of FCH/RCH PDUs for user terminals is generally a function of their QoS requirements. Idle PDUs are sent on the FCH/RCH to maintain the link whenever there is no data to send on the downlink/uplink. For variable rate services, closed loop power control is implemented on the FCH rather than the RCH.

2. Rate control

Rate control may be used for variable rate services operating on the FCH and RCH to adapt the rate of the FCH/RCH to changing channel conditions. The rates used by the FCH and RCH can be independently controlled. Furthermore, in spatial multiplexing mode, the rate of each wideband eigenmode of each dedicated transport channel can be independently controlled. Rate control is performed by the access point based on feedback provided by each active user terminal. A scheduler within the access point schedules data transmissions and determines rate assignments for active user terminals.

The maximum rate that can be supported on any link is a function of: (1) the channel response matrix for all data subchannels, (2) the noise level observed by the receiver, (3) the quality of the channel estimate, and possibly other factors. For a TDD system, the channel can be considered reciprocal for the downlink and uplink (after calibration has been performed to account for any differences at the access point and user terminal). However, the reciprocal channel does not mean that the noise floor is the same at the access point and the user terminal. Thus, for a given user terminal, the rates on the FCH and RCH can be independently controlled.

Closed-loop rate control may be used for data transmission on one or more spatial channels. Closed loop rate control may be implemented with one or more loops. The inner loop estimates the channel conditions and selects an appropriate rate for each spatial channel used for data transmission. Channel estimation and rate selection may be performed as described above. The outer loop may be used to estimate the quality of the data transmission received on each spatial channel and to adjust the operation of the inner loop. Data transmission quality may be quantified in terms of Packet Error Rate (PER), decoder metrics, and the like, or a combination thereof. For example, the outer loop may adjust the SNR offset for each spatial channel to achieve a target PER for the spatial channel. The outer loop may also instruct the inner loop to select a lower rate for a spatial channel if excessive packet errors are detected for the spatial channel.

Downlink rate control

Each active user terminal may estimate the downlink channel based on the MIMO pilot sent on the BCH in each TDD frame. The access point may also transmit a controlled reference in the FCH PDU sent to a particular user terminal. By using the MIMO pilot on the BCH and/or the steered reference on the FCH, the user terminal can estimate the received SNR and determine the maximum rate that can be supported on the FCH. If the user terminal is operating in spatial multiplexing mode, a maximum rate can be determined for each wideband eigenmode. Each user terminal may send back to the access point the maximum rate supported by each wideband eigenmode (for spatial multiplexing mode), the maximum rate supported by the primary wideband eigenmode (for beam-steering mode), or the maximum rate supported by the MIMO channel (for diversity mode) in the FCH rate indicator field of the RCH PDU. These rates may be mapped to received SNRs, which are then used to perform the above-described water-filling procedure. Alternatively, the user terminal may send back sufficient information (e.g., received SNR) to enable the access point to determine the maximum rate supported by the downlink.

The determination of whether to use diversity, beam steering or spatial multiplexing mode is made based on feedback from the user terminal. As the separation between the control vectors increases, the number of selected wideband eigenmodes also increases.

Fig. 14A illustrates a process for controlling the rate of downlink transmissions for a user terminal. A BCH PDU is transmitted in the first segment of each TDD frame and includes beacons and MIMO pilots that can be used by user terminals to estimate and track the channel. The controlled reference may also be transmitted in a preamble of an FCH PDU transmitted to the user terminal. The user terminal estimates the channel based on MIMO and/or a controlled reference and determines the maximum rate that can be supported by the downlink. If the user is operating in spatial multiplexing mode, one rate is supported for each wideband eigenmode. The user terminal then transmits the rate indicator for the FCH in the FCH rate indicator field of the RCH PDU it transmits to the access point.

The scheduler uses the maximum rate that the downlink can support for each active user terminal to schedule downlink data transmission in subsequent TDD frames. The rate and other channel allocation information for the user terminal is reflected in the information element sent on the FCCH. The rate assigned to one user terminal may affect the scheduling of other user terminals. The minimum delay between the user-determined rate and its use is about a single TDD frame.

By using the Gram-Schmidt ordering procedure, the access point can accurately determine the maximum rate supported on FCH directly from the RCH preamble. This can then greatly simplify rate control.

Uplink rate control

Each user terminal transmits a controlled reference on the RACH during system access and transmits the controlled reference on the RCH after being allocated to the FCH/RCH resources. The access point may estimate the received SNR for each wideband eigenmode based on a controlled reference on the RCH and determine the maximum rate supported by each wideband eigenmode. First, the access point may not have good channel estimates to allow reliable operation at or near the maximum rate supported by each wideband eigenmode. To improve reliability, the initial rate used on the FCH/RCH may be significantly lower than the maximum supported rate. The access point may integrate the controlled reference over multiple TDD frames to obtain an improved channel estimate. As the channel estimate increases, the rate may also be increased.

Fig. 14B illustrates a process for controlling the rate of uplink transmissions for a user terminal. In scheduling for uplink transmission, the user terminal sends an RCH PDU that includes a reference that the access point uses to determine the maximum rate on the uplink. The scheduler then schedules uplink data transmission in subsequent TDD frames using the maximum rate that the uplink can support for each active user terminal. The rate and other channel allocation information for the user terminal is reflected in an information element sent on the FCCH. The minimum delay between the access point determining the rate and its use is about a single TDD frame.

3. Power control

For fixed rate services, power control may be used for uplink transmissions on the RCH (rather than rate control). For fixed rate services, the rate is negotiated at call setup and remains fixed during the connection. Some fixed rate services may be associated with limited mobility requirements. In one embodiment, however, power control is implemented for the uplink to combat interference between user terminals, but not for the downlink.

A power control mechanism is used to control the uplink transmit power of each active user terminal such that the received SNR at the access point is maintained at a level that achieves a desired quality of service. This level is commonly referred to as the target received SNR, operating point, or set point. For a moving user terminal, the propagation loss is likely to vary as the user terminal moves. The power control mechanism tracks changes in the channel to keep the received SNR near the setpoint.

The power control mechanism may be implemented with two power control loops-an inner loop and an outer loop. The inner loop adjusts the transmit power of the user terminal so that the received SNR at the access point is maintained near the setpoint. The outer loop adjusts the set point to achieve a particular level of performance, which is quantified by a particular Frame Error Rate (FER) (e.g., 1% FER), Packet Error Rate (PER), block error rate (BLER), Message Error Rate (MER), or some other metric.

Fig. 15 illustrates an operation of internal power control of a user terminal. After the user terminals are assigned to the FCH/RCH, the access point estimates the received SNR on the RCH and compares it to the setpoint. The initial power to be used by the user terminal may be determined at call setup and is typically around its maximum transmit power level. For each frame interval, the access point can instruct the user terminal to reduce its transmit power by a particular amount (e.g., 1dB) in the FCCH information element sent to the user terminal if the received SNR exceeds a particular positive margin δ. Conversely, if the received SNR is lower than the threshold by a margin δ, the access point can instruct the user terminal to increase its transmit power by the specified amount. If the received SNR is within acceptable setpoint limits, the access point will not request a change in transmit power to the user terminal. The uplink transmit power is given as the initial transmit power level plus all power adjustments received from the access point.

The initial set point used at the access point is set to achieve a particular level of performance. The setpoint is adjusted by the outer loop based on the FER or PER of the RCH. For example, if no frame/packet errors occur over a particular period of time, the setpoint may be lowered by a first amount (e.g., 0.1 dB). If the average FER is exceeded due to the occurrence of one or more frame/packet errors, the setpoint may be increased by a second amount (e.g., 1 dB). The set point, hysteresis margin, and outer loop operation are specific to the power control design used by the system.

4. Timing control

Timing control is preferably used in TDD-based frame structures where the downlink and uplink share the same frequency band in a time division duplex manner. The user terminals may be spread throughout the system and thus associated with different propagation delays to the access point. To maximize efficiency on the uplink, the timing of uplink transmissions on the RCH and RACH from each user terminal may be adjusted to compensate for its propagation delay. This would then ensure that uplink transmissions from different user terminals arrive at the access point within a certain time window and do not interfere with each other on the uplink, or for downlink transmissions.

Fig. 16 illustrates a process for adjusting uplink timing of a user terminal. First, the user terminal transmits a RACH PDU on the uplink to enable access to the system. The access point derives an initial estimate of a round trip delay (TDD) associated with the user terminal. The round trip delay may be estimated based on: (1) a sliding correlator used by the access point to determine a transmission start point, and (2) a slot ID included in a RACH PDU transmitted by the user terminal. The access point then calculates an initial timing advance for the user terminal based on the initial RTD estimate. The initial timing advance is sent to the user terminal prior to its transmission on the RCH. The initial timing advance may be sent in a message on the FCH, in a field of the FCCH information element, or by some other means.

The user terminal receives an initial timing advance from the access point and then uses the timing advance on all subsequent uplink transmissions on the RCH and RACH. If the user terminal is allocated FCH/RCH resources, its timing advance can be adjusted by a command sent by the access point in the RCH timing adjustment field of the FCCH information element. The user terminal can then adjust its uplink transmission on the RCH based on the current timing advance, which is equal to the initial timing advance plus all timing adjustments sent to the user terminal by the access point.

The portions of the MIMO WLAN system and various techniques described herein may be time slotted by various means. For example, the processing at the access point and the user terminal may be implemented in hardware, software, or a combination thereof. For a hardware implementation, the processing may be implemented within the following components: one or more Application Specific Integrated Circuits (ASICs), Digital Signal Processors (DSPs), Digital Signal Processing Devices (DSPDs), Programmable Logic Devices (PLDs), Field Programmable Gate Arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, other electronic units designed to perform the functions described herein, or a combination thereof.

For a software implementation, the processes may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The software codes may be stored in a memory unit (e.g., memory 732 or 782 in fig. 7) and executed by a processor (e.g., controller 730 or 780). The memory unit may be implemented within the processor or external to the processor, in which case it can be communicatively coupled to the processor via various means as is known in the art.

The headings included herein facilitate indexing and help locate particular chapters. These headings are not intended to limit the scope of the concepts described therein under, which concepts may be employed in other sections throughout the specification.

The previous description of the preferred embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims (22)

1. A method of exchanging data in a wireless Time Division Duplex (TDD) multiple-input multiple-output (MIMO) communication system, comprising:

receiving a pilot from a user terminal on an uplink;

deriving at least one control vector for a downlink of a user terminal based on the received pilots; and

spatially processing a first data transmission sent to a user terminal on a downlink with the at least one control vector.

2. The method of claim 1, wherein a single control vector is derived for a downlink of a user terminal, wherein the first data transmission is beam-controlled spatially processed with the single control vector for sending the first data transmission via a single spatial channel of the downlink.

3. The method of claim 1, wherein a plurality of control vectors are derived for a downlink of a user terminal, wherein the plurality of control vectors spatially processes the first data transmission for sending the first data transmission via a plurality of spatial channels of the downlink.

4. The method of claim 1, further comprising:

deriving a matched filter for the uplink of the user terminal based on the received pilot; and

and performing matched filtering on a second data transmission received on the uplink from the user terminal by using the matched filter.

5. The method of claim 4, wherein the matched filter comprises at least one eigenvector for at least one eigenmode of the uplink, wherein the at least one eigenvector for the uplink is equivalent to the at least one steering vector for the downlink.

6. An apparatus in a wireless Time Division Duplex (TDD) multiple-input multiple-output (MIMO) communication system, comprising:

means for receiving a pilot on the uplink from the user terminal;

means for deriving at least one control vector for a downlink of a user terminal based on the received pilots; and

means for spatially processing a first data transmission sent to a user terminal on a downlink with the at least one control vector.

7. The apparatus of claim 6, wherein a single control vector is derived for a downlink of a user terminal, wherein the first data transmission is beam-controlled spatially processed with the single control vector for sending the first data transmission via a single spatial channel of the downlink.

8. The apparatus of claim 6, wherein a plurality of control vectors are derived for a downlink of a user terminal, wherein the plurality of control vectors spatially processes the first data transmission for sending the first data transmission via a plurality of spatial channels of the downlink.

9. The apparatus of claim 6, further comprising:

means for deriving a matched filter for the uplink of the user terminal based on the received pilot; and

means for matched filtering a second data transmission received on the uplink from the user terminal with the matched filter.

10. The apparatus of claim 9, wherein the matched filter comprises at least one eigenvector for at least one eigenmode for an uplink, wherein the at least one eigenvector for the uplink is equivalent to the at least one steering vector for the downlink.

11. A method of transmitting and receiving pilots in a wireless multiple-input multiple-output (MIMO) communication system, comprising:

transmitting a MIMO pilot from a plurality of antennas and over a first communication link, wherein the MIMO pilot comprises a plurality of pilot transmissions from the plurality of antennas, wherein the pilot transmission from each antenna is identifiable by a communication entity receiving the MIMO pilot; and

receiving a steered pilot from the communication entity via at least one eigenmode of a second communication link, wherein the steered pilot is generated based on the MIMO pilot.

12. The method of claim 11, wherein the first communication link is an uplink, the second communication link is a downlink, and the communication entity is a user terminal.

13. The method of claim 11, wherein the first communication link is a downlink, the second communication link is an uplink, and the communication entity is an access point.

14. The method of claim 11, wherein the pilot transmission from each antenna is associated with a different orthogonal code.

15. The method of claim 11, wherein the steered pilot is received via a single eigenmode of a second communication link and transmitted at full transmit power from multiple antennas at a communication entity.

16. The method of claim 11, wherein the steered pilot is received via a plurality of eigenmodes of a second communication link.

17. The method of claim 11, wherein the steered pilot is transmitted by a communication entity for a system configurable time duration.

18. An apparatus in a wireless multiple-input multiple-output (MIMO) communication system, comprising:

means for transmitting a MIMO pilot from a plurality of antennas and over a first communication link, wherein the MIMO pilot comprises a plurality of pilot transmissions transmitted from the plurality of antennas, wherein the pilot transmission from each antenna is identifiable by a communication entity receiving the MIMO pilot; and

means for receiving a steered pilot from a communication entity via at least one eigenmode of a second communication link, the steered pilot generated based on a MIMO pilot.

19. The apparatus of claim 18, wherein pilot transmission from each antenna is associated with a different orthogonal code.

20. The apparatus of claim 18, wherein the steered pilot is received via a single eigenmode of a second communication link and is transmitted from multiple antennas at a communication entity at full transmit power.

21. The apparatus of claim 18, wherein the steered pilot is received via a plurality of eigenmodes of a second communication link.

22. A processor that executes instructions for exchanging data in a wireless Time Division Duplex (TDD) multiple-input multiple-output (MIMO) communication system, comprising:

means for receiving a pilot on the uplink from the user terminal;

means for deriving at least one control vector for a downlink of a user terminal based on the received pilots; and

means for spatially processing a first data transmission sent to a user terminal on a downlink with the at least one control vector.