HK1071932B - Method for open loop tracking gps signals - Google Patents

Description

Method for open loop tracking of GPS signals

Technical Field

The present invention relates to global positioning systems, and more particularly, to a method and apparatus for receiving and tracking satellite signals in a highly sensitive and accurate receiver.

Background

Conventional Global Positioning System (GPS) receivers acquire, track, and demodulate synchronized signals transmitted from multiple GPS satellites in order to compute the position of the receiver.

Each GPS satellite transmits a direct sequence spread spectrum signal that is modulated by a repeating code represented by a sequence of binary phase states called chips. The particular chip sequence corresponding to a code is referred to as a pseudo-random or pseudo-noise (PN) sequence. Each GPS satellite broadcasts a signal with a unique code, i.e., a unique PN sequence. The american civil code, known as the C/a code, comes from a family of codes known as Gold codes. A C/a code consists of 1023 chips in a 1 millisecond (ms) frame period. Thus, one code is repeated every millisecond in the GPS signal.

In addition, 50 baud Binary Phase Shift Keying (BPSK) data is superimposed on the pseudo-noise sequence. The bit boundaries of BPSK data are aligned with the start of the PN sequence so that there are 20 complete PN sequences in each data bit period (i.e., 20 milliseconds). BPSK data contains ephemeris data representing the position of the satellites and the clock timing of the GPS signals.

Thus, an ideally received GPS signal without additional noise and interference is in the form of:

S(t)＝AP(t-τ)d(t-τ)cos[2πf₀(t-τ)+θ_t] (1)

wherein a is the magnitude of the signal; p (t) is the pseudo-noise modulation at time t, which has a value of 1 or-1; d (t) is BPSK data, also having a value of 1 or-1; f. of₀Is the nominal carrier frequency in hertz; theta_tIs a phase angle, the phase angle θ being due to Doppler (Doppler) effect, phase noise, etc_tMay change gradually; and τ is the delay to be estimated by the receiver. In practice, the size a generally changes gradually over time. The fact that the time variable t may be stretched or shrunk in time due to the doppler effect can be combined with the quantity f₀τ and θ_tIn (1).

The main task of a GPS receiver is to measure the parameters f of the received GPS signals originating from a plurality of GPS satellites₀τ and θ_tIn order to calculate the position of the receiver. The measurement of the parameter τ is generally performed in a continuous manner using a circuit called a "PN tracking circuit". As part of this measurement process, the carrier frequency f is measured in a circuit called a "carrier tracking circuit₀And residual phase angle theta_tMeasurements and compensation are performed.Traditionally, PN and carrier tracking circuits take the form of a feedback circuit known as a closed loop tracking loop.

In general, conventional tracking loops utilize feedback circuits that attempt to produce a locally generated reference signal that replicates the received signal, i.e., aligns the reference signal with the received signal. Fig. 1 shows a block diagram of a closed loop tracking method of one embodiment of the prior art. Signal evaluator 510 generates a correction signal indicating how to correct reference signal 514. The correction signal is fed back over line 515 to reduce the difference between the received signal and the reference signal. Since this is done in a continuous manner, the process can be viewed as a tandem or "closed loop" type of process.

For example, in a PN tracking loop, a local PN generator replicates the PN sequence in the received GPS signal. The locally generated PN sequence is compared in the circuit with the PN sequence in the received GPS signal to produce an output having a magnitude corresponding to the degree of synchronization between the local reference sequence and the received sequence. When the two sequences are synchronized, the size of the output reaches a maximum; when the two sequences are offset from each other, the output decreases in size. Thus, the change in the size of the output is used to adjust the timing of the local PN generator to increase the size of the subsequent output. This is a servo or feedback type technique. Similar techniques are also used to track the phase and frequency of the carrier of the GPS signal.

Closed loop tracking is suitably used when parameters in the received GPS signal, such as size and delay, change gradually over time, in which case the parameters of the GPS signal may be averaged over a long period of time. However, in many cases, the parameters of the signal change rapidly in an unpredictable manner. For example, when a GPS receiver enters a building, the received signal can fluctuate rapidly in magnitude, delay, and carrier due to unpredictable superposition between the direct signal and the reflected signal reflected off walls, large objects, and the like. Similarly, when the GPS receiver is operated in an urban environment with large buildings, so-called "urban canyons", the signal level fluctuates rapidly if the GPS receiver moves rapidly, such as when the receiver and its antenna move with the car. Also, the cause of the problem is the unpredictable interference between the direct and reflected signals. There is typically no direct signal from the GPS satellites and all received signals are reflected signals. This situation further exacerbates the speed and extent of fluctuations in the parameters of the received signal.

Closed loop tracking loops perform poorly when the received GPS signal fluctuates rapidly. In a closed loop tracking system, a previous measurement of a parameter p of the received signal and a correction signal generally associated with an estimation error in the previous measurement(s) are used to produce a current measurement of the parameter, namely:

wherein the content of the first and second substances,is the measured value of the parameter p at time n; e (n) is the estimation error of the previous measurement, and h is a linear or non-linear function. In general, n plus 1 may correspond to one or more PN frame periods.

In some formulas, the argument of function h contains a number of previous measurementsMagnitude ofAnd errors e (n), e (n-1), e (n-2), however, for simplicity and for illustrative purposes, we use simplified equation (2).

Also, many designers refer to the measured value of parameter p as an "estimate" because the measurement process includes noise that limits the accuracy of such measurements. In the present invention, the term "measurement" is synonymous with estimation of a parameter when referring to the parameter.

To characterize the closed loop tracking loop, consider the linear form of equation (2), namely:

where k is a constant or a gradually changing parameter. If the error e (n) is expressed asWhere p (n) is the exact value of parameter p at time n, then equation (3) can be rewritten as:

this equation is essentially a single pole filter type response with a time constant of 1/k. Thus, it is possible to prevent the occurrence of,the initial error in (1) is reduced by approximately exp (-1) ≈ 0.37 over a period of time equal to 1/k iterations.

In systems with k close to but less than 1, only the most recent measurements contribute to the correction; thus, the measurements can keep up with rapid changes in the received signal. However, when the received signal is weak, the system performs poorly because there is only a small number of averages of previous measurements. On the other hand, in a system with a small k value, many previous measurement values contribute to the correction, and thus the system performs well even when the received signal is weak. However, when the received signal is weak, the system performs poorly when the signal changes rapidly.

Therefore, closed loop tracking loops have inevitable limitations. A "loop out of lock" condition occurs when the parameters of the received signal change too quickly for the closed loop tracking loop to keep up with such changes quickly enough to produce accurate results. A "wide" loop with a strong (k close to 1) correction signal may keep up with the rapid changes in the received signal, but when the signal is weak, the loop behaves poorly because it cannot distinguish the received noise and interference from the desired signal. On the other hand, a "narrow" (small k) loop with a weak correction signal may often not be able to keep up with a rapidly changing signal.

Various approaches may be devised to adapt the loop design to different conditions and thus overcome some of the limitations just discussed. However, these "fit" approaches inevitably fail or perform poorly when signal dynamics and strengths vary sufficiently and are unpredictable.

Disclosure of Invention

Methods and apparatus for open loop tracking of Global Positioning System (GPS) signals are described herein.

In one aspect of the invention, an exemplary method comprises: (A) generating a set of at least three indicators based on processing a portion of a satellite positioning system signal received by a receiver; and (B) calculating a measured value of the parameter from the interpolation of the set of indicators. Each indicator represents the probability that a parameter of the signal (the time of arrival of the GPS signal or the carrier frequency of the GPS signal) is a predetermined value. Serial correlation, matched filtering, fast fourier transform, and fast convolution are used in the methods for generating these indicators.

In another aspect of the invention, an exemplary method comprises: despreading a portion of the satellite positioning system signals to produce despread data; and performing an open loop frequency measurement operation on the despread data to produce a frequency measurement of the signal. In some embodiments of the invention, the open loop frequency measurement operation comprises: (A) performing a non-linear operation on the despread data to produce first data; (B) performing a spectral analysis operation on the first data to generate spectral data; (C) the frequency measurement of the signal is calculated from the position of the peak in the spectral data. In some embodiments of the invention, the open loop frequency measurement operation comprises: (A) performing a non-linear operation on the despread data to produce first data; (B) averaging the first data to generate averaged data; (C) calculating the angle of the average data; and (D) calculating a frequency measurement of the signal using the angle. In some embodiments of the invention, the non-linear operation comprises a squaring operation. In certain other embodiments of the present invention, the non-linear operations include conjugate, delay, and multiply operations.

The invention includes apparatus for performing these methods, including a GPS receiver for performing these methods and a machine-readable medium for execution in the GPS receiver for causing the receiver to perform these methods.

Other features of the present invention will be apparent from the accompanying drawings and from the detailed description that follows.

Drawings

The present invention is illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings and in which like reference numerals refer to similar elements.

Fig. 1 shows a block diagram of a closed loop tracking method according to one embodiment of the prior art.

Fig. 2 shows a simple block diagram of a closed loop pseudo noise tracking method according to one embodiment of the prior art.

Figure 3 shows a detailed closed loop pseudo noise tracking method according to one embodiment of the prior art.

Fig. 4 illustrates signal waveforms of the processing stages in fig. 3.

Fig. 5 shows a block diagram of an open loop tracking method according to an embodiment of the invention.

Fig. 6 shows a simple block diagram of an open loop pseudo noise tracking method according to an embodiment of the invention.

Fig. 7 illustrates interpolation of correlation indicators.

Figure 8 illustrates a detailed open loop pseudo noise tracking method according to one embodiment of the present invention.

Fig. 9 shows a block diagram of a closed loop carrier tracking method according to one embodiment of the prior art.

Fig. 10 shows a detailed closed loop carrier tracking method according to one embodiment of the prior art.

Fig. 11 illustrates a multiplier-based open loop carrier frequency measurement method according to an embodiment of the present invention.

Figure 12 shows another open loop carrier frequency measurement method based on a frequency discriminator according to an embodiment of the invention.

Fig. 13 shows another open-loop carrier frequency measurement method based on a block phase estimator according to an embodiment of the invention.

Fig. 14 shows another open loop carrier frequency measurement method based on a channelization filter in accordance with one embodiment of the present invention.

FIG. 15 shows a flow diagram for open loop tracking of satellite positioning system signals to determine the position of a receiver according to one embodiment of the invention.

Fig. 16 illustrates a flow diagram for open loop carrier frequency tracking according to some embodiments of the invention.

Fig. 17 shows a block diagram representation of a receiver implementing the open loop tracking method according to the present invention.

Fig. 18 shows a block diagram representation of a remote satellite positioning system implementing an open loop tracking method according to the present invention.

Detailed Description

The present invention will be described with reference to various details set forth below, and the accompanying drawings will illustrate the present invention. The following description and drawings are illustrative of the invention and are not to be construed as limiting the invention. Numerous specific details are described to provide a thorough understanding of the present invention. However, in certain instances, well-known and conventional details are not described in order not to obscure the present invention in detail.

At least one embodiment of the present invention seeks to measure parameters of a GPS signal using an open loop tracking method. In an open loop tracking loop, the measurement of parameter p at time n +1 can be expressed as follows:

wherein u is_i(m) is the measured value of the quantity related to p at the instant m. Thus, the current measurement of p is a function of the determined number of related measurements. In the formula (5), the first and second groups,to pair(orEtc.) have no direct correlation.

In the case of open loop tracking, previous measurements may be used, e.g.OrThe calculation of the subsequent measurement value is effected in a form other than formula (2). Generally, a prior measurement is used to give a constrained range or "window" in which a subsequent measurement may be taken. The center position of the window and/or the width of the current window is typically adjusted as a function of the previous measurement or measurements.

Several open loop measurements can be combined to improve the accuracy of the measurement. For example, averaging together successive open loop measurements, or forming an intermediate of the measurements, or fitting the measurements to some type of curve (e.g., a linear fit) may be performed to obtain improved measurements. In the curve fitting process, noise and interference can be filtered out, while still being able to follow rapid changes in the received signal.

In general, a GPS receiver has two modes of operation, namely (1) an acquisition mode, and (2) a tracking mode. In acquisition mode, the GPS receiver seeks to roughly align the locally generated reference PN signal with the chip level of the received signal in order to detect the presence of and roughly synchronize with the GPS signal. In tracking mode, the GPS receiver tries to accurately synchronize the reference PN signal with the GPS signal so that the timing difference between the signals is much less than one chip period.

In acquisition mode, the nominal rate of the locally generated PN chips is set to be slightly different from the nominal rate of the received PN chips, so that the reference code "slips" relative to the received signal. A correlation process compares the two signals with each other. When the two signals are aligned within one chip, a large correlation output is produced. This causes the receiver to enter a tracking mode in which the nominal rate of the locally generated PN chips is set to the rate of the received PN chips. Typically, capture is accelerated by performing the correlation operations in parallel using multiple circuits.

Fig. 2 shows a simplified block diagram of a closed loop pseudo noise tracking method according to one embodiment of the prior art. In fig. 2, PN discriminator 520 compares the PN sequence in baseband input 522 obtained from the GPS signal with a reference PN sequence generated by local PN reference generator 524. PN discriminator 520 includes correlation indicator generators 531 and 532, which are typically serial correlators. The correlator is configured such that the closer the two input PN sequences are aligned, the greater the output produced by the correlator. The output of the correlator is thus an indicator of how closely the two input sequences are aligned. The result of subtractor 535 indicates whether the delayed reference PN sequence is better aligned with the PN sequence in the baseband input than with the reference PN sequence on line 537. The correlation signal from subtractor 535 adjusts the timing of the reference PN sequence so that both indicator generators produce the same output. When the PN sequences on lines 537 and 538 are both identically aligned with the PN sequences in the baseband input, the time of occurrence of the received PN sequence is midway between the time of occurrence of the PN sequences on lines 537 and 538.

Subtracting circuit 535 generates a correlation signal that is a signed number. The sign of this number indicates that the direction of the PN reference should be adjusted (earlier or later) to achieve more alignment. The magnitude of this difference is proportional to the degree of current misalignment.

Figure 3 illustrates a detailed closed loop pseudo noise tracking method of one embodiment of the prior art. In fig. 3, the baseband input on line 401 contains in-phase (I) and quadrature (Q) components, which are typically converted from the input signal by an I/Q down-converter. Typically, the signal on line 401 is a signal toolSuch as waveform 430 illustrated in fig. 4. The oscillator 419 is controlled by a correlation signal fed back from line 421 to align the PN sequence generated by the local generator 403 with the PN sequence in the baseband input on line 401. The PN discriminator 402, which includes two serial correlators and a subtractor 417, generates a basic error signal. Each of the serial correlators performs multiplication, accumulation and nonlinear detection operations to produce a correlation output. The early (E) signal on line 425 having waveform 431 illustrated in fig. 4 is fed to a first correlator comprising multiplier 404, accumulator 407 and nonlinear detector 410 to produce an early correlation output. The late (L) signal on line 428 is fed as an output from delays 413 and 414 to a second correlator comprising multiplier 405, accumulator 408 and non-linear detector 411 to produce a late correlation output. The late signal has the waveform 433 shown in fig. 4, which illustrates that each of the delays 413 and 414 causes a delay t_d. Thus, the on-time signal on line 426 having waveform 432 shown in FIG. 4 occurs midway between the occurrence of the early and late signals. Subtractor 417 subtracts the late output from the early output to form a base error signal, which is further filtered by accumulator 418 and loop filter 415.

In general, the non-linear detectors 410 and 411 perform absolute value operations or absolute value square operations; accumulators 407 and 408 accumulate over a time period equal to a number of PN frame periods. Accumulator 418 reduces the data rate provided to loop filter 415 and is typically used when loop filter 415 is implemented with a microprocessor.

When the received signal is closely aligned with the early signal, the early output is much larger than the later output. Thus, the early output minus the late output is positive and the positive correlation signal increases the frequency of oscillator 419. Thus, the clock provided by the oscillator to the reference generator occurs earlier in time, i.e., all waveforms 431, 432, 433 in FIG. 4 are shifted to the left to more closely align the late outputs to the received signal. The opposite occurs when the received signal is aligned more closely with the late signal than with the early signal. Thus, when the early signal and the late signal are both equally well aligned with the received signal, the loop reaches an equilibrium state, i.e. the time of occurrence of the received signal is the same as the time of occurrence of the on-time signal, midway between the time of occurrence of the early signal and the late signal. Generally, to maintain tracking, the on-time signal must occur close to the time of occurrence of the input signal (e.g., within about one chip).

When the loop reaches an equilibrium state, the received signal is aligned with the on-time signal on line 426 of fig. 3. The on-time output on line 427 contains the despread signal, i.e., the baseband signal with the spurious noise removed. The punctual output is mainly used to retrieve ephemeris data (BPSK data d in equation (1)) modulated on the GPS signal. The phase of the oscillator 419 and the code position (or code phase) of the PN reference generator 403 are often combined to produce a time of arrival (or pseudorange) measurement of the received signal.

Fig. 5 shows a block diagram of an open loop tracking method according to one embodiment of the invention, in which a signal difference quantizer 560 quantizes the parameter difference between the received signal 562 and a local reference signal 564 to determine a measure of a parameter of the received signal. Thus, measurements of the parameters can be obtained without adjusting the local reference signal to obtain the same parameters as the received signal.

Fig. 6 shows a detailed block diagram of an open loop pseudo noise tracking method according to an embodiment of the invention. The reference PN sequence generated by generator 574 is delayed by a set of delays 592, 593,. and 599 to establish a set of reference sequences, wherein each reference sequence corresponds to a different time of occurrence. The shuffled reference sequence is compared with the PN sequence in the baseband input 572 by a set of correlation indicator generators 581, 582, 583. Each indicator is a digital value that shows how well the corresponding delayed reference PN sequence is aligned to the PN sequence in the baseband input. Thus, each indicator can be considered to be indicative of a probability or likelihood that the time of occurrence of the corresponding delayed reference sequence is the time of occurrence of a PN sequence in the received signal. The peak locator 580 uses an interpolation scheme to accurately determine the amount of delay required to synchronize the reference PN sequence with the PN sequence in the baseband input. Thus, the time of arrival is determined from measurements of a set of correlator indicators without having to repeatedly synchronize the reference sequence to the received signal. No servo method is required.

Another way to observe fig. 6 is that the total delay from the reference PN generator to a given correlator indicator is proportional to the time offset between the PN reference and the input signal. Thus, the open loop process performs a set of tests that verify that the input signal parameter is one of several assumed or predetermined values. Then, a set of numbers or "indicators" is given, indicating that the parameter is equal to these predetermined values. The indicator may be interpreted as a probability or likelihood of agreement. Based on the values of these indicators, one indicator may then undergo an interpolation process, as in 580, to further refine the measured values of the parameters.

Fig. 7 illustrates interpolation of correlation indicators. The vertical lines at positions 450 to 458 indicate the magnitude of the outputs of 9 correlation operations, where the PN occurrences of successive correlator outputs differ by one-half chip from each other. The dotted line represents the interpolation between the indicator values provided by the correlator; "X" denotes the position of the peak found using the interpolation scheme. It will be appreciated that the interpolation scheme is not a common scheme in which the selection of the correlation indicator having the largest value simplifies the peak finding operation.

Figure 8 illustrates a detailed open loop pseudo noise tracking method according to one embodiment of the present invention. It is seen that retarders 108, 109,. and 110 correspond to retarders 592, 593,. and 599 of fig. 6; the correlators 111, 112, 113, 114 correspond to the correlation indicator generators 581, 582, 583,. The peak detector and time interpolator 115 corresponds to the peak locator 580. The correlator 111, which implements a correlation indicator generator, comprises a multiplier 105, an accumulator 106, a non-linear detector 107 and an accumulator 120. In general, the other correlators (e.g., correlator 113) have the same structure as correlator 111.

After receiving the time information from the peak detector and time interpolator 115, the microprocessor 116 may optionally adjust the oscillator 102 so that the PN sequence in the baseband is closely synchronized with one of the reference PN sequences. Although this feedback on line 117 may be considered part of the closed loop operation, the feedback is not necessary for accurate pseudorange measurements, as accurate measurements are made before this information is fed back on line 117 to adjust the timing of the oscillator. The method of fig. 8 may also be used for the acquisition mode. 115 may be performed by the microprocessor 116.

The output of the peak detector and time interpolator 115 represents the time of arrival of a single GPS signal from one satellite. Circuitry 101-115 may be viewed as a single "channel" allocated to processing the signal. Other channels (typically 8-12 channels) from other GPS signals assigned may be provided simultaneously to the microprocessor 116 to determine a measurement of the position of the receiver. Generally, simultaneous measurements of time of arrival from at least three GPS satellites are required in order to determine a two-dimensional position (four signals are required for three-dimensional position determination).

The successive measurements of the received position found using the open loop tracking method according to the present invention may be combined using various averaging or tracking algorithms to obtain improved position measurements over time. The successive measurements may be combined using a variety of algorithms including Least Mean Square (LMS) filtering and Kalman filtering, a form of LMS filtering. Other algorithms, such as median filtering type algorithms, that can discard measurements that appear very poor are more suitable when the received signal is very weak.

The combination of the position estimation data is called the averaging of the position fields. Better performance may be obtained by combining various time of arrival (TOA) or pseudorange measurements themselves from several consecutive time periods. This is called averaging of the measurement fields. The position is then computed by combining pseudoranges computed simultaneously from different GPS signals and multiple time instants. Various popular algorithms, such as LMS and kalman filtering algorithms, may be used for averaging of the measurement domain.

In some cases, prediction of the current value of the parameter from a previous measurement may indicate that the current measurement is very poor or unrealistic. In this case, the current measurement value may be deleted and such prediction may be used. Thus, it will be appreciated that combining a set of measurements may include operations to delete a current measurement and use an earlier measurement.

Fig. 9 shows a block diagram of a closed loop carrier tracking method according to one embodiment of the prior art. In fig. 9, a carrier discriminator 550 compares a received signal 542 with a reference signal generated by an oscillator 544. Carrier discriminator 550 includes a phase discriminator 551 and a frequency discriminator 552. The phase detector 551 generates a correction signal to reduce the phase difference between the received signal and the local carrier signal. Similarly, the frequency discriminator 552 generates a correction signal to reduce the frequency difference between the received signal and the local carrier signal. Since frequency is a time derivative of phase, the phase detector and frequency detector correction signals may be combined to adjust the oscillator 544 to generate a local carrier signal having the same phase and frequency as the received signal.

Fig. 10 shows a detailed closed loop carrier tracking method according to one embodiment of the prior art. The carrier tracking circuit according to the method in fig. 10 is generally combined with the PN tracking circuit according to the method of fig. 3. Generally, the signal on line 401 in fig. 10 is input as a baseband signal to line 401 in fig. 3 for tracking the PN sequence, and the on-time output on line 427 in fig. 3 is sent to line 427 in fig. 10 for generating the carrier correction signal.

To provide the baseband signal on line 401 in fig. 10, the Intermediate Frequency (IF) signal received on line 200 is digitized by an analog-to-digital (a/D) converter 201 and then passed through an I/Q down converter comprising multipliers 203 and 204 and low pass filters 205 and 206. Phase Locked Loop (PLL) discriminator 211 and Automatic Frequency Control (AFC) discriminator 220 use the on-time output on line 427 to produceA carrier correction signal is generated and the on-time output is the baseband input despread signal. The loop filters 214 and 226 adjust the rate and magnitude of the correction signal to replicate the carrier term cos [2 π f ] of equation (1) before it is combined into a signal on line 227 to adjust the frequency of the carrier local oscillator 210₀(t-τ)+θ_t]。

When oscillator 210 is adjusted to accurately reproduce the term cos [2 π f [ ] f₀(t-τ)+θ_t]The I baseband signal on line 207 contains the GPS baseband signal and the Q baseband signal on line 208 does not contain any GPS baseband signal. Corresponding to phase detector 551, PLL discriminator 211 measures the instantaneous phase difference between the carrier of the received signal on line 202 and the output of oscillator 210. Corresponding to discriminator 552, AFC discriminator 223 measures the instantaneous frequency difference.

Some embodiments of the GPS receiver utilize only a PLL discriminator, some embodiments utilize only an AFC discriminator, and other embodiments use a combination of both either simultaneously or at different times. The feedback loop using the PLL discriminator is generally more sensitive than the feedback loop using the AFC discriminator; however, PLL discriminator based loops are more susceptible to transients caused by platform dynamics. The choice of the best method is therefore dependent on the signal strength and the dynamics. Many receivers utilize both types of discriminators and use them for varying signal conditions.

A closed loop carrier tracking loop such as that in fig. 10 performs poorly when the signal becomes too weak or the carrier phase parameter changes too quickly. In these cases, a 180 degree error is often caused because the discriminator cannot distinguish phase errors that are multiples of 180 degrees. These types of errors are called "cycle slips" and are common in closed-loop carrier tracking loops when signal conditions are poor.

Certain embodiments of the present invention seek to measure the frequency and/or phase of a carrier using an open loop carrier tracking loop. The despread signal, typically provided by an on-time output on line 427, is analyzed to obtain a measurement without having to generate a reference carrier of the same frequency and phase as the received signal. Thus, the detrimental effects of a closed loop tracking system can be avoided. For these open-loop frequency/phase measurements, various methods will be described, including (a) a frequency multiplier-based method, (B) a frequency discriminator-based method, (C) a block phase estimator-based method, and (D) a channelization filter-based method. These will now be described.

The open loop carrier frequency tracking method according to the present invention measures the carrier frequency directly from the despread signal from the PN tracking circuit. After removing the pseudo noise from the baseband input, the despread signal in the on-time output is of the form:

s₁(t)＝Ad(t-τ)exp(j2πf_et+jθ_t) (6)

wherein, the frequency f_eIs the residual carrier frequency of the despread signal after the doppler cancellation operation typically performed by mixers 203 and 204 and low pass filters 205 and 206 of fig. 10. For the sake of simplicity, corresponds to-2 π f in equation (1)₀The constant phase term of the tau term is incorporated into theta_t。

Fig. 11 illustrates a multiplier-based open loop carrier frequency measurement method according to an embodiment of the present invention. In operation 302, the on-time output signal is squared, producing a nearly unmodulated sinusoidal signal:

s₂(t)＝A²exp(j4πf_et+j2θ_t) (7)

because of d²1. The M samples of the sinusoidal signal are collected via operation 302 and analyzed via operation 303 using a Fast Fourier Transform (FFT) method or a Discrete Fourier Transform (DFT) method to produce a spectral output. Operation 304 finds the peak of the spectral output that corresponds to the frequency and phase of the signal in equation (7), i.e., 2f_eAnd 2 theta_t. Thus, a divide-by-2 operation is required to generate the frequency f_eAnd phase theta_t. The accuracy of the measurement is high, the method is computationally intensive and somewhat less sensitive than other methods such as the method of FIG. 14.

Figure 12 shows a discriminator based open loop carrier frequency measurement method according to one embodiment of the invention. The discriminator comprises a delay 305, a conjugator 306 and a multiplier 307. Assume that with each BPSK data bit period T_dThe two samples are input from the on-time output sample and multiplier 307 produces an output at a rate equal to the data rate (i.e., at 50 bauds). For samples that occur in the same bit period, multiplier 307 produces:

s₃(t)＝A²exp(j2πf_eT_d/2+j(θ_t-θ_t-Td/2)) (8)

because, d (T- τ) d (T- τ -T)_dAnd/2) 1. Assuming that the residual phase modulation is negligible, i.e. (θ)_t-θ_t-Td/2) When the value is 0, s₃The phase angle of (t) is proportional to the residual carrier frequency. Thus, s can be determined by₃(T) phase angle divided by π T_dTo obtain f_e. The averaging process of 308 is used to improve the signal-to-noise ratio.

The discriminator based loop is quite sensitive and quite easy to implement, but it is less accurate than the multiplier based loop shown in fig. 11.

The same operation may also be achieved when the input sampled from the on-time output occurs at a rate other than twice the BPSK data rate, i.e., at a 1kHz rate. When the input of samples is output from on-time such that the delay of one sample caused by operation 305 is T_cSuch a rate may occur by dividing the phase angle by T_cTo obtain a residual carrier frequency f_e. However, the sensitivity is reduced because the signal-to-noise ratio (SNR) per on-time sample is reduced when the integration time per on-time sample is reduced. It can be seen that conjugator 306 may be used for signals on line 352 in addition to signals on line 351. Likewise, various other multiplication structures may be used in place of the four quadrant multiplier 307.

FIG. 13 illustrates block-based phase estimation according to an embodiment of the present inventionOpen loop of the device another open loop carrier frequency measurement method. Operation 310 squares the on-time output to produce the signal in equation (7). The signal being at time t₀A centered interval of length D (i.e., (t)₀-D/2，t₀+ D/2)) is:

s₄(t)＝A²exp(j4πf_et₀+jθ_t0)sinc(2f_eD) (9)

wherein sinc (x) is [ sin (pi x)]/(π x), and we assume θ_tAt (t)₀-D/2，t₀Variation over the range of + D/2) is ignored. Operation 312 measures s₄Half of the angle of (2 pi f)_et₀+θ_t0. Note that this is similar to equation (6) at time t₀Are the same. When theta is_tNot being constant, provided that theta_tThe change is not too fast within the interval D, the process is for 2 π f over the interval D_et₀+θ_tThe averaging of (d) yields a good estimate. The averaging process of 311 is used to improve the signal-to-noise ratio.

Due to s₄Is set to phi_tAt [0, 2 π]Within the range of (3), the half angle produced by 312 is defined as eta_tIn fact, it is uncertain, i.e. it may be equal to 1/2 φ_t+ n pi, where n may be 0 or 1. Thus, there is uncertainty in determining the phase angle as a function of time. The uncertainty can be determined by tracking phi_tTo solve the problem. When phi is_tPassing through 0 degree in positive direction_tAdding pi; when phi is_tPassing 0 degrees in the negative direction, from the phase angle η_tMinus pi. Operation 313 performs this operation, referred to as "sector tracking".

There are also various variations of the method of fig. 13. Operation 310 may have non-linearity for the magnitude portion of the signal to produce signal a^rexp(j4πf_et₀+jθ_t) R is not less than 0. The overall process remains the same. For example, instead of the squaring operation shown in FIG. 13, the signal samples may be converted to a polar table including magnitude and phase componentsShown in the figure. The resulting phase component is doubled, i.e., modulo 2 pi, to effectively square the phase component of the signal. The size term may be kept constant or raised to a small power (i.e., between 0 and 2). The signal may then be converted back to a rectangular coordinate system to facilitate subsequent averaging.

The above procedure provides for a phase angle of 2 π f_et₀+θ_t0Is estimated. By making successive such measurements, the frequency f can be determined_e. For example, if at time t₁Performing another measurement to obtain a phase angle of 2 π f_et₁+θ_t1. If theta is greater than theta_t1From t₀To t₁The variation is small, then the difference between these phase angles can be considered as 2 π f_e(t₁-t₀) And, due to (t)₁-t₀) Is known, and therefore f can be easily calculated_e。

Fig. 14 shows another open loop carrier frequency measurement method based on a channelization filter in accordance with one embodiment of the present invention. Local Oscillator (LO)314 generates signals having different frequencies (i.e., f)₁、f₂、...、f_M) Respectively, to a set of frequency channelizers. Each channelizer produces an output that quantifies the degree of coincidence between the frequency of the LO supplied to it and the carrier frequency of the signal in the on-time output. For example, the frequency channelizer 319, which includes multiplier 315, accumulator 331, nonlinear detector 332, and accumulator 333, measures f₁And the degree of coincidence between the carrier frequencies of the on-time output signals. The frequency of the channel giving the largest output is a rough estimate of the carrier frequency of the output signal when aligned. The peak detector 322 interpolates the output of the channelizer 319, 320, 321,. and 330 to calculate a more accurate measure of the frequency of the on-time output signal. In many cases, the calculated measurement is 10 times more accurate than the coarse estimate (i.e., the frequency of the channel that gives the largest output). In some cases, a single output from each detector may be sufficient, and the accumulator function (e.g., accumulator 333) following the detector may be disabled.

Once the PN component of the signal in equation (1) is removed, the punctual signal is in the form of equation (6) (i.e., Ad (t- τ) exp (j2 π f)_et+jθ_t)). The multiplier 315 multiplies the signal by the local reference signal exp (-j2 π f_mt) to generate a signal:

s₅(t)＝Ad(t-τ)exp(j2π(f_e-f_m)t+jθ_t)＝Ad(t-τ)exp(j2πtδ+jθ_t) (10)

wherein δ ═ f_e-f_m. If the accumulator 331 accumulates over a time period T that is less than or equal to the bit period (but does not cross a bit boundary), the output of the non-linear detector 332 can be approximated as an integral:

wherein, let θ be_tIs constant (or slowly varying over time). When δ is 0, i.e. when f_m＝f_eThe function is maximal.

Channelizer to different frequency values (e.g. f) corresponding to the channelizer₁、f₂A δ value of..) yields a sample of the function output of equation (11). When these samples of the function of equation (11) are close enough in frequency (e.g., within 0.5/T), they can be used in an interpolation process to estimate the peak position of the function of equation (11), which is located at f in the absence of noise_m＝f_eTo (3). Even if these sampling spacings are wider than 0.5/T, it is still possible to interpolate on the basis of the function of equation (11), since the functional form is known. One way to perform such an interpolation is to use splines (or other types of polynomials) to utilize three or more channelizer outputs in a curve fit. Due to the specific nature of the function of equation (11), it is possible to perform the final interpolation using only the outputs of two such channelizers. However, more than two channelizers are required to first determine which two channelizers' outputs fall on either side of the peak position. Once determined, a final interpolation process can be performed using the two channelizer outputs to calculate the location of the peak to determine f_e. Thus, as a whole, this interpolation algorithm requires at least three channelizers.

The channelization method is very sensitive but not as accurate as the frequency doubler and spectral analysis methods. However, it is easy to implement, especially when the number of channels is small.

In fig. 14, each channelizer, such as channelizer 319, can be considered to be a correlation indicator generator. For example, the punctual output feeds multiplier 315, accumulator 331, nonlinear detector 332, and accumulator 333. The output of accumulator 333 can be considered the output of the correlation operation. These processing steps are the same block-by-block as the correlation operations performed in fig. 8: multiplier 105, accumulator 106, non-linear detector 107 and accumulator 120. Another way to observe the frequency channelizer is as a spectrum analyzer, but without the prior non-linear characteristic of the frequency doubler of fig. 11. The channel of each channelizer can be viewed as producing independent spectral components corresponding to the frequency provided by LO generator 314. The absolute value and squaring operation of 332 is typically used for spectrum analyzers. The post-detection integration of 333 is used to provide an improved estimate of the energy associated with a certain spectral line. Since the integration time associated with 331 is typically limited to the time of the data bit period, the frequency accuracy of the channelizer approach is typically much less accurate than the squaring and spectral analysis approach of fig. 11. It is possible to utilize FFT or DFT algorithms for LO generation 314 of fig. 14, as well as the initial multiplication and summation functions of each channelizer.

The various methods of fig. 11-14 may be used to estimate carrier phase and/or frequency. Continuous estimation can be utilized in the tracking of the GPS signal to facilitate continuous processing of the signal. In general, it is the case that a prior estimate of the frequency is used to adjust the parameter estimation window for additional estimates performed as part of the overall tracking process. The adjustment typically takes the form of a window center adjustment and/or a window width adjustment. The frequency estimation operation is continued and then the window is used to limit the search range acceptable for efficient frequency estimation. Alternatively, a window may be used as at least one validity criterion, or a measurement may be accepted or deleted. The use of such a window helps to reduce the amount of computation required, as well as reducing the occurrence of erroneous results. The maximum window adjustment is generally based on a priori knowledge defined by physical principles such that a certain parameter does not change more than a predetermined amount from one measurement time to the next.

It should be understood that while the block diagrams of fig. 11-14 may illustrate a hardware implementation, it is often practical to implement at least some of these methods using software-based methods, particularly when a microprocessor with good signal processing capabilities is employed.

Various methods (e.g., matched filters, fast fourier transforms, fast convolution methods, etc.) may be used to generate the correlation indicators as discussed above. Some GPS signal processors utilize matched filters rather than serial correlators for PN acquisition and despreading operations. These matched filters periodically produce as output a value that is substantially the same as the value produced by a bank of serial correlators. In particular, it is possible to use a matched filter instead of a serial filter to produce early, late or on-time correlation outputs. Thus, the foregoing discussion applies to the case where a matched filter is utilized rather than a serial correlator. In addition, matched filtering operations may be performed using a Fast Fourier Transform (FFT) or other fast convolution method, such as the method described in U.S. patent No. 5,663,734. The invention is also preferably applicable to the case where these alternative methods of matched filtering are employed to generate the correlation indicator.

FIG. 15 shows a flow diagram of an open loop tracking satellite positioning system signal for determining a position of a receiver, according to one embodiment of the invention. After receiving a GPS signal from one of a plurality of GPS satellites in operation 602, a set of correlation indicators is generated for a set of predetermined values for a parameter of the GPS signal (e.g., time of arrival or carrier frequency) in operation 604. By interpolating the set of indicators, a measurement is obtained in operation 606. Optionally, an average of the measurement domain may be made in operation 608 to combine the measurement with prior measurements of the GPS signal into a combined measurement. The measurements may be combined using various methods such as least mean square filtering, Kalman filtering, median filtering, and the like. Operations 602-608 are repeated for the GPS signals from each of the plurality of GPS satellites. It should be appreciated that steps 602-608 may be performed sequentially or in parallel for each GPS satellite. After the GPS signals of all the satellites have been processed, as determined in operation 610, the position of the receiver is obtained in operation 612 from individual or combined measurements of the GPS signals corresponding to the plurality of GPS satellites. Optionally, an averaging of the location fields may be performed in operation 614.

Fig. 16 illustrates a flow chart for open loop tracking of carrier frequencies, in accordance with certain embodiments of the present invention. Operation 702 despreads a portion of the GPS signal to produce despread data. The despreading operation may be performed using a serial correlator or a matched filter. Next, a determination is made whether to perform spectral analysis in 708. If a spectral analysis operation is used, then the prior non-linear operation of 710 may optionally be employed. This non-linear operation is used, for example, in the frequency multiplier/spectral analysis method of fig. 11. If the channelisation filter method of figure 14 is used (as previously described, figure 14 is a form of spectral analysis), then the non-linear characteristic is not used. When spectral analysis is used, operation 722 uses a Discrete Fourier Transform (DFT) or a Fast Fourier Transform (FFT) to generate the spectral data. At 724, a measure of the carrier frequency of the GPs signal is determined by adding the peak location of the spectral data to the magnitude of the spectral data near the peak. Finally, the frequency measurements are used to determine the position of the receiver in operation 732.

If the decision 708 determines that spectral analysis is not to be used, then a non-linear operation (such as a squaring operation, or the delay, conjugate, and multiplication operations shown in FIG. 12) is always used on the despread data. An averaging operation is then performed at 712 to generate averaged data. Operation 712 generally operates to improve the signal-to-noise ratio. The phase angle of the averaged data is calculated in operation 714. In operation 716, a measurement of the carrier frequency of the GPS signal is calculated using the phase angle obtained in operation 714. For example, operations 712 through 716 are illustrated in both the frequency discriminator of fig. 12 and the block phase estimator of fig. 13. Regardless of the frequency measurement method used, the operation of block 734 is a control step in which the results of past frequency measurements (perhaps along with prior measurements) are used in order to adjust the frequency measurement range, or window. The window may be used for subsequent frequency measurements by limiting the range over which processing is performed or providing a range over which future frequency estimates are deemed valid.

Fig. 17 shows a block diagram representation of a receiver implementing the open loop tracking method according to the present invention. The input circuit 802 is coupled to a GPS antenna 811 to receive GPS signals from a plurality of satellites. Circuitry 804 is coupled to processor 806 to acquire, track, and demodulate GPS signals. The various methods of the present invention may be implemented in 804 and 806 (using hardware, software, or a combination of hardware and software) to calculate measurements of parameters of GPS signals from a plurality of satellites. The processor 806 uses these measurements to calculate the position of the receiver. An optional display 808 is coupled to the processor 806 to display the location of the receiver.

Fig. 18 shows a block diagram representation of a remote satellite positioning system implementing an open loop tracking method according to the present invention. The remote satellite positioning system includes a portable receiver that combines a communications receiver with a GPS receiver for one embodiment of the present invention. The combined mobile unit 910 includes circuitry for performing the functions required for processing GPS signals and for processing communication signals received over the communication link. A communication link, such as communication link 960, typically connects to another component, such as base station 952 having a communication antenna 951.

The portable receiver 910 is a combined GPS receiver and communication receiver and transmitter. The receiver 910 includes a GPS receiver stage that includes an acquisition and tracking circuit 921 and a communications receiver 905. The acquisition and tracking circuit 921 is coupled to a GPS antenna 901 and the communication receiver 905 is coupled to a communication antenna 911. GPS signals are received by GPS antenna 901 and input to acquisition and tracking circuit 921, which acquires the PN code for each received satellite. Data generated by circuitry 921, such as correlation indicators, is processed by processor 933 for transmission by transceiver 905. The communication transceiver 905 includes a transmit/receive switch 931 that switches communication signals to and from the communication antenna 911 and the transceiver 905. In some systems, band-splitting filters or "duplexers" are used instead of the T/R switches. Received communication signals are input to communication receiver 932 and passed to processor 933 for processing. Communication signals to be transmitted from processor 933 are passed to modulator 934 and frequency converter 935. Power amplifier 936 increases the gain of the signal to an appropriate level for transmission to base station 952.

In one embodiment of the combined GPS/communications system of the receiver 910, the data generated by the acquisition and tracking circuit 921 is communicated to the base station 952 via the communication link 960. The base station 952 then determines, based on the data from the remote receivers, the location of the receiver 910, the time at which the data was measured, and ephemeris data received from the data's own GPS receiver or other source of such data. The location data may then be transmitted back to the GPS receiver 910 or other remote location. The communication link 960 between the receiver 910 and the base station 952 may be implemented in various embodiments, including a direct link or a cellular telephone link. More details regarding portable receivers utilizing a communication link are described in commonly assigned U.S. patent No. 5,874,914(Krasner), the entire contents of which are incorporated herein by reference.

The previous disclosures are directed to determining code phase using an open loop method and frequency using an open loop method, respectively. It is understood that the two methods may be used separately or together. For example, a bank of correlators may be assigned to cover both a phase range and a frequency range (e.g., using the channelization method of fig. 14). Alternatively, the open loop frequency tracking method may be performed after tracking and determining the code phase with a relatively coarse carrier frequency estimate. Such an approach may compromise the time complexity of performing the entire tracking operation.

Open loop tracking methods may also be employed to improve the performance of open or closed loop code phase tracking. It is well known that carrier tracking can be advantageously used to estimate the velocity of the platform, which can then be used to estimate the rate of change of the PN code phase. For example, in one approach, carrier frequency tracking is used for so-called "code-carrier smoothing", in which successive estimates of PN code phase are filtered using a filter whose parameters are determined by the carrier frequency estimate. This works well because the carrier frequency is much higher than the frequency of the PN code (1540 times greater for the us GPS system) and therefore the rate of change derived from the carrier is much more accurate than the rate of change derived from the PN code (1540 times more accurate for the us GPS).

Although the method and apparatus of the present invention have been described with reference to GPS satellites, it should be understood that the teachings are equally applicable to positioning systems utilizing pseudolites or a combination of satellites and pseudolites. A pseudo-site is a ground-based transmitter that broadcasts a PN code (similar to a GPS signal) that is typically modulated on an L-band carrier signal, typically synchronized with GPS time. Each transmitter may be assigned a unique PN code to allow identification by a remote receiver. Pseudolites are useful in situations where GPS signals from an orbiting satellite might be unavailable, such as in tunnels, mines, buildings, or other enclosed environments. The term "satellite" as used herein is intended to include pseudolites or equivalents of pseudolites, and the term GPS signals as used herein is intended to include GPS-like signals from pseudolites or equivalents of pseudolites.

In the preceding discussion, the invention has been described with reference to an application based on the united states Global Positioning Satellite (GPS) system. It is apparent, however, that these methods are equally applicable to similar satellite positioning systems, particularly the russian global orbiting navigation satellite system (Glonass system) and the proposed european Galileo (Galileo) system. The main difference between the Glonass system and the GPS system is the use of slightly different carrier frequencies rather than different pseudo-random codes to distinguish between transmissions from different satellites. In this case, all the circuits and algorithms described previously are usable in essence. The term "GPS" as used herein includes alternatives to satellite positioning systems, including, for example, the Russian Glonass system.

In the foregoing specification, the invention has been described with reference to specific exemplary embodiments thereof. It will be evident that various modifications may be made thereto without departing from the broader spirit and scope of the invention as set forth in the following claims. Accordingly, the specification and drawings are to be regarded in an illustrative rather than a restrictive sense.

Claims (21)

1. A method of processing satellite positioning system signals, the method comprising:

generating a set of at least three indicators based on processing a portion of a satellite positioning system signal received by a receiver, each indicator representing a probability that a parameter of the signal is equal to a predetermined value; and

the measured value of the parameter is calculated by interpolation of the set of indicators.

2. The method of claim 1, wherein the parameter is one of:

(a) time of arrival, and

(b) the carrier frequency.

3. The method of claim 1, wherein one of the indicators is generated by one of:

(a) the correlation is performed in series and the correlation is performed,

(b) the filtering is carried out in a matched manner,

(c) a fast Fourier transform, and

(d) and (4) fast convolution.

4. The method of claim 1, further comprising:

a position is calculated using the measurements.

5. The method of claim 1, further comprising:

a position is calculated using the measurements and one or more prior measurements.

6. The method of claim 1, further comprising:

combining the measurement value with one or more prior measurement values into a combined measurement value.

7. The method of claim 6, wherein the combining comprises one of:

(a) a least mean square filtering is performed on the filtered signal,

(b) kalman filtering, and

(c) and (6) median filtering.

8. The method of claim 4, further comprising: combining the location with one or more previous locations into a combined location.

9. The method of claim 8, wherein the combining comprises one of:

(a) a least mean square filtering is performed on the filtered signal,

(b) kalman filtering, and

(c) and (6) median filtering.

10. A method of processing satellite positioning system signals, the method comprising:

receiving a set of at least three indicators based on a portion of a satellite positioning system signal received by a receiver, each indicator representing a probability that a parameter of the signal is equal to a predetermined value; and

the measured value of the parameter is calculated by interpolation of the set of indicators.

11. A receiver of satellite positioning system signals, the receiver comprising:

a first circuit configured to be coupled to an antenna for receiving satellite positioning system signals; and

a second circuit configured to be coupled to the first circuit, the second circuit configured to generate a set of at least three indicators based on processing a portion of the satellite positioning system signals received by the first circuit, each indicator representing a probability that a parameter of the signal is equal to a predetermined value, the second circuit configured to calculate a measurement of the parameter based on an interpolation of the set of indicators.

12. The receiver of claim 11, wherein the first and second circuits comprise a single integrated circuit.

13. The receiver of claim 11, wherein the second circuit comprises:

a memory configured to store an indicator; and

a programmable digital signal processor coupled to the memory, the programmable digital signal processor configured to generate the indicator and calculate the measurement.

14. The receiver of claim 11, wherein the parameter is at least one of:

(a) time of arrival, and

(b) the carrier frequency.

15. The receiver of claim 11, wherein the second circuit is configured to generate at least one indicator using one of:

(a) the correlation is performed in series and the correlation is performed,

(b) the filtering is carried out in a matched manner,

(c) a fast Fourier transform, and

(d) and (4) fast convolution.

16. The receiver of claim 11, wherein the second circuit is configured to calculate a position using the measurement values.

17. The receiver of claim 11 wherein the second circuit is configured to calculate a position using the measurements and one or more prior measurements.

18. The receiver of claim 11, wherein the second circuit is configured to combine the measurement and one or more prior measurements into a combined measurement.

19. The receiver of claim 17, wherein the second circuit is configured to combine the position and one or more previous positions into a combined position.

20. A satellite positioning system receiver, the receiver comprising:

a first circuit configured to be coupled to an antenna for receiving satellite positioning system signals;

a second circuit coupled to the first circuit, the second circuit configured to generate a set of at least three indicators, each indicator representing a probability that a parameter of a signal is equal to a predetermined value, based on processing a portion of the satellite positioning system signal received by the first circuit; and

a third circuit coupled to the second circuit, the third circuit configured to be coupled to an antenna to communicate the set of indicators received from the second circuit to the base station.

21. A base station for processing signals associated with a satellite positioning system, the base station comprising:

a first circuit configured to be coupled to an antenna to receive a set of at least three indicators from a receiver of satellite positioning system signals, each indicator representing a probability that a parameter of a satellite positioning system signal is equal to a predetermined value; and

a second circuit coupled to the first circuit, the second circuit configured to calculate a measured value of the parameter based on an interpolation of the set of indicators.