EA031050B1 - Transpositional modulation systems, methods and devices - Google Patents

Abstract

Systems, methods and devices for transmitting, receiving and demodulating transpositional modulated signals, for increasing information bandwidth of defined communication channels, and for time-delay shifting an input signal in accordance to an input control signal are provided. One such method of increasing the information bandwidth of a defined communication channel includes receiving a first modulated signal having a first carrier signal frequency; receiving a second modulated signal having a second carrier signal frequency, the second modulated signal being modulated with information independent of information modulating the first carrier signal, the second carrier signal frequency being harmonically or sub-harmonically related to the first carrier signal frequency; and combining said first and second signals according to said method.

Description

The present invention relates generally to signal processing and, more specifically, to systems, methods and devices for transmitting, receiving and demodulating transpositionally modulated signals to increase the width of the information band of specified communication channels and to shift the time delay of the input signal in accordance with the input control signal .

The known transmission systems, whether voice, video or data, have frequency band limits established by national and international regulatory bodies that regulate the use of the frequency spectrum. Carrier modulation methods have evolved from the initial amplitude modulation to modern methods, combining two or more carriers with amplitude, frequency or phase modulation in various combinations. Advanced carrier modulation techniques have been developed to maximize the energy transmitted in the frequency band of a dedicated channel, and therefore to ensure the maximum available information bandwidth of a dedicated communication channel.

A new method of modulating the main carrier was developed and first patented (see, for example, US Patent No. 4,613,974, fully incorporated into the present description, by Vokac et al.); In this method, a new type of modulation of the carrier is applied, which does not conflict with the amplitude, frequency and / or phase modulation used in the same carrier signal.

The idea of transpositional modulation (TM) was based on an earlier idea of how to impose information on a carrier signal without affecting its amplitude, frequency or phase (see, for example, US Pat. No. 4,613,974, fully incorporated into the present description, by Vokac et al.).

By creating an inflection, as will be shown below, it is possible to transmit information using a carrier signal. This method is not detected by known signal demodulators with amplitude, frequency, or phase modulation.

Using the previously patented method of forming, the oscillating signal described below is generated in the time domain with kinks enlarged for clarity. In practical application, kinks are visually imperceptible.

The disadvantage of the previous methods of forming an oscillating signal of this type consists in a small change in amplitude, which must be eliminated with the help of an adjustment circuit. For example, in FIG. 1 shows an illustrative example of a TM-modulated signal 100, formed in accordance with the prior art disclosed in US Pat. No. 4,613,974. As you can see, there is an error in the form of the difference between the negative peaks 101 and 102 due to the amplitude change.

Thus, there is a need to eliminate these disadvantages and inconsistencies of the prior art.

In embodiments of the present disclosure, systems, methods and devices are proposed for transmitting, receiving and demodulating transpositionally modulated signals to increase the width of the information band of specified communication channels, as well as to shift the time delay of the input signal in accordance with the input control signal. In one embodiment, a method is proposed for increasing the information bandwidth of a given communication channel, wherein according to this method, a first modulated signal is received having a frequency of a first carrier signal;

receiving a second modulated signal having a frequency of the second carrier signal, wherein the second modulated signal is modulated with information independent of the information modulating the first carrier signal, and the frequency of the second carrier signal is harmonically or subharmonically associated with the frequency of the first carrier signal; and summarize the specified first and second signals with obtaining the resulting signal.

In another embodiment, a time shift modulator is proposed for shifting the time delay of an input signal in accordance with an input control signal. A time shift modulator contains an all-pass filter modified by a voltage-controlled time delay.

In another embodiment, a method is proposed for increasing the communication bandwidth in a given communication channel, wherein according to this method, the second transpositionally modulated signal is summed with the resulting signal containing the first transpositional modulated signal and the first main carrier signal, the second transpositionally modulated signal being summed with the resulting signal using the second main carrier signal having the same frequency as the first main carrier signal and having an angle cd yoke in phase 90 ° with respect to the first main carrier signal.

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In another embodiment, a method is proposed for providing an increased width of the information bandwidth of the ultrasonic communication, whereby according to this method, a transposition modulation of the ultrasonic communication signal is performed by performing direct amplitude modulation of a single ultrasonic transducer.

In another embodiment, a method is proposed for providing an increased width of an ultrasonic communication information bandwidth, whereby according to this method, a transposition modulation of an ultrasonic communication signal is performed by performing direct amplitude modulation of a first ultrasonic transducer using a component of a main carrier signal of transposition modulation and performing a direct amplitude modulation of the second ultrasound transducer using the carrier component tr network harmonics transposition modulation.

In another embodiment, a method is proposed for providing an increased information bandwidth of ultrasonic communication, and according to this method, the components of the main carrier signal of the transposition modulation and the third harmonic carrier signal with the ultrasonic communication signal are summed up by performing a direct amplitude modulation of a single ultrasonic transducer using wideband modulation technology.

In another embodiment, a method is proposed for providing an increased width of an ultrasonic communication information bandwidth, whereby according to this method, a transposition modulation of an ultrasonic communication signal is performed by performing a direct angular modulation of the first ultrasonic transducer using a main carrier signal of transposition modulation and by performing a direct angular modulation of the second ultrasonic transducer using the carrier signal component rd harmonic transposition modulation.

In another embodiment, a method is proposed for providing an increased information bandwidth of ultrasonic communication, and according to this method, the components of the main carrier signal of the transposition modulation and the third harmonic carrier signal with the ultrasonic signal are summed up by performing a direct angular modulation of a single ultrasonic transducer using wideband modulation technology.

In yet another embodiment, a system is proposed for providing an increased width of the information band of optical communication. This system contains an optical beam and an optical modulator and is configured to perform direct modulation of the optical beam using a transposition modulation signal.

In another embodiment, a method is proposed for providing an increased information bandwidth of optical communication, and according to this method, a first frequency optical modulation of a first frequency beam is performed using a component of a transverse modulation main carrier frequency.

In another embodiment, a method is proposed for providing an increased information bandwidth of optical communication, and according to this method, the second frequency of the component is directly modulated with a component using a third harmonic signal of transposition modulation.

Other systems, methods, features and advantages of the present disclosure will be clear to those skilled in the art after reviewing the drawings and the detailed disclosure. It is assumed that all additional systems, methods, features and advantages are included in this disclosure, are within the scope of this disclosure and are protected by the attached claims.

Many aspects of the present disclosure may be better understood by referring to the drawings below. The components in the drawings are not necessarily to scale; instead, the focus is on illustratively illustrating the principles of the present disclosure. In addition, in the drawings, the same reference numbers correspond to the same components throughout the drawings.

FIG. 1 shows a TM-modulated signal generated according to the prior art.

FIG. 2 is a block diagram illustrating a modulation method of a carrier signal according to a first illustrative embodiment of the present disclosure.

FIG. 3 shows a signal formed as a quarter of a period, according to an embodiment of the present disclosure.

FIG. 4 shows the signal shown in FIG. 3, after adding up a quarter of a period, according to an embodiment of the present disclosure.

FIG. 5 shows a modulation input, which may be used in the embodiments provided by the present disclosure, to generate the signal shown in FIG. four.

FIG. 6 is a graph illustrating the frequency spectrum of the signal shown in FIG. four.

FIG. 7 shows a graph illustrating the frequency spectrum obtained by heterodyning the third harmonic component using the second harmonic of the signal shown in FIG. 6,

- 2 031050 according to the embodiment provided by the present disclosure.

FIG. 8 shows a filter that may be applied in the embodiments provided by the present disclosure.

FIG. 9a shows a block diagram illustrating a software-based forward spectrum system for generating a signal according to an embodiment provided by the present disclosure.

FIG. 9b shows a block diagram illustrating a hardware-based forward spectrum system for generating a signal according to an embodiment provided by the present disclosure.

FIG. 10 shows a block diagram illustrating a sub-period calibration system for demodulating a signal according to an embodiment provided by the present disclosure.

FIG. 11 shows a block diagram illustrating a system for determining a phase of a third harmonic for performing demodulation of a signal according to an embodiment provided by the present disclosure.

FIG. 12 shows a block diagram illustrating a system based on a fast Fourier transform for demodulating a signal according to an embodiment provided by the present disclosure.

FIG. 13 shows a block diagram illustrating a TM transmitter for generating and transmitting a signal representing the sum of the TM signal and the original signal, according to an embodiment provided by the present invention.

FIG. 14 shows a block diagram illustrating an example implementation of a carrier signal generating unit in a TM transmitter, according to an embodiment provided by the present disclosure.

FIG. 15 shows a block diagram illustrating an example implementation of a TM-modulated signal processing unit in a TM transmitter according to an embodiment provided by the present disclosure.

FIG. 16 shows a block diagram illustrating a TM receiver for receiving a signal representing the sum of a TM signal and an original signal, as well as for extracting and demodulating this TM signal according to an embodiment provided by the present invention.

FIG. 17 shows a block diagram illustrating an example implementation of a carrier signal recovery and harmonics unit in a TM receiver according to an embodiment provided by the present disclosure.

FIG. 18 shows a block diagram illustrating an example of the implementation of a block for the separation and demodulation of a TM signal according to an embodiment provided by the present disclosure.

FIG. 19 shows a graph illustrating the behavior of the frequency response of the filtering circuit based on the time delay of the split function according to the embodiment provided by the present disclosure.

Many variations of the present disclosure may take the form of computer-executable instructions, including algorithms executed by a programmable computer or microprocessor. However, the present disclosure may be practiced using other computer system configurations as well. Some aspects of the present disclosure may be implemented in a special purpose computer or in a specially programmed data processor made or built with the ability to implement one or more of the methods and algorithms disclosed below.

Aspects of the present disclosure described below may be stored or distributed on computer-readable media, including magnetic, optical and removable computer disks, floppy disks, magnetic disk drives, optical disk drives, magneto-optical disk drives, magnetic tape, hard disk drives disk (HDD), semiconductor drives (SSD), compact flash memory or non-volatile memory, and they can also be electronically distributed across networks, including the cloud. Specific data structures and their transmission forms according to aspects of the present disclosure are also included in the scope of the present disclosure.

FIG. 2 shows a block diagram 200 illustrating a modulation method of a carrier signal according to a first illustrative embodiment of the present disclosure. It should be noted that any process descriptions or blocks in the flowchart should be understood as representative modules, segments, parts of codes, or steps that include one or more commands to implement specific logic functions in the process, and alternative implementations are included in the scope of the present invention. in which functions can be performed in a different order than that shown or disclosed, including, essentially, simultaneous execution in the reverse order, depending on the functions involved, as will be clear from professionals with the proper qualifications in the technical field to which the present invention relates. This method solves the problem of amplitude modulation in the prior art (for example, as shown in FIG. 1 above) and can be implemented using hardware, software, or any combination thereof. Shown in FIG. 2, which may be called a method for combining quarter-periods (QC-method), may include using a lookup table (LUT) 210 as a quick way to get results, and the same can be done in a different way using mathematical functions, without the need for continuous implementation of mathematical operations. This QC method is based on the time domain.

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As shown in FIG. 3, the modulated output 300 obtained by the method shown in FIG. 2, contains four separate quarter-period segments for each full signal period. FIG. 3 shows three full periods (for example, periods a, b and c) which can be obtained at the output using the quarter-period method shown in FIG. 2. Each period consists of four quarter-period segments (for example, 301, 302, 303, and 304). For illustrative purposes only, gaps between quarter-period segments are shown. In addition, also for illustrative purposes, the places of kinks are highlighted (a1, a2, b1, b2, c1, c2). These kinks are formed between adjacent quarter-period segments, as shown in the drawing.

As shown in FIG. 3, the first quarter of each period (301a, 301b and 301c) may have different amplitudes depending on the value of the applied modulation. The same applies to each of the remaining quarters of each period shown. In other words, the second (302a, 302b, 302c), the third (303a, 303b, and 303c) and the fourth (304a, 304b, and 304c) quarters of each period can have different amplitudes, depending on the value of the applied modulation. If the first quarter (for example, 301a, 301b, 301c) of the period has a low amplitude, then the second quarter (for example, 302a, 302b, 302c) has a higher complementary amplitude, so that between the negative peak (Pk-) of the entire period and the positive peak ( Pk +) of the same period always has a constant amplitude. The same is true for the third and fourth quarters of each period. As a result, the values (Pk +) of a positive peak will always be the same. The values (Pk-) of the negative peak are also made the same in order to exclude amplitude fluctuations under the influence of the values of the applied modulation.

As further shown in FIG. 3, the first (301a, 301b, 301c) and the third (303a, 303b, 303c) quarters of the corresponding period have the same amplitudes. Similarly, the second (302a, 302b, 302c) and fourth (304a, 304b, 304c) quarters of the corresponding period also have the same amplitudes. The purpose of this is to provide the same area under the curve for each period, regardless of the value of the applied modulation. Thus, it is ensured that the average value of each period is zero, which prevents any shift of the DC value (direct current) in the carrier signal due to the applied modulation values.

However, it should be noted that in some applications, the DC offset may be acceptable, and because of this, a non-constant area under the curve may occur, i.e. no need for symmetry between periods. In this case, information or symbols can be transmitted at a rate of two characters per minute, or in each period there can be two different inflection points (for example, one point located in the growing half of the period between the negative peak and the positive peak, and another point located in the falling half of the period between the positive peak and the negative peak).

Each quarter of the period can be formed using a constant clock frequency or time step, and, thus, there will be no change in frequency from one period to another under the influence of the values of the applied modulation. Each bend (a1, a2, b1, b2, c1, c2) in each period is separated from the bend in the next half of the period by a distance equal to 180 ° in angular equivalent. Thus, there is no phase change due to the value of the applied modulation.

By summing the quarter periods (for example, those shown in FIG. 3), a smooth and continuous signal 300 is obtained, as shown in FIG. four.

FIG. 5 shows a TM-modulation signal 500, which is used to generate the modulated signal 300 shown in FIG. 4. As shown in FIG. 4 and 5, there is one value of 500 TM modulation per carrier period. However, as noted above, there may be two TM-modulation values over the carrier period if these periods can have different areas under the curve, i.e. if there is no need for symmetry between the periods, this ensures the possibility of transmitting two symbols for each period. In this case, there can be two TM modulation values for the carrier period, which corresponds to two different symbols (or information) for the carrier period. This technology may be suitable, for example, for transmission over an optical fiber, since there will be no other signals occupying the transmission frequency band; however, the DC offset is generally not suitable for transmission through other means of transmission.

The variable assigned as the TM modulation period, t _TMM , is the retention time of the TM modulation value, and this time is a multiple of the carrier period. In this case, this implies that the maximum frequency of TM-modulation, f _TMM , is H from the carrier frequency, f _C. In other words, the modulation frequency band is limited to от from f _C , since it is known that the Nyquist frequency or the lower limit of the sampling frequency in the case of discretization of a signal without distortions caused by the superposition of spectra is equal to twice the frequency band of the signal limited in frequency band. However, in the case of the presence of two values of TM modulation for the period of the carrier signal, the maximum frequency of the TM modulation, fTMM, is equal to the carrier frequency, fC. The minimum value of fTMM is missing, including the DC characteristic.

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Turning again to FIG. 2. Reference table. 210 stores quarter periods unique to each TM modulation value. In each period, the carrier contains four quarters of the period (for example, as shown in FIG. 3). If there is an assignment of one digital bit (N = 1), each period of TM modulation will require only two unique levels of TM modulation or two unique pairs of two quarters of the period stored in the reference table. 210 - one level, showing logical 0, and one level, showing logical 1. If there are two digital bits (N = 2) for each, there will be four levels of potential TM modulation. Similarly, if there are three bits (N = 3) for each t _{TMM, there} will be eight levels of TM modulation, etc.

Reference Table 210 contains 2 ^N different quarter-period waveforms, and their total number is 4x2 ^N , since each full waveform consists of four quarter-wavelength waves. The number of time steps or periods of a clock signal (for example, a processor clock signal or a central processing unit for reading reference table 210) for a quarter period will depend on the maximum possible number of disturbances in the waveform that electronic means of implementing the method can withstand. With carrier frequencies of the order of 300 MHz, this may require subnanosecond time steps. Lower carrier frequencies may be more suitable for both TM methods (i.e., a method based on a lookup table and a mathematical method described here) and may be subjected to up-linking to a carrier frequency.

Through block 202, a TM modulation signal is inputted to reference table 210. The TM modulation signal may be a signal containing any number or represented by any number of digital bits (for example, an N-bit signal). Reference table 210 contains values or representations for period quarters that could otherwise be formed using mathematical block 220. For example, for each value of TM modulation that can be represented using lines 210a (for example, from 1 to 2 ^N ), a quarter the period, which can be associated with the TM modulation value and saved, is presented in columns 210b as coordinate data (for example, x, y) for an increasing period of time (for example, from the initial time to 1/4 of the period). Through block 204, a carrier signal is inputted having a carrier frequency f _C. The carrier signal can be an RF signal and can be used as a clock signal. In block 206, a decision is made on how TM modulation will be performed using reference table 210 or using mathematical block 220. Reference table 210 or mathematical block 220 can be used to generate a modulated output signal. If reference table 210 is used , then the quarters of the period associated with the accepted TM-modulated values will be output from reference table 210 to analogue gate 208.

If mathematical block 220 will be used, for example, if block 220 is selected in decision block 206, a TM modulation signal is input to mathematical block 220. Mathematical block 220 generates essentially the same quarter-period signals that would be issued in case using reference table 210 for the same accepted values of TM modulation. However, instead of storing the associated quarter-period values for each value of TM modulation, mathematical block 220 generates a quarter of the period for each received value of TM modulation. Mathematical unit 220 generates modulated quarter periods by forming 180 ° cosine segments with a doubled carrier frequency (2f _C ) and in equivalent quadrants of a carrier frequency of 0-90 °, 90-180 °, 180-270 ° and 270-360 °. Thus, these shaped cosine segments form quarter-period segments at the carrier frequency. The amplitude is specified using the accepted TM modulation values for the 0-90 ° and 180-270 ° quadrants (i.e., the first and third quarters of the period) and the complementary modulation values for the 90-180 ° and 270360 ° quadrants. Specialists in this field of technology is easy to understand that any sinusoidal signal can be formed using well-known mathematical relationships that can be implemented using circuitry and / or software. Thus, in the mathematical block 220, cosine-shaped segments having an amplitude defined by the received value of the TM modulation can be formed.

Mathematical unit 220 performs mathematical calculations to form quarter-period segments using a processor having a clock frequency several times higher than the carrier frequency for executing a program code or for driving a signal-based signal generator; this generator may be any of the known signal generators. Probably, mathematical block 220 will need a higher clock frequency than reference table 210. The output signal of reference table 210 or mathematical block 220 is fed to analog gate 208, which combines a quarter of the period into a single continuous signal and feeds it to local oscillator 212.

- 5 031050

The frequency domain provides insights into aspects of the present disclosure in terms of transmission and heterodyning. FIG. 6 shows a plot of the frequency spectrum of the TM modulated signal 300 shown in FIG. 4, where f _C is the carrier frequency, 2f _C , 3f _C , etc. - the second, third, etc. carrier frequency harmonics. The signal 300 has a spectrum as shown in FIG. 6, at the starting point with visually noticeable kinks in some cases.

In addition to the primary carrier component 610, the third harmonic component 620 of the signal 300, which includes phase modulation, is present. The components of the TM modulation are present only at the third harmonic, i.e. the TM modulation components are third harmonic components 620. The second harmonic signal is missing. By generating a second harmonic signal in block 214 as a local oscillator, using a mixing circuit for heterodyning the third harmonic component, two output frequencies will occur: (3f _C -2f _C ) and (3f _C + 2f _C ). This is shown in FIG. 7. The TM modulation component, i.e. component 620 of the third harmonic, will be shifted down to the main carrier frequency (signal 710). An additional component of heterodyning, i.e. fourth harmonic component 730 may be excluded (for example, using filter 810 shown in FIG. 8) in block 214, and may also be filtered to interface with the output of a dedicated communication channel for transmission.

Unlike the known modulation technologies, in the present disclosure the phase shift of the third harmonic is provided, but this phase shift takes place with respect to the main carrier, and not the third harmonic. With conventional FM and FM transmission, the phase shift occurs in the carrier itself. TM-modulation does not change the main carrier, and only the phase of the third harmonic is correlated with the main carrier.

This distinction is important for several reasons. Each half of the period of the main carrier (i.e. each TM-modulated symbol) corresponds to 1.5 periods of the third harmonic without modulation. The change of the 3rd harmonic occurs only in the case of a change in data (i.e. in the case of a change in the signal of 500 TM-modulation). Thus, there is a very weak effect on power and spectrum, and this is another reason why we have transparency with normal modulation, since in most practical applications there can be 100 or more carrier periods per TM symbol (the number is limited by communication channel), similar to the AM and FM radio broadcast, during which there are no changes (i.e., modulation changes) of the 3rd harmonic. It is simply shifted in phase (in time) relative to the fundamental frequency.

The implementation of the QC method requires an analog signal frequency band, which is three or more times wider than the carrier frequency when the third harmonic is used (for example, 3f _C ). In addition, the QC method requires a clock frequency that is 16 times higher than the frequency of the carrier signal for only four time steps per quarter period. The QC-signal can be formed at a lower frequency of the carrier signal and subjected to up-linking to the desired carrier frequency. A lower carrier frequency will determine the upper frequency limit for the TM modulation value.

FIG. 9a and b show flowcharts illustrating a system and a direct spectrum shaping method (DS method) in yet another embodiment of the present disclosure. DS mode can provide a simpler implementation of TM modulation. In the DS method, the frequency spectrum of the sideband is formed directly and imposes spectrum energy on that which occurs within the bandwidth of the communication channel. The DS method is based on the frequency domain.

According to FIG. 6, a known transmitter performs a certain type of complex modulation. Practical standard types of complex modulation include QAM, QPSK, OFDM, etc. The sideband energy in the known modulation types is represented by component 610 in FIG. 6. The addition of TM-modulation leads to the appearance of the third harmonic, and the side-band energy of the TM-modulation is represented by component 620. It should be borne in mind that the component of the second harmonic may be present, but does not contain modulation.

The second harmonic signal is valuable in that it can be used to reduce the sideband energy of 620 TM modulation to the main carrier frequency 610. This is done by heterodyning using the mixing function, which multiplies the two sinusoidal output signals and generates the difference and additive output signals frequencies. FIG. 7, the shaded areas show the energy that was obtained by converting the third harmonic 720 to the primary 710 and fifth fifth 730 harmonics.

The use of the second harmonic is optional. As is known in the art, a PLL can provide a stable second harmonic. In addition, a possible non-linearity may in practice lead to a down conversion of some part of the sideband energy, but this may be an unstable or unreliable down conversion.

Communication standards require that all transmitters use an output filter to ensure that no energy is radiated beyond the frequency band of a dedicated communication channel. As shown in FIG. 8, the output filter 810 can be used to eliminate harmonics during transmission over a dedicated communication channel. This filter can be a bandpass filter.

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812.

FIG. 9a and 9 show two systems and two methods of directly shaping the spectrum using the ideas disclosed above. FIG. 9a shows a software-based system and method for directly generating spectrum, and FIG. 9b shows a hardware-based system and method for directly generating spectrum. FIG. 9a, a clock signal 910 and a digital modulation signal 920 are input to a microprocessor 901. FIG. 9b, the carrier signal 915 and the analog modulation signal 925 are inputted to the non-linear analog circuit 902. Third-harmonic sidebands (eg, the TM-modulation component 620) are directly generated by the microprocessor 901 and / or circuit 902 based on these input signals. The microprocessor 901 and / or circuit 902 can directly heterodyne the third harmonic sideband 620 using an input clock signal 910 (FIG. 9a) or carrier 915 (FIG. 9b) to directly generate the sideband energy (eg, 710) at the fundamental frequency. The DS-method is based on the software formation of general mathematical expressions or on non-linear analog circuits that implement mathematical expressions. In other words, microprocessor 901 (FIG. 9a) and / or circuit 902 (FIG. 9b) directly calculate and form the third harmonic side band 620 based on the input signals using known mathematical dependencies. Then, the third harmonic sideband 620 is heterodyned using a microprocessor 901 and / or a nonlinear analog circuit 902 to shift the third harmonic side band 620 to the fundamental frequency.

Next will be disclosed systems and methods for receiving and demodulating a signal with transposition modulation. FIG. 10 shows a flowchart illustrating a system and method for demodulating TM signals, which may be referred to as sub-cycle calibration (SCC). The SCC method of demodulating a TM-modulated signal is performed in the time domain by reconstructing the signal, for example, as shown in the section on the QC method (in particular, the signal 300 in FIG. 4).

In the SCC method, the third harmonic is summed with the received signal 1001 in a broadband environment. The PLL 1010 provides an accurate and modulated third-harmonic signal that is summed with the received signal 1001 in element 1020 or multiplied with it. The voltage levels of the positive and negative peaks are then determined using a positive peak detector 130 and / or a negative peak detector 1040, and these levels are used to form a reference linear signal (using a reference linear generator 1050), with a pairing of positive and negative peaks. Thus, in each half-period (i.e., each time a peak is reached) of the received signal 1001, the system is calibrated at the time when a new reference linear signal is generated. This linear signal is re-formed in each half-period of the carrier signal 1001. Peak moments are used using the peak synchronization element 1060 to set the moments of the reference linear signal. The kinks are determined by the detectors 1030 and 1040, and the kink moments are used to sample the reference linear signal, output this signal by the generator 1050 of the reference linear signal, and store the sample values of the linear signal. The resulting voltage is the analog value of TM-modulation; it is issued by the element 1070 sampling and storage and can be used both directly and after conversion into digital form. The reference linear signal has a positive slope in the negative-positive half-period of the carrier. In the next half-period of the carrier (i.e., in the positive-negative half-period), the reference linear signal has a negative slope.

The advantage of the SCC system and demodulation method is that they provide reliable demodulation technology. This is due to the fact that SCC-demodulation is based only on the appearance of positive and negative peaks and on the presence of kinks between these peaks. Because of this, SCC demodulation is much less prone to errors due to noise than other demodulation technologies.

FIG. 11 shows a block diagram illustrating a demodulation system and method that may be referred to as a third-harmonic phase detection (Third-Harmonic Phase Detection (3PD)), in accordance with another embodiment of the present invention. The method of demodulation by determining the phase of the third harmonic (3PD-method) with TM-demodulation is carried out by restoring the component of the third harmonic and demodulating the phase-modulated signal located on this component.

As shown in FIG. 11, the received TM-modulated signal 1101 is used by the PLL 1110 to generate a stable unmodulated signal of the main carrier frequency, which is subtracted from the received signal 1101 by the subtracting element 1120. The output signal of the subtractive element 1120 can be filtered by the notch filter 1130 of the main frequency to exclusion of any spurious emissions at the main carrier frequency. The rest of the signal, therefore, is the energy of the sideband (for example, a component of TM-modulation), by means of which the third-harmonic phase detector 1140 is excited. The third harmonic phase detector 1140 may be any known or standard phase detector. The resulting output signal 1150 is an analog value.

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TM modulation.

FIG. 12 shows a block diagram illustrating another system and method of demodulation, according to another embodiment of the present disclosure. The demodulation system and method shown in FIG. 12 are based on the Fast Fourier Transform (Fast Fourier Transform, TMFFT) with TM demodulation and analyze the spectrum of the sideband.

The TMFFT method can provide the simplest hardware implementation; however, it may also be the most difficult from the point of view of signal processing at the same time. The TM-modulated received signal 1201 is analyzed using the FFT function 1210 after quantization using an analog-to-digital converter. After the signal is amplified by the receiver to a level suitable for converting to digital bits, the signal is sent to element 1210, which may be a processor, such as a CPU computer or a more specialized processor, such as a user-programmable gate array or any custom integrated circuit specifically designed to calculate the Fourier transform. The output of the FFT element 1210 is a set of data values corresponding to the voltage of the received TM signal 1201 at discrete frequencies. The TM spectrum is known because it refers to the TM mode of operation — the number of bits per character (that is, the number of bits allocated during the TM-modulation period) and the transmission rate of characters.

The symbol rate is equal to the quotient from dividing the carrier frequency by the number of periods of the carrier frequency per symbol. In mathematical terms, as an illustrative example: 1 MHz carrier frequency / 10 carrier frequency periods per symbol = 100,000 characters per second.

The frequency of characters is 100,000 characters per second / 2 = 50,000 character periods per second.

Thus, the frequency of interest in this example is 50 kHz, i.e. represents the frequency of symbol periods that is 50 kHz above and below the carrier frequency. A more accurate FFT demodulation process will also be oriented at 100 and 150 kHz to include additional sidebands related to the Bessel function in the case of multiple modulation levels, such as 6 bits per symbol or 64 modulation levels. In addition, in the case of only a small number of periods per symbol, a larger number of sideband frequencies reduces the frequency of demodulation errors. In some receivers, the carrier frequency is heterodyne to an intermediate frequency (IF) in order to amplify or to the base frequency band in which the carrier frequency corresponds to zero.

The output FFT signal at 50 kHz will have a value that corresponds to the TM modulation. If TM-modulation has 4 bits per symbol, then as a result of converting the digital value of the output signal FFT, classified into 16 levels and converted into 4 binary bits, the value of TM-modulation is formed.

FIG. 13 shows a block diagram illustrating a TM transmitter 1300 for generating and transmitting a signal that is the sum of a TM signal and an original signal (eg, a modulated RF signal). The TM transmitter 1300 includes a carrier signal generation unit 1310 and a TM modulation signal processing unit 1320. An illustrative implementation example of a carrier signal generating unit 1310 is shown in block diagram form in FIG. 14, and an illustrative implementation example of the TM modulation signal processing unit 1320 is shown in block diagram form in FIG. 15.

The carrier signal generation unit 1310 is used to obtain low-level samples (eg, using a directional connector 1312) of the original signal 1301 (which may or may not be modulated) and remove any existing known modulation (eg, AM, FM or modulation of any other known type) to obtain a single carrier signal of the main frequency (FC - the main carrier). Existing known modulation can be removed from samples of the original signal 1301 using a band-pass filter stage 1314, which can have a narrow bandwidth selected to exclude known modulations from the carrier signal. The second harmonic generator 1316 generates a second harmonic signal (H2 is the second harmonic), for example, by multiplying the FC signal by itself. Similarly, a third harmonic generator 1318 generates a third harmonic signal (H3), for example, by multiplying FC by H2. It should be understood that the second and third harmonic generators 1316, 1318 can be (or use) any known circuits (or methods) to form harmonics, including, for example, a PLL.

As shown in FIG. 14, the band-pass filter cascade 1314 may comprise a first SAW filter 1313 having a very narrow frequency band, an amplifier cascade 1315, a comparator 1317 for stabilizing the amplitude, and a second SAW filter having a narrow frequency band. Cascade 1311 phase adjustment can be included in the block 1310 of the formation of the carrier signal in order to coordinate in phase and synchronization with the output main TM-carrier using the original main signal (ie FC). The second and third carrier generators can be implemented as multipliers 1316, 1318 signals.

As shown in FIG. 14 and 15, the primary TM carrier and primary carrier of the original signal are marked as subject to phase synchronization (for example, using the phase adjustment stage 1311). This can be done using a feedback loop that has a TM modulation scheme.

- 8 031050 within the large PLL circuit with the aim of matching the phase of the main carrier signal of the output TM signal with the main carrier of the original signal in the adder 1338.

The circuit shown in FIG. 14 is important because it generates signals at frequencies that are completely dependent on the frequency accuracy of the original signal. Signals that are shifted in frequency will not functionally affect TM.

The TM-modulation signal is processed in the TM-modulation signal processing block 1320. The TM modulation is placed on the third harmonic, the frequency is shifted to the main carrier (FC) and added to the original signal 1301.

The TM-modulation signal supplied from input 1302 to the TM-modulated signal processing unit 1320 of the TM Transmitter 1300 is analog in nature and limited in bandwidth (for example, using a modulation low-pass filter 1322 with Nyquist limit) to obtain energy sideband corresponding to the bandwidth of the communication channel. The TM modulation signal is then processed using an inverting optimizer 1324 and a TM modulator (or time shift modulator) 1326. As shown in FIG. 15, an amplifier stage 1321 may be included in the block, and the inverting optimizer 1324 may include a sample and hold function, along with an optimization function. A low pass filter 1322, an amplifier stage 1321 and an inverting optimizer 1324 are used to limit the bandwidth of the TM modulation to the bandwidth of the communication channel. Optimization on / off may be performed depending on the presence / absence of an input signal on the inverting optimizer 1324.

The third harmonic signal (H3) excites the TM modulator (or modulator with a time shift) 1326 by temporal shift of the third harmonic. Thus, the group of side bars of the Bessel function is formed. Only one pair of upper and lower sidebands is required for TM-demodulation. These sidebands are limited in width with respect to the third harmonic by filtering the TM modulation signal prior to the start of the TM modulation process in order to match the bandwidth of the communication channel.

The author of the present invention has confirmed by modeling that the modulation disclosed by the time shift, unlike phase modulation, ensures the formation of only a single pair of sidebands. This was confirmed laboratory using equipment, including an oscilloscope and a spectral analyzer. Phase modulation, as expected, ensures the formation of sidebands in the form of Bessel rows. In contrast, modulation with a time shift ensures the formation of only one pair, consisting of upper and lower sidebands.

The TM modulator (or time shift modulator) 1326 may modulate the time shift using an all-pass filter modified with a voltage-controlled time delay. The control voltage is provided with a transformed TM modulation signal (with or without optimization). The time shift takes place on the third harmonic signal (H3). Although the TM modulator 1326 is disclosed here primarily with respect to modulation by a time shift, it will be clear to those skilled in the art that the TM modulator 1326 may simply be a phase modulator.

Further details regarding the time delay shift schemes, principles and functions that can be used in the TM modulator (or time shift modulator) 1326 will be disclosed. For a single-frequency input signal such as a sine wave, the time delay is similar to the phase shift.

The all-pass filter may contain an operational amplifier with a feedback resistor connected between the output of the operational amplifier and the negative or inverting input of the operational amplifier, a second resistor of the same rating connected between the negative or inverting input of the operational amplifier and a signal input and a positive or non-inverting input connected to the middle contact of the circuit of a series-connected capacitor and resistor, one end of which is connected to the input for the signal, and the other end of it dinen to the ground.

The value of a series-connected capacitor or a series-connected resistor can be changed by a control signal using a four-quadrant multiplier with an output replacing the ground connection of a serial RC circuit, with the first input of a four-quadrant amplifier connected to the middle contact of the serial circuit, and the second input is used as a control input signal (i.e., TM modulation signal).

A time delay shift of the input signal in accordance with the input control signal may result in a phase shift as a time shift.

The system may include a TM on / off selector 13, which selects a third-harmonic unmodulated signal (H3) (eg, the output signal of a third harmonic generator 1318) or a third-harmonic TM-modulated signal (eg, time shift modulator 1326 output). This function retains the full power supplied to the transmitting antenna 1340, depending on whether or not TM is used.

- 9 031050

The TM-modulated signal H3 is down-converted in the down-converter 1330 (or in the local oscillator frequency transfer unit 1330 shown in FIG. 15) by multiplying the TM-modulated signal H3 by the second harmonic signal (H2). As a result, the sideband energy of the TM-modulated signal H3 is shifted to the FC frequency, and then band-pass filtering of the FC is performed using a band-pass filter 1332.

The resulting FC-based TM signal is then passed through amplifier 1334 to provide the usual gain to obtain an acceptable power level. The amplifier 1334 may be, for example, an RF power amplifier. The FC-based TM signal is passed through an end bandpass filter 1338, which limits the bandwidth, and an adder 1338, where the output TM signal is summed with the original signal 1301. The resulting signal is fed to a transmitting antenna 1340 for transmission. There is phase synchronization between the output TM signal and the original signal 1301.

FIG. 16 shows a block diagram illustrating a TM receiver 1600 for receiving a resultant signal 1641 (for example, the sum of a TM signal with an original signal) and extracting and demodulating a T msignal.

The TM receiver 1600, which receives the resulting signal, is located as close as possible to the antenna 1640 or as close as possible to the source of the first IF (intermediate frequency) to be generated (for example, to the output of a known receiver, which is common in some types of communication equipment).

The receiver 1600 includes a carrier signal and harmonics recovery unit 1610 and a TM signal extraction and demodulation unit 1620. An illustrative implementation example of a carrier signal recovery and harmonics block 1610 is shown in the block diagram of FIG. 17, and an exemplary implementation example of the TM signal extraction and demodulation block 1620 is illustrated in the block diagram of FIG. 18.

Block 1610 recovery carrier signal and harmonics in the TM-receiver 1600 contains a circuit that performs the following: (a) restores the original main carrier signal (FC) in the form of a modulated signal; (b) generates a second harmonic signal; and (c) generates a third harmonic signal of the reconstructed main signal. All these signals act as local oscillator signals, except that they are obtained with high accuracy from the received signal. The schemes for performing these actions are similar to those used in the transmitter 1300.

In block 1610, carrier and harmonic recovery, received signal 1641 (for example, the resulting RF signal, which is the sum of the TM signal and source signal) from a receiving antenna 1640 or from the output of the IF (intermediate frequency) of a known receiver (common in some types of communications equipment) filtered using a filtering cascade 1614 with a very narrow band of frequencies in order to eliminate the existing modulation, resulting in the formation of a pure main carrier signal (FC is the main carrier). This FC signal is multiplied by itself in the second-harmonic generator 1616, as a result of which the second-harmonic signal (H2) is generated. The FC signal and the second harmonic signal (H2) are multiplied together in the third harmonic generator 1618, as a result of which the third harmonic signal (H3) is generated.

As shown in FIG. 17, the receiver front end 1600 may have a gain adjustable by an AGC circuit (automatic gain control), and may include a SAW filter 1613 and an amplifier stage 1615. Similar to a carrier signal generating unit 1310 in a transmitter 1300, a carrier recovery unit 1610 and the harmonics in the receiver 1600 may include a comparator 1617 and a second SAW filter 1619. The phase adjustment cascade 1611 may be included in the carrier signal recovery and harmonics block 1610 to compensate for the phase shift in the SAW filter. The second and third harmonic generators can be implemented as multipliers 1616, 1618 signals.

In block 1620, the extraction and demodulation of the TM signal, the received signal 1641 (after processing by the receiver input stage with a gain adjusted by the AGC scheme) is subjected to bandpass filtering using a band-pass filter 1636 with the same bandwidth as the communication channel.

The broadband received signal is then processed by extraction and extraction. The first function is to determine the difference (using an amplifier 1634 with a time delay) between the received signal and the same signal, but with a time delay. The delay is equal to a quarter of the third harmonic period. A time delay amplifier 1634 uses a filter circuit based on a time delay (for example, a delay stage 1633 and a differential amplifier stage 1635), which emits received TM energy.

The selected signal (for example, the difference signal of the fundamental frequency) is up-converted in the boost converter 1630 by multiplying with the signal (H2). In other words, the selected signal is heterodyne to the frequency of the third harmonic of the received main signal (FC). As a result, a third-harmonic signal is generated (after filtering with a band-pass filter 1632 to eliminate the component due to the fundamental product) with TM-modulation. At this point, a change in amplitude may occur due to the modulation of the carrier present and the influence of the transmission medium. The received signal is fed to an analog comparator 1650 with

- 10 031050 common reference level, which generates a signal without amplitude change. By filtering using a band-pass filter 1631, a third-harmonic carrier frequency with TM modulation is selected and other harmonics are eliminated.

The output of the bandpass filter 1632 is not only fed to the analog comparator 1650, as mentioned above, but also used as an input signal in the detector 1628 TM signal, which determines the presence of TM based on the correlation function (i.e., by comparing or determining the correlation between the signal received as the output from the bandpass filter 1632, and the third harmonic signal (H3, adopted as the output from the third harmonic generator 1618). The detector 1628 TM-signal generates a signal indicating whether or not TM is used, i.e. is there a tm signal in the received signal 1641.

The extraction process completes the processing of extracting and extracting the TM signal from the received signal 1641. The extracted signal (for example, the output signal of the bandpass filter 1631) contains a TM modulation that is time-shifted relative to the reconstructed third-harmonic reference signal, derived from the received source signal used as a reference), necessary for the demodulation of the TM information.

The TM-demodulator 1626 demodulates the TM signal by determining the time shift between the third harmonic signal (H3) (taken as the reference input from the third harmonic generator 1618) and the third harmonic signal with TM-modulation (adopted as the input signal from the bandpass filter 1631 ). The TM-demodulator 1626 may determine the time shift between the input signals using the result of multiplying these two signals as a function of the correlation. In another embodiment, the TM-demodulator 1626 may use an exclusive OR function to determine the difference between the reference signal (H3) and the TM-modulated signal obtained as a result of the extraction and extraction processing.

The TM signal extraction and demodulation unit 1620 may, if necessary, contain an inverting optimizer 1624 for reconstructing a signal transmitted by a transmitter having an inverting optimizer 1324, as indicated above.

The demodulated signal is passed through the modulation 1622 low-pass filter in order to eliminate any carrier and other sources of interference, resulting in the formation of the output signal 1602 TM modulation.

FIG. 18 shows the main set of functions for recovering a received TM signal from the resulting signal. The differentiating unit or time delay filter (i.e. delay stage 1633) may have an optimal time delay (.25) / (3 xf _fC ). Deviation from this value simply reduces the level of the allocated TM-signal, although certain values actually reduce the level of the allocated TM-signal to zero. The extraction function only eliminates the amplitude changes of the selected signal.

The time delay based filtering circuit of the separator function in FIG. 18 shows the unique behavior of the frequency response, as shown in FIG. 19. There is a suppression of a periodic signal at DC frequencies, the sixth harmonic, etc.

The receiver of the original signal does not respond to the energy of the sideband of the TM signal. Adding TM to the original signal leads to a decrease in the signal-to-noise ratio (SNR) of the received original signal. Similarly, the effect of the noise of the original signal on the received TM signal occurs.

TM-receiver 1600 uses the signals of the second and third harmonics related to the received main signal. The frequency shift due to the Doppler effect or due to a change in the length of the signal path (movement of the receiver or transmitter) does not affect the demodulation of the TM signal, since all processing is based on the frequency of the original signal.

As described above with reference to FIG. 13-19, TM modulation can be applied to the carrier signal, which can be obtained based on the original signal of the transmitter. The modulated TM signal can then be summed with the original transmitter signal (modulated or unmodulated), resulting in an increase in the width of the information bandwidth of a predefined communication channel. Although FIG. 13-19 were explained above with respect to specific TM signals, this is not intended to be limiting, since the same distinctive features and principles can be applied to any modulated signals. Information can be transmitted as a difference in time or phase angle between two signals of different frequencies.

Thus, methods are provided for increasing the width of the information band of any given communication channel by adding a second carrier signal having a frequency that harmonically refers to the frequency of the first carrier signal and is modulated with information independent of the information modulating the first carrier signal. The second signal and the modulation frequency bands may be heterodyned to the frequency of the first carrier signal, and the modulation frequency band may be equal to or less than the frequency band of the communication channel.

The first carrier signal and the second modulated carrier signal can be transmitted without any changes.

The first carrier signal can be modulated (using any type of modulation) or

- 11 031050 is demodulated, and the second carrier signal can have time shift modulation or angular modulation. The second carrier signal may have a frequency that is in a known relationship with the frequency of the first carrier signal. Similarly, the second carrier signal may have a known ratio in phase angle or time to the first carrier signal. The information to be transmitted may initiate time shift modulation or angular modulation of the second carrier signal, and the modulation means of the second carrier signal may change the ratio in time or phase angle with the first carrier signal.

The time shift modulation or angular modulation of the second carrier signal provides sideband energy that is frequency-shifted to occupy the same frequency band as the modulation (if present) of the first carrier signal, or to fit within the channel frequency band communication that can be used for communication. The combination of the first carrier signal and sidebands of the second carrier signal can be transmitted within the frequency band bounded by the limit frequencies of the communication channel and received by the receiver. Additionally, a combination of two carrier signals can be transmitted without limitation of a frequency band and received by a receiving device.

The receiver may demodulate the modulating information of the second carrier signal using the first carrier signal as a reference signal for demodulation.

Also in the present disclosure, quadrature transpositional modulation techniques or methods of summing the second transpositionally modulated (TM) signal with the original transpositionally modulated signal are provided, resulting in an increase in the width of the information bandwidth within a given communication channel, in addition, provided by transpositional modulation, as disclosed above.

In one embodiment, the second signal of transposition modulation can be summed with the original transposition modulated signal using the frequency of the main carrier signal, which has the same frequency as the original transposition modulated signal, but differs in phase by 90 °. Thus, the property of mutual transparency of transpositional modulation with known types of modulation — amplitude, frequency, and phase — is provided. Adding a quadrature transposed modulation signal also provides mutual transparency between two transpositionally modulated carriers, both of which have transparency with the original signal modulated in the usual way.

Transposition modulation can be used for many purposes, including, for example, optical coupling. Here, methods are provided for applying a broadband transposition modulation signal directly to an optical frequency beam for communication with an increased frequency band of data.

The use of optical rays due to their broadband properties. They are modulated in various ways. Transposition modulation can be applied to an optical beam for transmitting information in various ways, all of which are addressed by and within the scope of the present disclosure.

One such example may be disclosed as follows. The carrier signal is required in any modulation method for transmitting information. Transpositional modulation can use the original modulated carrier signal as a carrier for transpositional modulation. Transposition modulation can also provide a carrier signal if none of these signals are initially present. The transposition modulated signal, regardless of whether the original signal is used or not, is designed to excite an optical modulator.

There are optical modulators ranging from laser diode and optical diode exciters to optical beam modulators that change the brightness or phase of the light beam. In the present disclosure, it is possible to use a transposition modulated carrier signal in a device with one modulator. Thus, it is possible to increase the initial width of the information frequency band, which is used in the known amplitude, frequency or phase modulation.

In the absence of any other modulation, an unlimited bandwidth transposition modulation signal is transmitted, which will contain a third-harmonic component. As a result, the fundamental frequency component will be placed at a lower frequency, at which less attenuation usually occurs. Transpositional demodulation uses this component as a reference. The higher frequency component of the third harmonic has a wider frequency band, which makes it possible to obtain the widest modulation information band.

In another embodiment, two separate optical beams can be used, of which the lower frequency beam is modulated in the usual way (for example, using amplitude or phase modulation) and is also used as a reference carrier for transposition modulation. A higher frequency optical beam is used for transpositional modulation of the third harmonic component.

Another possible application area for transposition modulation is ultrasound.

- 12 031050 communication, such as underwater wireless. For example, in one embodiment, a transposition modulated signal can be fed to an ultrasonic transducer in order to generate an acoustic signal, and this acoustic signal can be received and demodulated in order to restore the original modulating information. The modulation process involves the use of a pathogen with an impedance close to Kulyu, which initiates the operation of the converter beyond its own resonant frequency.

In addition to the wide modulation bandwidth, the receiver can use a natural broadband transducer based on technologies that convert acoustic energy into electrical energy without maximum response at a resonant frequency. Such a receiving transducer is manufactured using MEMS (micro-electro-mechanical systems) technology, which provides a broadband sensitivity characteristic.

One carrier frequency of transposition modulation can be used. Two separate ultrasonic frequencies can be used for separate transmission of the signal component of the main carrier and the signal component of the third harmonic.

It should be emphasized that the variants of the present disclosure described above, in particular, any preferred variants, are merely possible examples of implementation and are intended only to better understand the principles of the present disclosure. Many changes and modifications can be made to the above described variants of the present disclosure without going beyond the ideas and principles of disclosure. All of these changes and modifications are subject to inclusion here within the scope of the present disclosure, represent the disclosure and are protected by the following claims.

Claims (19)

CLAIM 1. The method of increasing the bandwidth of the signal, according to which take the count of the first modulated signal, including the carrier signal, modulated by a signal of non-transposition modulation; remove the signal of non-transposition modulation from the reference of the first modulated signal to obtain a carrier signal; modulate the carrier signal with a transposition modulation signal to produce a second modulated signal; summarize the second modulated signal with the first modulated signal with the creation of the resulting signal and transmit the resulting signal. 2nd harmonic Existing modulation FIG. 7 TM Modulation Geodynamic TM Modulation Signal 730 Amplitude BUT Band-pass filter, necessary to suppress harmonics and other spurious emissions from the transmitter, in front of the antenna. This is a common requirement for all types of modulated carrier transmitters. --------------------- Frequency fc FIG. eight 2. The method according to claim 1, according to which the modulation of the carrier signal by the signal of transposition modulation with obtaining the second modulated signal includes the steps according to which the third harmonic of the carrier signal is formed; modulate the third harmonic of the carrier signal with the creation of a signal of transposition modulation and shift the frequency of the signal of transposition modulation to the frequency of the carrier signal. 3. The method according to claim 2, according to which the second harmonic of the carrier signal is additionally formed, and the frequency shift of the transposition modulation signal to the frequency of the carrier signal includes mixing the transposition modulation signal with the second harmonic of the carrier signal. 4. The method according to claim 2, according to which the modulation of the third harmonic of the carrier signal to create a signal of transposition modulation includes a time shift of the third harmonic of the carrier signal based on the data signal. 5. The method according to claim 1, according to which further match the phase of the second modulated signal with the phase of the first modulated signal. 6. The method according to claim 1, according to which the signal nontranspositional modulation is one of the following: a signal with amplitude modulation, a signal with frequency modulation or a signal with phase modulation. 7. The method of claim 1, further comprising adding a third modulated signal with a resultant signal, wherein the third modulated signal includes a second transposition modulation signal and a second carrier signal, the second carrier signal being shifted in phase by 90 ° with respect to the carrier signal. 8. A method of increasing a communication bandwidth in a given communication channel, according to which a second transpositionally modulated signal is summed with a resultant signal containing a first transpositionally modulated signal and a first main carrier signal, the second transpositionally modulated signal being summed with the resultant signal using the second main carrier signal having the same frequency as the first main carrier signal and having a phase angle of 90 ° relative to the first main carrier signal. 9. The method for increasing the communication bandwidth of claim 8, characterized by one or both 10. A system for increasing the signal bandwidth, comprising at least one processor and a data storage device, connected to said at least one processor and containing instructions stored thereon which, when executed by said at least one processor, condition the execution of said at least one processor operations, according to which take the count of the first modulated signal, including the carrier signal, modulated by a signal of non-transposition modulation; remove the signal of non-transposition modulation from the reference of the first modulated signal to obtain a carrier signal; modulate the carrier signal with a transposition modulation signal to produce a second modulated signal; summarize the second modulated signal with the first modulated signal with the creation of the resulting signal and transmit the resulting signal. 11. The system of claim 10, in which the modulation of the carrier signal by a transposition modulation signal to obtain a second modulated signal includes the steps according to which the third harmonic of the carrier signal is formed; modulate the third harmonic of the carrier signal with the creation of a signal of transposition modulation and shift the frequency of the signal of transposition modulation to the frequency of the carrier signal. 12. The system of claim 11, wherein the operations also comprise the steps of forming the second harmonic of the carrier signal, wherein the frequency shift of the transposition modulation signal to the frequency of the carrier signal involves mixing the transposition modulation signal with the second harmonic of the carrier signal. 13. The system according to claim 11, in which the modulation of the third harmonic of the carrier signal to obtain a signal of transposition modulation includes a time shift of the third harmonic of the carrier signal based on the data signal. - 13 031050 of the following features: (a) double the information bandwidth without increasing the bandwidth of the communication channel; (b) increase the spectral efficiency of the transposition modulation. - 14 031050 202-7 LUT reference table An analog valve that sends the output signal of the reference table further or combines a quarter of a period into a single continuous signal and sends it further. The filter limits the bandwidth of the output signal of the analog gate in order to match the bandwidth of the dedicated communication channel. Yes X move N-bit cash TM-moduli1 Block reference table combinator FIG. 2 clocks on the carrier) _______ frequency f _c _____ Is Λ the correction table used?) Form 180-degree cosine segments with a double carrier frequency of 2fc in equivalent quadrants 0 “-90 °, 90“ -180 ”, 180“ -270 ”and 270’-360’ carrier frequency The amplitude is set using the TM modulation value for the quadrants 0 ° -90 ° and 180 “-270, and the complementary Formation of the second harmonic used for heterodyning the energy of the sideband of the TM modulation to the main carrier frequency (see Fig. 6-8) 216 period and period b period from FIG. 3 14. The system of claim 10, in which the operations also comprise the steps that match the phase of the second modulated signal with the phase of the first modulated signal. - 15 031050 FIG. 6 710 Formed 15. The system of claim 10, in which the signal of non-transposition modulation is one of the following: a signal with amplitude modulation, a signal with frequency modulation or a signal with phase modulation. - 16 031050 FIG. 9 FIG. ten FIG. eleven Received TM-modulated digital signal 1201 Fast Fourier Transform (FFD ------------------------, ---------------- 1 1210 FIG. 12 The output of the digital value of TM-modulation BUT 1220 Decoder for converting FFT to binary 1230 16. The system of claim 10, in which the operations also comprise the steps of summarizing the third modulated signal with the resulting signal, the third modulated signal comprising the second transposition modulation signal and the second carrier signal, the second carrier signal being shifted in phase by 90 ° according to relative to the carrier signal. - 17 031050 1312 FIG. 13 1334 FIG. 14 ¹³ , ³² | 334 ¹³ , ³⁶ | Existing 1338 J j JI signal ^: 2V / V ^ 6 Filter Surfactant Surfactant filter Both band-pass filters have the same bandwidth. as the communication channel TM antenna output (1340)> "- combinator Phase synchronized between two signals 1324 FIG. 15 17. A system for increasing a communication bandwidth in a given communication channel, comprising at least one processor and a data storage device connected to said at least one processor and containing instructions stored thereon which, when executed, are indicated by at least one the processor determines the execution of the specified at least one processor operations, according to which the second transpositionally modulated signal is summed with the resulting signal containing the first transpositional the modulated signal and the first main carrier signal, the second transposition modulated signal being summed with the resulting signal using the second main carrier signal having the same frequency as the first main carrier signal and having a phase angle of 90 ° with respect to the first main signal carrier signal. - 18 031050, --- 1640 1628 FIG. sixteen FIG. 17 Second harmonic from a harmonic recovery circuit Third harmonic from a harmonic recovery circuit The signals in this area occur at the frequency of the 3rd harmonic. 1650 Highlight function The optimizer, filters, analog-to-digital converters and output drivers are excited by the output of the TM demodulator. FIG. 18 - 19 031050 Frequency domain FIG. nineteen