DK158165B - APPLIANCE FOR ELECTRIC MEASUREMENT OF HEAT QUANTITY CONSUMED IN A HEAT CONSUMER. - Google Patents

Description

in

DK 158165 B

The present invention relates to an apparatus for electrically measuring the amount of heat consumed in a heat consumer by measuring the temperature of a heat medium in the heat consumer's inputs and outlets, thereby determining the temperature difference and comparing the voltages proportional to the temperature and partly between a proportional voltage and a reference voltage, said apparatus comprising an analog portion and a digital portion, which parts can be fed from a single battery, and wherein measuring cycles comprising at least one measurement phase and a reference phase are triggered when a pulse which produced by a volume meter determined for the heating medium, and which apparatus further comprises two temperature sensing measuring resistors, an analog / digital converter with integrator and comparator, and where switches are included in the analog part, and a clock generator, counter and control logic are included in the digital share.

Such heat flow meters are known, for example, from German publication publications 28 16 611, 28 01 938 and 27 10 782 or British patent specification 1,546,507. At each of these known circuits 20, a separate reference resistor is coupled in series with each of the two measuring resistors for measuring ten 1 and the outlet temperature, respectively, so as to obtain a measuring circuit. In addition, a voltage divider is arranged to generate the reference voltage. For further processing of the measuring voltage, differential amplifiers including resistors are required. Failure and operation of all these resistors, as well as the deviations of the differential amplifiers, integrator and comparator, are directly included as zero point and / or steepness errors in the measurement result. In particular, due to the long-term operation of the electronic components, errors of significant sizes are produced.

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In addition, the complication is that a resistor circuit with resistors produces system-related measurement errors. This is described in British Patent Specification No. 1,546,507. It is therefore proposed therein that the two reference resistors must be replaced by two constant current generators.

35 However, since each constant current generator consists of faulty and operational electronic circuit elements, the theoretically possible improvement of the measurement can nevertheless not be achieved.

From German Publication Publication No. 28 22 467 a circuit is known

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2 for digital reproduction of the wire value of an unknown measurement resistor. The measurement resistor and the reference resistor are in series via a third resistor so that they are passed through the same current. The voltage drop obtained over the measuring and reference resistors is digitized 5 by means of a double loop circuit, the voltage drop across the reference resistor being first stored in a capacitor. For the measurement of temperature differences, as is necessary for appliances for measuring heat consumption, this publication does not provide any information, and even less so as to correct non-linear characteristics.

The object of the present invention is to improve the apparatus for electrically measuring the heat consumption initially described, so as to minimize error and operating sources, optimum accuracy and durability with respect to the measurement result. This is achieved by an apparatus of the kind mentioned in the preamble, which according to the invention is characterized in that the two measuring resistors during the measurement phase are connected in series with one and the same reference resistor, so that they are passed through the same current, and that the voltage drop across the measuring resistors and the reference resistor is of the switches consecutively leads to an input of the analog / digital converter which, as a double loop circuit, forms the temperature difference of the mutual relationship between the resistors and the resulting voltage drop, the potential existing at a junction between the reference resistor and the measuring resistors the zero potential of the circuit during the measurement phases.

Thereby, the advantages are obtained that the two measuring resistors for sensing the supply and return line temperatures are connected in series with one and the same reference resistor, whereby only this reference resistor is responsible for the accuracy and stability of the measurement results. The minimum amount of error sources obtained by the invention cannot be reduced since at least one reference source is needed in all measurement circuits. Changes in the electrical values of the other 35 circuit elements as well as decreasing battery voltage over the years do not affect the accuracy and stability of the measurement result.

According to an advantageous further development of the invention there is an impedance converter coupled amplifier whose output forms

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3 zero potential shine for the circuit. The possible low-ohm connection of the zero potential rail is particularly advantageous when the double loop circuit is designed with an automatic displacement comparison device, since the displacement storage capacitor, which can be charged in a shorter time.

A further acceleration of the zero comparison phase is achieved by temporarily paralleling a lavohmetic resistor to the integration resistor. The acceleration of the execution process achieved by this measure is a prerequisite for a sparse battery for use.

A preferred embodiment of the invention is the subject of claim 6. Here, the two measuring resistors and the reference resistor are constantly connected in series and thus always pass through the same current. The resulting voltage drop is intervened in each capacitor in a sample and holding circuit and is subsequently evaluated sequentially in the double loop circuit to generate the measurement result. Here, too, the reference resistor is solely responsible for the accuracy and temporal stability of the measurement result.

For correcting the measurement values depending on the density and enthalpy of the heating medium and the characteristics of the measuring resistors, a preferred embodiment of the invention is claimed in claim 8, wherein during the reference phase a correction current is driven in the integration capacitor which correction current is dependent on the difference of the measuring resistor. Thus, the voltage drops taken over the measurement resistors also serve to form the correction current.

The invention will now be explained in more detail with reference to the drawing, in which fig. 1 is a circuit diagram of a heat flow meter according to the invention; FIG. 2 shows the voltage-time diagram of the integrator output of FIG. 1, FIG. 3 is a further circuit example of a heat flow measuring apparatus according to the invention; FIG. 4 shows the associated voltage-time diagram for the integrator output 4 in FIG. 3, FIG. 5 shows another circuit example; FIG. 6 shows another circuit example where the deviations cancel each other out; FIG. 7a and 7b show the associated voltage-time diagrams; 8 shows an additional circuit using storage capacitors.

In the circuit shown in FIG. 1 there are two measuring resistors 1,2, where the measuring resistor 1 can be in the return line and the measuring resistor 10 2 in the flow. The conduit in a heating system. In series with the two measuring resistor stands 1 and 2 a reference resistor 3 is connected. A switching group 101,102,111,112 switches the measuring resistors 1,2 alternately to a supply line 64 for the battery voltage Ug minus pole. Additional switches 103,113 provide a corresponding connection 15 of the reference resistor 3 to the supply line 63 for the battery voltage Ug plus pole, respectively, to the input of an analog / digital converter.

At the switching point 61 between the reference resistor 3 and the measuring resistors 1,2 is connected the input of an impedance converter coupled 20 amplifier 5. The output of the impedance converter 5 forms the zero potential of the measuring circuit 62.

The analog-to-digital converter consists of an integrator 7 with an integration resistor 41 and an integration capacitor 27 as main components, a zero voltage amplifier 8 with an operational amplifier connected via a resistor 47 and a comparator 9. Additional digital circuit means are connected to the comparator 9 in a known manner. , which processes the measurement result until it is finally reproduced on a reproducing means 17.

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For automatic zero correction of the analogue / digital converter 7,8,9 there is a zero correction capacitor 49 which during the zero correction phase at the closed switch 114 is charged to such a voltage that, under compensation for the shear voltages of all 35 operational amplifiers, the output of the zero voltage amplifier 8 as at the zero potential of the rail 62, i. zero.

After the zero-correction of the analog / digital converter 7,8,9 now happens

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5 first an upward integration of the measuring voltage of the measuring resistor 1, the control logic 11 making sure that this upward integration is terminated after N pulses generated by a clock generator 12. In addition, a downward integration of the reference voltage of 5 the temperature-independent reference resistor 3. occurs. always with the same increase and terminate after nine pulses as soon as the voltage of the integration capacitor 27 has reached the value zero and the comparator 9 stops the control logic 11.

The impulses N and n, respectively, are counted by a target time counter 13, for example by a forward / reverse counter. After a complete integration process, a certain impulse number remains on the measurement time counter, which is proportional to the difference between the resistance values of the measurement resistor 1 and the reference resistor 3.

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After a further zero correction phase, an upward integration of the voltage drop across the measurement resistor 2 occurs for a period corresponding to N pulses and a subsequent downward integration of the voltage drop across the reference resistor 3 for a period of n2 pulses. The impulse-difference n2 - n1 is a measure of the resistance difference R2 - Rj and thus the temperature difference between the supply and return lines.

In order to compensate the temperature with respect to the non-linearities of the measurement resistances 1,2 and the dependence on the singular pi and density of the heat carrier, the numeral n2 - rij is corrected. To this end, the voltage is applied to the output 67 of the integrator 7 via a resistor 43 and is fed to an impedance converter 10.14. When the pulse number n2 is equal to the pulse number np, the switch 110 is closed so that a correction current corresponding to the difference of the measurement resistance values via the resistor 45, 30 is fed to the integrator 7's summing point 66. This correction current is not constant and allows a near complete linearization of the relation between the actual amount of heat and the number of talents.

35 The zero correct ion can be used in the circuit of FIG. 1 due to the relatively long charging time required for the capacitor 49 due to the oscillation of the amplifier coupling 7,8 and the suppression of the oscillation, cause some difficulties which can be avoided with that of FIG. 3. This circuit has it

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6 further advantage that control logic 11 can be performed simpler and more safely. In this circuit, the zero correction capacitor 49 and thus the resistors 46,27 and the switches 113,114 lapse. Resistors 29,30,31,32,33 and a switch 107 then replace them.

5 Before the beginning of the measurement, the switch 107 is parallel or connected to the integration capacitor 27 to avoid incorrect couplings.

At the beginning of a measurement, which is triggered by the arrival of a pulse on the control logic over the quantity counter contact 52 and the resistor 51, the measuring resistor 1 in the return line is measured with the drive voltage -and from the supply line 64 via the switch 101. The switch 107 is opened and the switch 111 is closed. The upward integration of the measuring voltage resulting from the reflux temperature takes place. When the output voltage of the output of integrator 7 after time t ^ (MR = measured voltage in return line) reaches the voltage at the summation point 66 (Fig. 4) determined by the magnitude of the resistors 29,30,31 on the threshold voltage switch 4 and the voltage at potential rail 62 and in lead 63, the voltage at the threshold of the voltage switch 4 is switched from the potential of the lead 64 to the potential of the lead 63 and supplies an impulse to the control logic 11. Thus, the switch 101 is interrupted and the switch 103 circuited. The reference voltage 3 above the reference resistor 3 is applied to the integrator 7, which now integrates downwards. After time t ^, the voltage at summation point 66 goes through zero, the voltage at the output of comparator 9 is switched from the potential of the lead 64 to the potential of the lead 63, and in turn gives an impulse to the control logic 11.

Now switch 103 is again disconnected and switch 101 is closed, whereby the regular upward integration of the reflux temperature derived from the measured voltage is made under N clock pulses. Following the impulse transmitted from the measurement duration counter 13 to the control logic 11, switch 101 is interrupted and switch 103 is closed. Thus, the downward integration of the reference voltage associated with the reflux or return measurement occurs, which, for example, results in a new switching of the comparator 9 giving nine clock pulses. These nine pulses are stored again in the memory 14.

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Now the switches 101,103 are switched off and the switches 112,102 are closed. The voltage resulting from the ambient temperature of the measuring resistor 2 is integrated upwards. When the voltage at the summation point 66 reaches the voltage again at the output 67 of the integrator 7, the threshold switch 4 switches, and at its output an impulse arises at the transition from the potential of the lead 64 to the potential of the lead 63, which impulses the control logic 11 and causes it to interrupt switch 102 and terminates switch 103. Here, again, the voltage at the summation point 66 to the integrator 7 again becomes integral, which at zero throughput (potential rail 62) again causes switching of comparator 9 and influence of control logic 11 and switching of switch 103 and switching on. of the switch 102.

15 It now integrates proportional measuring voltage with the flow temperature. After N clock pulses, from the measurement duration counter 13, a pulse is output to the control logic 11, thereby switching switch 102 and closing switch 103. Downward integration is performed on the reference voltage associated with the flow temperature measurement. Since the measuring resistor 2 in the forward direction is greater than the measuring resistor 1 in the return direction, the numerical value of the clock pulses n2 becomes greater than nj. When the value n ^ stored in the memory 14 is exceeded, the comparator 15 is actuated and activates the control logic 11 so that the clock pulses n2 - n ^ are counted into the result counter 16. At each overflow in the counter 16, the reproducing means 17 is counted further.

In the apparatus described in FIG. 1 and FIG. 3, the integration times are still relatively long. However, it would be desirable, especially in battery operation, that only the result counter 16 and the receive portion 16 controlled by the 30 count counter 52 in the control logic 11 must be continuously engaged while the analog portion and the remaining digital portion 11,12,13,14,15 need only be switched on during the actual measurement time via a switch 115 controlled by the control logic 11 which is in place of the wiring connection 116 at the end of the quantity counter -35 switch 52 to make the integration times as short as possible.

This is made possible by the one shown in FIG. 5 shows further development of the invention.

The upward integration therefore lasts so long because one integrates over

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the whole resistance value. For example, to form a difference of 12 ohms between assumed 120 ohms and 132 ohms for n2 - nj, as already stated in connection with FIG. 1/2 and FIG. 3/4 the one with the "dead" n ^ proportional time n ^. r, which corresponds to the resistance 120 ohms, is passed twice, (r = the time distance between two pulses). If instead of zero to the measured value, conversely, the difference from the highest expected value to the measured values can be measured, the integration time can be made considerably shorter.

10 This idea is realized with the circuit of FIG. 5. The measuring resistors 1,2 are connected in series over the resistors 23 or 24 connected to each resistor belonging to each resistor, e.g. in the form of field effect transistors, one after the other by means of the switch 102 or 105 with the reference resistor 3. This measuring chain is fed with the voltage Ug of the leads 63,64. The measuring chain is connected in parallel with the voltage divider 21,22, which is also connected to the voltage Ug. The zero potential point 61 gives the reference voltage zero which is passed over the voltage amplifier 5 to the potential rail 62 via the voltage amplifier 5. The differential amplifier 6 is connected with its non-inverting input directly to the potential rail 62 and with its inverting input to the potential point 68, i.e. to the connection point between the reference resistor 3 and the controlled resistors 23,24, each of which is switched on each time during a measurement run. Since the differential amplifier 6 has a high gain, e.g. 2.

5 25 10, and since the required control voltage at its output 69 for the controlled resistors 23,24 varies only a few hundred mV, the error between the voltages on the potential rail 62 and the potential point 68 on the input of the differential amplifier 6 is only on the order of fractions. of / tV to some / iV. At the voltages proportional to the temperatures over the measuring resistor 1 or 2 respectively over the controlled resistors 23,24 of several mV per meter. Thus, this gives rise to errors only in fractions of 10 "3 ° C.

The start of the measurement cycle is similar to that described in connection with FIG. 3 and 4.

If temperatures are to be used between 20 ° C and 100 ° C, and if e.g. the ratio of resistors 21.22 is dimensioned so that even 150 ° C could be measured (at R2 = 160 ohms and R24 = 0

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9 ohms), in the above example, only resistor differences of 160 - 120 = 40 ohms and 160 - 132 = 28 ohms (relative to 120 and 132 ohms respectively) are obtained, so that the integration time is only one quarter of the integration time of the circuit of FIG. 3. The voltage divider 21,22 is not critical and does not cause any measurement error. It is only required that the ratio ^ 1 ^ 22 ^ 0Γ ^ ινθΓ be stable during the measurement process.

FIG. 6 shows a greatly truncated measurement method by which all 10 shear sizes are avoided. The analogue / digital converter is here a saw converter. The measurement process is explained in connection with FIG. 6 and 7.

A quantity pulse from switch 52 engages control logic 11 over resistor 51. Here, it contains the receiving means for the switching orders from comparator 9, control means for switches 301-307 and 315, and a clock generator for analog / digital conversion. Switch 315 brings the analog portion to voltage + Ug. Above resistors 21.22, a voltage split occurs. The partial voltage 61 is fed to the amplifier 5 "coupled as a constant current source. From the output, the current flows across the switch 301, the measuring resistor 1 and the reference resistor 3 to -Ug.

At the output of the amplifier 5, the voltage 62, which serves as the reference voltage for the integrator 7, is adjusted. The various potential levels are shown in FIG. 7a.

The integration of the reference voltage takes place across the resistor 41 and the capacitor 27. The output voltage 67 of the integrator 7 reaches the magnitude of the travel of the voltage 69 connected to the comparator over the switch 302 and the comparator 9 is switched from -Ug to + Ug. This signal is fed to the control logic 11 upon which the switches 301 and 302 open and the switches 305 and 306 are closed. The voltage across the reference resistor 3 does not change because the current is constant. Above switch 306, voltage 70 is now fed to comparator 9. At the same time, the clock generator is switched on and connected to the result counter 16. Further, over switch 307 and resistor 45, the correction voltage generated by resistors 43 and 44 and amplifier 10 is applied.

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10 the integration summation point 66. A further upward integration occurs until the voltage 67 becomes as large as the voltage 70. The comparator 9 is again switched from -Ug to + Ug and the control logic 11 completes the process.

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The number of n clock pulses applied to the result counter 16 is proportional to the difference between the resistors 1,2 divided by the reference resistor 3.

10 In a modified embodiment of the described process, during the first phase, the switches 303,304 may be terminated by the control logic 11 and thus in parallel connect the resistors 53 and 41 and in series connect the resistor 54 with the measuring resistor 1. The resistor 54 reduces the voltage 69 to a voltage Ugg - AU.

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The comparator 9 is switched when the voltage at the output of the integrator reaches the value Ugg - AU. This occurs at the time ty, which is only a fraction of the time t, because the integration time at closed switch

A

303 is shortened by resistor 53.

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Upon the first switching of comparator 9, control logic 11 causes switches 303,304 to break again. Everything else happens as already described above and shown in fig. 7b.

FIG. 8 shows a circuit in which the two measuring resistors 1,2 and the reference resistor 3 are continuously connected in series and are directly connected between the supply 63 of the battery voltage Ug plus pole and the zero reference potential of the rail 62 in the measuring circuit. Parallel to each resistor 1,2,3, a capacitor 221,222,223 is connected over a switch 203 - 207 and 208 - 211, respectively, in a known manner as a sample and holding circuit. The capacitors 221, 222 parallel to the measuring resistors 1,2 can be coupled so that the voltage difference is formed directly.

35 The zero potential of the rail 62 is increased by means of an impedance converter coupled amplifier 5 and a voltage divider with the resistors 21,22 in relation to the negative battery voltage -and on the supply 64. This potential displacement is necessary to prevent operational amplifiers 5 ', 7, 9 works with DK 153165 B π unauthorized operating mode.

The circuit of FIG. 8 works as follows. Each quantity pulse from switch 52 is passed over resistor 51 as starting pulse to control logic 11. This controls switch 115, which connects the on-coil (+ υβ) of the battery voltage υβ with the operational amplifiers 5.5 ', 7 and 9 in the analog portion.

The control logic 11 controls a series of analog switches 201-215. The 10 contains a clock generator and a measurement duration counter.

A measurement cycle occurs in three stages. Phase 1: Switches 201-207 and 214 and 216 are disconnected. Switches 201 and 202 cause the automatic zero adjustment. Switch 214 shortens integrator 7's time constant for the duration of the zero adjustment. By parallel coupling a resistor 241 across switches 203-207, the voltage drops across resistors 1,2,3 are stored in capacitor 221,222 and 223.

Phase 2: In this phase, the proportional voltage and the associated integration voltage are formed with the temperature difference. Switches 201-207 as well as 214 and 216 are opened, and now switches 208-210 and 215. The negative terminal of capacitor 222 is connected to the reference potential rail 62. The positive terminal of capacitor 222 is connected to switch 209. the positive pole of the capacitor 221. The negative pole of the capacitor 221 is connected above the switch 210 to the input of the amplifier 5 '. The potential of the input of the amplifier 5 'is the negative difference of the voltages across the measuring resistors 1 and 2 and thus proportional to the difference between the two measurement resistance values.

The amplifier 5 'depicts this differential voltage at its output 1: 1, and over the integration resistor 41 it is integrated into the integration capacitor 227. This integration voltage at the end of phase 35 2 is consequently proportional to the measurement resistance difference.

Switches 215 and 216 serve to prevent counterfeit effects of any creep currents on the shear voltage stored in capacitor 225.

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Phase 3: In this phase, the reference voltage is integrated. At the same time, a correction of the integration current with a current dependent on the resistance between resistors 1 and 2 is made. Switch 210 is opened and switches 211,212,213 are closed. Switches 208,209 and 215 remain closed. The voltage (-Uy -UR) at the footpoint of capacitor 221 and switch 212, respectively, is maintained. Furthermore, the result counter 16 is interconnected with the clock generator in the control logic 11. The voltage URe ^ across capacitor 223 is passed across switch 211 to the input of amplifier 5 'and causes downward integration in the integration capacitor 227.

On the positive current driven from the integration resistor 41 into the integration capacitor 227, a further negative (correction) current is superimposed on the switches 212 and 213 as well as the resistors 245 and 246. Thus, the overall integration current becomes, the greater the resistance difference between the measuring resistors. 1 and 2, respectively, the greater the temperature difference becomes.

20 When the voltage across the integration capacitor 227 or at the output of the integrator 7, respectively, becomes zero at the integration of the reference and correction currents, the comparator 9 kicks in and outputs a stop signal to the control logic 11. The switch 115 again disconnects the voltage + Ug from the analog part and the clock generator is switched off. Consequently, in the score counter 16, the number of beats has been inserted in phase 3.

At each overflow of the result counter 16, the control logic 11 switches the indicator 17 a step further.

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Claims (10)

1. Apparatus for electrically measuring the amount of heat consumed in a heat consumer by measuring the temperature of a heating medium in the heat consumer's inputs and outlets, thereby determining the temperature difference and comparing the voltage proportional to the temperature and partly between the temperature the trip difference is proportional voltage and a reference voltage, which apparatus comprises an analog portion and a digit portion, which portions 10 can be fed from a single battery, and wherein measuring cycles comprising at least one measurement phase and a reference phase are triggered when an impulse generated of a volume meter determined for the heating medium, and which apparatus further comprises two temperature sensing measuring resistors (1,2), an analog / digital converter with integrator (7) and comparator (9), and switches (101,102,103) included in the analog part, and a clock generator, counter and control logic (11) are included in it. digital part, characterized in that the two measuring resistors (1,2) during the measurement phase are connected in series with one and the same reference resistor (3), so that they pass through the same current and that the voltage drop across the measuring resistors (1,2) and the reference resistor (3), by means of the switches (101,102,103), is successively fed to an input of the analog / digital converter which, as a double loop circuit (7,27,41,46,9), forms the temperature difference of the relationship between the resistors and the above voltage drop, the potential present at a node (61) between the reference resistor (3) and the measuring resistors (1,2), generates the zero potential of the circuit during the measurement phases. Apparatus according to claim 1, characterized by an amplifier (5) coupled to an impedance converter, the output of which forms the zero potential rail of the circuit (62). Apparatus according to claim 2, characterized in that the input of the amplifier (5) is connected to the node (61) of the measuring and reference resistors (1,2,3). Apparatus according to claim 1,2 or 3, characterized in that the double loop circuit (7,27,41,46,9) is supplemented with a device for automatic balancing and comprising a DK 158165 B balancing capacitor (49), a zero amplifier ( 8) with associated resistors (47,48) as well as a switch (114) for charging the balancing capacitor (49) during a zero equalization phase. Apparatus according to any one of claims 1-4, characterized in that the two measuring resistors (1,2) are alternately connected in series with the reference resistor (3) and to the input of the analogue / digital converter by means of the switches (101,102,111,112). Apparatus according to any one of claims 1-4, characterized in that the two measuring resistors (1,2) and the reference resistor (3) are continuously connected in series, that each of these resistors (1.2.3) is connected to a capacitor (221,222,223) operating in a manner known per se as a sample and holding circuit by alternately parallel coupling to the resistors (1,2,3) via the switches (203-207 and 208-211, respectively) or connected to zero potential (62) analog / digital converter input stage (5 ') and during the sample phase when switches (203-207) between resistors (1.2.3) and capacitors (221,222,223) are closed and below 20 zero equalization phase is partly the input stage (5 ') of the integrator (7) is connected to the zero potential rail (62) via a switch (201), partly the zero conductor of the integrator (7) is connected via a switch (202) and partly an integration resistor (41) is connected in parallel to a with a resistor (241) by means of a switch (214). 25 Apparatus according to claim 6, characterized in that the capacitors (221, 222) associated with the measuring resistors (1,2) are connected in series to the measuring resistors (1,2) but connected to zero potential (62) and analogue / the input stage (5 ') of the digital converter 30, on the other hand, antiser series coupler so that the resulting voltage (on the switch 210) has opposite polarity relative to the voltage across the capacitor (223) of the reference resistor (3) (on the switch 211). Apparatus according to any one of claims 1-7, characterized in that for the correction of the measurement result depending on the density and singular pi of the heating medium and depending on the characteristics of the respective measurement resistance, there is at least one resistance (45; 245,246; 28 ), which can be switched on and off via at least one switch DK 158165 B (110,212,213,307,108) during the reference phase to drive a correction current dependent on the difference of the measuring resistors (1,2) through the integration capacitor (27,227). Apparatus according to any one of claims 1-8, characterized in that in the double loop circuit between the integrator (7) and the comparator (9) a threshold voltage switch (4) is switched on, which, when exceeding a voltage threshold, emits a impulse to the control logic (11), which thereby connects the measuring resistors (1,2) to the input of the integrator (7) instead of the reference resistor (3) via the switches (101,103,102,112) 10. Apparatus according to any one of claims 1-9, characterized in that in the balancing branch of the analogue / digital converter, a further switch (215) is connected to the switch (202) which supplies balancing current, which further switches. (215) during the measurement phase, the balancing step switches to zero potential (62), and a third switch (216) is arranged to break the connection to the balancing capacitor (225). 20 Apparatus according to any one of claims 1-10, characterized in that a differential amplifier (6) with controllable resistors (23, 24) is switched on to switch the integration direction in the measuring input of the circuit. 25 30 35