DE4407369A1 - Transition time measurement method - Google Patents

Abstract

The method involves using a signals transmitter and receiver (1) with a envelope slope detector (5) and a quadrature modulator (20). The transition time is first determined approximately and finally a correction value, taking into account the exact transition time, is determined. The phase angle of the pulse is measured and the correction value derived from a fraction of the carrier frequency (fT) defined by the phase angle. The defined time point can be defined by a preceding pulse modulated pulse with the same carrier signal. The phase angles of the two pulses can determined and their phase difference formed. The correction value can then be determined from a fraction of the carrier frequency defined by the phase difference.

Description

The invention relates to a method for measuring the transit time according to the features of the preamble of claim 1 and a circuit arrangement for performing the runtime measurement solution and its use.

Methods for measuring the transit time are generally known and are becoming a non-contact distance mood and used to locate objects. Essential With these pulse runtime methods, the transmission is a pulse-modulated signal of a certain frequency and its reception after reflection on a target object. When The duration is determined as a measure of the distance to be determined telt that only for distance determination with the corresponding propagation speed, which depends on Transmission medium is, must be multiplied. Depending on Frequency range of the modulated carrier signal of the outside A differentiation is made between different forms of Pulse transit time measurement, such as B. ultrasonic transit time solution or microwave transit time measurement or radar transit time Measurement.  

Such a transit time measurement is used, for example se for determining fill levels in containers, for removal measurement in cameras, medical diagnostic devices as well as for positioning tasks in automation technology nik.

To extract the runtime information from the existing ones There are different methods for sending and receiving signals. The simplest procedure is to monitor one determined amplitude threshold. Will this threshold exceeded in the transmission phase, for example starts a counter which, after exceeding the threshold value the received signal is stopped again and thus a measure for the searched distance. However, since most Systems with relatively low signal bandwidths work and thereby the increase or decrease in amplitude in the transmit / emp Catch pulse relatively slowly over several periods of Trä vibration occurs, but at the same time the amplitude de of the received signal depends on the target distance and Shaft damping can change greatly due to the Use a fixed threshold in the Runtime determination often significant errors.

It is therefore preferred to use methods that consist of the envelope for the electrical transmit and receive pulses recover and on the rising or falling edge this envelope when exceeding or falling below one Detection threshold start or stop the time measurement. If the amplitude value of the detection threshold in a fixed relationship to the flank associated maximum of Envelope held, so the problem described above can can be solved with fluctuating reception amplitude.

However, a problem with this method is a behavior  moderately slow slope increase of the envelope due to the desired low system bandwidth. Because of the slow Rising or falling edges of the envelope have an effect namely low interference signals, e.g. B. in the form of noise the envelope flank immediately as a measurement error by the Time of exceeding the detection threshold ver push.

One way to keep the system belt constant broad signal components that are higher in frequency than the modula tion frequency of the transmitted and received pulses are to be used for more precise determination of the term, is the additional exploitation of the carrier vibration, the Frequency regularly several times higher than the amplitude is the modulation frequency.

Such a pulse transit time measurement method, which also Features of the preamble of the present patent claim 1 is known from EP 0 324 731 B1. At that one The procedure for measuring the transit time is described first the beginning of the falling edge of the envelope of the pulse recorded in order to define a reference point in time. As a reference The point in time serves to occur after the maximum of the envelope end first peak value of the pulse-shaped modulated pulse ses. The reference point to be determined is the reference point Roughly roughly run time at a given time Right. To determine the runtime exactly, is about also provided the occurrence of the first zero crossing to be recorded after this reference date. At first only the time therefore becomes approximately predetermined span between the reference time and the occurrence of the zero Passage added. Will be as a given time the first zero crossing of a transmission pulse after the first Peak value in the falling edge of the envelope of the  Transmission pulse selected, an exact runtime or Distance measurement done.

This supports the ver described in EP 0 324 731 B1 drive on a transit time measurement that is a zero crossing the carrier oscillation of the pulse-modulated pulse is detected, which was previously limited in time. Because of this The measuring accuracy can be used to measure the transit time increase.

However, it has been found that this is known Procedure then leads to incorrect runtime measurements if the zero crossing to be evaluated by interference signals, such as. B. Noise or those occurring when measuring distance False echo signals is falsified or not at all is to be detected.

The present invention is therefore based on the object de to specify a method for measuring the transit time which just if the carrier oscillation of the pulse-modulated pulse exploits, but then also a high measuring accuracy returns when the zero crossings of the carrier wave are not Detect more or no longer exactly due to interference signals are animal. In addition, a circuit arrangement to carry out such a procedure and a use be specified for such a runtime measurement.

This task is carried out for the process by the characteristics of Claim 1 solved.

The invention is essentially based on both Amplitude information of the pulse modulated pulse as well take advantage of its phase information. In the invention Procedure for measuring the transit time between a given one  Time and one with a carrier frequency signal pulsmodu gated impulse, similar to the prior art, initially the runtime is roughly predetermined and then a correction value taking into account the exact running time determined. In contrast to the known transit time measurement however not a single zero crossing of the pulse summarizes, but measured the phase angle of the pulse and the Correction value according to the invention from a by the measured Phase angle determined fraction of the carrier frequency of the Carrier frequency signals determined.

Although this method according to the invention is suitable in principle for determining the transit time of a pulse-modulated pulse at a predetermined zero time, the method according to the invention is ideally suited for determining the transit time between two pulse-modulated pulses, as occurs in the distance measurement. For this purpose, the pulse-modulated transmit pulse with a carrier signal having the carrier frequency via a Kopplungsein direction, for. B. an antenna, applied to a test section. The transmitted pulse reflected on a target object arrives as a received pulse in its amplitude due to the transmission path attenuated and delayed to a receive device. First, the transit time is roughly predetermined in any way, advantageously to ± ¼ λ _{T of} the carrier frequency, and then the correction value to be taken into account for the exact transit time is determined. To determine this correction value, the phase angles of both pulses are determined and a phase difference angle is calculated from the two phase angles. The correction value is finally determined from a fraction of the carrier frequency determined by the phase difference angle.

The initially rough predetermination of the term takes place in  a development of the invention based on an envelope tection of the pulse or pulses, whereby for envelope detection in a manner known per se the pulse is rectified and an envelope evaluation circuit is supplied.

In addition, it is provided according to the invention for covering curve evaluation a quadrature demodulation, which is also in the In connection with the phase angle detection can perform. For the envelope detection a digital maximum search of the pulse by the pulse undergoes quadrature demodulation, the Maximum by the sum of the squares in the Qua draturdemodulation resulting 0 ° output signals and 90 ° output signals is determined. Calculating the maxi The values are then expediently carried out by a micro computer. You only need to do this in suitable memories the quadrature output signals previously stored digitally demodulation of the pulse filed for a measurement cycle become. This allows simple quadrature demodulation the rough distance required between send and receive determine pulse.

Although the phase angle to be determined is the one to be evaluated pulse modulated pulse in any way be can be voted, it has proven to be appropriate Phase angle of a pulse over the entire pulse length measure and average out. This allows the Meßge accuracy can be further increased.

To determine the phase angle is in a training the invention a quadrature demodulation of the to be evaluated Provided with subsequent low-pass filtering, where reference carrier signals for quadrature demodulation can be selected that exactly the carrier frequency of the modulated  Have impulses. The phase angle you are looking for is determined in this quadrature demodulation from an arctangent bil the quotient of the quadrature demodulation and Low pass filtering resulting 0 ° output signals and 90 ° - Output signals. The low-pass filtering means that the high frequency components resulting from quadrature demodulation suppressed so that on the output side for the duration of the pulses a DC voltage is available, the amplitude of which is only still of the phase shift between the carrier vibrations supply of the pulse and the reference carrier signal of the Quadra turdemodulation depends.

According to the invention, the frequencies are the reference carrier Quadrature demodulation and carrier signals equal to the impulse. For example, a ge common oscillator device are provided, from the Output signal both the carrier signal or pulses as well as the reference carrier signals for the quadrature demodul lation can be derived. This ensures that the frequency of the carrier signal of the pulses and the frequency of the two for mixing in quadrature demodulation used reference carrier signals are exactly the same. Through the Use of a common oscillator device also achieved that the phase relationship of the two Reference carrier signals related to quadrature demodulation remains the same for all measuring cycles.

An increase in the signal sensitivity of the transit time measurement results according to the invention by the 0 ° output signals as well as the 90 ° output signals of several successive gender measuring cycles are averaged. Through this middle of the bar ment or integration of the two output signals each Both the phase and the remain separate Get amplitude information of the echo signal to be evaluated  Subsequent envelope formation after the relationship

over the entire measuring cycle, with Q being the 0 ° output signal and I the 90 ° output signal of the Quadra turdemodulation leads to the well-known envelope ven echo signal, but compared to one by usual Envelope formation, such as. B. Two-way rectification, won an echo signal has a higher signal-to-noise ratio, making echoes with a very small amplitude easier to detect animals are. Compared to share averaging or integer gration of z. B. obtained by two-way rectification Envelope, which also increases the signal-to-noise ratio that causes the gain in signal sensitivity through the averaging of the 0 ° output signal and 90 ° output signal separated and final envelope determination according to the relationship mentioned with the same number of telten measurement cycles significantly higher because of the fiction according to the procedure also to evaluate the phase information of the the echo signal using quadrature demodulation for the Mit tele value formation is retained.

The method according to the invention for measuring the transit time can be advantageously for distance measurement and in particular use them for level measurement in containers, whereby a pulse-modulated transmission pulse into an interior of a container ters and after reflection on a target object as Receive pulse or echo pulse in a receiving device Will be received. The according to the inventive method exactly determined transit time between transmit pulse and receive impuls is used to determine the distance with a given Propagation speed by the transmission medium is multiplied. By the exact registration of the Runtime between the two pulses is an exact distance measurement possible.  

Finally, the inventive method can be in also advantageously used for distance measurement, at which false echoes occur. So according to the invention Interference echo signal with amplitude, distance and phase values ten stored and from a received echo pulse whose envelope and phase angle are detected. The real thing che useful echo pulse can by comparing the known False echo signal and the received echo pulse in simp be reconstructed in a safe manner.

The method according to the invention can thus be run at runtime measurement in an advantageous manner in pulse transit time systems use, the fields of application of the existence of false echoes likely to appear. This is particularly true Measurements for level measurement in containers, where next to the There are still numerous echoes from the product surface che further reflections can occur, for example se through struts arranged inside the container or other internals are necessary. To make a clear distinction to be able to meet between useful and false echo impulses therefore the false echoes regi existing when the container is empty striert and this with amplitude and distance values saved. A comparison between any one received echo profile with filled container and the ge stored false echo information then permits identification decoration of the interferers and easier locating of the useful echo pulses.

Since according to the invention both the amplitude and Phase information of the received echo pulse is determined and, as required, the amplitude and phase information tion of the false echo is known from the received Echo signal easily on the amplitude and phase angle of the Useful echo pulse can be inferred, even if  Interference and useful echo partly superimposed on each other.

A circuit arrangement according to the invention for implementation the procedure for measuring the transit time is the subject of the contract Proverbs 12

Further developments of this circuit arrangement are in the Subclaims 13 to 18 specified.

The circuit arrangement according to the invention consequently has a Sending and receiving device for sending and receiving of pulse-modulated pulse with the same carrier frequency as well as an envelope evaluation device for determination of the envelopes of the impulses. In addition, is a Quadrature demodulator provided to out of the pulses each generate a 0 ° output signal or 90 ° output signal, the quadrature demodulator with the carrier frequency of the pulse-modulated pulses can be operated. An evaluation scarf tion first forms from the maxima of the envelopes of both Impulse a measure of the roughly predetermined transit time between between the two pulses and generated from the 0 ° output signals len and 90 ° output signals for the runtime visible correction value.

The evaluation circuit has distance determination between two pulses via a device to get out of the determined duration taking into account a given distance of the impulses the distance to calculate the target object.

The invention and its advantages are hereinafter collectively menhang explained with three figures. Show it:

Fig. 1 is a timing diagram of a pulse-modulated pulse of a Laufzeitmeßeinrichtung for explaining the method according to the invention,

Fig. 2 shows a circuit arrangement for carrying out the transit time or distance measurement according to the invention in a level measuring device, and

Fig. 3 shows signal waveforms for the circuit arrangement of FIG. 2.

In the following FIGS. 1 to 3, the same reference symbols denote the same parts and the same signals, unless stated otherwise.

In Fig. 1, a timing diagram of a receive pulse E is shown how this is received, for example, in the level measurement solution in a receiving device. This received pulse E consists of a carrier frequency signal which has a carrier frequency f _T , and is additionally pulse-shaped amplitude modulated, the frequency of the amplitude modulation being many times lower than the carrier frequency f _T. The amplitude modulation of the received pulse E is selected such that the received pulse E has an envelope H with an initially rising and then a falling edge. The receive pulse E in the exemplary embodiment of FIG. 1 has seven local maxima M1 to M7, the receive pulse E being symmetrical to the local maximum M4. The local maximum M4 is also the maximum value ME of the receive pulse E.

In order to determine the transit time t1 or distance x1 of a predetermined point t, for example the maximum value ME, of the receive pulse E at a predetermined point in time, in the exemplary embodiment of FIG. 1 the zero point, the transit time or distance is first determined roughly predetermined and then a correction value is taken into account for the exact running time or the exact distance. This correction value is determined by the phase angle of the relevant point of the received pulse E and determined from a fraction of the carrier frequency determined by the phase angle for the transit time and from a fraction determined by the phase angle of the carrier wavelength λ _T for the distance to be determined.

The term phase here means the rotation of the phase pointer of a specific point of the carrier oscillation of the received pulse E based on a fixed time or phase point at the beginning of each measurement, here the zero point. A phase angle change of 360 ° corresponds to a total change in path by a carrier wavelength λ _T or a change in total transit time by 1 / f _T. Since the phase angle repeats after one revolution, that is to say after 360 °, there is uniqueness between the phase angle and the running time or the distance to be determined only over this 360 ° or within a distance of λ _T. The running time t of a specific point within the receive pulse E based on the zero point is accordingly determined from the sum of an integer number k of the reciprocal of the carrier frequency f _T and a fraction of this reciprocal of the carrier frequency f _T determined by the phase angle Φ. The term is calculated using the following formula:

t = k / f _T + Φ / (360 ° f _T ).

The distance x of a certain point within the receive pulse E based on the zero point, on the other hand, is calculated from the sum of an integer number k of wave lengths λ _T and a fraction of this wavelength determined by the phase angle Φ, with level measurement devices still to be taken into account that a sent pulse is first sent to the target object, reflected there and sent back to the receiving device, so that the double distance must be taken into account as the outward and return path and a multiplication factor of 0.5 must therefore be used in determining the distance. The reflector di punch in such a level measuring system is calculated accordingly

x = 0.5 (k · λ _T + Φ · λ _T / 360 °).

As illustrated in FIG. 1, the transit time t1 or distance x1 of each point in the received pulse E can be determined according to the above formulas via the number k and the phase angle Φ.

Since all the points in the received pulse E λ at a distance of a wavelength _T same phase angle Φ own, this group can be determined for each group of points at a distance of a wavelength λ _T from a single point of the phase angle. As a group of points, for example, all zero crossings of the receive pulse E with a positive slope or all local maxima M1 to M7 can be selected. The choice of the group of points, which is to represent their phase angle Φ as the phase angle amt of the total pulse, is arbitrary. In the exemplary embodiment of FIG. 1, the group of local maxima M1 to M7 is selected. In the example shown, all points of this group have the phase angle Φ = 90 °, so that, according to this definition, this phase angle can represent the entire received pulse E.

In order to determine the transit time or distance of the maximum value ME of the received pulse from the zero point, apart from the phase angle of this point, all that is required is to determine the integral part k of wavelengths λ _T or reciprocal values of the carrier frequency f _T between the zero point and this Point necessary.

According to the invention, the transit time or distance from the point in question to the zero point is approximately pre-determined. To determine the multiplier k, an accuracy of the distance measurement within the error limits -λ _T / 4 to + λ _T / 4 or an accuracy for the transit time measurement within the error limits -0.25 / f _T to + 0.25 / f _{T is} sufficient . Such an approximate determination of the transit time or the distance is possible, for example, by means of an evaluation method with a detection threshold for the flanks of the envelope H of the received pulse E. In the present case, this rough determination of the transit time or distance of the maximum value ME of the received pulse E can be carried out in a simple manner by averaging two distance values which result when a detection threshold is exceeded and undershot, since the received pulse E is symmetrical as required. By adding the phase angle Φ of the received pulse E, the roughly determined transit time or distance can be corrected to the exact value. Due to the permitted error mentioned, a distinction must be made between the following cases after determining the integer part k:
For

-λ _T / 4 <1/2 (g · λ _T + Φ · λ _T / 360 °) - x _V λ _T / 4

applies k = g,
For

λ _T / 4 <1/2 (g · λ _T + Φ · λ _T / 360 °) - x _V λ _T / 2

applies k = g - 1,
For

-λ _T / 2 <1/2 (g · λ _T + Φ · λ _T / 360 °) - x _V - λ _T / 4

applies k = g + 1, where x _{V is} the roughly determined distance and g is the integer component of wavelengths λ _T within the roughly determined distance x _V.

The exact distance x _{G is} calculated accordingly

x _G = 1/2 · (k · λ _T + Φ · λ _T / 360 °).

For the exact determination of the transit time, λ _{T must be} replaced by 1 / f _T.

In this way, the transit time or distance to the zero point can be exactly determined for each point of the received pulse E by utilizing the phase information contained in the carrier frequency signal. The measurement is not limited to the local maxima M1 to M7. With a different definition of the phase angle, this can also be, for example, all zero crossings with a positive slope or another group of distinctive points within the receive pulse. It is only essential that the corresponding point can be determined approximately, in particular to within ± λ _T / 4 or ± 0.25 / f _T. If this is the case, the accuracy of the measuring method only depends on the phase measurement.

Although the phase angle of the point in question can be determined in a variety of ways, it has proven to be advantageous to measure and average the phase angle over the entire pulse length, so that the measurement accuracy of the phase measurement is increased. If, for example, a phase measurement error of ± 10 ° is assumed, this results in a measurement error for the distance with a level measurement of ± 1/72 · λ _T.

The inventive method for runtime or distance determination between a fixed zero point and a Receive pulse makes sense where between the zero point and the phase relationship of a transmission pulse is a known combination menhang prevails. If this connection is not known, then can both by using the inventive method on transmit as well as on receive pulse and determination of the Difference between the running time or the distance value of the Transmit pulse and the transit time or the distance value of the Receive pulse their runtime difference or their Ab status can be determined.

The method according to the invention is explained below using a specific exemplary embodiment in connection with a circuit arrangement shown in FIG. 2 and the associated signal profiles in FIG. 3.

The circuit arrangement of FIG. 2 is, for example, part of a level measuring device. The circuit arrangement has a transmitting and receiving device 1 for transmitting and receiving pulses with the same carrier frequency f _T pulse-modulated. The transmitting and receiving device 1 is connected to a coupling element 2 , which serves on the one hand for coupling the electrical transmission pulse to the measuring section and converting it into an electromagnetic wave, and on the other hand after reflection of the emitted wave on a reflector 3 , for example a product surface in a container , to convert back the received electromagnetic wave into an electrical signal and thus a receive pulse is hen. At a signal output 4 of the transmission and reception device 1 , a transmission pulse and then a reception pulse can be tapped, which are the same as regards their carrier frequency f _T. However, due to the damping of the transmission path, the received pulse E is damped in its amplitude. In the transmitting and receiving device 1 , additional devices can be seen to strengthen the transmit and / or receive pulse and, if necessary, to filter it.

An example of a transmitted by the transmit and receive device 1 transmit pulse and received receive pulse is shown in Fig. 3 in the timing diagram a. The transmit pulse is designated by the reference symbol S and the receive pulse by the reference symbol E. As can be clearly seen, the transmit pulse S and receive pulse E have the same pulse-modulated carrier frequency f _T , the receive pulse E having a lower amplitude due to the transmission path. The transit time of transmit pulse S and receive pulse E is determined by the time interval between their two maximum values MS and ME. With the circuit arrangement to be described below, the transit time t and thus the distance x between transmit pulse S and receive pulse E can be exactly determined by applying the method according to the invention.

For this purpose, the arrangement shown in FIG. 2 has an oscillator device 26 which provides an oscillator output signal with an oscillator frequency f₀. The oscillator output signal is fed to a first divider 27 which divides the oscillator output signal by a factor N, so that at the output of the divider 27 a signal having a frequency f _S is available, which determines a measuring cycle. In addition, the oscillator output signal reaches a further divider stage 28 , which divides the oscillator output signal by a factor P. At the output of the divider stage 28 , the carrier frequency signal for the pulse transmission in the transmitting and receiving device can be tapped. The carrier frequency signal has the carrier frequency f _T , which is many times greater than the frequency f _S. The outputs of both divider stages 27 and 28 are connected to the transmitting and receiving device 1 .

For the approximate predetermination of the transit time t or distance x between the transmission pulse S and the reception pulse E and subsequent correction value determination, the circuit arrangement in FIG. 2 has an envelope evaluation device 5 and a quadrature demodulator 20 , each of which has the input and signal outputs 4 of the transmission and reception direction 1 are connected.

The envelope curve evaluation device 5 serves to determine the envelope curves H of the transmit and receive pulse S, E. For this purpose, the envelope curve evaluation device 5 has on the input side a rectifier arrangement 6 , for example a two-way rectifier, with a downstream low-pass filter 7 . The output of the low pass 7 is connected to a comparator 8 , which has an adjustable threshold value. The clock connection of a JK flip-flop 9 , whose Q output connection q is connected to the input of a binary counter 12, is connected to the output of this comparator 8 . The output of the comparator 8 is also connected via an inverter 11 to a clock connection of a further JK flip-flop 10 , the Q connection q of which is connected to a further binary counter 13 . The two binary counters 12 , 13 each have a reset connection R and a clock connection T. The reset connections R are connected to the output of the divider stage 27 , while the clock connections T are connected to the output of the oscillator device .26.

The output connections 32 , 33 of the two binary counters 12 , 13 are connected to an evaluation circuit 14 . This evaluation circuit 14 generates from the maxima MS, ME of the transmission pulse S and reception pulse E determined in the envelope evaluation device 5 a measure of the approximately predetermined transit time or distance between the transmission pulse S and reception pulse E. For this purpose, the evaluation circuit 14 has a microcomputer 15 . The microcomputer 15 also determines the correction value of the preliminary approximate transit time or distance measurement that takes into account the exact transit time or exact distance. The evaluation circuit 14 also has two analog-digital converter stages 18 , 19 , the output connections of which are each connected to a memory 16 , 17 . The memories 16 , 17 are connected to the microcomputer 15 . The analog-to-digital converter stages 18 , 19 are each connected to an output connection 30 , 31 of the quadrature demodulator 20 .

The quadrature demodulator 20 is constructed in a manner known per se. The quadrature demodulator 20 has a first multiplier 21 and a second multiplier 22 , the first input connection of which is each connected to the signal output 4 of the transmitting and receiving device 1 . The second signal inputs of the two multipliers 21 and 22 are connected to the output terminal of the divider stage 28 , a phase shifter 25 being arranged in front of the second input of the second multiplier 22 , which phase-shifts the output signal of the divider stage 28 by -90 °. The outputs of the two multipliers 21 and 22 are each connected to an output terminal 31 , 30 of the quadrature demodulator 20 via a low-pass filter 23 , 24 .

The input signal at the second input of the first multiplier 21 is denoted by the reference symbol u and the input signal phase-shifted by 90 ° to the second input of the second multiplier 22 is denoted by v. The 0 ° output signal at the output terminal 31 of the quadrature demodulator 20 is designated by Q and the 90 ° output signal at the output terminal 30 by I.

The operation of the circuit arrangement shown in Fig. 2 is clear from the waveforms a to k in Fig. 3. The signal curves a to k shown in FIG. 3 are marked in FIG. 2 at the points that occur.

As already explained, the signal curve a represents the transmission pulse S and the reception pulse E which occurs at intervals therefrom. The transmission and reception device 1 is element by a flank of the signal present at the output of the divider 27 with the frequency f _S at the coupling 2 triggered (cf. b in Fig. 3). Envelope evaluation circuit 5 is used for preliminary rough determination of the transit time or distance.

By rectification and low-pass filtering of the gear on Signalaus 4 pending signal at the output of low-pass filter 7 of the Hüllkurvenauswerteeinrichtung 5, the signal shown in waveform c of Figure 3. Picked off, is freed of the high-frequency components of the carrier signal. In the comparator 8 , the threshold value SW indicated by dashed lines c is set. This detection threshold SW can be fixed or can be set via the control and evaluation circuit 14 . At the output of the comparator 8 , a square-wave signal can be tapped, the rising flanks of which are determined to be exceeded and the falling edges of which are below the detection threshold SW of the signal present at the output of the low pass 7 .

The rising edges of this square wave signal in the signal run d trigger the JK flip-flop 9 , which was previously reset like the JK flip-flop 10 and the binary counters 12 and 13 by the rising edge of the signal f _S at the beginning of the pulse transmission . The JK flip-flop 9 releases the binary counter 12 at its input on the first rising edge of the square-wave signal in the signal curve d and stops it on the next rising edge, as can be seen in the signal curve. The JK flip-flop 10 and the binary counter 13 work in an analogous manner, whereby by the provision of the inverter 11 at the clock input of the JK flip-flop 10 now not the rising edges of the rectangle signal, but its falling edges are decisive.

As can also be seen in FIG. 3 from the signal profiles e and f, the counting cycle is many times greater than the carrier frequency f _T. The counting clock corresponds to the frequency of the oscillator output signal and thus the oscillator frequency f₀.

At the end of a measuring cycle, which is determined by the falling edge of the signal curve b, the binary counter 12 thus contains a number Z1, which is a measure of the distance between the leading edge of the transmission pulse S and the leading edge of the received pulse E. Similarly, the binary counter 13 contains a number Z2, which is a measure of the distance between the trailing edge of the transmit pulse S and the trailing edge of the receive pulse E. Since the amplitudes of the transmit and receive pulses S, E are usually different, the counter readings in the binary counters 12 , 13 and thus the numbers Z1 and Z2 stored there are not the same. The microcomputer 15 in the control and evaluation circuit 14 forms an average value from these two counter readings, which is to be regarded as a measure of the distance between the maximum value MS of the transmission pulse S and the maximum value ME of the reception pulse E. The preliminary rough determination of the distance or transit time between transmit pulse S and receive pulse E is thus carried out. The preliminary rough determination of the distance x _V between transmit pulse S and receive pulse E results from the following formula:

x _V = 1/2 (Z1 + Z2) 1 / f₀v v 1/2,

where v denotes the speed of propagation of the wave net.

In order to determine the exact transit time or distance, the phase angle of the transmit pulse S and receive pulse E is also determined using the circuit arrangement shown in FIG. 2. The quadrature demodulator 20 serves this purpose. In the quadrature demodulator 20 , the transmit pulse S and receive pulse E are multiplied by the signals u and v and then low-pass filtered. The signals u and v have the same frequency as the carrier signal of the transmit pulse S and receive pulse E. This frequency is the carrier frequency f _T. As can be seen from the signal profiles g and h in FIG. 3, the signals u and v are rectangular signals with the carrier frequency f _T , the signal v being out of phase with the signal u by -90 ° because of the phase shift arrangement 25 . In the two channels of the quadrature modulator 20 , the transmit pulse S and receive pulse E are converted into the intermediate frequency position zero by mixing or multiplying by the signals u or v. The low-pass filters 23 and 24 behind the two multipliers 21 , 22 suppress the high frequency components which arise during multiplication, so that a DC voltage is present at the outputs of the two low-pass filters 23 , 24 for the duration of the pulses, the amplitude of which only depends on the phase shift between the Corresponding transmit pulse S or receive pulse E and the signal u or v depends.

The output signals belonging to the transmit and receive pulse S, E shown in FIG. 3 at the connections 30 and 31 of the quadrature demodulator 20 are shown in FIG. 3 on the basis of the signal profiles i and k. The output signals Q and I of the quadrature demodulator 20 can be calculated as follows:

I = a · cos α
Q = a · sin α.

By appropriate selection of the signals u and v in relation to Zero point is reached that the displayed phase angle a equal to the phase angle of the transmission pulse described above S or receive pulse E is. The constant of proportionality te a depends on the parameters of practical realities and is mostly unknown. By division leaves however, they eliminate themselves because

This relationship can be used by the control and evaluation circuit 14 . In the analog-digital converter stages 18 , 19 , the analog output signals I and Q of the Quadra turdemodulators 20 are digitally converted and stored in the subsequent memories 16 , 17 . In the waveforms i and k of FIG. 3, QS and QE denote the 0 ° output signals of the quadrature demodulator 20 with the transmit or receive pulse applied on the input side and IS or IE the corresponding 90 ° output signals. The microcomputer 15 can by means of division and arctangent formation, the phase angle gesuchten in each case for the transmit pulse S and receive pulse E determined. The ambiguity of the arc tangent between 0 ° and 360 ° can be avoided by including the signs of the signals I and Q. In the embodiment of Fig. 3 is obtained for the phase angle Φ _S Φ of the transmitted pulse S _S = 180 ° and Φ the phase angle of the received pulse _E E _E Φ = 90 °.

The phase angles Φ _S and Φ _{E determined in this way} result in their mutual phase shift Φ _G by forming the difference.

Φ _G = Φ _E - Φ _S = 270 °.

For the exact path length between send and receive pulse S, E therefore applies:

x _G = 1/2 (k · λ _T + Φ _G · λ _T / 360 °),

where k is determined according to the above case distinction is.

It is thus shown that with the circuit arrangement presented order a highly accurate transit time and distance measurement is possible at which the runtime or distance roughly predetermined and then the exact one Correction value taking into account the running time or distance is determined. The phase win kel of the pulse or pulses measured and the correction value  a fraction of the Trä determined by the phase angle gerfrequency or carrier wavelength determined. Through the Application of quadrature demodulation to determine the phase angle tion is also ensured that the phase angle of the pulse measured over the entire pulse length and is thus averaged out. This makes a highly accurate Measurement possible.

Reference list

1 transmitting and receiving device
2 coupling element
3 reflector
4 signal output
5 envelope evaluation device
6 rectifier arrangement
7 low pass
8 comparator
9 JK flip-flop
10 JK flip-flop
11 inverters
12 counters
13 counters
14 control and evaluation circuit
15 microcomputers
16 memories
17 memory
18 A / D converter stage
19 A / D converter stage
20 quadrature demodulator
21 first multiplier
22 second multiplier
23 low-pass device
24 low-pass device
25 phase shift arrangement
26 oscillator device
27 part level
28 part level
30 exit
31 exit
32 output connector
33 Output connection
t1 period of time
t time
Z1 first counter reading
Z2 second counter reading
K correction value
E receive pulse
QE 0 ° output signal of the receive pulse
QS 0 ° output signal of the transmission pulse
IE 90 ° output signal of the receive pulse
IS 90 ° output signal of the transmission pulse
H envelope
I 90 ° output signal
M maximum
N factor
Q 0 ° output signal
P factor
S transmit pulse
R reset connection
T clock connection
SW threshold
f _T carrier frequency
fs frequency
fo oscillator frequency
q Q connection
u signal
v signal
x, x1 distance
λ _T wavelength
Φ _E phase angle
Φ _S phase angle
Φ phase angle

Claims (19)

1.Procedure for measuring the transit time between a predetermined point in time and a pulse-modulated pulse (E) with a carrier signal having a carrier frequency (f _T ), in which the transit time initially predetermines and then a correction value which takes the exact transit time into account ( K) is determined, characterized in that the phase angle (Φ) of the pulse (E) is measured and the correction value (K) is determined from a fraction of the carrier frequency (f _T ) determined by the phase angle (Φ). 2. The method according to claim 1, characterized in that the predetermined point in time is determined by a previous pulse pulse modulated (S) with the same carrier signal, that the phase angle (Φ _S , Φ _E ) of both pulses (S, E) are determined, that a phase difference angle (Φ _S -Φ _E ) is formed from the two phase angles (Φ _S , Φ _E ), and that the correction value (K) from a fraction determined by the phase difference angle (Φ _S -Φ _E ) is the carrier frequency (f _T ) is determined. 3. The method according to claim 1 or 2, characterized in that the phase angle (Φ _S , Φ _E ) of a pulse ses (S, E) over the entire pulse length is measured and averaged. 4. The method according to any one of claims 1 to 3, characterized in that the transit time is initially predetermined to ± 4 / f _T. 5. The method according to any one of claims 1 to 4, characterized characterized in that the term is initially based on a Envelope detection of the pulse or pulses (S, E) approximately is predetermined. 6. The method according to claim 5, characterized in that a digital maximum search for envelope detection of the pulse (S, E) is performed by the pulse (S, E) subjected to quadrature demodulation and the Maximum (MS, ME) by the sum of the squares of the 0 ° - Output signals (Q) and 90 ° output signals (I) Quadrature demodulation is determined. 7. The method according to any one of claims 1 to 6, characterized in that a pulse (S, E) for determining its phase angle (Φ _S , Φ _E ) a Quadraturde modulation with subsequent low-pass filtering is Unterzo gene, reference carrier signals selected for the quadrature demodulation that have the carrier frequency (f _T ) and that the phase angle (Φ _S , Φ _E ) from an arctangent formation of the quotient of the 0 ° output signal (Q) and 90 ° output signal (quadrature demodulation and low-pass filtering) I) is determined. 8. The method according to any one of claims 1 to 7, characterized in that from a common Oszillatorein direction ( 26 ) by division both the carrier signal with the carrier frequency (f _T ) of the pulse (E) or the pulses (S, E) as well the reference carrier signals for the quadrature demodulation are derived. 9. The method according to any one of claims 1 to 8, characterized in that the 0 ° output signals (Q) formed by the quadrature demodulation among each other and the 90 ° output signals (I) among each other are sharply averaged and subsequently the envelope formation after in several successive measurement cycles the relationship is carried out, the averaged values of the 0 ° output signal (Q) and the 90 ° output signal (I) being used for the envelope formation. 10. Use of the method according to one of claims 1 to 9 for distance measurement, in particular for level measurement solution in containers, the preceding pulse (S) a transmission pulse sent into the container and the further pulse (E) is an echo pulse and the determined Runtime between the two pulses (S, E) for removal determination with a predefined dispersion range speed is multiplied. 11. Use according to claim 10, in which at least one false echo signal with amplitude, distance and phase values is stored and a useful echo pulse is reconstructed from the received echo pulse with detected envelope values and phase angle (Φ _E ). 12. Circuit arrangement for carrying out the method according to one of claims 1 to 10, characterized by the following features:

• - A transmitting and receiving device ( 1 ) for transmitting and receiving pulses modulated with the same carrier frequency (f _T );
• - an envelope evaluation device ( 5 ) for determining the envelopes (H) of the pulses (S, E);
• - a quadrature demodulator (20) to produce from the pulses (S, E) each having a 0 ° output signal (Q) or 90 ° -Aus input signal (I), wherein the quadrature Demo dulator (20) with the carrier frequency (f _T ) Send reference carrier signals is operable;
• - An evaluation circuit ( 14 ) to generate a measure of the approximately predetermined transit time between the two pulses (S, E) from the maxima (MS, ME) of the envelopes (H) of the two pulses (S, E) and to the 0 ° output signals (Q) and 90 ° output signals (I) to form the correction value (K) for the running time.

13. Circuit arrangement according to claim 12, characterized in that the evaluation circuit ( 14 ) has a Einrich device to a distance between the two pulses (S, S) from the ascertainable transit time, taking into account a predetermined speed of the pulses (S, E), E) to be calculated. 14. Circuit arrangement according to claim 12 or 13, characterized in that an oscillator device ( 26 ) is provided, which is optionally connected via divider stages ( 27 , 28 ) to the transmitting and receiving device ( 1 ) and the quadrature demodulator ( 20 ). 15. Circuit arrangement according to one of claims 12 to 14, characterized in that the quadrature demodulator ( 20 ) is provided on the output side with a low-pass device ( 23 , 24 ). 16. Circuit arrangement according to one of claims 12 to 15, characterized in that the evaluation circuit ( 14 ) has a microcomputer ( 15 ), that the microcomputer ( 15 ) with the envelope evaluation device ( 5 ) and with a memory device ( 16 , 17 ) is connected , and that between the memory device ( 16 , 17 ) and the quadrature demodulator ( 20 ) an analog-to-digital converter device ( 18 , 19 ) is connected. 17. Circuit arrangement according to one of claims 12 to 16, characterized in that the Hüllkurvenauswertein direction ( 5 ) on the input side a rectifier arrangement ( 6 ) with a downstream low-pass filter ( 7 ) and comparator ( 8 ) that two at the output of the comparator ( 8 ) Flip-flops ( 9 , 10 ) with subsequent counters ( 12 , 13 ) are connected, an inverter ( 11 ) and the output connections ( 32 , 33 ) of the counters being arranged in front of one of these two flip-flops ( 9 , 10 ) ( 12 , 13 ) are connected to the evaluation circuit ( 14 ). 18. Circuit arrangement according to claim 17, characterized in that the flip-flops ( 9 , 10 ) are JK flip-flops, a clock connection of a first flip-flop ( 9 ) with the comparator ( 8 ) directly and a clock connection of the second flip-flops ( 10 ) is connected to the comparator ( 8 ) via the inverter ( 11 ) and Q-output connections (q) of the JK flip-flops ( 9 , 10 ) are each connected to one of the counters ( 12 , 13 ) are. 19. Circuit arrangement according to one of claims 12 to 16, characterized in that the envelope curve evaluation device ( 5 ) is part of the evaluation circuit ( 14 ) and the envelope curve values within the stored in the storage device ( 16 , 17 ) of the evaluation circuit ( 14 ) ° and 90 ° output signal values (I, Q) using the microcomputer ( 15 ) according to the relationship can be calculated.