DE1562341C - Telephone conference call - Google Patents

Description

1 2

The invention relates to a conference circuit for overall reinforcement along the four-wire loop a large number of two-wire subscriber lines must be available to ensure vibration

To avoid double talk with a large number of associated amplification in the four-wire circuit,

However, if such a four-wire circuit in

devices, each switched on in a loop 5 one of the conference subscriber lines

lying amplifier stage with a gain of, the well-known conference circuit makes the

approximately one, but less than one, attenuation of the nearby hybrid circuit

with a bias circuit for each amplifier, so that the four-wire line has an oscillation

Kungseinrichtung, which shows this tendency in the conductive state. The signals from the incoming

holds, and with the amplification devices on the pulling side of the four-wire circuit namely run up

arranged coupling circuits, each of which has a two-wire side with only very little attenuation

Subscriber line to the conference call via the loop of the known conference call

couples. to the outgoing side of the four-wire circuit.

A conference call can be defined as the boundary surface at the same time as that switching connection of several two-wire telephone lines 15 point of the loop at which a subscriber line is used, in such a way that each participant is connected to the conference circuit, so all others can be connected to the conference circuit. you can also formulate this in such a way that participants can speak to the acquaintance. For one satisfied - the back absorption. is small at an interface. The conference call must be in constant operation. This leads to echoes, ie to a return of certain properties; this includes a verbal signal to a conference participant around the relatively small entry loss, same loop around, as well as instabilities.
Transmission properties between any two. The object of the invention is to provide a participant conference call connected to the conference call. mern with a loop gain and the possibility of creating an alternating number of 1 and high back attenuation. To activate the. Of conference participants. The invention is based on a conference circuit known conference circuits consist of a device of the type mentioned and solves this simple parallel connection of subscriber lines task by the features specified in the characterizing part of or a complicated arrangement of fork claim 1. _:
Transformers, multipath resistor bridge circuits This makes a con and amplifier without much effort. 30 reference circuit with low insertion loss and there is already an improved conference-the same transmission properties between belieschaltung known (German Auslegeschrift 1182304), created conference participants that have a high return attenuation at all interfaces for one of the number of two-wire participants Large number of transistors - so that regardless of the type of amplifier stages connected to a closed loop, instabilities do not occur.
is interconnected. The amplification of each development of the invention are the subject of a stronger stage is set so that the total of the dependent claims.

Strengthening the loop is about 1 at most. The invention is hereinafter based on the

Number of transformers couples two-wire part drawings described in detail; it shows

subscriber lines between each two amplifier stages 40 F i g. 1 a connection between two subscribers

än the loop on. lines to explain the definitions for the

In this well-known conference circuit, return loss and insertion loss,

Stuff the amplifier used there a phase F i g. 2 a known conference call,

repentance at every stage. If an even number F i g. 3 a first embodiment of the invention

of stages is present in the loop, there is no connection with transformers,

Phase reversal or cancellation. Such a F i g. 4 an alternating current equivalent circuit diagram for the arrangement tends, however, to instabilities, i.e. to refine the working method of a single Schwingingen or whistling, since the loop as a positive stronger level according to Fig. 3,

Feedback amplifier circuit acts. The reinforcement F i g. 5 an alternating current equivalent circuit diagram for

The loop around the loop, like the cleaning of the entire conference call, must be followed up

the phase must be critically monitored in order to identify such F i g. 3,

Avoid instabilities. If this happens to the fig. 6 a further alternating current equivalent circuit diagram

in connection with the known arrangement to explain the conference call after

way, the circuit works as long as F i g. 3,

as satisfactory as none of the 55 F i g. 7 is an alternating current equivalent circuit diagram for the connection of connected subscriber lines, a clarification of the advantages of the conference circuit on a four-wire line with noticeable amplification. F i g. 3 also for the extreme case that the impedances, however, if one of the subscriber lines is one of the different lines connected to the conference circuit according to the four-wire line with greater gain, FIG. 3 connected telephone subscriber lines again problems. It is well known that a quadrangle extends over a large area
wire cable at both ends with a fork scarf F i g. 8 completed a second embodiment of the invention, which prevents incoming manure with coupling capacitors,
Signals are transmitted back to the outgoing side F i g. 9 a first type of impedance determined by the. In each hybrid circuit, a connection network is usually used that can be set up so that the connection network connects the subscriber lines to the conference with considerable attenuation between the incoming circuit,

mendiiii and tier outgoing line is present. F i g. 10 a third embodiment of the Erfm-.

This attenuation at both ends, which is related to compensation gain,

F i g. 11 shows an alternating current equivalent circuit diagram to explain the conference call according to FIG. 10, if the impedance of the switching network according to FIG. 9 in each with the conference call connected subscriber line is introduced,

F i g. 12 a fourth embodiment of the invention,

F i g. 13 a second type of impedance that can be introduced by the switching network that connects the participant lines to the conference call,

F i g. 14 shows an alternating current equivalent circuit diagram to explain the conference call according to FIG. 12, if the impedance of the switching network according to FIG. 13 in each with the conference call connected subscriber line is inserted,

F i g. 15 different transmission properties that are necessary for the conference call according to FIG. 12 off the equivalent circuit according to Fig. 14 were calculated,

F i g. 16 shows a fifth embodiment of the invention,

F i g. 17 shows a resistor arrangement which is used at each stage of the conference call of FIG. 18 used will,

F i g. 18 a sixth embodiment of the invention,

F i g. 19 shows a seventh embodiment of the invention,

F i g. 20 a stage of the conference call according to FIG. 19, into the various current values, voltages and values of switching elements for explaining the conference call are entered in FIG.

Only in the first embodiment of the invention are transformers required to the subscriber lines to connect to the conference call. In other embodiments is in each stage A gain is provided in the loop to reduce attenuation in the remaining parts of the telephone system to compensate. In further exemplary embodiments of the invention, arrangements are provided which improve the input impedance seen by each subscriber line and provide a uniform gain provide between the subscriber lines.

In Fig. 1 shows a connection between two subscriber lines. Each subscriber line can be imagined as consisting of a signal source and an impedance Z connected in series. The subscriber lines are connected to one another via a switching network. The insertion loss, measured in db, is defined as the return loss is a measure of how much of the signal generated by one participant is transmitted to the other participant and then transmitted back to the first participant. If Zl is the impedance of one line and Zl is the impedance of the other line, the return attenuation is

Zl J-Z2

20 lo _gll

ZX-Zl

If, in the ideal case, Zl is equal to Zl , the back loss becomes infinite. The same characteristics that apply to a connection between two participants also apply to a conference connection

iS to. For any combination of two lines should the return loss as high as possible and the insertion loss as small as possible.

F i g. Figure 2 shows a typical prior art conference call. Six subscriber lines are connected in parallel. Consider a single subscriber line, for example line 1. For ideal insertion and return loss, line 1 should see an effective impedance that is equal to its own line impedance R (the main component of a line impedance Z is real). Line 1, however, sees five other lines connected in parallel, that is, one

Total resistance -j-. For this reason, the

State of the art a negative impedance -j- in parallel

switched to the six lines. A parallel connection of two resistors with the values

-r- and -τ- result in an effective impedance R.

As a result, the return loss and the insertion loss have the desired values.

Using a negative resistor in

however, this species is dangerous if certain precautionary measures are not taken.

For example, assume that R is 2000 ohms. The negative resistance contained in the conference call

- R

then has the

20 lo _gll

Vl

Vl

where Vl is equal to half the voltage produced by the subscriber line generating the signal when the subscriber line is idle, and Vl is the voltage across the impedance in the other subscriber line when both subscriber lines are connected together. In the ideal case, Vl = Vl. This is the case when the two subscriber line impedances are the same and the switching network and the conference circuit do not introduce attenuations and impedance transformations. In this case the insertion loss is zero and maximum power is transmitted from each subscriber line to the other.

Value -500 ohms. It is also assumed that the impedance of each line is not close to that The ideal value is 2000 ohms, but is 3006 ohms (possible state in a telephone system with long lines connected to other systems and with individual deviations for the subscriber lines). Under these conditions, the parallel connection of the six line impedances results —2— = 501 ohms. If now the

negative resistance of -500 ohms is switched on, the total value for the parallel connection can be calculated Calculate the six line impedances with the negative resistance:

(501) (-500) _ - (501) (500)
(501 - 500) ~~ 1

This value is negative and the system will oscillate. For this reason it was previously at Use of negative resistors required to provide complicated circuitry to ensure that the total resistance between the two common wires does not become negative.

The circuit according to the prior art according to FIG. 2 has only been described to the To identify problems that have arisen so far when negative resistances to improve the transmission properties used by conference calls. Some of the ones to be considered below Embodiments of the invention also use elements of negative impedance. It however, it will be seen that the critical settings required in previous conference calls are not necessary in the conference call according to the invention.

F i g. 3 shows a first embodiment of the invention with transformers. There can be six subscriber lines are connected to each other, with only three transistors 10, 11 and 12 for the conference call required are. Zener diodes 14, 15 • and 16 are used for biasing purposes. For example, if the voltage of source 30 is 24 volts, then there is a suitable breakdown voltage for the zener diodes at 12 volts. In the idle state there is a voltage of 24 volts and a voltage of 12 volts at each base. The emitter currents are determined by the value a resistor 18 in each emitter circuit is determined. These resistances are each through one Capacitor 19 bridged so that there is no attenuation of the alternating current signals by the resistors 18 occurs.

Each collector is connected to the base of the following transistor via a Zener diode. On two subscriber lines are inductively coupled to each transistor. If from a subscriber line a signal is generated, the current changes through the corresponding transistor and directly controls the Transmission of a signal to the subscriber line coupled to the same transistor. The collector voltage changes, and since it is coupled to the base of the subsequent transistor, changes the current in this transistor depends on the signal. The corresponding current controls again the transmission of the signal to the two subscriber lines connected to the transistor, and the change in the collector voltage controls due to the collector-base coupling again the transmission to the last two subscriber lines.

The operation of the circuit according to FIG. 3 can best be seen with reference to FIG. 4 to 7 understand. F i g. 4 shows an alternating current equivalent circuit diagram to explain the mode of operation of an individual amplifier stage in the conference circuit. The three collector impedances Z are to be neglected for the moment. Every transistor can be thought of as having an effective emitter resistance r _e . If one looks into the base of one of the transistors, one finds an impedance β (r _e + Rle), where β is a property of the transistor. Rle and Rlc are the reflected line impedances seen from the emitter and collector of each transistor, respectively. Assume that Rle and Rlc are both 1000 ohms, that r _{e is} 10 ohms, and β is 100. Since Rle is thus much greater than r _e , the impedance, seen in the base of the transistor furthest to the right, is approximately β Rle or 100,000 ohms.

Let us first consider the gain from the line at the collector to the line at the emitter.

Assuming that almost all of the signal current goes from the line at the emitter to the collector of the leftmost transistor flows (ideally the base current of a transistor is negligible), is the ratio of the collector load voltage to the emitter load voltage

Rlc

Rle + te

- or

1000 1010

So the gain is 0.99. Now is the gain from the line at the emitter to the line at the collector considered. The emitter current is /. The current

due to the collector load is approximately. If V ρ, 1 is the voltage at the base of the leftmost transistor, since ideally the emitter and base voltages are about the same, is

Emitter current τττττ: · The one from the collector of the furthest 1010

The impedance seen on the left-hand side of the transistor is Rlc parallel to β Rlc or 990 ohms, and the collector voltage, i.e. the voltage across the collector load, is

Since β is 100, the ratio of the collector voltage V is about 0.97. The gain from the line at the emitter to the line at the collector is about 0.97. Consequently, the gain from one of the two lines of a transistor or amplifier stage to the other line of the same amplifier stage is approximately 1.

The equivalent circuit according to FIG. 4 has been described primarily to show why the Gain of each amplifier stage is about 1. The operation of the entire conference call leaves can best be found on the basis of the alternating current equivalent circuit diagrams in FIG. Understand 5 through 7. For the further Description, certain assumptions are made for the analysis of the alternating current equivalent circuit diagrams. It it is assumed that the base and emitter voltages of each transistor are the same. It is further assumed that no current flows through the base of a transistor. Consequently, in the circuit of FIG. 3 the collector and emitter currents of each transistor and the base and emitter voltages equal. These two Assumptions are often made when analyzing a transistor circuit. They are not accurate though correct, but are sufficient to approximately determine how the circuit works.

The collector impedance Z is in the collector circuit according to F i g for the following reason. 3 switched on: First, note that the conference call is using a three stage amplifier represents a negative feedback. Therefore, the known stability requirements for feedback Amplifier must also be met in this conference call. First and foremost, the Nyquist curve is allowed do not include the complex point 1 + / 0 for gain and phase position when idling.

Second, there are the impedances in the emitters and collectors - these are telephone subscriber lines - Particularly outside of the voice frequency band are now controlled, certain Funds required to ensure stability. For this reason, it has proven to be

7 8

moderately emphasized, an impedance Z in each collective or minus sign indicates a voltage. Using a number gate circle to get the required stability unsigned and without an arrow is a resistor. Typically, the impedance Z can be value in kiloohms (kQ). It is assumed that the result of a simple parallel capacitor or a reflected impedance in each emitter and collector series connection of a capacitor with a resistor circuit in FIG. 3 is 1 kOhm,
As explained above, each will lie in circles for the investigation. It can also be assumed that more complicated arrangement equivalent circuits are used in the voltages. However, it is only necessary that transistors are ideal components. The base that the impedance Z the open-circuit curve for the supply and emitter voltage of a transistor are then amplification and phase position changed in such a way that the io is the same, and no current flows across the base of a stability requirements for the feedback transistor, ie, the emitter and Collector current are. are also the same. In Fig. 5 is the line in

The impedance Z becomes the supply source in the emitter circuit of the first transistor 10 now to be considered. AC equivalent circuit diagrams and in the following- The feed signal has +2 volts, and the line pulses Embodiments are neglected, and it is 15 then 1 kOhm. That assumed for the investigation of the substitutes, The method used to make the collector and emitter load alike is the following. It will are. One must remember, however, that it is believed to be that of the feeding line practically executed conference call required - seen impedance the desired value of can be borrowed, has such a 1 kOhm in each collector circuit. The different ones in the circuit Turn on impedance to self-excitation to 20 resulting voltages are then calculated, and avoid. the actual impedance is shown to be the

Before going to an analysis the alternating current substitute; - has assumed value. If one assumes that the Circuit diagrams are ignored, it should be noted that the impedance seen is 1 kOhm, i.e. the impedance, that when using the conference call, each coupler moves in the direction of the emitter of the first Tranpeltransfer should be completed with a load. 25 sistors 10 results, is 1 kOhm, so it works If a conference with six participants required - supply source to a resistance of 2 kOhm, and is borrowed, each transformer is with one of the participants - a current of 1 mA flows. As the whole lines completed. The conference call allows emitter current to flow across the collector of the transistor however, it can also be used if less than six and no proportion of this current goes into the base of the second Participants request a conference connection. In 30 transistors, the current is across the collector load In this case, some transformers are not with part of the first transistor 1 mA. Then there is one subscriber lines completed. This transformer voltage of + 1VoIt. This tension feeds the should however be connected to impedances, the base of the second transistor 11. Since the emitters are equal to the impedance of a subscriber line. voltage of this transistor is equal to its base span-Wenn If this does not happen and it is assumed that it is +1 volt, the voltage is above that If it were an ideal transformer, each would not have +1 volts at the emitter load of transistor 11 and it would flow a subscriber line connected transformer a current of 1 mA through the transistor 11. The same Infinitely large impedance for the corresponding current also flows through the collector load of the tran-emitter or represent collector. In this case would have sistor 11, and the collector voltage of the transidie The amplification of the corresponding stage is by no means 40 stors 11 is therefore -1 volt. The is accordingly about the desired value 1. For this reason, the emitter voltage of the third transistor must be 12-1 volts, any transformer that is not connected to a subscriber line and a current of 1 mA flows through the transistor, is switched on, with an equivalent impedance equal. Then a collector voltage of +1 volt is applied Size to be completed. The equivalent circuit remains generated by the third transistor 12. That tension lies then regardless of whether there is a 45 on the transformer, thus also on the base of the first transistor 10. Subscriber line is connected or not, the same, on the other hand, the emitter of the first transistor 10 is located since the to the emitter and collector circuits also reflected on + 1VoIt. The transistor 10 works Impedance is the same. therefore not as an amplifier, but only as a guiding principle

An alternating current equivalent circuit of the conference switching path between emitter and collector.) If

circuit according to FIG. 3 is in FIG. 5 shown. It will be 5 ° but the emitter of transistor 10 is at +1 volt,

Assuming that the feeding line is the one, a current of only 1 mA flows through the emitter resistor.

which is connected to the emitter of transistor 10. was standing. This is the current that is used for the assumed

It is further assumed that, as shown, the value of the input impedance yields

AC signal generated on this line has 2 volts. So the assumption has been confirmed. Besides that

As explained above, the requirements for the con-55 shows that the two requirements are met,

reference circuits the following: The impedance seen from the feeding line

1. The impedance distance seen by the feeding line is 1 kOhm. The one to every other line should be equal to the impedance of the line itself, transmitted voltage is 1 volt. This is the tension and which are related to a connection with two participants

2. the voltage on every other line should be equal to 60 if the supply voltage were from half the supply voltage. 2 volts evenly applied to both line impedances

An analysis of the equivalent circuit according to FIG. 5 shares. It is true that only one of the emitter lines is

shows that these requirements have been met with the circuit, but you can see that due to the circuit

F i g. 3 can be achieved. symmetry also achieves the desired effect

In all equivalent circuits 65 to be described below, if one of the other two emitter lines

If a number together with an arrow gives a current that is the source of supply.

in milliamps (mA), the arrow indicating the current In F i g. 6 it is assumed that the feeding line

indicating direction. A number with a prefixed plus is attached to the collector of the first transistor 10.

is switched. The voltage of the supply source is again +2 volts. In addition, it is again assumed that the impedance seen from the feeding line has the desired value of 1 kOhm has. As a result, a current of 1 mA flows through the collector of the transistor 10, and it becomes a voltage of +1 volt above the emitter load. The base of the transistor 11 is at +1 volt, and there the Emitter of transistor 11 is at the same voltage, a current of 1 mA flows through the emitter load. This current also flows across the collector load and creates a collector voltage of -1 volts. the The emitter of the third transistor 12 is consequently at -1 volt, and a current of 1 mA flows through the Transistor. A voltage of 1 volt is created across each load, and those to the base and emitter of the The voltage transmitted to the first transistor 10 is + 1 volt. As a result, a current of 1 mA flows through the emitter load of the first transistor. This was also the value that corresponds to the assumed input impedance of 1 kOhm. The initial assumption is thus confirmed. Again it appears that, as desired, the voltage across each of the other five loads is 1 volt and that from the feeding collector line seen input impedance is 1 kOhm. Due to the circuit symmetry, the same applies for the other two collector lines.

When examining the equivalent circuit diagrams according to FIG. 5 and 6 it was assumed that all six loads are equal. In practice, however, the line impedances often deviate from the mean value. In known conference calls, for example the one shown in FIG. 2, such a deviation can lead to vibration. However, this does not apply to the conference circuit according to FIG. 3. In the case of different line impedances, the insertion and return attenuation do not have the ideal value, but the conference circuit does not oscillate in any case. This can be seen from the equivalent circuit diagram according to FIG. 7 show. This represents one of the worst cases. Instead of all loads having the same value R , the collector loads are all equal to ajR and the emitter loads are all equal to Rjot, where cc is a change factor. It is assumed that the feeding line is connected to the emitter of the first transistor 10. The impedance seen by the feeding line is Zi _n . To examine the circuit, a voltage or a current must be assumed at any point in the circuit. It is expediently assumed that the emitter of the first transistor 10 is at +1 volt. This assumed value is shown in FIG. 7 indicated in a box. If the emitter is connected to +1 volt and viewed in the emitter impedance Zt is _n, a stream of _n IjZi mA flows through the transistor 10. The collector voltage generated is then aR + / Z _in volts. This voltage is transmitted to the emitter of the second transistor. Since the emitter load is R / <x , the emitter current OiRjZi _n divided by Rja, or Oi ² JZi _n mA. This current flows through the collector load of the second transistor 11, and the collector voltage generated is -oc ³ R / Zi _n volts.

The various voltages and currents of the circuit can be derived in a corresponding manner. The collector voltage of the third transistor 12 is Oi ⁵ R) Zi _n volts. This voltage is the same as at the base and at the emitter of the first transistor 10. A value of +1 volt was assumed for this voltage. Consequently, Oi ⁵ RjZi _n = 1 and Zin = (X ⁵ R. If κ is equal to 1, the input impedance has the desired value R. But even if α is not equal to 1, this is the impedance seen from the feeding line positive, so there are no oscillations in the circuit.

It is also possible to use the various gains in the circuit according to FIG. 7 to be calculated for the worst case scenario. If Zi _n = <x ⁵ R , all collector and emitter voltages can

ίο can be expressed by κ alone. These are the voltages that are generated when the emitter of the first transistor is at +1 volt. It is possible to determine the value for the voltage V which is required to produce a voltage of +1 volt at the emitter of the first transistor. The supply source operates on the line impedance Rja in series with the equivalent input impedance K ⁵ R. A voltage V of 1 volt appears across the input impedance Oi ⁵ R , and consequently a current of

ao Ij (X ⁵ R mA flow from the supply source. This current generates a voltage of 1 / oc ⁶ volts above the line impedance Rjoi. The supply voltage V must therefore be 1 + 1 / « ⁶ or 1 + oc ⁶ /« ⁶ volts The absolute value for the ratio of each load voltage to the voltage V of the supply source can now be calculated. The following results are obtained:

load voltage ratio 1st collector
2. Emitter
2nd collector
3. Emitter
3rd collector +1
+ α
-1
Oi ²
-1
+ 1 ft ² χ ~ [~ OC
a ² I + « ⁶ I + « ⁶
Oi * 1 + Λ ^β
α ⁶ Ι + « ⁶

It turns out that for the ideal case with α = 1 these ratios all have the value 1. This is the desired value, since for the case with two participants according to FIG. 1 half of the supply voltage appears on each line. Even if the gains are not uniform or have ideal values, if <x is not equal to 1, this is not due to the conference call itself, but to deviations in the subscriber lines. The point is that the impedance seen by a feeding line in any of the emitter circuits is never negative. A similar investigation can be carried out in the event that the feeding line is in the collector circuit of one of the transistors. Again, although the gains are not uniform, the input impedance seen by the feeding line is never negative.

F i g. 8 shows a second embodiment of FIG Invention. The main difference between the embodiments according to FIG. 3 and 8 consists in that in the second case there are no transformers. According to FIG. 3 is each subscriber line via one Transmitter coupled to the conference call.

11 12

In Fig. 8, one side of each subscriber line is connected to standest ID. The voltage on the collector

Ground or, if desired, to a voltage source of the transistor QIA is connected directly to the transistor

applied, and the other side is transferred via a condenser QID to the connection point, since the base

Sator either at the emitter or collector of one of the and the emitter voltage of the transistor QID equal

Transistoia QlA, Q2A and Q3A. There are in the scarf 5. The direction towards the base of the transistor QID

according to FIG. 8 no windings required, however, impedance is very large, so this

are, it can not use miniaturization transistor the collector of the transistor QIA

management procedures are established. It is therefore encumbered.

the possibility of using the arrangement according to FIG. 8 The resistors are also used to

to set up a small and cheap conference call. io a bias voltage for the transistor β 1C ready-

Each stage of the circuit contains four transistors. The base of this transistor is namely on and a Zener diode. The three transistors that are present in the voltage, the connection point of the resistor basic loop , are held by the transistor states RLE and RID . The capacitor Cl renQlA, QlA and Q3A. The B and C transistors are ac shunted. When in each stage they are used for biasing purposes. A signal current flows to 15 and the collector voltage of the voltage transistor QIA applied to the base of the transistor QlB changes, the voltage source 810 changes to have a voltage of 4 volts. The value at the connection point of the emitter from the transistor of RIA is 2 kOhm. As a result, a current flows from QID and the Zener diode Z1. This voltage changes -2 mA via the transistor QIB. This current is fixed, since the base and emitter terminals of the transistor QIB 20 at the connection point of the resistors RID and are held at a voltage which is caused solely by RIE if the capacitor C1 is not the source 810 is determined. As a result, the burden that would be present. Such a change would then QLB transistor, the emitter line is not because the signal in turn affect the conductivity of the transistor Q 1C and changes the current through the transistor QIA the current through the transistor QIA, go alone to the emitter load. The transistor 25 has QLB This is undesirable, however, and merely by turning on the purpose, a bias current of 2 mA for the capacitor Cl will deliver all of the signal voltage transistor QLA. changes that otherwise affect the connection

The same applies to transistor Q1C. The voltage point of the resistors RID and RLE appear voltage at the base of this transistor which is short-circuited through the capacitor. The resistors RLE, RID and RlC and the conductivity of the transistor Q 1C is not determined by the breakdown voltage of the Zener diode Zl. Which is influenced by the signal changes.
Reason that the base of transistor Q 1C is undesirable for the base of transistor Ql C not to be connected to a separate source, corresponding to that directly from a separate source, such as source 810, will be shown later. 810, to be preloaded. Instead, a negative base voltage of the transistor β 1C and the value 35 direct current feedback, which result from the resistor RIB, are chosen so that the current of the connection between the base of the transistor Q 1C and the transistor Q1C is the same as the one above the junction of the resistors RlE and the transistor QlB. So there are 2 mA flowing through the RID . The current through transistor QIB transistors Q 1 C, QlA and QlB. Because the base is 2 mA. It is assumed that the base of the voltage of the transistor QlC is relatively constant, 4 ° transistor QlC biases the collector of the transistor by a separate source so that a quiescent current of 2 mA through gate QlA is not. Since the transistor QID does not flow through the transistor Q1C either. Due to the temperature signal current pulls (the impedance, seen in the base temperature changes and other changes can be that of the transistor, is very large), the total signal current via the transistor QIC will be slightly larger, and current will be transferred to the collector load. 45 for example 2.1 mA. The current across the

The B and C transistors of each stage control the sistorßl / 1 is fixed at 2 mA, as this is the value of the

Quiescent current through the corresponding ^ transistor. There is current through transistor QIB . If the

however, a base voltage for each ^ transistor current through the transistor β 1C must be above this value

to be provided. It is wanted by the Zenerdoid to increase if the tran-

and the E, D and C resistances of the cyclically pre-50 sistor β 1C result. That then in turn leads to

walking step generated. The different resistances an alternating current load of the collector from

stands and the Zener diode form a voltage transistor QlA. So if a separate source for

divider and hold the base of the subsequent A-Tian- the bias of the base of the transistor β 1C

sistors at a voltage lower than the one that would be used could reduce the DC current instability

Source 810. 55 of the circuit to an AC load of the

In Fig. 3 is the collector of each transistor to lead transistor β IA. The use of the cons base loop directly on the Zener diode 15, 16 or 14 state network for biasing the transistor β 1C in series with the base of the transistor of the next causes the required DC stability. When the stage is switched on. However, for the circuit, the quiescent current through the transistor β 1C increases according to FIG. 8 not appropriate. If the collector 60 wants, the emitter voltage of the transistor of the transistor β IA increases directly to the junction point QID. As a result, the base voltage of the Zener diode Zl and the resistor RlD an- transistor β 1C increases and holds the current through which it would be switched if the resistors would set the collector transistor β 1C to the desired value of 2 mA. load gate of the transistor β IA. The only load The negative DC feedback brings so on the collector should however be the collector line. 65 the required DC stability.
For this reason, a fourth transistor is β ID. A corresponding biasing arrangement is for between the collector of transistor QIA and the other two stages of the circuit of FIG. 8 connection point of the Zener diode Zl and the resistor provided. If one assumes ideal transistors, so

13 14

the insertion and return attenuation of the addition and return attenuation only if the circuit according to FIG. 8 all the ideal value, since the collector load does not consist of subscriber lines. Alternating current equivalent circuit for the arrangement This results from the following investigation according to FIG. 8 is the same as for the hand arrangement of FIG. 11, which represent the alternating current substitutes of FIG. 3. 5 circuit for F i g. 10 shows when the emitter is the supply source. Attenuation and back attenuation are due to the fact that the AC equivalent circuit according to FIG. 11 assumed that each subscriber line is derived as follows: Each line impedance is assumed to be a signal source in series with an equivalent 2 kOhm. The impedance R _{s of} the switching line impedance can be imagined and that each line network consisting of the switching equipment directly to the conference circuit to an emitter, which each subscriber line is connected to the confeter or collector connection. In a remote circuit that connects to 0.5 kOhm of real telephone system, however, this is not the case. The G resistors all have the value Fall. If you also think of each line as a signal - 1 kOhm and the .F resistors all source the value in series with an equivalent impedance - 15 0.25 kOhm. It should be noted that the B and C terminals can, but also the switching network sistors of each stage in the equivalent circuit, do not show that the line with the conference circuit is in charge. These transistors are omitted, connects certain impedances. since they are only intended for prestressing purposes. At F i g. 9 it is assumed that the switching and an infinitely large impedance for AC supply network brings a series resistance R _s . 20 represent. The collector load for each ^ transistor during the conference calls is Rlc and has the value of the line impedance, F i g. 3 and 8 ideal transmission properties for namely 2 kOhm. The emitter of each D-transistor ideal telephone systems are connected directly to the F-resistor of the next stage in a practical case the transmission properties, since the Zener diodes in the AC are not ideal even if the conference circuit 25 substitute circuit can be omitted. When is ideal. also each D emitter still with a series circuit. The third embodiment of the invention is connected to a resistor with a capacitor the loop is turned on, so the C and D are counteracted. This gain is the possibility of 30 stands each stage large enough to be omitted from the Wechseldie ideal values for the insertion and Rückdämp- current investigation may levies also be achieved if the mediation at the junction of the E- and D- Resistors network that connects the subscriber lines with the conference voltage generated in each stage is only required for pre-limit switching, a series resistor is required for voltage purposes, so that each subscriber line inserts in one sub. 35 investigation of the alternating current behavior of the C-, E- and Z There are two fundamental differences) resistors and the associated shunt between the circuit shown in F i g. 8 and the switching capacitor in each stage are neglected according to FIG. 10. Once each stage contains two cans. In Fig. 11 only three stages are shown. The additional resistors, for example RIF and RIG, different voltages in the fourth and and an additional transistor, for example QIE. 40 fifth as well as in the sixth and seventh stage he- This combination provides the required amplification to be generated, are identical to those in the second kung, in order to determine the values of both the insertion loss and third stage voltages,
as well as the back attenuation for each subscriber line. The impedance of each subscriber line is to be improved to 2 kOhm. and the supply voltage is assumed to be 4 volts. The second difference is that the voltage generated at each collector of the ^ transistors in Fig. 10 should not be connected to any other subscriber line, i.e. subscriber lines. Instead, it should also be 2 volts. The feeding subscriber consists of the collector load of each transistor only line sees a switching network impedance of a resistor. Therefore the circuit needs 0.5 kOhm in series with Zj _n , where _{Zt n is} the Impen according to F i g. 10 is a larger number of stages than the 50 dänz that one moves in the direction of the emitter of the circuit according to FIG. 8, since only one subscriber line sees transistor Ql / i. Can be coupled to each step for an infinitely large rear. For attenuation, the feeding subscriber line should see a conference with six subscribers are seven 2 kOhms, and since the switching network requires stronger stages. The basic loop is only 0.5 kOhm brings in Zi _n for ideal return then stable if they have an odd number of steps 55 attenuation 1.5 kOhm. That at the entertains. Each stage provides a phase shift from the method used is to calculate an odd number of stages in that Zi _{n has} the desired value and the circuit is included, the total phase-different voltages are to be calculated that are involved in the shift around the loop 180 °, so that the other subscriber lines are generated. When tung is stable. An even number of stages, on the other hand, would lead to a group that fulfills the requirement from spans to positive feedback and to oscillations in the loop, which is indicated. For this reason, the values obtained for Zi _{n in} seven stages are confirmed,
Required if a conference with six participants is assumed that Zi _{n is} 1.5 kOhm, then see participants. (The circuit according to Fig. 10, the supply source in its own line impedance, on the other hand, of course, if necessary, 65 series with R _s (0.5 kOhm) and Zi _n . The total impedance used for a conference with seven participants is 4 kOhm , and there will be a current flowing.) The compensation effect of the additional 1 mA in the emitter of the transistor QiA. These borrowed elements in each stage improve the on-current flowing through from the collector of transistor QIA

15 16

the resistance with 2 kOhm (Rlc) to earth, there is a voltage on the right side - 2.5 volts, no current in the base of the transistors QlE and so that the assumed value is confirmed.
QID occurs. There is a voltage of 2 volts over Since the base of the transistor Q3D is generated on a spandem resistor and is connected to the emitter of the two voltage of 2 volts, its emitter has also transmitted the transistors QIE and QID. Because of the 5 same voltage. The emitter of this transistor is the emitter of transistor QlE which is at 2 volts, flow connected to the fourth stage. The fourth and fifth 2 mA through the transistor and the emitter resistor stage are identical to the second and third stage, with 1 kOhm. This current of 2 mA must and since the input voltage of the second stage via R ₁ F, whose value is assumed to be 0.25 kOhm is also 2 volts, the values in the fourth and sei, flow, since no current flows into the base of the transistor io fifth stage generated voltages identical to the QlA occurs. The voltages generated across the resistor R ₁ F with in the second and third stage. The voltage generated by 0.25 kOhm is therefore 0.5VoIt. The same applies to the sixth and seventh levels. The emitter of the transistor QIA and thus also the emitter of the D transistor in the last stage, whose base is at a voltage of 1.5 volts, is at the same voltage as the emitter of the because the voltage drop across Rle and R _s 2.5 Volts load 15 .D transistor in the third stage, namely to 2 volts, carries. As a result, the left side of the resistor Da this emitter is connected to the left side of the resistor with 0.25 kOhm at a voltage of 2 volts. with 0.25 kOhm is switched on in the first stage. If the calculated value for Zj »is confirmed, since a voltage in the other stages shows that the group of span-collector voltage of the seventh stage which corresponds to the requirements Is 2 volts, 20 voltages is generated in the circuit. It thus shows, equal to the voltage on the left side of the, that the circuit according to FIG. 10 is the ideal value for the resistance R ₁ F , the assumed value provides insertion and return loss, although this is confirmed for Zin. Telephone system itself, in which the conference

Since the base of transistor QID is at 2 volts, circuitry used is not ideal.

its emitter has the same voltage. At each stage of the circuit according to FIG. 10 is only

Investigating the second stage of the circuit, a subscriber line is coupled. So that you can

assumed that the base of the transistor Q 2 A ideal value for the return and insertion loss

is at a voltage of 2.5 volts. Since the emitter holds, the collector load of each ^ transistor must be the same

is at the same voltage, a current flows from the line impedance (2 kOhm). If partial

1 mA through the series connection of Rle and R _s 30 slave lines to the collectors of the ^ 4 transistor

with 2.5 kOhm. This current produces a voltage ren coupled would be the collector load

from 2VoIt over Rle, namely the required value due to the

for an ideal insertion loss between the switched impedances is 2.5 kOhm. Out

feeding subscriber line and the one to the second for this reason becomes the ideal value for the return and

Level connected subscriber line. The current 35 insertion loss is only obtained if partial

from ImA through the transistor Q 2 A generates a slave line only to the emitter of the ^ 4 transi-

Voltage drop of 2 volts across the collector load are switched on.

of 2 kOhm. The voltage of -2 volts becomes. A similar result would be obtained if

The emitter of the transistor Q2E is transmitted, and there subscriber lines only to the collectors of these

2 mA flow through this transistor. The current flows 40 transistors were coupled. The one with a scarf

through the resistance with 0.25 kOhm and through direction like the one according to FIG. 10 related problem

the transistor QID to earth. Hence one stands in that the number of stages is equal to the number of

Voltage of 0.5 volts across the resistor with participant lines for which the conference switch

0.25 kOhm generated. It had been shown that the direction must be built up when the number is odd

left side of the resistor is at 2 volts. Hence 45 is, and must be 1 greater than this number if the

the right side is at 2.5 volts, so that he assumed the maximum number of subscriber lines for which the

mene value is confirmed. Conference connection is wanted.

The voltage of -2 volts at the base of the Tran-In ^{- in} a practically executed system, every sistor Q2D does not have the ideal phase shift of 180 ° sistor at the emitter of this Tran-stage either. When examining the third stage, 50 will effect. If each stage brings a phase shift for the base of transistor Q3A with a different voltage value of 180 °, no oscillations would be assumed voltage value. That is -2.5 volts. The possible. However, if each stage has a phase-ver emitter of transistor Q3A on the same shift, which produces slightly larger or slightly smaller voltage, so that 1 mA over the entire emitter load is than 180 °, oscillations are possible. They arise from a flow of 2.5 kOhm. Again, a voltage 55 is created when an additional 180 ° phase shift of 2 volts is created across the emitter load. The one in the loop is generated. So if every stage has a loss of communication, it is ideal. The current of 2 mA additional phase shift of 180/7 or 25.71 ° flows through the collector load of the transistor Q3A, and or a phase shift that is 25.71 ° smaller because the collector of this transistor is on a span, the circuit oscillates. For this reason a voltage of 2 volts. Since the emitter of transistor Q3E 60 may be advisable in certain applications to have two at 2 volts, 2 mA will flow through the transistor. Subscriber lines instead of just one to each stage - These 2 mA have to be coupled from transistor Q 2 D via the to. Even if the return and insertion attenuation connected to the collector of transistor Q3E is not ideal, the smaller resistance does come with 0.25 kOhm. The voltage generated over the number of steps required in the circuit to resistance with 0.25 kOhm is 65 of greater stability. The fourth embodiment thus 0.5 volts, with the left side being positive with the example of the invention, with reference to the right side. Since the left side of the Fig. 12 differs from the circuit according to FIG. 10 resistor is at a voltage of -2 volts, to the effect that only three stages are provided

and two subscriber lines, rather than just one, are coupled to each stage.

The investigation for the behavior of the circuit according to FIG. 12, the fourth embodiment of the invention, is that above for the circuit of FIG. 10 similar investigation. The insertion and return attenuation can be examined for the two cases in which the feeding subscriber line is coupled to the collector or emitter of a ^ transistor. The investigation just carried out assumed that the switching network introduced a series impedance into each line. In certain telephone systems, it can be shown that the impedance introduced is equivalent to the circuit shown in FIG. Here the introduced impedance consists of a T-link with two small series resistances r and one large transverse resistance R '. In a well-developed system, the T-link has a wave resistance which is equal to the connection impedance. Then the T-link introduces damping with a flat profile. The attenuation is usually measured in dB. When examining a circuit in which each subscriber line is connected to the conference circuit via the network according to FIG. 13, the T-element consisting of resistors can be neglected in the calculation of the voltage measured in dB which is transmitted from the supply source to each subscriber line will. After the signal level for each subscriber line has been calculated, the actual signal level in db can be derived by subtracting the flat attenuation introduced by the T-element from the calculated value.

F i g. 14 is the AC equivalent circuit for F i g. 12. There each load is shown in the form of a line impedance of 2 kOhm. Impedances R _s are not present. The equivalent circuit of each stage in Fig. 14 is basically the same as that for each stage in FIG. 11. The collector load for each Λ transistor is again 2 kOhm, which corresponds to the impedance of a subscriber line. The emitter loads are 2 kOhm instead of 2.5 kOhm. In Fig. 14 it is assumed that the emitter subscriber line of the first stage is the supply source. The input impedance seen by the feeding subscriber line can be calculated along with the voltage generated on each of the other five subscriber lines. The actual subscriber line to subscriber line gains can then be determined by subtracting from each calculated value the attenuation introduced by the T-links. Since the T-elements do not noticeably influence the currents and voltages in the circuit, they are neglected in FIG.

In FIG. 14, each line impedance is 2 kOhm. The voltage V of the supply source thus supplies a current to the emitter of the transistor QIA via an impedance of 2 kOhm. The emitter of transistor QID is assumed to be 3 volts. Then there is the same voltage at the base of this transistor, and a current of 1.5 mA flows through the collector load of the transistor QIA. (This current also flows through the emitter load of transistor QIA). A voltage drop of 3VoIt occurs at the collector load. The emitter of transistor QIE is also at a voltage of 3 volts, and a current of 3 mA flows through the transistor, which also flows from left to right through the resistor with 0.25 kOhm in series with the collector of transistor Q1E . The voltage drop across the resistor with 0.25 kOhm can be calculated, but the voltage at the connection point between the collector of transistor Q1E and the base of transistor Q1A is not yet known.

To relieve tension at different points in the

To calculate the second stage, it is assumed that the base of transistor Q2,4 is at 4 volts. The same voltage prevails at the emitter of the transistor, and 2 mA flows through the transistor and the collector and emitter loads. The base of transistor Q2E is therefore at -4 volts, and since the emitter of this transistor is at the same voltage, 4 mA flow from right to left through the resistor with 0.25 kOhm in series with the collector of transistor Q1E. This 4 mA current flowing through the resistor and transistor Q1D creates a voltage drop of 1 volt. Since the emitter of transistor QID is 3 volts, there is a voltage of 4 volts at the base of transistor QIA, as assumed.

The base and emitter voltage of transistor QID is -4 volts. It is assumed that the base of transistor Q3A is at -16/3 volts. The currents in the third stage can be calculated in a corresponding manner and the assumed voltage value can be confirmed. The base and emitter voltage of transistor Q 3 D is -16/3 volts. This is the voltage on the left side of the leftmost resistor with 0.25 kOhm. Since the current through this resistor is 3 mA, the voltage across it is 0.75 volts. The base of the transistor QIA is thus at a voltage of (-16/3) + (3/4) or 55 / 12VoIt. The emitter of the transistor QIA is at the same voltage. The voltage drop across the emitter load of transistor QIA is 3 volts. Consequently, the voltage V required to generate the various voltages and currents derived in the circuit, assuming that the emitter of transistor QID is at a voltage of 3 volts, is equal to (55/12) + 3 or 7.58 Volt.

The voltage on the five passive lines can be calculated by multiplying each line impedance of 2 kOhm by the current flowing through it. The ideal voltage for each subscriber line is equal to half of 7.58 volts, or 3.79 volts. The actual voltages that are generated for the individual subscriber lines are as follows: A voltage of 3 volts occurs at the collector load of the transistor QIA. Both loads in the second stage have a voltage of 4 volts and both loads in the third stage have a voltage of 16/3 volts.

In Fig. 15 is the absolute value for the ratio between each subscriber line voltage and the desired voltage shown. The five ratios are also given in db. As obsn said, is one additional attenuation in each subscriber line due to the equivalent consisting of resistors T-members are present which are introduced through the network which the subscriber lines connects to the conference call. If the attenuation introduced by each T-link is 1 db b, the calculated gain of participant

line to subscriber line can be reduced by 2 db to reflect the actual gain in the system derive. These values are in the last column of the table according to FIG. 15 shown. Two subscriber lines actually have a reinforcement. Another two have an attenuation of only 1.6 db, and only a subscriber line has an attenuation of 4 db.

So if only three stages are used for a conference call with six participants, the gains are different from the case according to FIG. 10 not even. However, the unevenness is not great. It is essential, however, that the input impedance seen by the respective feeding subscriber line is always positive so that no oscillations occur. The input impedance Zi _n can be calculated by dividing the voltage at the emitter of transistor QIA, 55/12 volts, by the 1.5 mA current entering the terminal. The input impedance is then 3.06 kOhm. The back attenuation can now be calculated. According to F i g. 15 is the impedance Zle of the feeding subscriber line 2 kOhm and the back attenuation 14 db. This value is high enough to avoid noticeable vibrations.

One must remember that the insertion of the combination with a negative resistance in each The circuit stage is not due to the way the conference circuit itself works. The insertion and back attenuation for the conference call of FIG. 8 is ideal when there is no impedance is introduced into the subscriber lines through the switching network that runs the subscriber lines connects to the conference call. In practice, however, the switching network carries a certain amount Impedance on. The reinforcement provided in each stage compares the faulty connections out. If only one subscriber line is coupled to each stage, such as in FIG. 10, and the Impedance of the switching network is of the type shown in Fig. 9, the ideal values for achieve the insertion and return attenuation, with the conference call coming from the switching network inserted attenuations are fully compensated. However, a complete compensation can be made not reach if, as can happen in a practical installation, two subscriber lines are connected each stage are coupled. However, the transmission properties of the entire system can still can be improved if reinforcement is provided in each stage.

The immediately preceding investigation related to the case that the introduced impedance of the switching network corresponds to that shown in FIG. 13 and that the feeding subscriber line is one of the three emitter loads. A corresponding investigation can be carried out in the event that a collector line is the supply source. The impedance R ₃ according to FIG. 9 can be added to any load if the impedance of the switching network is a series impedance rather than a T-junction. Then similar results are obtained. It can in fact be shown that if in the circuit of FIG. 14 the feeding line is one of the collector loads, the values for the insertion loss are identical. Only the input impedance Zt _{n changes} , which is now 1.31 kOhm instead of 3.06 kOhm. The same value of 14 db is obtained for the back attenuation.

F i g. 16 shows a fifth embodiment of the invention. The basic circuit is that according to FIG. 12th similar. Each stage contains the additional T resistor. However, there is a fundamental difference. In Figure 12 (and in Figure 10) the base of each D transistor is taken directly from the collector of the corresponding one ^ 4 transistor (which is at the same voltage as the emitter of the IT transistor) fed, the D transistor connecting the collector of each ^ transistor to the base of the yi transistor in the subsequent stage connects. According to FIG. However, 16 is the collector of the ^ transistor in each stage not connected to the base of the corresponding D transistor. Instead, the D resistance is in each stage now consists of two separate resistors, and the connection point of the two resistors is connected directly to the base of the D transistor to feed the next stage. The D resistors in Fig. 12 each have a value of 1 kOhm (cf. Fig. 14). The total resistance in the emitter circuit of the .Ε transistor in each stage according to F i g. 16 is 1 kOhm. The number / is between 0 and 1. The upper resistance of each pair has a value / kOhm and the lower resistance of each Pair of (1 - /) kOhm. In Fig. 12, the base of each D transistor is drawn from the collector voltage of the corresponding ^ -Transistor fed. The collector voltage of the ^ transistor is equal to the base and emitter voltage of the Is transistor. In the circuit according to FIG. So every D transistor becomes 16 instead of the full collector voltage of the Λ-transistor only fed by a fraction of this voltage, namely the voltage multiplied by /.

The reason for introducing the term / in the circuit is that the additional variable enables a greater number of settings. For example, a value for / can be selected such that the return loss for each line becomes infinitely large, ie Zi _n is 2 kOhm for each line. It is assumed that the impedance of the switching network introduces an attenuation with a flat profile. The various reinforcements are uneven, but all improved. Other values of / result in improved gains, but less back attenuation. In a practically implemented system, a compromise must be found, since the values for the insertion and return attenuation can only be improved at the expense of the other value. Investigations corresponding to the above can be used for the circuit according to FIG. 16 to derive the actual values of the return and insertion loss for given values of /. In all cases, vibrations can be avoided and the circuit is stable. It should be remembered, however, that the resulting non-ideal values for the return and insertion loss are not due to imperfections in the conference circuit itself. On the contrary, the "simple" circuit according to FIG. 8 provides the ideal values when the switching network is ideal. The at F i g. 10, 12, and 16, to varying degrees, compensate for imperfections in the remaining parts of the telephone system.

F i g. Fig. 18 shows the sixth embodiment of the invention, the perfect one with only three stages Compensation leads, d. that is, the return and insertion loss values are ideal. It is remembered

21 22

that the ideal values with the circuit according to FIG. 8 ^ transistor connected. The emitter of each D-Tran can be reached when the load seen by the conference connection is no longer seen via a Zener diode. The impedance of the subscriber line is the base of the ^ 4 transistor in the subsequent stage. Each port in FIG. 18 contains a T-link with switched on. Instead of the Zener diode, a capacitor is provided in each negative resistor that completely compensates for the impedance introduced by the switching stage, a parallel circuit of a resistor and network. In the first stage, if one assumes that the impedance of the is the resistance RIG and the Konin F i g. 13 is the type shown, since the switching capacitor ClB. The capacitors in FIG. The impedance introduced in the network completely eliminates a short circuit for the alternating current signals. As will be obtained, the ideal values for the insertion io the Zener diodes in Fig. 8. The G resistors attenuation and back attenuation are obtained. act a voltage drop for bias purposes

Each port in FIG. 18 contains a T-element like the Zener diodes. The parallel connection of the counter-negative resistor, which has two resistors with a stand with the capacitor in each stage, represents the value - r and a resistor with the value —R ' merely provides another means of biasing. All known types of negative 15 of the yi transistors can be used. The ^ resistor can be used in every tive resistor. According to FIG. 17 stage is also used for biasing purposes and each T-element with negative resistances, which in this case improves the dynamic current range, is connected upstream of a connection, in series with that of the D-stage.

Impedance of the switching network. The two Each stage contains the network Z, which in connection

Resistances r and -r in the middle of the overall network with F i g. 3 has been described. The network cancel each other and therefore short-circuited factory Z are each enclosed in a conference call. The parallel that then results when the stability of the system is improved. Connection of R and - R leads to an overall target. Corresponding transistors in FIG. 8 and shunt impedance 19 are not all of the same type. For example

(R) . c__ R) KR H- R) ^{25 s} ' ^{n <} ^ ^ ^e ^ transistors according to F i g. 19 of the pnp type.

These differences in Fig. 19 are primarily

This transverse impedance is infinitely large and can be provided to show alternative possibilities, therefore be neglected. All that remains is that it should be remembered that when examining the

nor the two external resistors —r and r, the various exemplary embodiments of the invention are in series. These resistors also cancel each other. It was assumed that ideally the transistors are off, and as a result everyone is conference on and no current flows into the base of a transistor directly across an effective impedance 0 to one. However, as at the beginning of the description, he-subscriber line is connected. Consequently, the base of a transistor sees, refines, draws in practice, although the switching network has a certain impedance, a small current, and for this reason the danz is inserted into every subscriber line, every conference, every stage is not 1, even though it is this connection an ideal subscriber line, and the value comes close. The resistor network with the ideal values for the insertion and return attenuation resistances H, / and L in each stage is dimensioned to be achieved. If the effective impedance introduced by the switching network that positive feedback with such a form is realized other than the amount that the gain of each of the values shown in FIG. 13, an equivalent 40 stage corresponding to the middle transistor B of the element with negative resistances can come closer to each ideal value. The positive feedback terminal connection should be switched on in order to achieve the ideal values based on FIG. 20 which are used to achieve the insertion loss and return loss. first stage of the circuit according to FIG. 19 with ver The exemplary embodiment according to FIG. 19 shows various current, voltage and switching elements showing another way of improving the transmissions.

properties of the conference call. The ideal case would be the entire collector current of the

Strengthening each stage in the circuit of FIG. 8 is transistor QIA to flow collector load. That would be about 1, but due to the fact that this is actually the case when the transistor QIB transistors are not ideal, the gain at its base with a separate voltage source is somewhat smaller than this ideal value. Embodiment 50 would be biased and the base of transistor QID of FIG . 19 shows a method of slightly drawing no current. For the following investigation, increasing the gain of each stage so that the ver increases, it is assumed that a negative signal is due to the gain from one stage in the loop to another - base of the transistor QIA to an increase in the ren closer to the ideal value 1. Collector current leads and that a small part of this

The bias voltage generation for the circuit after 55 current Δ i flows into the base of the transistor QID. F i g. 19 corresponds to the basic method. As will be shown later, this current generates the according to FIG. 8. However, there are certain differences in positive feedback. The feedback is small between the two circuits. The base of this because only a small amount of reinforcement is required. In J5 transistor in each stage it is no longer directly indicated in connection therewith that that of a separate source is biased. Instead, 60 resistance RlL is measured in ohms, while a voltage divider with resistors RlH, RlJ, resistors RIA, RlH and RlJ in kOhms and RlL are used in the first stage to give the basis. The current Al going to bias into the base of the Tranjedes ^ transistor. This voltage transistor QID enters, is amplified, and it flows a divider is used to a small positive return current ßAi to the collector of the transistor. This coupling in each stage of the conference circuit to 65 current comes almost exclusively via the realizer, so that the gain is closer to the RIL, since this is small compared to the other ideal value 1. The base of the D transistor in each resistor in the circuit is. If Δ i is measured in ampere level again with the collector of the corresponding, it results across the resistor RlL

generated voltage ziv to 38.3 (βA i) volts as shown.

The resistor RIL is parallel to the series connection of the resistors RlJ and RlH. The one at the base of the transistor β Ii? generated tension at therefore protrudes

18.7OzIv

(18.70 + 2.74)

or 0.87 / 1 v. Since the base and the emitter of the transistor gli? are about the same voltage, the additional voltage appears across the resistor RIA. The current Zh ", which flows up through this resistor, is therefore

0.87.4 ν

_: 1960 amps

Almost all of this current flows out of the collector of transistor QIB since the base of transistor QIB draws almost no current. The current zh "flows through the collector load, and if the value for β is chosen such that Ai '= Δ i , the gain of the stage is increased to 1, since the reduction in the load current is compensated for by the transistor QID . For one stage with the gain 1 it results

Zl / '=

0.87zl v _ 0.87 (38.3) (βΔ i)
1960 ~ 1960

and β = 58. When β the value for denTransi- stoxQlD slightly larger than 58 selects, for example (in order to create a balance dadür that the collector current of the transistor QLB slightly smaller than the emitter current), the gain of the stage can actually the Have an ideal value of 1.

Claims (15)

Claims: 40 1. Conference circuit for a plurality of two-wire subscriber lines for two-way communication with an associated plurality of amplification devices interconnected to form a loop, each of which is contained in an amplifier stage lying in the loop with a gain of approximately one, but less than one, with a bias circuit for Each amplification device which keeps it in the conductive state and with the amplification devices associated coupling circuits, each of which couples a subscriber line to the conference circuit, characterized in that to improve the stability and to increase the back attenuation, the loop has an odd number of 2 "-1 (n any positive integer greater than 1) has amplifier stages and each coupling circuit coupling a subscriber line (transformer in FIG. 3 to 7 or coupling capacitors in FIG. 8, 10, 12, 16 and 18 to 20) does not appear the control electronics lying in the loop de (base), but is directly connected to one of the two remaining conductive electrodes of the same amplifier stage, so that the signals impressed there, which have passed through the amplifier loop to the control electrode of the same amplifier stage, are no longer amplified in it. 2. Conference call according to claim 1, characterized in that each amplifying device two coupling circuits each (transformer in FIG. 3, coupling capacitors in FIG. 8) assigned that are attached to each amplifier stage of the conference call Connect two subscriber lines each (1 to 6 in FIG. 3). 3. Conference circuit according to claim 1, characterized in that each amplifier stage is assigned an auxiliary amplifier arrangement (ßl £ ', RlF, RlG in Fig. 10) which neutralizes an insertion loss or an attenuation in the subscriber line. 4. Conference circuit according to claim 1, 2 or 3, characterized in that each amplifier stage is assigned a network for positive feedback (Q 1B, RlH, RlJ, RlL in Fig. 19) which compensates for attenuation occurring between the amplifier stages. 5. Conference call according to claim 1 or 2, characterized in that each amplifying device consists of a transistor (10 in Fig. 3; oil ^ in Fig. 8). 6. Conference call according to claim 5, characterized in that the bias circuit a Zener diode (15 in F i g. 3), which between the collector of the transistor one Amplifier stage and the base of the transistor of a subsequent amplifier stage is connected. 7. Conference circuit according to claim 5, characterized in that the bias circuit contains an i? C parallel circuit (RIG, ClB in Fig. 19). 8. Conference circuit according to claim 6, characterized in that the bias circuit has a first circuit with a transistor (QID in FIG. 8) connected between the collector of the transistor (Q IA in FIG. 8) of an amplifier stage and the Zener diode Compensation of the signal attenuation due to the Zener diode, and a second circuit with a transistor (Q 1C, RLE, RID in FIG. 8) which controls the quiescent operating point of the transistor (RIA) of the amplifier stage so that the gain the loop is held at the value 1. 9. Conference circuit according to claim 8, characterized in that the second circuit has a capacitor (Cl in Fig. 8) which derives voltage changes caused by the impressed signal or speech alternating voltage from the control electrode of the transistor (Q 1 C) of the second circuit, so that the transistor (Q 1 C) can maintain the quiescent operating point of the transistor (QIA) of the amplifier stage. 10. Conference call according to claim 2 and 5, characterized in that each amplifier stage has two transformers (F i g. 3), which are directly connected to the emitter or collector of the transistor Amplifier stage are switched on, and that each transmitter has a subscriber line to the conference call coupled. 11. Conference call according to one of claims 5, 7, 8 or 9, characterized in that that the coupling circuit has a capacitor (capacitors in FIG. 8) which is directly connected to the emitter or collector of the transistor is coupled to the amplifier stage. 009 547/178 12. Conference circuit according to claim 11, characterized in that the coupling circuit has a member with negative resistances ( -r, -r, -R ' in FIG. 18) in order to neutralize the load properties of a connected subscriber line. 13. Conference circuit according to claim 3 and 8, characterized in that the auxiliary amplifier arrangement has a transistor (QlE in F i g. 16) whose collector is connected to the base of the transistor of the amplifier stage and whose emitter has a real impedance (RlG in F i g. 12) is connected to a bias voltage source. 14. Conference circuit according to claim 13, characterized in that the real impedance consists of a series-connected first and second resistor (RIG in FIG. 16) that the The first resistor has the value / (i?) and the second resistor the value (1 - /) (R) , where / is a predetermined value greater than 0 and less than 1 that the second resistor is coupled to the transistor (QIE) and that the base of the transistor (QID) of the first circuit of the bias circuit is coupled to the junction between the first and second resistors. 15. Conference circuit according to claim 4 and 5, characterized in that the network for the positive feedback has a transistor (Q IB in Fig. 19) which is connected to the emitter of the transistor of the amplifier stage, and that a biasing device (Aiii, RlJ , RIL in FIG. 19; 810, RIA in FIG. 8) biases the transistor in such a way that it brings about positive feedback (positive feedback). For this purpose 4 sheets of drawings