CN201430532Y - A zero-voltage switch flyback DC-DC power conversion device - Google Patents

Abstract

The utility model relates to a DC/DC power conversion device, in particular to a zero-voltage switch flyback DC-DC power conversion device with high efficiency conversion, high efficiency conversion under light load and low standby power consumption. An auxiliary switch and absorption capacitor are added to the flyback circuit. The auxiliary switch and the absorption capacitor are connected in series to form an auxiliary branch. The auxiliary branch can be connected in parallel at both ends of the transformer primary winding or at both ends of the primary switch. Turns on for a set period of time only before the primary switch turns on. Compared with the prior art, the utility model transmits the energy of the leakage inductance of the line to the output terminal after being absorbed and is used to realize the soft switch of the primary side switch, and the efficiency of the circuit can be greatly improved; the parasitic oscillation caused by the leakage inductance is suppressed, and the circuit EMI characteristics can be improved; the control of the circuit is simpler, which can greatly improve the efficiency of the circuit at light load and reduce the loss at no load.

Description

A zero-voltage switch flyback DC-DC power conversion device

technical field

The utility model relates to a DC/DC power conversion device, in particular to a zero-voltage switch flyback DC-DC power conversion device with high efficiency conversion, high efficiency conversion under light load and low standby power consumption.

Background technique

DC/DC conversion is one of the most basic forms of electrical energy conversion. Due to its simple topology and few components, the flyback converter is widely used in low-power DC/DC conversion, usually below 100-200W. The loss of the flyback converter mainly includes the loss of the primary switch, the loss of the transformer, the loss of the snubber circuit and the loss of the secondary rectifier. These losses are also closely related to the control mode of the flyback converter.

Usually, the transformer of a flyback converter is the main component for storing and transmitting energy. Due to the existence of leakage inductance, a large spike voltage is generated when the switch tube on the primary side is turned off, and an additional absorbing circuit is required to absorb and consume the leakage. Inductive energy, such as the commonly used RCD clamp absorption circuit, but the energy consumed by the absorption circuit reduces the efficiency of the circuit.

In recent years, with the continuous improvement of the efficiency requirements of power supply products in the world, such as the Energy Star of the United States and the energy efficiency standards of the European Union, the power conversion efficiency has become a focus of power supply design. At present, the efficiency requirements are not only for the full load condition, but also for the efficiency under various other load conditions. Usually, under the load conditions of 25%/50%/75%/100%, the average value of the measured efficiency needs to be fulfill the standard. Therefore, how to improve the efficiency under light load conditions has also become a key. Moreover, under no-load conditions, the standby loss of the converter is required to be extremely small, usually less than 0.3W, or a smaller value.

The traditional fixed-frequency flyback converter cannot meet the current standards under light-load and standby conditions due to its fixed switching frequency. More control methods are researched and proposed to improve efficiency and reduce no-load loss.

In terms of improving efficiency, one method is the active clamping technology, which mainly absorbs the energy of the leakage inductance, and uses the energy of the magnetizing inductance or the energy of the leakage inductance to realize the soft switching of the primary side switch and improve the efficiency. However, this type of switch needs to add an auxiliary switch, which is realized through the complementary control of the auxiliary switch and the primary switch, and both adopt constant frequency control.

Another method is to use the quasi-resonance (QR) method to improve the conversion efficiency of the power supply, and use the flyback converter to work in the current critical discontinuous mode (Critical DCM) or current discontinuous mode (DCM), the excitation inductance and the original The parasitic oscillation of the side switch realizes that the primary switch is turned on at the lowest point of its drain-source voltage (or collector-emitter voltage) or when the drain-source voltage is the input voltage to reduce switching loss. The continuous or discontinuous current referred to here refers to the continuous or discontinuous current of the excitation inductance of the transformer. The flyback converter adopts the quasi-resonant control method, and its switching frequency will change with the change of the load and the input voltage. A relatively common control is to use the load terminal (output side) feedback to adjust the peak value of the primary current, so as to control the output power, that is, to control the conduction time. Generally, the switching frequency becomes higher as the load becomes smaller, which is not conducive to the improvement of light-load efficiency and the improvement of electromagnetic interference. Usually, additional frequency clamping or hiccup mode (Burst Mode) is required to improve light-load and standby power consumption, control Electromagnetic interference caused by high frequency. Another way is to keep the peak value of the primary current unchanged, and adjust the output power by adjusting the off time, that is, to control the off time, so that the frequency decreases with the decrease of the load, which is beneficial to light load efficiency and standby power consumption. However, the switching frequency usually drops to the audio range (below 20kHz) at very light loads, so noise in the audio range needs to be avoided.

Although the quasi-resonant control method can reduce the switching loss, in the case of relatively large input range (for example, the AC range suitable for global use is 90V ~ 265VRMS, and the DC bus voltage formed after rectification is usually 100V ~ 380VDC), usually at low input The soft switching of the primary side switch can be realized under the condition of , but there is still a large switching loss under the condition of high input. Moreover, the energy of the leakage inductance of the transformer still needs to be clamped and absorbed by the absorbing circuit. Likewise, with off-time control, switching losses can be reduced by taking advantage of frequency drops, but switching losses still exist at high input voltages or when load conditions change.

In the case of using a synchronous rectifier on the secondary side, the current on the primary side of the transformer can be reversed by using the synchronous rectifier on the secondary side, and the soft switching of the primary side switch can be realized. The leakage inductance energy can be absorbed and returned to the input terminal by connecting a capacitor in parallel with the switch tube. , to achieve non-destructive absorption of leakage inductance energy, but the design of the capacitance is related to the withstand voltage of the switch tube, the leakage inductance of the transformer, and the size of the circulating energy, so the design is difficult. Moreover, due to the need to use the output energy to make the current of the exciting inductor reverse to realize the soft switching of the primary switch, the cycle energy of the circuit increases, resulting in an increase in the conduction loss of the line, so that the saved loss of the absorption circuit is replaced by the increased conduction loss. compromise. In addition, the control circuit of the synchronous rectifier will be more complicated, and the prior art usually adopts an external driving method.

Attached Figure 1 is a traditional flyback circuit, Vin represents the DC input terminal, and the load Load is connected across the output port. When the switch Q1 is turned on, the transformer stores energy (that is, the magnetizing inductance Lm of the transformer stores energy), and the rectifier diode Q1 in the output rectifier circuit is reverse-biased and cut off. During the off period of Q1, the energy stored in the transformer is released to the output through D1, providing a DC current for the load. The above working principle of the flyback converter is a kind of common knowledge, and will not be repeated here. The flyback transformer marked in Figure 1 is a commonly used model, which is equivalent to a leakage inductance Lk, a magnetizing inductance Lm and an ideal transformer. By adjusting the turn-on time, duty cycle or turn-off time of switch Q1, the energy storage of the transformer in each switching cycle can be adjusted, thereby adjusting the output of the DC side. Through the well-known negative feedback method in the industry, the voltage/current or Power feedback can stabilize the output voltage, current or power. In the traditional flyback circuit, due to the influence of the transformer leakage inductance Lk, the RCD clamping absorption circuit is required to absorb the spike voltage and prevent the overvoltage of the switching tube Q1, no matter the circuit adopts the fixed frequency or variable frequency control mode.

In order to further improve efficiency, the active clamp flyback circuit shown in Figure 2. By adding an auxiliary switch Qa, when the primary side switch Q1 is turned off, the energy of the leakage inductance is absorbed into the clamp capacitor Cr, and then released to the load or the input terminal. Usually the switch Q1 and the auxiliary switch Qa work in a complementary state, as shown in the gate drive waveforms of Figure 3 and Figure 4, that is, after Q1 is turned off, the auxiliary switch Qa is turned on, and there is actually a small gap between them. The dead time is to prevent the shoot-through phenomenon caused by non-ideal switching characteristics and damage the circuit. This period of time is relatively short and can be ignored. This is also common knowledge. According to the working mode of the magnetizing inductance current of the transformer, there are two working modes, that is, the unidirectional [1][4] of the magnetizing inductance current and the bidirectional [2][3] of the magnetizing inductance current, as shown in attached drawings 3 and 4 respectively. Show. Since the switching states of Q1 and Qa are complementary, the transformer excitation inductance current is in continuous mode (CCM) regardless of unidirectional or bidirectional. Therefore, in practical applications, a fixed frequency control method is usually used. Although the active clamp flyback circuit can avoid the RCD clamp snubber circuit and reduce circuit loss. However, when the load becomes lighter, since the magnetizing inductor current is continuous, the circulating energy is large, resulting in low efficiency. Furthermore, during the conduction period of the auxiliary switch tube, the leakage inductance of the transformer resonates with the clamp capacitor, and the difference between the leakage inductance current and the excitation current is transmitted to the output. Compared with the traditional flyback circuit, in the case of equal output current , the peak value and effective value of the secondary current are large, resulting in a large secondary conduction loss. In the case of AC input, when the DC voltage is provided by rectification or pre-stage power factor correction circuit (PFC), when the load is light, due to the continuous excitation inductor current and constant switching frequency, the light load efficiency is low and cannot meet the requirements. The currently required standby power consumption must be less than 0.3W without a PFC front stage. The "burst" (Burst) mode can reduce standby power consumption. To ensure normal operation, the control signals of the two switches need to be blocked at the same time, so the control circuit is complicated, resulting in a reduction in the cost and reliability of the circuit.

Figure 5 is a flyback converter [5] with synchronous rectification technology on the secondary side. Since the secondary side adopts synchronous rectification technology, the soft switching of the primary side switch Q1 can be realized by using the output energy to make the exciting inductor current reverse. Therefore, a snubber capacitor Cds can be connected in parallel at both ends of the switch tube Q1 on the primary side to absorb the energy of the leakage inductance (including the parasitic capacitance in the line), and the RCD clamp snubber circuit on the primary side is unnecessary. The waveform is shown in Figure 6. This circuit needs to use the output energy to realize the soft switching of the primary side switch, so the circulating energy of the circuit is relatively large. Similarly, when the primary switch is turned off, the leakage inductance of the transformer generates parasitic oscillation in the parallel capacitance Cds of the primary switch. Since there is no external loss resistance, the attenuation of the oscillation depends on the parasitic resistance of the line, and the amplitude and time of the parasitic oscillation are relatively long , resulting in poor electromagnetic interference (EMI) performance of the line. Therefore, usually an additional RCD clamp snubber circuit as shown in Fig. 1 is still required, but additional loss is introduced. Therefore, how to further improve efficiency, especially light load and average efficiency, and reduce standby loss has always been the focus of research.

references

[1] Robert Watson, et al, "Utilization of an Active-Clamp Circuit to Achieve Soft Switching in Flyback Converters", IEEE Trans.On Power Electronics, vol.11, No.1, Jan.1996, pp.162-169( Robert Weston et al., "Using Active Clamp Circuits in Flyback Converters to Realize Soft Switching", IEEE Journal of Power Electronics, Volume 11, Issue 1, January 1996, pp. 162-169);

[2] Koji Yoshida, et al, "Zero Voltage Switching Approach for Flyback Converter", IEEEINTELEC'92, pp.324-329 (Kongji Yoshida et al, "Zero Voltage Switching Approach for Flyback Converter", IEEE 1992 INTELEC Conference, pp. 324-329);

[3] E.H.Wittenbreder, "Zero Voltage Switching Pulse Width Modulated Power Converters", US Patent 5402329, March 1995 (Wittenbreder, "Zero Voltage Switching Pulse Width Modulated Power Converters", US Patent 5402329, March 1995 );

[4] David A. Cross, "Clamped Continuous Flyback Power Converter", USA patent No.5570278, Oct.29, 1996 (David A. Cross, "Clamped Continuous Flyback Power Converter", US Patent No. 5570278, 1996 October);

[5] M.T.Zhang, Milan M.Javanovic, F.C.Lee, "Design Consideration and Performance Evaluations of Synchronous Rectification in Flyback Converters", IEEE Trans.on PE, Vol.13, No.3, May 1998, pp538～546 (Michael Zhang et al., "Design and Performance Evaluation of Synchronous Rectifiers in Flyback Converters", IEEE Journal of Power Electronics, May 1998, Vol. 13, No. 3, pp. 538-546).

Utility model content

The technical problem to be solved by the present invention is to provide a zero-voltage switch flyback DC-DC power conversion device that can reduce the switching loss of the flyback circuit, the loss of the leakage inductance absorbing circuit, improve the light load efficiency and reduce the standby loss.

In order to solve the above problems, the utility model provides a zero-voltage switch flyback DC-DC power conversion device, including:

One input port accepts DC input voltage, and one output port provides DC power to the load;

A transformer comprising at least one primary winding and one secondary winding;

The primary side switch is connected in series with the primary side winding of the transformer; during the conduction period of the primary side switch, the DC input voltage is applied to the primary side winding of the transformer, and the transformer stores energy; when the primary side switch is turned off During this period, the DC input voltage is disconnected from the primary winding of the transformer, and the energy stored in the transformer during the conduction period of the primary switch is released to the load through the secondary winding of the transformer;

The output circuit is coupled to the secondary side of the voltage transformer, and generates a direct current at the output port from the energy released by the transformer during the turn-off period of the primary side switch, and provides it to the load;

It also includes a clamping circuit, including a capacitor, an auxiliary diode and an auxiliary switch, the auxiliary switch is connected in series with the capacitor, the auxiliary diode is connected in parallel with the auxiliary switch, and provides the voltage for the transformer when the primary switch is turned off. a current path, the clamping circuit is connected in parallel with the primary winding of the transformer or in parallel with the primary switch;

The auxiliary switch is turned on for a set time before the primary-side switch is turned on, and the auxiliary switch is turned off by a set time before the primary-side switch is turned on. At the moment of conduction, the voltage borne by the primary switch is close to zero.

Further, the primary side switch or the auxiliary switch is a metal oxide semiconductor field effect, an insulated gate bipolar transistor or a bipolar transistor.

Further, the auxiliary diode is a parasitic diode of the auxiliary switch.

Further, the time when the auxiliary switch is turned on is ahead of the primary switch is fixed or set.

Further, the turn-on time of the auxiliary switch is less than the turn-off time of the primary switch; the auxiliary switch is not turned on during the conduction period of the auxiliary diode in the clamping circuit;

Further, the exciting current of the transformer works in a discontinuous state or a continuous state;

Further, the output port includes a rectifier device, which is reverse-biased and turned off during the on-time of the primary-side switch, and allows current to pass through during the off-time of the primary-side switch.

Further, the energy absorbed by the capacitor in the clamping circuit during the conduction period of the auxiliary diode is released to the load and the DC input terminal by the auxiliary switch conduction device through the leakage inductance of the transformer.

Furthermore, the rectifying device is a diode.

Further, the output is adjusted by controlling the turn-on time of the primary switch, controlling the turn-off time of the primary switch or controlling the conduction duty ratio of the primary switch.

The utility model is further described below:

A zero-voltage switch flyback DC-DC power conversion device, comprising:

One input port accepts DC input voltage, and one output port provides DC power to the load;

A transformer comprising at least one primary winding and one secondary winding;

The primary side switch is connected in series with the primary side winding of the transformer; during the conduction period of the primary side switch, the DC input voltage is applied to the primary side winding of the transformer, and the transformer stores energy; when the primary side switch is turned off During this period, the DC input voltage is disconnected from the primary winding of the transformer, and the energy stored in the transformer during the conduction period of the primary switch is released to the load through the secondary winding of the transformer;

The output circuit is coupled to the secondary side of the voltage transformer, and generates a direct current at the output port from the energy released by the transformer during the turn-off period of the primary side switch, and provides it to the load;

A clamping circuit including a capacitor, an auxiliary diode and an auxiliary switch, the auxiliary switch is connected in series with the capacitor, the auxiliary diode is connected in parallel with the auxiliary switch, and provides a current for the transformer during the turn-off period of the primary switch path. The clamping circuit is connected in parallel with the primary winding of the transformer or in parallel with the primary switch;

The auxiliary switch is turned on for a set time before the primary-side switch is turned on, and the auxiliary switch is turned off by a set time before the primary-side switch is turned on. At the moment of conduction, the voltage borne by the primary switch is close to zero.

Further, the primary switch or auxiliary switch may be a metal oxide semiconductor field effect (MOSFET), an insulated gate bipolar transistor (IGBT), or a bipolar transistor (BJT); the auxiliary diode may be is the parasitic diode of the auxiliary switch;

The turn-on time of the auxiliary switch is fixed or set before the turn-on time of the primary switch, and does not change with the turn-on time or turn-off time of the primary switch, nor with the turn-on time of the primary switch. The on-duty ratio changes; the turn-on time of the auxiliary switch is less than the turn-off time of the primary switch; the auxiliary switch is not turned on during the turn-on period of the auxiliary diode in the clamp circuit;

Wherein, the excitation current of the transformer works in a discontinuous state or a continuous state; the transformer can be a plurality of transformers connected in series or in parallel.

Further, the output circuit includes a rectifier device, which is reverse-biased and turned off during the turn-on time of the primary switch, and allows current to pass through during the turn-off time of the primary switch.

The energy absorbed by the capacitor in the clamping circuit during the conduction period of the auxiliary diode is released to the load and the DC input terminal through the leakage inductance of the transformer at the conduction device of the auxiliary switch.

The rectifying device is a diode, which may be a metal oxide semiconductor field effect (MOSFET). Further, the output is adjusted by controlling the turn-on time of the primary switch, controlling the turn-off time of the primary switch or controlling the conduction duty ratio of the primary switch.

The DC input voltage of the input port can be obtained from the AC of the power grid through a diode rectifier circuit, or it can be a DC voltage output by other conversion circuits. The DC input voltage can be constant or have a large range of variation, such as 3 ~5-fold change.

Compared with the prior art, the circuit structure and control method adopted by the utility model have obvious advantages. The energy of the leakage inductance of the line is absorbed and then transmitted to the output terminal and used to realize the soft switch (zero voltage switch) of the primary side switch. , ZVS), the efficiency of the circuit can be greatly improved; the parasitic oscillation caused by the leakage inductance is suppressed, and the EMI characteristics of the circuit can be improved; similarly, since the exciting inductance can work in the continuous current state, it can also work in the current discontinuous mode, making The control of the circuit is simpler, which can greatly improve the efficiency of the circuit at light load and reduce the loss at no load.

Description of drawings

Figure 1 is a flyback circuit diagram with an RCD clamp snubber circuit;

Figure 2 Active clamp flyback circuit diagram;

Fig. 3 The working waveform diagram of the active clamp flyback circuit in the single direction of the exciting inductor current;

Figure 4 is the working waveform diagram of the active clamp flyback circuit under the bidirectional excitation inductance current;

Figure 5 is a flyback circuit diagram with synchronous rectification on the secondary side (there is no RCD clamping and absorbing circuit on the primary side);

Figure 6 The working waveform diagram of the flyback circuit using synchronous rectification on the secondary side (there is no RCD clamping and absorbing circuit on the primary side);

The circuit structure diagram of Fig. 7 the utility model;

Fig. 8 is a gate pulse sequence diagram and a working waveform diagram under a specific embodiment of the utility model CCM mode;

Another specific embodiment of the utility model in Fig. 9 is a gate pulse sequence diagram and a working waveform diagram under the DCM mode;

Figure 10 Schematic diagram of the waveform of the flyback circuit in the VF DCM working mode (QR working mode);

Fig. 11 is the schematic diagram of the waveform of an embodiment of the utility model under the VF DCM mode of operation (QR mode of operation);

Fig. 12 is a schematic diagram of the waveform of an embodiment of the utility model realizing the opening of the peak point under the VF DCM working mode;

The circuit diagram of Fig. 13 peak point detection;

A schematic diagram of an embodiment in which the auxiliary branch is connected in parallel with the switch tube in the utility model in Fig. 14;

Fig. 15 is applied to a plurality of transformer series structure diagrams (output series connection) in the utility model;

Fig. 16 is a structural diagram of multiple transformers used in series in the utility model (output in parallel).

Detailed ways

Below in conjunction with accompanying drawing, the utility model is described in detail. Through the description of the specific embodiments of the utility model, it is easier to understand the features and details of the utility model. This article does not describe the known implementations and means of operation in detail, so as not to confuse the various technical implementations of the present utility model. New understanding and implementation.

The "embodiment" or "an embodiment" described in this specification refers to the specific features, structures, implementation methods and characteristics included in at least one embodiment of the present utility model described in conjunction with the embodiment. Therefore, when "in one embodiment" is mentioned in different places in the specification, they do not necessarily refer to the same embodiment. The features, structures or characteristics may be combined in any suitable manner in one or more embodiments.

A schematic circuit diagram (accompanying drawing 7) of a specific embodiment of the present invention and its unique control strategy (accompanying drawing 8), in which a primary side switch Q1 and a primary side auxiliary switch Qa are included in the figure. The schematic diagram of the circuit structure shown in Fig. 7 is equivalent to Fig. 2, and the auxiliary switch Qa is an N-type field effect transistor MOSFET. Different from the control method shown in Figure 3 or Figure 4, here the primary switch Q1 and the auxiliary switch Qa are not complementary, and Qa is only turned on for a short period of time before the primary switch Q1 is turned on, as shown in Figure 8 and the attached Figure 9 shows.

As shown in Fig. 8, before time t0, the primary switch Q1 is turned on, the primary current rises, the transformer stores energy, and the output rectifier diode is cut off. At time t0, the primary switch Q1 is turned off, at this time, the energy stored in the transformer leakage inductance Lk is stored in the snubber capacitor Cr through the body diode of the auxiliary switch Qa. Usually, the capacitance value of Cr is relatively large, which can be regarded as a constant voltage source. Since the energy of the leakage inductance is absorbed into the capacitor Cr, the parasitic oscillation caused by the leakage inductance is suppressed, which helps to improve the EMI characteristics of the circuit. At the same time, the energy stored in the magnetizing inductance begins to release to the output side through the rectifier diode.

At time t1, the energy in the leakage inductance is completely absorbed by the capacitor Cr, and the primary current Ip drops to zero. At this time, the energy stored in the magnetizing inductance Lm of the transformer is released to the output side, and the magnetizing inductance current decreases linearly. This stage lasts until time t2.

At time t2, the auxiliary switch Qa is turned on, the absorption capacitor Cr reversely charges the leakage inductance, the primary current Ip becomes negative, and the difference between the slope of the current reverse increase and the value of the absorption capacitor voltage V _Cr and the output voltage converted to the primary side In direct proportion, the energy in the snubber capacitor is released to the load through the transformer and stored in the leakage inductance. Similarly, since the difference between the absorption capacitor voltage V _Cr and the output voltage converted to the primary side is not very large, when the auxiliary switch Qa is turned on, its drain-source voltage is very small, so the turn-on loss of the auxiliary switch Qa is very small.

At time t3, the auxiliary switch Qa is turned off. At this time, the energy stored in the leakage inductance discharges the equivalent parasitic capacitance Cds between the drain and source of the primary switch Q1. The equivalent parasitic capacitance Cds between the drain and source mentioned here is Contains other parasitic capacitances in the line that play the same role. At time t4, the voltage Vds across Q1's drain-source drops to zero. At this time, the primary side switch realizes zero-voltage turn-on (soft switching). During the time interval of t3-t4, if the current of the magnetizing inductance becomes negative, the energy stored in the magnetizing inductance can also help realize the soft switching of the primary switch Q1. The time [t3-t4] is mainly used to realize the soft switching of Q1 and to prevent the cooperation between Q1 and Qa, which plays a role similar to the dead time.

After time t4, the primary switch is turned on, and the input voltage charges the magnetizing inductance to store energy. This stage is the same as the traditional flyback converter.

In the waveform shown in accompanying drawing 8, the magnetizing inductance current I _Lm is in the continuous mode (CCM mode), and in the time period t2～t4, the magnetizing inductance current I _Lm can become negative or positive all the time without affecting The operating properties of the circuit.

Accompanying drawing 9 shows the working waveform (DCM mode) of the excitation inductance current I _Lm in discontinuous mode.

Before the t1 moment, the work of the circuit is the same as before. During the t1-t2 period, the magnetizing inductor current drops to zero at t2a. During the time period from t2a to t2, the magnetizing inductance resonates with the equivalent parasitic capacitance Cds of the primary switch Q1, and the auxiliary switch Qa is turned on until the time t2.

At time t2, the auxiliary switch Q2 is turned on, and the subsequent working process is completely consistent with the continuous state of the exciting inductance. It is worth noting that at time t3, the magnetizing inductance current I _Lm is negative, which can help the primary side switch realize zero-voltage switching. Because the magnetizing inductance current can work in DCM mode, its light load efficiency can be greatly improved.

From the above analysis, it can be seen that the circuit structure and its control method adopted by the utility model have obvious advantages compared with the prior art. The soft switching of the switch can greatly improve the efficiency of the circuit; the parasitic oscillation caused by the leakage inductance is suppressed, and the EMI characteristics of the circuit can be improved; similarly, since the exciting inductance can work in the continuous current state, it can also work in the current discontinuous mode. It makes the control of the circuit simpler, can greatly improve the efficiency of the circuit at light load, and reduce the loss at no load.

The circuit shown in Fig. 7 and the auxiliary switch switch control mode proposed by the utility model can be combined with various existing flyback circuit control schemes. A fixed-frequency control method can be used, that is, the switching frequency of Q1 is fixed (set by the user). The frequency conversion control method can also be used, that is, the switching frequency of Q1 is not fixed, but changes with changes in load, input voltage, and parameters, including changing the on-time and off-time.

In the traditional frequency conversion control method of changing the conduction time, it is usually hoped that the circuit works in the critical discontinuous mode of the inductor current. In order to reduce the switching loss, after the excitation inductor current drops to zero, the oscillation of the excitation inductance and parasitic capacitance is used to realize the valley bottom Opening, also known as VF DCM working mode, or quasi-resonant flyback circuit (QR Flyback). In the traditional way, the circuit structure is shown in Figure 1, and the circuit waveform is shown in Figure 10. At time t0, the magnetizing inductor current drops to zero, and after an appropriate delay or detection method (usually half the resonant period of the parasitic oscillation), the switch Q1 is turned on at time t1, and the minimum voltage (that is, the bottom) is turned on. The turn-on time of Q1 is determined by the feedback link, which is common knowledge and will not be described here. The detection method of the valley voltage moment is also realized by the prior art, which will not be described here. Sometimes, in order to prevent the switching frequency from being too high under light load or high input voltage, the turn-on of Q1 can be at the bottom of the second resonance, as shown in Figure 10, or it can be more, usually by limiting the turn-off of Q1 time, that is, the time from t1a to t3 in Fig. 10 cannot be less than a certain set value. At present, it has been realized by many control chips, such as ONSEMI's NCP1207A, NXP's TEA1552 and other series chips.

The control mode that the utility model adopts can adopt VF DCM work mode (or quasi-resonance QR mode) equally. As a specific embodiment, as shown in FIG. 11, at time t0, the magnetizing inductance drops to zero, and at time t1, the auxiliary switch Qa is turned on for a predetermined time, and then switch Q1 is turned on again. The turn-on time of Qa is set by the circuit, and the turn-on time of Q1 is determined by the feedback loop. Those skilled in the art can easily realize the control technology disclosed by the utility model by the existing chip. The waveform is shown in Figure 11. In order to further improve the performance, by changing the appropriate delay or detection method, detect the peak value of the parasitic oscillation, and further reduce the switching loss. A simple way is to compare the parasitic oscillation with the zero potential through the auxiliary winding, and the generated waveform is properly delayed, and the falling edge moment is the detection of the oscillation peak, as shown in the schematic diagram in Figure 13. Those skilled in the art can obtain various peak point detection methods on the basis of the prior art, which does not affect the implementation of the present invention.

In the frequency conversion control method of changing the off time, the on time of Q1 is fixed or changes according to the preset law (determined according to the peak current passing through the switch Q1, that is, the switch current reaches the set value to turn off the switch), and the feedback link adjusts the original The off time of the side switch Q1 is used to adjust the output voltage or current, that is, it can work in CCM or DCM. In this way, the switching frequency decreases as the load becomes lighter, which can significantly improve light-load efficiency, and the circuit operates at its highest frequency at full load. Working in CCM mode is similar to CCM mode of constant frequency control. In the DCM mode, the working waveform is similar to that of Figure 12, the difference is that the on-time of Q1 is preset, and the off-time (time from t1a to t3) is determined by the feedback link, therefore, the on-time of Qa Might not be at the peak.

To sum up, the utility model can realize the soft switching of the primary switch and lossless absorbing leakage by adding an auxiliary switch through a simple circuit and controlling the auxiliary switch to be turned on for a set time before the primary switch Q1 is turned on. Inductive energy, and transfer it to the output and input, improve the efficiency of the circuit, prevent the parasitic oscillation caused by the leakage inductance, improve the EMI characteristics of the circuit, suitable for various control methods of the current flyback circuit, such as fixed frequency, variable frequency ( The scheme of adjusting the on-time or controlling the off-time) has obvious advantages compared with the prior art.

The auxiliary switch shown in accompanying drawing 7 is an N-type MOSFET, and the auxiliary branch formed by the auxiliary switch Qa and the absorption capacitor Cr is connected in parallel at both ends of the transformer, and those skilled in the art can easily obtain various equivalent circuits, as shown in the accompanying drawing In the equivalent circuit shown in 14, the auxiliary branch is connected in parallel at both ends of the switch tube, and a P-type MOSFET is used. The auxiliary switch can also be other types of switches. The conduction control requirements of the auxiliary switch Qa and the primary switch Q1 remain unchanged, which is consistent with that described in Fig. 8/4C and Fig. 6 .

In the above embodiment, a transformer is used as an example. Similarly, the transformer can be a plurality of transformers connected in series, and its primary currents are equal. The energy of each transformer leakage inductance can be absorbed by the capacitor Cr shown in Figure 7. Edges can be connected in series or in parallel. Accompanying drawing 15, 16 has provided the embodiment that 2 primary sides of transformer are connected in series. Those skilled in the art can also find an embodiment in which multiple transformers are connected in series.

The above detailed descriptions of the embodiments of the present invention are not intended to be exhaustive or to limit the present invention to the above-mentioned specific forms. While specific embodiments of, and examples for, the invention are described above for illustrative purposes, various equivalent modifications are possible within the scope of the invention, as those skilled in the relevant art will recognize.

The enlightenment provided by the utility model here is not necessarily applied to the above-mentioned system, but can also be applied to other systems. The elements and actions of the various embodiments described above can be combined to provide further embodiments.

Modifications can be made to the invention in light of the above detailed description, and while the above description describes particular embodiments of the invention and describes the best mode contemplated, no matter how detailed description appears above, the invention is capable of being practiced in many ways. utility model. The details of the above-mentioned circuit structure and its control method can be changed quite a lot in its implementation details, but it is still included in the utility model disclosed here.

As mentioned above, it should be noted that the special terms used when describing some features or solutions of the present utility model should not be used to indicate that the term is redefined here to limit some specific features of the utility model related to the term, features or schemes. In conclusion, the terms used in the following claims should not be construed to limit the invention to the particular embodiments disclosed in the specification, unless the above detailed description expressly defines those terms. Accordingly, the actual scope of the invention includes not only the disclosed embodiments, but also all equivalents of implementing or implementing the invention under the claims.

Claims (10)

1, a kind of Zero-voltage switch flyback-type DC-DC power conversion equipment comprises:

An input port is accepted DC input voitage, and an output port provides direct current to load;

A transformer comprises a former limit winding and a secondary winding at least;

Former limit switch is in series with the former limit of described transformer winding; During the switch conduction of described former limit, DC input voitage is added to the former limit of described transformer winding, described transformer stored energy; At described former limit switch blocking interval, described DC input voitage and the former limit of described transformer winding disconnect, and the energy that described transformer is stored during the switch conduction of described former limit discharges to load by the secondary winding of described transformer;

Output circuit, the secondary coupling with described voltage device produces a direct current at the energy that described former limit switch blocking interval discharges at described output port with described transformer, offers load;

It is characterized in that: also comprise a clamp circuit, comprise an electric capacity, a booster diode and an auxiliary switch, described auxiliary switch and capacitances in series, described booster diode is in parallel with auxiliary switch, the switch blocking interval provides a current path for described transformer on described former limit, described clamp circuit in parallel with the former limit winding of described transformer or with described former limit switch in parallel;

Described auxiliary switch is in the time of described former limit switch conduction setting of conducting before the moment, described auxiliary switch turn-offs the time that shifts to an earlier date the setting constantly of described former limit switch conduction constantly, at described former limit switch conduction constantly, the voltage that described former limit switch bears is close to zero.

2, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 1, it is characterized in that: described former limit switch or auxiliary switch are metal oxide semiconductor field-effect, igbt or bipolar transistor.

3, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 1, it is characterized in that: described booster diode is the parasitic diode of described auxiliary switch.

4, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 1 is characterized in that: described auxiliary switch conducting is leading constantly, and the described former limit switch conduction time constantly is fixing or sets.

5, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 1 is characterized in that: described auxiliary switch ON time is less than the described former limit switch turn-off time; Described auxiliary switch is booster diode not conducting of conduction period in described clamp circuit.

6, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 1, it is characterized in that: the exciting curent of described transformer is operated in on-off state or continuous state.

7, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 2, it is characterized in that: described output circuit, comprise a rectifying device, break anti-Pianguan County in the time, allow electric current to pass through in the turn-off time at described former limit switch at described former limit switch conduction.

8, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 2 is characterized in that: the energy that the electric capacity in the described clamp circuit was absorbed in described booster diode conduction period is discharged to load and direct-flow input end by described transformer leakage inductance by described auxiliary switch conduction device.

9, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 8, it is characterized in that: described rectifying device is a diode.

10, Zero-voltage switch flyback-type DC-DC power conversion equipment according to claim 1, it is characterized in that: the adjusting of output is by the ON time of control described former limit switch, or the duty ratio of controlling the turn-off time of described former limit switch or controlling the conducting of described former limit switch is regulated.

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