CN1792665A - AC Feed System for Electrified Railway - Google Patents

Abstract

An AC power feeding system of the present invention is intended to simplify electric train control without a switching area, and one of a plurality of substations includes: a 1 st three-phase power supply having a 1 st frequency; a transformer for converting the three-phase ac of the 1 st three-phase power supply into a two-phase ac and connecting only one of two-phase output terminals to the single-phase ac feeder; a 1 st power converter having ac output terminals connected to M blocks of two-phase output terminals of the transformer; a 2 nd power converter having an ac output terminal connected to a T-socket of the two-phase output terminal of the transformer; a dc smoothing capacitor connected to a common dc terminal of the 1 st and 2 nd power converters; a compensation current control unit for controlling the compensation current generated from the 1 st and 2 nd power converters; a 2 nd three phase power supply having a 2 nd frequency; and a diode rectifier for converting the three-phase AC of the 2 nd three-phase power supply into DC and supplying DC power to the DC smoothing capacitor.

Description

AC Feed System for Electrified Railway

technical field

The invention relates to an AC feed system for an electrified railway.

Background technique

Currently, as documents related to such electrified railway AC power feeding systems, techniques described in JP-A-2001-47894 (Patent Document 1) and JP-A-2001-71820 (Patent Document 2) are known.

In the previous AC feed system for electrified railways, the three-phase AC power supply was converted into two phases by using the Scott connection transformer, and it was used as the power supply for two single-phase AC feed lines, and each AC feed line passed through the pantograph The rack supplies power to the tram load. The two single-phase AC feeders are bounded by the substation and divided into M block and T block in different directions. The voltages of the M block and the T block are respective single-phase voltages among the 2-phase voltages output from the above-mentioned Scott connection transformer, and have a phase difference of 90°.

The load (tram) of the M block is different from the load (tram) of the T block. If viewed from the transformer, it becomes a two-phase unbalanced load. For example, the load (tram) of the M block performs power operation, and the load (tram) of the T block When regenerative operation is performed. From the perspective of three-phase power supply, it also becomes a three-phase unbalanced load, causing voltage distortion, and has a bad influence on other electrical equipment connected to the same power system.

Japan's power system is divided into the 50Hz system in East Japan and the 60Hz system in West Japan with Fujikawa as the boundary. For an AC feeder with a frequency of 60 Hz, a 50 Hz/60 Hz frequency converter such as an M/G device is required to receive power from a 50 Hz power system. Recently, a stationary type frequency converter (PWM converter + PWM inverter) is installed in parallel with a rotary type frequency converter (M/G device).

The aforementioned three-phase unbalanced load leads to an increase in the capacity of the aforementioned M/G device (or static frequency converter), resulting in a problem that it becomes an uneconomical system.

On the other hand, the voltage supplied to the train is switched from the power source of the M block to the power source of the T block in the vicinity of the substation, and a switching area is required for this switching. Every time the train passes through the switching area, there is no power supply (blackout) for a while, and the operation of the converter or inverter mounted on the train is stopped, and then the converter/inverter needs to be restarted after the power supply is restored. . During this period, the train runs in a snaking manner, and acceleration or braking force cannot be obtained, which not only lowers the acceleration and deceleration performance of the vehicle, but also forcibly drives with a bad ride.

Fig. 62 is a block diagram showing a configuration example of a conventional AC power feeding system for an electrified railway. In the figure, SUP1 represents the first three-phase AC power supply (60Hz), SUP0 represents the second three-phase AC power supply (50Hz), M/G represents the rotary frequency converter (50Hz/60Hz frequency converter), M-TR1, M -TR2 means three-phase transformer, SS1~SS3 means substation, CB1~CB11 means three-phase AC switch, CBm1~CBm6, CBt1~CBt6 means single-phase AC switch, S-TR1~S-TR3 means Scott connection transformer, Fa , Fb means pull through single-phase AC feeder, DS1~DS3 means the regional transition zone (switching area) between M block/T block, KS1~KS4 means the area switch between substations, and Train means train load.

The rotary frequency converter M/G generates the first three-phase AC power supply SUP1 (60Hz) from the second three-phase AC power supply SUP0 (50Hz), for example, a 10-pole synchronous motor M and a 12-pole synchronous generator G are mechanically Combined composition. When the motor M is driven by a 50 Hz power supply, the rotational speed becomes N=600 rpm, and the generator G generates a three-phase voltage with a frequency of 60 Hz. This M/G device is installed in a frequency conversion station. The three-phase -60Hz voltage source generated by the M/G device is transmitted to the substations SS1-SS3 of the electrified railway through the AC transmission line.

For example, in the substation SS1, the three-phase -60 Hz is distributed through the AC switches CB3 and CB4, and then transmitted to the distribution lines in the substation. Furthermore, the Scott connection transformer S-TR1 converts the three-phase AC voltage into a two-phase AC voltage to generate the two-phase AC of the M-seat and the T-seat. In the two-phase output voltage of the Scott connection transformer S-TR1, the M-base output is connected to, for example, the AC feeders Fa and Fb in the Tokyo direction via single-phase AC switches CBm1 and CBm2. In addition, the T-block output is connected to, for example, AC feeders Fa and Fb in the direction of Osaka via single-phase AC switches CBt1 and CBt2. An area transition section (switching area) DS1 is set between the AC feeders of the M block/T block in the substation SS1.

In addition, in the substation SS2, through the AC switches CB5 and CB6, the power distribution three-phase -60Hz is sent to the distribution line in the substation. Furthermore, the Scott connection transformer S-TR2 converts the three-phase AC voltage into a two-phase AC voltage to generate the two-phase AC of the M block and the T block. In the two-phase output voltage of the Scott connection transformer S-TR2, the output of the T block is connected to the AC feeders Fa and Fb in the direction of Tokyo through the three-phase AC switches CBt3 and CBt4. In addition, the M block output is connected to the AC feeders Fa and Fb in the direction of Osaka via the single-phase AC switches CBm3 and CBm4.

An area transition section (switching area) DS2 is set between the AC feeders of the M block/T block in the substation SS2. In addition, the T-seat AC feeder from the substation SS1 and the T-seat AC feeder from the substation SS2 are electrically combined or separated through the area switch KS2.

That is, when the power supply from the substation SS2 cannot be supplied due to some kind of accident, by closing the area switch KS2, the power supply from the substation SS1 to the T-seat voltage can be extended, and the power supply to the train can be ensured. In addition, under normal operation, by closing switch KS2, parallel operation of substation SS1 and substation SS2 can be performed, and the voltage of the AC feeder can be stabilized.

Furthermore, in the substation SS3, through the AC switches CB7 and CB8, the power distribution three-phase -60 Hz is sent to the distribution line in the substation. Furthermore, the Scott connection transformer S-TR3 transforms the three-phase AC voltage into a two-phase AC voltage to generate the two-phase AC of the M block and the T block. In the two-phase output voltage of the Scott connection transformer S-TR3, the output of the M block is connected to the AC feed Fa and Fb in Tokyo through the single-phase AC switches CBm5 and CBm6. In addition, the output of the T block is connected to AC feeders Fa and Fb in the direction of Osaka through single-phase AC switches CBt5 and CBt6.

A regional transition section (switching area) DS3 is set between the AC feeders of the M block/T block of the substation SS3. In addition, the M-seat AC feeder lines from the substation SS2 and the M-seat AC feeder lines from the substation SS3 are electrically connected or separated through the area switch KS3.

That is, when the power supply from the substation SS2 cannot be supplied due to some kind of accident, by closing the area switch KS3, the power supply M-seat voltage can be extended from the substation SS3, and the power supply to the train can be ensured.

In such a conventional AC feed system for electrified railways, the M-seat load and the T-seat load of the Scott connection transformer are basically inconsistent, and usually become an unbalanced load. Depending on the situation, the power running train load may be connected to the M block, and the regenerative train may be connected to the T block. Depending on the unbalanced load, a large load may be forced on equipment such as the Scott connection transformer or the M/G device. . In addition, the deliberately regenerated power cannot be interchanged between the M block and the T block, which becomes a difficult problem for the power system.

In the case of Fig. 62, the unbalanced power is ultimately borne by the M/G device. Although it does not affect the second AC power supply (power system) SUP0, the M/G device needs to be manufactured with a margin, which is uneconomical. system. For example, in the case of a three-phase balanced power, if the maximum 50MW is used, it is sufficient to prepare an M/G device of 50MW, but it is necessary to manufacture an M/G device with a voltage-resistant single-phase load of 50MW, which is equivalent to three Phase × 50MW = 150MW M/G device.

In addition, in the case of FIG. 62 , the unbalanced power is ultimately borne by the power system, which distorts the voltage of the power system and adversely affects other electrical equipment connected to the power system. From such a situation, in the invention of Patent Document 1, in order to minimize the three-phase unbalanced current, a transformer for power feeding like a Scott connection transformer is connected in parallel to convert three-phase AC power into DC power , an AC-DC conversion system that converts the DC power into a single-phase AC power. In this case, the AC-DC conversion system provides compensation current to balance the unbalanced current on the system side generated in the transformer for feeding.

On the other hand, when viewed from a train, the conventional AC power feeding system has the following problems. FIG. 63 shows the structure of DS1 during the zone transition period (switching zone) between M-blocks/T-blocks in substation SS1 in the system of FIG. 62 . In the figure, SUP1 represents the first AC power supply (three-phase -60Hz), S-TR represents the Scott connection transformer, CBm1, CBm2, CBt1, CBt2 represent single-phase AC switches, SWm1, SWm2, SWt1, SWt2 represent switches, Fm1 , Fm2 represent M-seat AC feeders, Ft1, Ft2 represent T-seat AC feeders, Fd1, Fd2 represent AC feeders in the switching area, and Train1, Train2 represent trains. Here, it is assumed that the right side is the Tokyo side and the left side is the Osaka side.

The Scott connection transformer S-TR transforms the three-phase AC voltage into a two-phase AC voltage, and applies the M-seat voltage VM to the AC feeders Fm1 and Fm2 in Tokyo through the switches CBm1 and CBm2. In addition, through the switches CBt1 and CBt2, the T-seat voltage VT is applied to the AC feeders Ft1 and Ft2 in Osaka.

For example, when the train Train moves from the M-seat AC feeder Fm1 to the T-seat AC feeder Ft1, the following operations are performed in the non-electric segment Fd1. When the train Train is at Fm1, the switch SWt1 is released in advance, and the switch SWm1 is closed. After the train Train enters Fd1, the switch SWm1 is released to enter a power failure state. Then, the train Train detects a power outage, temporarily stops the operation of the driving device (PWM converter + VVVF inverter) on the train, and makes the train run in a snaking state (only by the inertia of the train).

Then, after a certain period of time, the switch SWt1 is closed, and the voltage VT of the T seat is applied to the AC feeder Fd1, so that the power supply is resumed. The train Train detects that the power supply is restored, and starts the operation of the above-mentioned driving device according to the control of the phase coincidence with the T-block voltage VT. Since the feed voltage to the train changes from the M-seat voltage VM to the T-seat voltage VT, it takes considerable time to switch the phase synchronous control.

Considering the time from detection of such a power failure to restart and the train speed, the length of the switching area Fd1 needs to exceed a distance of km. Acceleration control or deceleration control (regenerative braking) of the train in this section cannot be expected. That is, the train from Tokyo to Osaka must repeatedly stop and restart the above-mentioned operation every time it passes through the substation, which has problems such as spoiling the riding mood and reducing the acceleration and deceleration performance of the train.

As a solution to this problem, it is also conceivable to install power converters for converting three-phase AC to single-phase AC in all substations, and to connect the single-phase AC generated here, for example, between Tokyo and Osaka. However, this would necessitate the removal of existing equipment (M/G units or Scott connection transformers, etc.) and all replacement with new equipment. In addition, even if only one substation is replaced to generate single-phase AC, and the two-phase voltage of M block/T block is output from the adjacent substation, the single-phase AC cannot be pulled through, and new equipment must be replaced in multiple substations at the same time. In general, it is very difficult to stop the operation of trains in the public interest, and even performing equipment replacement at night has time constraints, making it very difficult to implement.

Contents of the invention

The present invention has been made in view of the above-mentioned problems of the prior art, and an object of the present invention is to provide an economical electrified railway AC feeder system in which single-phase AC feeder lines are drawn across a plurality of substations by effectively utilizing existing equipment. , There is no switching area, an electrified railway AC feed system that can simplify the control of the tram, improve the acceleration and deceleration performance of the vehicle, and improve the riding mood.

The purpose of the present invention is also to provide an electrified railway AC feed system capable of achieving power exchange between powered trains and regenerative trains, compensating unbalanced power accompanying single-phase loads, and balancing three-phase currents supplied from AC power sources.

Another object of the present invention is to reduce the load on the installed rotary frequency converter (M/G device) when the frequency of the power system (the second three-phase AC power supply) is different from the frequency of the AC feeder, An electrified railway AC feed system with a structure that can be easily replaced with a static frequency converter in the future.

The present invention uses a transformer that transforms three-phase power into two-phase power to obtain the power supplies of two single-phase feeders, and at the same time passes through the current collectors from the above-mentioned single-phase feeders to supply power to electric train loads. , is characterized by having: a device capable of continuously exchanging or continuously supplying power from one single-phase AC to the other single-phase AC of the above-mentioned transformer.

In addition, the present invention is an electrified railway AC feeder system that connects single-phase AC feeders across multiple substations. It is characterized in that at least one of the above-mentioned multiple substations is equipped with: transforming the three-phase AC voltage into a two-phase AC voltage , in the two-phase output terminals, only one-phase output terminal is connected to the transformer on the above-mentioned single-phase AC feeder; an AC output terminal is connected to one phase of one of the two-phase output terminals of the above-mentioned transformer, and the AC power is converted A first power converter for converting direct current power; a second power converter for converting alternating current power into direct current power by connecting an alternating current output terminal to one of the other phases of the two-phase output terminals of the above-mentioned transformer; 2. The power converter compensates the unbalanced current generated by the load of the above-mentioned transformer, etc., and controls the compensation current control unit for the compensation current as the balance current.

In addition, the present invention is an AC feed system for electrified railways that connects single-phase AC feeders across multiple substations. It is characterized in that at least one of the above multiple substations is equipped with: transforming a three-phase AC voltage into a two-phase AC voltage, Among the two-phase output terminals, only one of the one-phase output terminals is connected to the transformer of the above-mentioned single-phase AC feeder; an AC output terminal is connected to one phase of one of the two-phase output terminals of the above-mentioned transformer, and the AC power is converted A first power converter for converting DC power; connecting an AC output terminal to the other one phase of the two-phase output terminals of the above-mentioned transformer, and converting AC power into a second power converter for DC power; controlling from the above-mentioned first and second power converters 2 Compensation current control unit output by the power converter to compensate unbalanced current generated by the load of the above-mentioned transformer, etc.; an energy storage device that transmits and receives power between the common DC terminals of the first and second power converters above.

In addition, the present invention is an electrified railway AC feeder system for pulling single-phase AC feeders across a plurality of substations, characterized in that at least one of the plurality of substations includes: a first three-phase AC power supply having a first frequency Transform the three-phase AC voltage of the first three-phase AC power supply into a two-phase AC voltage, and in the two-phase output terminals, only connect the one-phase output terminal to the transformer on the above-mentioned single-phase AC feeder; The first power converter with AC output terminals is connected to one phase (M seat) of the two-phase output terminals of the transformer; the other one phase (T seat) of the two-phase output terminals of the transformer is connected to the AC output terminal A second power converter; a DC smoothing capacitor connected to a common DC terminal of the first and second power converters; a compensation current control unit for controlling compensation currents generated from the first and second power converters; having a first A second three-phase AC power supply of 2 frequencies; a diode rectifier for converting the three-phase AC of the second three-phase AC power supply into DC and supplying DC power to the above-mentioned DC smoothing capacitor.

In addition, the present invention is an electrified railway AC feeder system for pulling single-phase AC feeders across a plurality of substations, characterized in that at least one of the plurality of substations includes: a first three-phase AC power supply having a first frequency Transform the three-phase AC voltage of the first three-phase AC power supply into a two-phase AC voltage, and in the two-phase output terminals, only connect the one-phase output terminal to the transformer on the above-mentioned single-phase AC feeder; The first power converter with AC output terminals is connected to one phase (M seat) of the two-phase output terminals of the transformer; the other one phase (T seat) of the two-phase output terminals of the transformer is connected to the AC output terminal A second power converter; a DC smoothing capacitor connected to a common DC terminal of the first and second power converters; a compensation current control unit for controlling compensation currents generated from the first and second power converters; and the above-mentioned An energy storage device that transmits and receives power between DC smoothing capacitors; a charge and discharge current control unit that controls the charge and discharge current to the above energy storage device; a second three-phase AC power supply with a second frequency; the second three-phase AC power supply A diode rectifier that converts three-phase AC to DC and supplies DC power to the above-mentioned DC smoothing capacitor.

Furthermore, the present invention is an electrified railway AC feeder system for pulling single-phase AC feeders through a plurality of substations, characterized in that at least one of the plurality of substations includes: a first three-phase AC power supply having a first frequency Transform the three-phase AC voltage of the first three-phase AC power supply into a two-phase AC voltage, and in the two-phase output terminals, only connect the one-phase output terminal to the transformer on the above-mentioned single-phase AC feeder; The first power converter with AC output terminals is connected to one phase (M seat) of the two-phase output terminals of the transformer; the other one phase (T seat) of the two-phase output terminals of the transformer is connected to the AC output terminal A second power converter; a DC smoothing capacitor connected to a common DC terminal of the first and second power converters; a compensation current control unit for controlling compensation currents generated from the first and second power converters; having a first a second three-phase AC power supply of 2 frequencies; and a third power converter for converting the three-phase AC of the second three-phase AC power supply into DC and supplying DC power to the DC smoothing capacitor.

Furthermore, the present invention is an electrified railway AC feeder system for pulling single-phase AC feeders through a plurality of substations, characterized in that at least one of the plurality of substations includes: a first three-phase AC power supply having a first frequency Transform the three-phase AC voltage of the first three-phase AC power supply into a two-phase AC voltage, and in the two-phase output terminals, only connect the one-phase output terminal to the transformer on the above-mentioned single-phase AC feeder; The first power converter with AC output terminals is connected to one phase (M seat) of the two-phase output terminals of the transformer; the other one phase (T seat) of the two-phase output terminals of the transformer is connected to the AC output terminal A second power converter; a DC smoothing capacitor connected to a common DC terminal of the first and second power converters; a compensation current control unit for controlling compensation currents generated from the first and second power converters; having a first 2nd three-phase AC power supply of 2 frequencies; a diode rectifier for converting the three-phase AC of the second three-phase AC power supply into DC and supplying DC power to the above-mentioned DC smoothing capacitor; connected to the AC terminal of the diode rectifier via a reactor A third power converter with an AC terminal; make the third power converter operate according to a certain pulse pattern, by adjusting the phase angle of the AC side terminal voltage of the third power converter with respect to the voltage of the second three-phase AC power supply A control unit of a third power converter that controls the input current or active power supplied from the second three-phase AC power supply, and controls the voltage applied to the DC smoothing capacitor.

According to the present invention, while seeking to effectively use the installed AC feed system equipment, an electrified railway AC feed system that pulls through single-phase AC can be achieved without switching between M-seats/T-seats in the past Area, continuously supply power to the tram load and simplify the tram control, improve the acceleration and deceleration performance of the vehicle, and improve the riding mood. In addition, it is possible to achieve power exchange between power running trains and regenerative trains, compensate unbalanced power or reactive power and high-order harmonics accompanying single-phase loads, and balance three-phase currents supplied from AC power sources. Furthermore, the load of the already installed M/G device (rotary frequency converter) can be reduced, and an economical static frequency converter can be provided as a replacement unit when the device is aged.

In addition, according to the present invention, while effectively utilizing most of the existing equipment, single-phase AC can be pulled through multiple substations, and the control on the vehicle side can be simplified (especially It is the switching of the phase synchronization signal or the control stop and restart of the on-board converter/inverter, etc.), not only to improve the acceleration and deceleration performance of the train, but also to improve the riding mood.

Description of drawings

FIG. 1 is a block diagram showing the overall structure of an AC power feeding system for an electrified railway according to the present invention.

Fig. 2 is a block diagram of an AC power feeding system for an electrified railway according to the first embodiment of the present invention.

FIG. 3 is a block diagram of a specific main circuit configuration example of the two-phase power conversion device PPC in FIG. 2 .

FIG. 4 is a block diagram of a specific main circuit configuration example of the voltage-source converter CNVm1 in FIG. 3 .

FIG. 5 is a waveform diagram of an AC output voltage for explaining the operation of the voltage-source converter CNVm1 of FIG. 4 .

6 is a block diagram of a compensation current control unit in the first embodiment of the present invention.

Fig. 7 is a voltage-current vector diagram of the M-seat and the T-seat during power running.

Fig. 8 is a voltage-current vector diagram of the M-seat and the T-seat during regenerative operation.

FIG. 9 is a waveform diagram showing an example of an operation waveform of an analysis result of the apparatus of FIG. 2 .

Fig. 10 is a block diagram of an AC power feeding system for an electrified railway according to a second embodiment of the present invention.

FIG. 11 is an operation waveform diagram of each part for explaining the operation of the apparatus of FIG. 10 .

Fig. 12 is a block diagram showing a third embodiment of the compensation current control unit of the device of the present invention.

Fig. 13 is a block diagram showing a fourth embodiment of the compensation current control unit of the device of the present invention.

Fig. 14 is a voltage-current vector diagram when controlled by the control unit in Fig. 13 .

15 is a block diagram showing a fifth embodiment of the compensation current control unit of the present invention.

Fig. 16 is a voltage-current vector diagram when controlled by the control unit in Fig. 15 .

FIG. 17 is a block diagram showing a sixth embodiment of the compensation current control unit of the present invention.

Fig. 18 is a voltage-current vector diagram when controlled by the control unit in Fig. 16 .

FIG. 19 is a block diagram showing a seventh embodiment of the compensation current control unit of the present invention.

Fig. 20 is a voltage-current vector diagram when controlled by the control unit of Fig. 19 .

Fig. 21 is a block diagram of an AC power feeding system for an electrified railway according to an eighth embodiment of the present invention.

Fig. 22 is a block diagram showing another example of an energy storage device in the device of Fig. 21 .

23 is a block diagram of a compensation current control unit in an eighth embodiment of the present invention.

24 is a block diagram of a main circuit configuration and a control circuit of an energy storage device in an eighth embodiment of the present invention.

FIG. 25 is a characteristic example of the electric power command generator Fe(x) in the control circuit of FIG. 23 .

Fig. 26 is the M-seat and T-seat voltage-current vector diagrams when controlled by the control circuit of Fig. 23 .

Fig. 27 is a block diagram showing an AC power feeding system for an electrified railway according to a ninth embodiment of the present invention.

Fig. 28 is a block diagram of an AC power feeding system for electrified railways according to a tenth embodiment of the present invention.

29 is a characteristic diagram of the voltage command generator Fd(x) of the compensation current control unit in the tenth embodiment of the present invention.

Fig. 30 is another characteristic diagram of the voltage command generator Fd(x) of the compensation current control unit.

Fig. 31 is a voltage-current vector diagram of the M block and the T block during power running load of the electrified railway AC power feeding system according to the tenth embodiment of the present invention.

Fig. 32 is a block diagram of a compensation current control unit in a tenth embodiment of the present invention.

Fig. 33 is a diagram showing the operating characteristics of the compensation current control unit.

Fig. 34 is a block diagram of an AC power feeding system for an electrified railway according to an eleventh embodiment of the present invention.

Fig. 35 is a characteristic graph of the voltage command generator Fd(x) of the compensation current control unit in the eleventh embodiment of the present invention.

Fig. 36 is a characteristic diagram of the electric power command generator Fe(x) of the energy storage device in the eleventh embodiment of the present invention.

Fig. 37 is a voltage-current vector diagram of the M block and the T block during the power running of the electrified railway AC power feeding system according to the eleventh embodiment of the present invention.

38 is a block diagram of another example of compensation current control means in the eleventh embodiment of the present invention.

Fig. 39 is an operational characteristic diagram of the AC power feeding system for electrified railways according to the eleventh embodiment of the present invention.

Fig. 40 is a block diagram of an AC power feeding system for an electrified railway according to a twelfth embodiment of the present invention.

41 is a block diagram of a main circuit configuration and a control circuit of a third voltage source self-excited power converter in a twelfth embodiment of the present invention.

Fig. 42 is an equivalent circuit diagram for explaining the operation of the voltage source self-excited power converter.

Fig. 43 is a voltage-current vector diagram in the equivalent circuit of Fig. 42 .

44 is an operation waveform diagram of the third voltage-source self-excited power converter during power running according to the twelfth embodiment of the present invention.

Fig. 45 is an operation waveform diagram of the third voltage source self-excited power converter during regenerative operation according to the twelfth embodiment of the present invention.

46 is a block diagram of another example of the control circuit of the third voltage source self-excited power converter in the twelfth embodiment.

Fig. 47 is a block diagram of a compensation current control unit in a twelfth embodiment of the present invention.

Fig. 48 is a characteristic diagram of the power command generator Fs(x) of the compensation current control unit described above.

Fig. 49 is a voltage-current vector diagram of the M block and the T block during power running of the electric gas railway AC feeding system according to the twelfth embodiment of the present invention.

Fig. 50 is a block diagram of an AC power feeding system for an electrified railway according to a thirteenth embodiment of the present invention.

Fig. 51 is a block diagram of a compensation current control unit in a thirteenth embodiment of the present invention.

52 is a block diagram of a control circuit of a third voltage source self-excited power converter in a thirteenth embodiment of the present invention.

Fig. 53 is a block diagram of an AC power feeding system for an electrified railway according to a fourteenth embodiment of the present invention.

Fig. 54 is a block diagram of a main circuit configuration and a control circuit of a hybrid converter in a fourteenth embodiment of the present invention.

Fig. 55 is an explanatory diagram of the operation of the one-phase portion of the hybrid converter.

Fig. 56 is an operation waveform diagram of the hybrid inverter during power running according to the fourteenth embodiment of the present invention.

57 is a block diagram of another example of the control circuit of the hybrid converter in the fourteenth embodiment of the present invention.

Fig. 58 is a block diagram of a compensation current control unit in a fourteenth embodiment of the present invention.

Fig. 59 is a characteristic diagram of the power command generator Fs(x) of the compensation current control unit described above.

Fig. 60 is another characteristic diagram of the electric power command generator Fs(x) of the compensation current control unit.

Fig. 61 is a voltage-current vector diagram of the M block and the T block during power running of the electrified railway AC feed system according to the fourteenth embodiment of the present invention.

Fig. 62 is a block diagram of a conventional AC power feeding system for electrified railways.

Fig. 63 is a block diagram for explaining the operation in the area transition section between the M block and the T block in the conventional AC power feeding system.

Detailed ways

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

FIG. 1 is a block diagram showing the overall configuration of an electric railway AC power feeding system common to all embodiments of the present invention. In the figure, SUP1 represents the first three-phase AC power supply (60Hz), SUP0 and SUP2 represent the second three-phase AC power supply (50Hz), M/G represents the rotary frequency converter (50Hz/60Hz frequency converter), M-TR1 、M-TR2 means three-phase transformer, SS1～SS3 means substation, CB1～CB11 means three-phase AC switch, CBm1～CBm6 means single-phase AC switch, S-TR1～S-TR3 means Scott connection transformer, PPC1～PPC3 Indicates a two-phase power interchange device with the function of a frequency converter, Fa and Fb indicate single-phase pull-through AC feeders, Ks1-Ks4 indicate area switches, and Train indicates train load.

The purpose of the AC feed system of the present invention is to cancel the conventional zone transition zone (switching zone) by effectively utilizing the installed equipment (M/G device or Scott connection transformer, etc.) .

The rotary frequency converter M/G generates the first three-phase AC power supply SUP1 (60Hz) from the second three-phase AC power supply SUP0 (50Hz), for example, a 10-pole synchronous motor M and a 12-pole synchronous generator G are mechanically combined to form. When the motor M is driven by a 50 Hz power supply, the rotational speed becomes N=600 rpm, and the generator G generates a three-phase voltage with a frequency of 60 Hz. Here, two M/G devices are operated in parallel to secure the required capacity. This M/G device is installed in a frequency conversion station. The three-phase -60Hz voltage source generated by the M/G device is sent to the substations SS1-SS3 of the electrified railway through the AC transmission line.

For example, in the substation SS1, the three-phase -60 Hz is distributed through the AC switches CB3 and CB4, and then transmitted to the distribution lines in the substation. Furthermore, the Scott connection transformer S-TR1 converts the three-phase AC voltage into a two-phase AC voltage to generate the two-phase AC of the M block and the T block. Of the two-phase output voltages of the Scott connection transformer S-TR1, only one phase (M-base) is connected to the single-phase feeders Fa, Fb. CBm1 and CBm2 are single-phase AC switches, Fa is the AC feeder for the up train, and Fb is the AC feeder for the down train. At this time, the T-seat winding of the above-mentioned transformer S-TR1 becomes unloaded.

The two-phase power conversion device PPC1 consists of a three-phase transformer TRa1, a rectifier REC1, a first voltage-type self-excited power converter CNV11, a second voltage-type self-excited power converter CNV12, a DC smoothing capacitor Cd1, and a single-phase transformer TRm1, TRt1 is configured to perform 50Hz/60Hz frequency conversion while achieving power interchange between the M- and T-mounts of the above-mentioned Scott connection transformers S-TR1 and S-TR2. That is, the rectifier REC1 converts the three-phase AC (50 Hz) into DC, supplies the DC power to the DC smoothing capacitor Cd1, and then converts it into a single-phase AC through the first voltage-type self-excited power converter CNV11, and supplies power to the AC Feed lines Fa, Fb. In this way, it is possible to serve as a backup against insufficient capacity or failure of the M/G device.

The other substations SS2 and SS3 are configured in the same way, and only the M block is connected to the AC feeder line, and the T block is no-loaded. At this time, the two-phase power switching devices PPC2 and PPC3 perform power switching between the M and T blocks of the Scott connection transformers S-TR2 and S-TR3.

The regional switches KS1-KS4 separate the single-phase AC feeders in the substations SS1-SS3, and by closing the switches Ks1-Ks4, the parallel operation between the substations or the extended power feeding can be performed.

As mentioned above, in the electrified railway AC feeding system of the present invention, while effectively using most of the existing equipment, a single-phase pull-through AC feeding system can be realized, and the following effects can be obtained. First, when viewed from the train using the AC feeder, there is no need for switching between M-blocks and T-blocks as in the past, and there is no power failure between switching areas. As a result, the control on the vehicle side (especially switching of the phase synchronization signal or the control of stop and restart of the on-board converter/inverter, etc.) is simplified, not only the acceleration/deceleration performance of the train is improved, but also the ride comfort is improved. On the other hand, from the perspective of the ground side, most of the existing equipment can be effectively used, and in the case of aging of the M/G device or the Scott connection transformer, it can be replaced sequentially or made static. From the operational point of view An economical system can also be provided. Furthermore, since two-phase balance is sought, reactive power or higher harmonic current can be compensated, so it becomes an ideal load for Scott connection transformer or M/G device, and can provide a good electrified railway feeding system in the power supply system. In addition, in the interval where the frequency (60Hz) of the single-phase AC feeder line is different from the frequency (50Hz) of the power supply system, a rectifier (follower) that converts the 50Hz three-phase AC to DC and a two-phase power conversion device are combined Therefore, it is possible to reduce the capacity of the installed M/G device, and it is possible to replace the static frequency converter with respect to the aging of the M/G device.

first embodiment

Fig. 2 is a block diagram showing a first embodiment of an AC power feeding system for an electrified railway according to the present invention. In the figure, SUP1 represents a three-phase AC power supply, TR represents a Scott connection transformer, Fa represents a single-phase AC feeder, Load represents a tram load, TRm and TRt represent a single-phase transformer, and CNV1 and CNV2 represent voltage-type self-excited power conversion Cd represents a DC smoothing capacitor, and CONT1 represents a control unit that balances the above-mentioned two-phase unbalanced compensation currents (compensation current detection values) IMc, ITc outputted through power converters CNV1, CNV2 and single-phase transformers TRm, TRt. The compensation currents IMc and ITc can be detected by arranging current detectors, for example, at positions indicated by arrows in FIG. 2 , like a Hall CT not shown.

The Scott connection transformer TR transforms the three-phase AC power sources Vu, Vv, Vw into two-phase AC voltages VM, VT, and the two-phase voltages VM and VT have a phase difference of 90°. In addition to the Scott transformer TR, there are modified Woodbridge connection transformers, etc.

The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-seat terminal through the single-phase transformer TRm, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 is connected to the terminal through the single-phase transformer TRt T block terminal. The power converters CNV1 and CNV2 and the smoothing capacitor Cd constitute a two-phase power conversion device PPC.

The compensation current control unit CONT1 is composed of a DC voltage control circuit Vd-Cont, a compensation current command circuit Ic-Ref, a current control circuit IMc-Cont, ITc-Cont, and a pulse width modulation control circuit PWM1, PWM2.

The DC voltage control circuit Vd-Cont detects the voltage Vd applied to the DC smoothing capacitor Cd, compares the command value Vd ^* with the voltage Vd, and amplifies the difference to obtain a peak value command Ism ^* of the input current.

The compensation current value command circuit Ic-Ref generates input current command values IMs ^* and ^Its * by multiplying the peak value command Ism ^* by unit sine waves sinωt and cosωt synchronized with the two-phase voltages VM and VT.

IMs ^* ＝Ism ^* × sinωt

ITs ^* ＝Ism ^* ×cosωt

Next, the two-phase load currents IML, ITL are detected, and the input current command values IMs ^* , ITs ^* are respectively subtracted from the load current detection values IML, ITL to generate command values of compensation currents generated from the power converters CNV1 and CNV2. IMc ^* , ITc ^* .

IMc ^* = IML-IMs ^*

ITc ^* =ITL-ITs ^*

But in this case, since the load current ITL of the T-seat is 0, ITc ^* =-ITs ^* .

The current control circuit IMc-Cont compares the compensation current IMc with the command value IMc ^* , amplifies their difference, generates a voltage command value e1 ^* , and inputs it to the pulse width modulation control circuit PWM1 of the power converter CNV1. Power converter CNV1 outputs voltage VMc proportional to voltage command value e1 ^* , and is controlled so that compensation current IMc matches its command value IMc ^* . As a result, the input current IMs of the M block is controlled so that IMs=IML-IMc=IML-IMc ^* =IMs ^* .

Also control the current control circuit ITc-Cont of the T seat, compare the compensation current detection value ITc with the above-mentioned command value ITc ^* , amplify their difference, generate a voltage command value e2 ^* , and input it to the pulse of the power converter CNV2 Wide modulation control circuit PWM2. The power converter CNV2 outputs a voltage VTc proportional to the above-mentioned voltage command value e2 ^* , and is controlled so that the compensation current ITc matches the command value ITc ^* . ITc=ITc ^* , and the input current ITs of the T block also becomes ITs=ITs ^* .

That is, the peak values of the currents IMs and ITs of the M and T blocks of the Scott connection transformer TR are the same ISm ^* , and are controlled to be sinusoidal currents in phase with the respective voltages VM and VT. As a result, only the sinusoidal current of power factor=1 balanced in three phases is supplied from the AC power supply SUP1, and a good system can be constructed in the power supply system.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the above-mentioned peak value command Ism ^* is increased to increase the supply power Ps from the AC power supply SUP1 to become a part larger than the load power PL, Ps-PL It is stored as energy in the above-mentioned DC smoothing capacitor Cd. As a result, Vd increases and is controlled so that Vd=Vd ^* . Conversely, in the case of Vd>Vd ^* , the peak value command Ism ^* is reduced to become Ps<PL, and the stored energy of the DC smoothing capacitor Cd is reduced, which is also controlled so that Vd=Vd ^* .

FIG. 3 is a block diagram showing a specific main circuit configuration example of the two-phase power conversion device PPC. In the figure, CNVm1~CNVmn are voltage-type converters with n single-phase full-bridge connections constituting the first voltage-type self-excited power converter CNV1, Trm1~Trmn are n single-phase transformers, the primary windings are connected in series, connected to Scott connects the M-block winding terminals of the transformer TR. The secondary windings of the single-phase transformers Trm1 to Trmn are connected to the AC terminals of the respective voltage source converters CNVm1 to CNVmn.

In addition, CNVt1~CNVtn are n sets of single-phase full-bridge connection voltage-type converters constituting the second voltage-type self-excited power converter CNV2, Trt1~Trtn are n sets of single-phase transformers, the primary windings are connected in series, and connected to the Scott T-seat winding terminal of special connection transformer TR. The secondary windings of the single-phase transformers Trt1 to Trtn are connected to the AC terminals of the respective voltage source converters CNVt1 to CNVtn.

Cd1 to Cdn are DC smoothing capacitors, and are divided and connected to DC-side terminals of single-phase full-bridge-connected voltage source converters CNVm1 to CNVmn and CNVt1 to CNVtn. In addition, the DC-side terminals are connected in parallel.

By connecting n sets of single-phase output converters in series and multiplexing wiring, the capacity of the power converters CNV1 and CNV2 can be increased, and the current control response of the compensation current IMc and ITc can be improved with a lower on-off frequency, which can reduce the frequency associated with PWM. Controlled higher harmonic currents.

FIG. 4 is a block diagram showing a specific main circuit configuration example of the single-phase full-bridge connection voltage source converter CNVm1. In the figure, Sa, Sb, Sc, and Sd represent self-arcing components, Da, Db, Dc, and Dd represent high-speed diodes, Trm1 represents a single-phase transformer, and Cd1 represents a DC smoothing capacitor.

FIG. 5 shows the waveform (one pulse) of the AC output voltage Vc1 of the single-phase full-bridge-connected voltage-source converter CNVm1, as follows.

When Sa and Sd are connected (Sb and Sc are disconnected), Vc1=+Vd

When Sb and Sc are connected (Sa and Sd are disconnected), Vc1=-Vd

When Sa and Sc are connected (Sb and Sd are disconnected), Vc1=0

When Sb and Sd are turned on (Sa and Sc are turned off), Vc1=0

That is, voltages of three stages (+Vd, 0, -Vd) can be generated, and further, by increasing the number of pulses, it is possible to achieve finer control and reduce harmonics.

In addition, as shown in Figure 3, the capacity of the converter can be increased and high-order harmonics can be reduced by connecting multiplexed cables in series.

FIG. 6 is a block diagram showing a specific configuration example of the compensation current control unit CONT1 of the system of the present invention. In the figure, Fd(x) denotes the voltage command generator, C1~C3 denotes the comparator, Gv(S) denotes the voltage control compensation circuit, M1, M2 denotes the multiplier, AD1~AD4 denotes the adder/subtractor, Gi1(S) , Gi2(S) represent the current control compensation circuit, PWM1, PWM2 represent the pulse width modulation control circuit. In this embodiment, the voltage command generator Fd(x) outputs a constant DC voltage command Vd ^* .

The comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to proportional or integrate the deviation εv=Vd ^* -Vd Amplifies and generates the command Ism ^* of the peak value of the input current.

The multiplier M1 obtains the unit sine wave sinωt synchronized with the M-seat voltage VM of the Scott connection transformer TR, multiplies the above-mentioned input current wave peak command Ism ^* , and outputs the input current command IMs ^* =Ism ^* ×sinωt.

The multiplier M2 obtains the unit sine wave cosωt synchronized with the T-seat voltage VT of the Scott connection transformer TR, multiplies the above-mentioned input current wave peak command Ism ^* , and outputs the input current command ITs ^* =Ism ^* ×cosωt.

The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* .

Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to obtain the T-seat compensation current command value ITc ^* =ITL-ITs ^* .

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows. Of course, when the leakage inductance of the above-mentioned single-phase transformer TRm is small, a reactor Lsmo may be inserted in series in the primary or secondary winding of the transformer TRm.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, the T-seat input current ITs supplied from the Scott connection transformer TR is controlled so that ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=ITs ^* . This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. However, the T-seat load current becomes ITL=0.

The currents IMs and ITs of the M-seat and T-seat of the above-mentioned Scott connection transformer TR have the same amplitude value Ism ^* , and are two-phase balanced currents whose phases are shifted by 90°. As a result, the current supplied from the three-phase AC power supply SUP1 also becomes a sinusoidal current with power factor=1 balanced by the three phases.

As a result, not only the capacity of the above-mentioned Scott connection transformer TR1 can be reduced, but also the capacity of the equipment of the first AC power supply SUP1 or the N/G device can be reduced.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the above-mentioned peak value command ISm ^* increases, and the supply power Ps1 from the AC power supply SUP1 increases to become larger than the load power PL(av), and Ps1-PL( The part of av) is stored as energy in the above-mentioned DC smoothing capacitor Cd. As a result, Vd increases and is controlled so that Vd=Vd ^* . Conversely, when Vd>Vd ^* is satisfied, the peak value command Ism ^* is reduced to become Ps1<PL(av), and the stored energy of the DC smoothing capacitor Cd is reduced, so that Vd=Vd ^* is also controlled.

FIG. 7 shows the voltage-current vector diagrams of the M-seat and T-seat when the electric car connected to the AC feeder Fa is running. The T-seat load current ITL=0, and the M-seat load current IML has a phase θ slightly lagging behind the voltage VM. The load power PL=VM×IML×cosθ, if the loss is assumed to be very small, is equal to the input power Ps=VM×IMs+VT×ITs. The effective power PMs of the M-seat is equal to the effective power PTs of the T-seat, half of the load power PL is supplied from the M-seat winding of the Scott connection transformer, and the remaining half is supplied from the T-seat winding.

The power PTs=PL/2 supplied from the T-block winding is regenerated by the second voltage source self-excited power converter CNV2 and supplied to the DC smoothing capacitor Cd. That is, ITc=-ITs.

Furthermore, its power PL/2 is supplied to the single-phase AC power feeder Fa via the first voltage-source self-excited power converter CNV1. At this time, including the reactive power QL=VM×ILN×sinθ of the load, it is also supplied from the above-mentioned first voltage type self-excited power converter CNV1, and only the active power PMs is supplied from the M-seat winding of the Scott connection transformer TR. =PL/2.

FIG. 8 shows the voltage-current vector diagrams of the M-seat and the T-seat during the regenerative operation of the electric car connected to the AC feeder line Fa. Half of the regenerative power PL flows through the M-seat winding of the Scott connection transformer TR, and the remaining half passes through the first voltage type self-excited power converter CNV1→DC smoothing capacitor Cd→the second voltage type self-excited power converter Device CNV2 flows through the T-seat winding. At this time, the reactive power QL of the load is compensated by the first voltage source self-excited power converter CNV1.

Figure 9 shows the simulation results when the M seat adopts power to run the load. The waveform of (a) represents the voltage VM of the M seat and the voltage VT of the T seat, and the waveform of (b) represents the load current IML of the M seat and the load of the T seat. For the current ITL, the waveform of (c) represents the compensation current IMc of the M-seat and the compensation current ITc of the T-seat, and the waveform of (d) represents the input current IMs of the M-seat and the input current ITs of the T-seat. As shown by the waveform of (b), the load current ITL of the T-seat becomes ITL≈0, the M-seat compensation current IMc becomes the power running, and the T-seat compensation current ITc becomes the regenerative current. As a result, the winding current IMs of the M seat of the Scott connection transformer TR is equal to the winding current ITs of the T seat, and the phases of the voltage waveforms are staggered by 90°. That is, the sine wave current is controlled to balance the winding current IMs of the M base and the winding current ITs of the T base.

2nd embodiment

Fig. 10 is a block diagram showing an AC power feeding system for an electrified railway according to a second embodiment of the present invention.

In the figure, SUP1 represents a three-phase AC power supply, TR represents a Scott connection transformer, Fa represents a single-phase AC feeder, Load represents a tram load, TRm and TRt represent a single-phase transformer, and CNV1 and CNV2 represent voltage-type self-excited power conversion Cd represents a DC smoothing capacitor, Lf and Cf represent reactors and capacitors constituting an LC filter, and CONT1 represents a control unit for compensating currents IMc and ITc output from the above-mentioned power converters CNV1 and CNV2.

The Scott connection transformer TR transforms the three-phase AC power sources Vu, Vv, Vw into two-phase AC voltages VM, VT, and the two-phase voltages VM and VT have a phase difference of 90°.

The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-block terminal, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 is connected to the T-block terminal. Determine the value of the reactor Lf and the value of the capacitor Cf so that the LC filter (Lf, Cf) resonates at twice the frequency of the AC feeder (60 Hz).

The compensation current control unit CONT1 is composed of a DC voltage control circuit Vd-Cont, a compensation current command circuit Ic-Ref, a current control circuit IMc-Cont, ITc-Cont, and a pulse width modulation control circuit PWM1, PWM2.

The DC voltage control circuit Vd-Cont detects the voltage Vd applied to the above-mentioned DC smoothing capacitor Cd, compares it with the command value Vd ^* , and amplifies the difference to obtain the peak value command Ism ^* of the input current.

The compensation current value command circuit Ic-Ref generates input current command values IMs ^* and ^Its * by multiplying the peak value command Ism ^* by unit sine waves sinωt and cosωt synchronized with the two-phase voltages VM and VT.

IMs*=Ism*×sinωt

ITs*＝Ism*×cosωt

Next, the two-phase load currents IML, ITL are detected, and the input current command values IMs ^* , ITs ^* are respectively subtracted from the load current detection values IML, ITL to generate command values of compensation currents generated from the power converters CNV1 and CNV2. IMc ^* , ITc ^* .

IMc*=IML-IMs*

ITc*=ITL-ITs*

But in this case, since the load current ITL of the T-seat is 0, ITc ^* =-ITs ^* .

The current control circuit IMc-Cont compares the compensation current detection value IMc with the command value IMc ^* , amplifies their difference, generates a voltage command value e1 ^* , and inputs it to the pulse width modulation control circuit PWM1 of the power converter CNV1. Power converter CNV1 outputs voltage VMc proportional to voltage command value e1 ^* , and controls compensation current IMc so that compensation current IMc matches its command value IMc ^* . As a result, the input current IMs of the M block is controlled so that IMs=IML-IMc=IML-IMc ^* =IMs ^* .

Similarly, the current control circuit ITc-Cont of the T-seat is controlled so that ITc=ITc ^* , and the input current ITs of the T-seat also becomes ITs=ITs ^* .

That is, the peak values of the currents IMs and ITs of the M and T blocks of the Scott connection transformer TR are the same ISm ^* , and are controlled to be sinusoidal currents in phase with the respective voltages VM and VT. As a result, only the sinusoidal current of power factor=1 balanced in three phases is supplied from the AC power supply SUP1, and a good system can be constructed in the power supply system.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the above-mentioned peak value command Ism ^* is increased to increase the supply power Ps from the AC power supply SUP1 to become a part larger than the load power PL, Ps-PL It is stored as energy in the above-mentioned DC smoothing capacitor Cd. As a result, Vd increases and is controlled so that Vd=Vd ^* . Conversely, in the case of Vd>Vd ^* , the peak value command Ism ^* is reduced to become Ps<PL, and the stored energy of the DC smoothing capacitor Cd is reduced, which is also controlled so that Vd=Vd ^* .

11 shows a waveform example of voltage VM, current IML, power PL of Shinkansen load Load connected to single-phase AC feeder Fa, and voltage Vd applied to DC smoothing capacitor Cd.

The load current IML lags behind the M-seat voltage VM by the phase angle θ. The single-phase load power PL fluctuates at twice the frequency (120 Hz) of the AC feeder frequency f1 = 60 Hz.

That is, when the M seat voltage VM=Vsm×sinωt, and the load current IML=ILm×sin(ωt-θ), the power PL becomes:

PL＝Vsm×sinωt×ILm×sin(ωt-θ)

=(Vsm×ILm/2){cosθ-cos(2·ωt-θ)}

The first term is a constant value corresponding to the power Ps supplied from the aforementioned AC power supply SUP1 via the Scott connection transformer TR. The second term is the power fluctuation part ΔPL, which fluctuates at a frequency twice the power supply frequency.

When the DC smoothing capacitor Cd is used to absorb the power fluctuation ΔPL accompanying the single-phase load, the fluctuation ΔVd of the voltage Vd applied to the DC smoothing capacitor Cd is proportional to the load power PL, and is proportional to the DC smoothing capacitor Cd. Capacity is inversely proportional.

That is, the current Icap flowing through the DC smoothing capacitor Cd becomes: when the average value of the DC voltage is expressed as Vdo

IcapΔPL/Vdo

＝-{Vsm×ILm/(2·Vdo)}·cos(2·ωt-θ)

Therefore, the variation ΔVd of the DC voltage Vd becomes:

ΔVd=(1/Cd)∫Icap·dt

-{Vsm×ILm/(4·Vdo·ω·Cd)}×sin(2·ωt-θ)

For example, when the power supply frequency f1=60Hz, load power PL=20MW (power factor=0.95), DC voltage Vdo=8kV, and Cd=10mF, the peak value of the current Icap flowing through the DC smoothing capacitor Cd is Icap(peak)=20MW /0.95/8kV≈2632A, the voltage fluctuation ΔVd at this time is the peak value, and becomes ΔVd(peak)≈349V.

The fluctuation ΔVd of the DC voltage affects the compensation current control of the first and second self-excited power converters CNV1 , CNV2 , causing distortion of the compensation current. In order to suppress the DC voltage variation ΔVd to be small, it is necessary to increase the capacitance of the capacitor Cd, and if the capacitance of Cd is too large, it becomes an uneconomical system.

In the embodiment shown in FIG. 10 , an LC filter is connected in parallel to the DC smoothing capacitor Cd so that the resonant frequency matches approximately twice the frequency of the single-phase AC power feeder.

For example, when the power supply frequency f1 = 60 Hz, prepare an LC filter whose resonance frequency matches 2 x f1 = 120 Hz. That is, when Cf=4mF, Lf=0.44mH. The Icap is absorbed by the LC filter circuit, and the variation ΔVd of the DC voltage is suppressed. In addition, although the DC smoothing capacitor Cd cannot be eliminated in order to absorb the high-order harmonic current accompanying the PWM control of the voltage source self-excited power converters CNV1 and CNV2, the capacity of Cd can be greatly reduced, and the volume of the device can be reduced. Small, lightweight and cost-effective.

As described above, the DC voltage fluctuation ΔVd is suppressed by absorbing the power fluctuation portion ΔPL accompanying the single-phase load fluctuation at twice the frequency of the AC feeder line by the LC filter. As a result, the capacity of the DC smoothing capacitor Cd can be reduced, and the fluctuation ΔVd of the DC voltage can be significantly reduced. Although there is concern about the electrical oscillation phenomenon caused by the LC filter during a transition such as a sudden load change, since the DC voltage control is performed by the above-mentioned first and second voltage-source self-excited power converters CNV1 and CNV2, it is possible to provide Damping stabilized system for damping of electrical oscillations.

third embodiment

12 is a block diagram showing a third embodiment of a control unit for compensation currents IMc, ITc outputted from the first and second self-excited power converters CNV1, CNV2 in the AC power feeding system of the present invention.

In the figure, Kff represents a proportional element, C1~C3 represent a comparator, Gv(S) represents a voltage control compensation circuit, M1, M2 represent a multiplier, AD1~AD5 represent an adder/subtractor, Gi1(S), Gi2(S) Indicates the current control compensation circuit, PWM1 and PWM2 are pulse width modulation control circuits.

The power PL of the single-phase load Load fluctuates at twice the frequency f1 of the AC feeder line. The detected value of the load power PL is time-averaged to obtain the average value PL(av) of the load power. Next, a forward-compensated effective current peak value command Ismff ^* proportional to the load power average value PL(av) is generated through the proportional element Kff.

When half of the load power PL is supplied from the M-seat and T-seat windings of the Scott connection transformer TR, the above-mentioned peak value command Ismff becomes as follows when the voltage VM and the peak value of VT are Vsm. which becomes:

PL(av)=VM×IMs+VT×ITs

＝Vsm sinωt×Ismff sinωt

+Vsm·cosωt×Ismff·cosωt

＝Vsm·Ismf f

Find out from this:

Ismff*=PL(av)/Vsm

Therefore, Kff=1/Vsm.

On the other hand, the DC voltage command value Vd ^* is compared with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd by the comparator C1, and the deviation εv=Vd ^* -Vd is calculated by the subsequent voltage control compensation circuit Gv(S). Proportional or integral amplification is performed to generate another peak value command Ismo ^* of the input current.

Furthermore, the adder-subtractor AD5 adds the output signal Ismo ^* from the voltage control compensation circuit Gv(S) to the output signal Ismff ^* of the above-mentioned proportional element Kff to generate a new effective current peak value command Ism ^* =Ismo ^* +Ismff ^* .

The multiplier M1 solves the unit sine wave sinωt synchronized with the M-seat voltage VM of the Scott connection transformer TR, multiplies the above-mentioned input current wave peak command Ism ^* , and outputs the input current command IMs ^* =Ism ^* ×sinωt. In addition, the multiplier M2 calculates the unit sine wave cosωt synchronized with the T-seat voltage VT of the Scott connection transformer TR, multiplies the above-mentioned input current wave peak command Ism ^* , and outputs the input current command ITs ^* =Ism ^* ×cosωt.

The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* .

Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to obtain the T-seat unblocking compensation current command value ITc ^* =ITL-ITs ^* .

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and the signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, the T-seat input current ITs supplied from the Scott connection transformer TR is controlled so that ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=ITs ^* . This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. However, the T-seat load current becomes ITL=0.

The currents IMs and ITs of the M-seat and T-seat of the above-mentioned Scott connection transformer TR have the same amplitude value Ism ^* , and are two-phase balanced currents whose phases are shifted by 90°. As a result, the current supplied from the three-phase AC power supply SUP1 also becomes a sinusoidal current with power factor=1 balanced by the three phases.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the above-mentioned peak value command Ismo ^* increases, the supply power Ps from the AC power supply SUP1 increases, and becomes larger than the load power PL, Ps-PL It is stored as energy in the above-mentioned DC smoothing capacitor Cd. As a result, Vd increases and is controlled so that Vd=Vd ^* . Conversely, in the case of Vd>Vd ^* , the peak value command Ismo ^* decreases to become Ps<PL, and the stored energy of the DC smoothing capacitor Cd is reduced, and Vd=Vd ^* is also controlled.

In the case of a sharp change in load power PL, the effective current peak value command Ismff ^* of the forward compensation that is proportional to the average value of load power PL(av) also changes, and it can provide the same value as the average value of load power PL(av) The consistent two-phase balanced effective current command values IMs ^* , ITs ^* can suppress the variation of the DC voltage Vd at the time of transition.

4th embodiment

Fig. 13 is a block diagram showing a fourth embodiment of a control unit for compensating current in an AC power feeding system according to the present invention. In the figure, C1~C3 represent the comparator, Gv(S) represents the voltage control compensation circuit, INV represents the inverting circuit, M2 represents the multiplier, Km represents the proportional element, AD2 and AD4 represent the adder and subtractor, Gi1(S), Gi2(S) represents the current control compensation circuit, and PWM1 and PWM2 represent the pulse width modulation control circuit.

The comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to proportional or integrate the deviation εv=Vd ^* -Vd Amplified, through the inverting circuit INV, the effective current wave peak command ITcm ^* of the T seat is generated.

Find the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR, multiply the inverse value of the above-mentioned input current wave peak command ITcm ^* by the multiplier M2 behind, and output the T-seat compensation current command value ITc ^* =ITcm ^* ×cosωt.

On the other hand, the detected value of the M-seat load current IML is passed through the proportional element Km to obtain the M-seat compensation current command value IMc ^* . The proportionality constant Km can be selected between 0 and 1, for example, Km=0.5. That is, the M-seat compensation current command value becomes IMc ^* =IML/2.

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-IML/2=IML/2.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, since the T-seat load current is ITL=0, the T-seat input current ITs supplied from the Scott connection transformer TR is controlled to ITs=ITL-ITc=-ITc ^* =-ITcm ^** cosωt. This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the deviation εv becomes a positive value, and the peak value command ITcm ^* of the above-mentioned T-seat compensation current amplified and inverted takes a negative value. increase, the compensation current ITc=-ITs of the second voltage-type self-excited power converter CNV2 flows in antiphase with the T-seat voltage VT. As a result, the effective power PTs=VT×ITs is supplied to the DC smoothing capacitor Cd from the T-seat winding of the Scott connection transformer TR through the second voltage source self-excited power converter CNV2, and the DC voltage Vd is increased.

Conversely, when Vd>Vd ^* , the deviation εv becomes a negative value, and the peak value command ITcm ^* of the compensation current of the T seat that is amplified and inverted increases with a positive value, and the compensation current ITc=-ITs, and T The base voltage VT increases in phase, and the effective power PTs returns to the T base winding from the DC smoothing capacitor Cd. As a result, it is controlled so that Vd=Vd ^* .

Fig. 14 shows a voltage-current vector diagram at the time of power running load under control by the control unit of Fig. 13 . The M-seat load current IML flows with a phase angle θ lagging behind the M-seat voltage VM. When Km=0.5, the compensation current IMc from the first voltage source self-excited power converter CNV1 is controlled to be IMc=IML/2. As a result, the voltage Vd applied to the DC smoothing capacitor Cd decreases, and becomes Vd ^* >Vd. The compensation current wave peak command ITcm ^* of the T seat increases with a negative value, and the compensation current ITc having an opposite phase to the voltage VT flows. The current ITs of the T-seat winding is the inverse value of the compensation current ITc, so the power PTs is supplied from the T-seat winding to the DC smoothing capacitor Cd via the second voltage-type self-excited power converter CNV2, and is controlled to be Vd ^* =Vd.

The current IMs supplied from the M-seat winding is IMs=IML-IMc=IML/2. For the T-seat input current ITs, the phase of the M-seat input current IMs becomes (90°+θ). Although some current imbalance remains, the compensation current IMc can bear half of the ineffective component of the load current IML, which can suppress the power converter CNV1 capacity.

fifth embodiment

15 is a block diagram showing a fifth embodiment of another example of the compensation current control unit of the present invention. In the figure, C1~C3 represent comparators, Gv(S) represents voltage control compensation circuit, INV represents inverting circuit, M2 and M3 represent multipliers, Km and Kp represent proportional components, AD2, AD4 and AD5 represent adder and subtractor , Gi1(S), Gi2(S) represent the current control compensation circuit, PWM1, PWM2 represent the pulse width modulation control circuit.

The comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to proportional or integrate the deviation εv=Vd ^* -Vd Amplified, through the inverter circuit INV, the effective current wave peak command ITpm ^* of the T seat is generated.

Find the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR, multiply the above-mentioned effective current wave peak value command ITpm ^* by the multiplier M2 behind, and output the T-seat effective compensation current command value ITcp ^* =ITpm ^* ×cosωt.

In addition, the reactive power QL of the load is detected, and a T-seat reactive current peak value command ITqm ^* proportional to its time average value QL(av) is generated. Kp is a proportional constant at this time, for example, the proportional constant Kp is provided so as to bear half of the reactive power QL(av) of the load. Find the unit sine wave sinωt synchronous with the M-seat voltage VM of the Scott connection transformer TR, multiply the above-mentioned invalid current wave peak value command ITqm ^* by the multiplier M3 behind, and output the T-seat invalid compensation current command ITcq ^* =ITqm ^* × sin ωt.

Next, the T-seat valid compensation current command ITcp ^* and the T-seat invalid compensation current command ITcq ^* are added by the adder-subtractor AD5 to obtain the T-seat compensation current command value ITc ^* .

On the other hand, the detected value of the M-seat load current IML is passed through the proportional element Km to obtain the M-seat compensation current command value IMc ^* . The proportionality constant Km can be selected between 0 and 1, here Km=0.5. Therefore, the M-seat compensation current command value becomes IMc ^* =IML/2.

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* . The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-IML/2=IML/2.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned T-seat compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies the deviation εt=ITc ^* -ITc, Input to the adder-subtractor AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* . The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, since the T-block load current becomes ITL=0, the T-block input current ITs supplied from the Scott connection transformer TR is controlled to be ITs=ITL-ITc=-ITc ^* =-(ITpm ^* ×cosωt+Itqm ^* ×sinωt). The input current ITs only lags the load power factor angle θ with respect to the T-seat voltage VT.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the deviation εv becomes a positive value, and the peak value command ITpm ^* of the above-mentioned T-block compensation effective current that is amplified and inverted is negative. As the value increases, the active component of the T-seat input current ITs=-ITc increases, and the effective power PTs is supplied to the DC smoothing capacitor Cd from the T-seat winding of the Scott connection transformer TR through the second voltage-type self-excited power converter CNV2 , so that the DC voltage Vd rises.

Conversely, when Vd>Vd ^* , the deviation εv becomes a negative value, and the peak value command ITpm ^* of the above-mentioned T seat compensation current that is amplified and inverted increases with a positive value, and the effective component of the input current ITs=-ITc Negative, the effective power PTs returns to the T-seat winding from the DC smoothing capacitor Cd. As a result, it is controlled so that Vd=Vd ^* .

Fig. 16 shows a voltage-current vector diagram at the time of power running load under control by the control unit of Fig. 15 . The M-seat load current IML flows with a delay of the phase angle θ from the M-seat voltage VM. When Km=0.5, the compensation current IMc from the first voltage source self-excited power converter CNV1 is controlled to be IMc=IML/2. The current IMs supplied from the M-seat winding is IMs=IML-IMc=IML/2. IMc=IMs For M-seat voltage VM, only the load power angle θ lags behind.

On the other hand, the effective component of the T-seat compensation current ITc is supplied so that the DC voltage Vd matches the command value Vd ^* , and is equal to the inverse value of the effective component of the M-seat compensation current IMc in the steady state of Vd=Vd ^* . Furthermore, if the ineffective component of the T-seat compensation current ITc is set to 1/2 of the ineffective component of the load current IML as described above, the T-seat compensation current ITc has the same amplitude as the M-seat compensation current IMc, and the phase lag is 90°. °.

Since the T-seat input current Its is ITs=ITc, the M-seat input current IMs has the same amplitude and is 90° ahead of the phase angle, and the T-seat voltage lags only by the phase angle θ.

As a result, the currents of the M-block and T-block windings of the Scott connection transformer TR are balanced, and the current supplied from the three-phase AC power supply SUP1 is also a three-phase balanced current.

According to this control method, although the currents IMs and ITs of the M-base and T-base windings become phases lagging behind the load power angle θ, they become currents balanced by two phases. In addition, the amplitudes of the compensation currents IMc and ITc are equal, and the two voltage-source self-excited power converters CNV1 and CNV2 can be set to have the same capacity.

sixth embodiment

17 is a block diagram showing a sixth embodiment of the compensation current control unit in the AC power feeding system of the present invention. In the figure, C1~C3 represent comparators, GV(S) represent voltage control compensation circuit, INV represent inverting circuit, M1, M2 represent multipliers, KL represent proportional elements, AD2~AD4 represent adder and subtractor, Gi1(S ), Gi2(S) represent the current control compensation circuit, PWM1, PWM2 represent the pulse width modulation control circuit.

The comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to proportional or integrate the deviation εv=Vd ^* -Vd Amplify to generate the effective current wave peak command ITsm ^* of the T seat.

Find the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR, multiply the above-mentioned input current wave peak command ITsm ^* by the multiplier M2 behind, and find the T-seat input current command value ITs ^* = ITsm ^* × cos ωt. The M-seat compensation current command electric value ITc ^* =ITL-ITs ^* is generated by the adder-subtractor AD3. However, since T-seat load current ITL=0, it becomes ITc ^* =-ITs ^* .

On the other hand, detect load power PL=VM×IML×cosθ, and calculate its time average value PL(av). Through the proportional element KL, for example, when KL=0.5, the part equivalent to the load power average value PL(av)1/2 is used as the peak value command IMcm ^* of the M-seat compensation effective current, which is multiplied by the multiplier M1 Using the unit sine wave sinωt synchronized with the M-seat voltage VM, calculate the M-seat compensation current command value IMc ^* =IMcm ^* ×sinωt.

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. Conversely, in the case of IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also set.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* . Since the compensation current IMc does not include an ineffective component, the M-seat input current IMs includes half of the active component of the load current IML and all the ineffective components of IML.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, in the case of ITc ^* <ITc, the deviation εt is negative, the signal et ^* is controlled to decrease, the compensation current ITc is reduced, and ITc ^* =ITc is also set.

As a result, the T-seat load current becomes ITL=0, and the T-seat input current ITs supplied from the Scott connection transformer TR is controlled to ITs=ITL-ITc=-ITc ^* =-ITcm ^* ×cosωt. The T-seat input current ITs is controlled to be a sine wave current in phase with the T-seat voltage VT.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the deviation εv becomes a positive value, and the peak value command ITpm ^* of the above-mentioned T-seat compensation current amplified and inverted takes a negative value. Increase. At the same time, the T-seat input current ITs=-ITc increases, and the effective power PTs is supplied to the DC smoothing capacitor Cd from the T-seat winding of the Scott connection transformer TR through the second voltage-type self-excited power converter CNV2, so that the DC voltage Vd rises.

Conversely, when Vd>Vd ^* , the deviation εv becomes a negative value, and the peak value command ITcm ^* of the above-mentioned T-seat compensation current that has been amplified and inverted increases with a positive value, and the effective component of the input current ITs=-ITc Negative, the effective power PTs returns to the T-seat winding from the DC smoothing capacitor Cd. As a result, it is controlled so that Vd=Vd ^* .

Fig. 18 shows a voltage-current vector diagram at the time of power running load under control by the control unit of Fig. 17 . The M-seat load current IML flows with a phase angle θ lagging behind the M-seat voltage VM. The compensation current IMc from the first voltage source self-excited power converter CNV1 is controlled so as to be in phase with the M-seat voltage. The current IMs supplied from the M-seat winding is IMs=IML-IMc, and is controlled to include half of the active component of the load current IML and all the ineffective components of the IML.

On the other hand, supplying the active component of the T-seat compensation current ITc makes the DC voltage Vd coincide with the command value Vd ^* , and in the steady state of Vd=Vd ^* , the above-mentioned M-seat compensation current IMc has the same amplitude and a phase lag of 90 degree vector. The T-seat input current ITs is ITs=-ITc, which is controlled to be a sine wave in phase with the T-seat voltage VT.

According to this control method, the compensation currents IMc and ITc of the M-seat and T-seat become KL=0.5 times of the effective component of the load current IML, and the two voltage-type self-excited power converters CNV1 and CNV2 can be set to be the same The capacity does not need to compensate the reactive power QL of the load. As a result, the capacity of the self-excited power converters CNV1 and CNV2 can be reduced.

Seventh Embodiment

19 is a block diagram showing a seventh embodiment of still another example of compensation current control means in the AC power feeding system of the present invention. In the figure, C1~C3 represent comparators, Gv(S) represent voltage control compensation circuit, INV represent inverting circuit, M1, M2 represent multipliers, KL represent proportional elements, AD1~AD4 represent adder and subtractor, Gi1(S ), Gi2(S) represent the current control compensation circuit, PWM1, PWM2 represent the pulse width modulation control circuit.

The comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to proportional or integrate the deviation εv=Vd ^* -Vd Amplify to generate the peak value instruction ITsm ^* of the input current of the T seat.

Find the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR, multiply the above-mentioned wave peak value command ITsm ^* by the multiplier M2 behind, and find the T-seat compensation current command value ITs ^* = ITsm ^* ×cosωt. The M-seat compensation current command value ITc ^* =ITL-ITs ^* is generated by the adder-subtractor AD3. However, since T-seat load current ITL=0, it becomes ITc ^* =-ITs ^* .

On the other hand, detect load power PL=VM×IML×cosθ, and calculate its time average value PL(av). Through the proportional element KL, for example, when KL=0.5, the part equivalent to 1/2 of the load power average value PL(av) is used as the peak value command IMsm ^* of the M-seat input current, which is multiplied by the multiplier M1 Using the unit sine wave sinωt synchronized with the M-seat voltage VM, find the M-seat input current command value IMs ^* =IMsm ^* ×sinωt.

The input current command value IMs ^* is subtracted from the detected value of the load current IML by the adder-subtractor AD1 to generate the M-seat compensation current command value IMc ^* =IML-IMs ^* . The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . That is, the M-seat input current IMs is controlled to be a sine wave having the same phase as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* . The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, since the T-seat load current is ITL=0, the T-seat input current ITs supplied from the Scott connection transformer TR is controlled to ITs=ITL-ITc=-ITc ^* =-ITcm ^* ×cosωt. The input current ITs is controlled to be a sine wave current in phase with the T seat voltage VT.

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the deviation εv becomes a positive value, and the peak value command ITcm ^* of the above-mentioned T-block compensation effective current that is amplified and inverted is negative. value increases. Therefore, the T-seat input current Its=-ITc increases, and the effective power PTs is supplied to the DC smoothing capacitor Cd from the T-seat winding of the Scott connection transformer TR through the second voltage-type self-excited power converter CNV2, so that the DC voltage Vd rises.

Conversely, when Vd>Vd ^* , the deviation εv becomes a negative value, and the peak value command ITcm ^* of the above-mentioned T-seat compensation current that has been amplified and inverted increases with a positive value, and the effective component of the input current ITs=-ITc Negative, the effective power PTs returns to the T-seat winding from the DC smoothing capacitor Cd. As a result, the DC voltage Vd decreases, and is controlled so that Vd=Vd ^* .

Fig. 20 shows a voltage-current vector diagram at the time of power running load when controlled by the control unit of Fig. 19 . The current IMs of the M-seat winding and the current ITs of the T-seat winding are controlled to be in phase with the voltage VM of the M-seat and the voltage VT of the T-seat respectively. In the steady state of DC voltage Vd=Vd ^* , the amplitude values are also the same, and the two-phase balanced sine wave current becomes, and the three-phase current supplied from the three-phase AC power supply SUP1 also becomes a balanced sine wave current.

Eighth embodiment

Fig. 21 is a block diagram showing an AC power feeding system for an electrified railway according to an eighth embodiment of the present invention. In the figure, SUP1 represents a three-phase AC power supply, TR represents a Scott connection transformer, Fa represents a single-phase AC feeder, Load represents a tram load, TRm and TRt represent a single-phase transformer, and CNV1 and CNV2 represent voltage-type self-excited power conversion Cd means smoothing capacitor, ESS means energy storage device.

The Scott connection transformer TR transforms the three-phase AC power sources Vu, Vv, Vw into two-phase AC voltages VM, VT, and the two-phase voltages VM and VT have a phase difference of 90°. In addition, as a connection method of a transformer for converting a three-phase voltage into a two-phase voltage, there is a modified Woodbridge connection transformer and the like.

The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-block terminal, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 is connected to the T-block terminal. A smoothing capacitor Cd is connected to the DC-side terminals of the first and second voltage source self-excited power converters CNV1 and CNV2, and an energy storage device ESS for transferring energy to and from the DC smoothing capacitor Cd is connected in parallel.

The first and second voltage-source self-excited power converters CNV1 and CNV2 have the same configuration as those described in FIGS. 3 and 4 . In order to increase capacity and reduce high-order harmonics, output transformers TRm and TRt are used for series (or parallel) compound operation.

The energy storage device ESS is composed of, for example, a bidirectional circuit breaker CHO, a DC reactor Ld, and an electric double layer capacitor EDLC.

FIG. 22 shows an embodiment of the main circuit structure of the energy storage device ESS. In the figure, Cd denotes a DC smoothing capacitor, CHO denotes a bidirectional circuit breaker, Ld denotes a DC reactor, and EDLC denotes an electric double layer capacitor.

The bidirectional circuit breaker is composed of self-arc-extinguishing components Sx, Sy, and high-speed diodes Dx, Dy, and controls the output voltage Vcho through PWM control. That is, when the voltage applied to the DC smoothing capacitor Cd is denoted as Vd, it becomes

When Sx is turned on (Sy is turned off), Vcho＝+Vd

When Sy is turned on (Sx is turned off), Vcho=0.

The average value Vcho(av) of the output voltage Vcho becomes Vcho(av) when the on-period of the self-arc-extinguishing element Sx is denoted as Ton (the on-period of Sy becomes T-Ton) for the switching period T of the circuit breaker. )=(Ton/T)×Vd.

The electric double layer capacitor EDLC is a large-capacity capacitor, which has the characteristics of fast charge and discharge and long life. The voltage Ved applied to the electric double layer capacitor EDLC is lower than the voltage Vd applied to the above-mentioned DC smoothing capacitor Cd.

A difference voltage between the output voltage Vcho of the bidirectional circuit breaker CHO and the voltage Ved of the electric double layer capacitor EDLC is applied to the DC reactor Ld, and the current Ied of the DC reactor Ld can be controlled by adjusting the difference voltage (Vcho-Ved).

23 shows a compensation current control circuit CONT1 for controlling the first and second voltage source self-excited power converters CNV1 and CNV2 in the device of FIG. 21 , and FIG. 24 shows a charge and discharge current control circuit CONT2 for controlling the bidirectional circuit breaker CHO. In the figure, C1～C5 represent comparators, Gv(S) and H(S) represent voltage control compensation circuits, M1 and M2 represent multipliers, Fe(x) represent voltage command generators, DV represent dividers, AD1～AD4 , AD6-AD8 represent adder-subtractors, Gi1(S), Gi2(S), Gi3(S) represent current control compensation circuits, PWM1-PWM3 represent pulse width modulation control circuits.

The compensation currents IMc, ITc output from the first and second voltage source self-excited power converters CNV1, CNV2 are controlled as follows.

The comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to proportional or integrate the deviation εv=Vd ^* -Vd Amplifies and generates the command Ism ^* of the peak value of the input current.

The multiplier M1 obtains the unit sine wave sinωt synchronized with the M-seat voltage VM of the Scott connection transformer TR, multiplies the above-mentioned input voltage peak command Ism ^* , and outputs the input current command IMs ^* =Ism ^* ×sinωt.

The multiplier M2 obtains the unit sine wave cosωt synchronized with the T-seat voltage VT of the Scott connection transformer TR, multiplies the above-mentioned input voltage wave peak command Ism ^* , and outputs the input current command ITs ^* =Ism ^* ×cosωt.

The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* .

Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to generate the T-seat compensation current command value ITc ^* =ITL-ITs ^* .

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows. Of course, when the leakage inductance of the above-mentioned single-phase transformer TRm is small, a reactor Lsmo may be inserted in series on the primary or secondary winding side of the transformer TRm.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and the signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* . The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current IMc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, the T-seat input current ITs supplied from the Scott connection transformer TR is controlled so that ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=ITs ^* . This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. Wherein, the T seat load current ITL=0.

The currents IMs and ITs of the M-seat and T-seat of the above-mentioned Scott connection transformer TR have the same amplitude value Ism ^* , and are two-phase balanced currents whose phases are shifted by 90°. As a result, the current supplied from the three-phase AC power supply SUP1 also becomes a sinusoidal current with power factor=1 balanced by the three phases.

On the other hand, the bidirectional circuit breaker CHO of the energy storage device ESS is controlled as follows by the control circuit CONT2 shown in FIG. 24 . First, detect the load power PL, calculate its time average value PL(av), and input it to the voltage command generator Fe(x). The voltage command generator Fe(x) provides the effective power command Pso ^* supplied from the AC power supply SUP1 according to the load power PL(av).

Also, the voltage Ved applied to the electric double layer capacitor EDLC is detected, and the comparator C4 calculates the deviation εed=Ved ^* −Ved from the command value Ved ^* . By integrating this deviation εed, the compensation power command ΔPs ^* is obtained and input to the adder-subtractor AD6.

In the adder-subtractor AD6, the output signal Pso ^* from the above-mentioned power command generator Fe(x) is added to the above-mentioned compensation power command ΔPs ^* , and the command Ps ^* =Pso of the active power supplied from the above-mentioned AC power supply SUP1 is generated. ^* +ΔPs ^* .

Furthermore, the following adder-subtractor AD7 obtains the difference between the detected load power value PL(av) and the above-mentioned effective power command value Ps ^* , and generates the effective power command value Ped ^* =PL(av )-Ps ^* . The divider DV divides the effective power command value Ped ^* by the voltage Ved applied to the EDLC to obtain the command value Ied ^* of the current flowing through the DC reactor Ld.

The deviation between the current command value Ied ^* and the detected value of the current Ied flowing through the above-mentioned DC reactor Ld is generated by the comparator C5 εied=Ied ^* -Ied, and the current control compensation circuit Gi3(S) in the back converts the deviation εied is inverting and amplified, and its output signal e3 ^* is input to the adder-subtractor AD8. In the adder-subtractor AD8, the signal Eed ^* equivalent to the voltage Ved of the electric double layer capacitor EDLC is added to the above-mentioned signal e3 ^* , and the control signal echo is input to the pulse width modulation control circuit PWM3 of the circuit breaker device CHO. The pulse width modulation control circuit PWM3 transmits a gate signal to the self-arcing elements Sx and Sy of the circuit breaker device, and generates a voltage Vcho proportional to the input signal echo ^* from the circuit breaker device CHO.

When Ied ^* >Ied, the deviation εied becomes a positive value, and the inverted amplified signal e3 ^* becomes a negative value to decrease the control signal echo ^* to the PWM control circuit PWM3. As a result, the output voltage Vcho of the circuit breaker device CHO decreases, the voltage Ved-Vcho applied to the DC reactor Ld increases, and the current Ied increases.

Conversely, when Ied ^* <Ied, the deviation εied becomes a negative value, and the inversely amplified signal e3 ^* becomes a positive value to increase the control signal echo ^* to the PWM control circuit PWM3. As a result, the output voltage Vcho of the circuit breaker device CHO increases, the voltage Ved-Vcho applied to the DC reactor Ld decreases, and the current Ied decreases.

In this way, the current Ied flowing through the DC reactor Ld is controlled so as to match its command value Ied ^* .

For example, if the output signal Pso ^* from the voltage command generator Fe(x) is constant, and when the compensation signal ΔPs ^* =0, the command value Ps ^* =Pso ^* +ΔPs ^* of the active power supplied from the AC power supply SUP1 becomes constant, If the power running load power PL(av) is increased, the power command value Ped ^* =PL(av)-Ps ^* supplied from the energy storage device ESS increases, so that the current Ied=Ied ^* of the DC reactor Ld follows the The direction of the arrow (discharge direction) increases. As a result, the energy stored in the electric double layer capacitor EDLC decreases, and the applied voltage Ved also decreases.

As a result, Ved ^* >Ved, the deviation εed becomes a positive value, the output signal ΔPs ^* of the voltage control compensation circuit H(S) is gradually increased, and the command value Ps ^* of the active power supplied from the AC power supply SUP1 is increased. Therefore, the power command value Ped ^* =PL(av)-Ps ^* supplied from the energy storage device ESS decreases and becomes a negative value. The so-called Ped ^* <0 means that Ied=Ied ^* is also negative, and the direction of the current is opposite to the arrow in the figure. That is, a charging current is supplied to the electric double layer capacitor EDLC to gradually increase the voltage Ved. Finally, Ved=Ved ^* is controlled. When the capacity of the electric double layer capacitor EDLC is large, the change in the applied voltage Ved accompanying the charging and discharging described above is small, approximately Ved≈Ved ^* .

FIG. 25 shows an example of the characteristics of the power command generator Fe(x) in FIG. 24. Before the power running load power PL(av) reaches PLa, the power command value Pso ^* is given as Pso ^* =k·PL(av). Wherein, k is a proportional constant, which can be selected within the range of k=0-1. If given as k=1, then Pso ^* =PL(av), which becomes the total effective power of the load power PL(av) supplied from the AC power supply SUP1.

When PL(av) exceeds the set value PLa, the effective power command value Pso ^* =Psa ^* becomes constant. At this time, power Ped=PL(av)-Pso, which is the difference between load power PL(av) and power Pso supplied from AC power supply SUP1, is supplied from energy storage device ESS.

In addition, in the regenerative operation, until the load power PL(av) reaches -PLb, the power command value Pso ^* is given as Pso ^* =k·PL(av). When k=1, Pso ^* =PL(av), control is performed so that all the effective power of the regenerative load power PL(av) is regenerated in the AC power supply SUP1.

When the regenerative power PL(av) exceeds the set value -PLb, the effective power command value Pso ^* =-Psb ^* becomes constant. At this time, the electric power of Ped=PL(av)-Pso is regenerated to the energy storage device ESS.

Generally, in electrified railways, the time for powering the load is long and the time for regenerative load is short. In the characteristics of the power command generator Fe(x) in FIG. 25, the upper limit value Psa ^* on the power running side is set larger, and the lower limit value Psb ^* on the regenerative side is set smaller. Thereby, the charging and discharging energy W=Ped×t to the energy storage device ESS becomes 0 on average, and the applied voltage to the electric double layer capacitor EDLC can be Ved=Ved ^* . When the applied voltage Ved to the EDLC deviates from the command value Ved ^* , as described above, the corrected power command ΔPs ^* acts and is controlled so that Ved=Ved ^* gradually becomes.

In addition, as mentioned above, by setting the proportionality constant k=1, when the load power operates within the range of -PLb ^* <PL(av)<PLa ^* , Ps=PL(av) becomes Ps=PL(av), and there is no need to supply power from the energy storage device ESS Power Ped. That is, energy transfer to and from the electric double layer capacitor EDLC occurs only when the above-mentioned set value is exceeded, and since the time is short, the capacity of the electric double layer capacitor EDLC can be suppressed to be small. advantage.

Fig. 26 shows the voltage and current vector diagrams of the M seat and the T seat during power running when controlled by the control circuit of Fig. 23 and Fig. 24 . The T-seat load current ITL=0, and the M-seat load current IML is delayed by a certain phase θ with respect to the voltage VM. The load power PL=VM×IML×cosθ is equal to the sum of the supply power Ps from the AC power supply SUP1 and the supply power Ped from the energy storage device ESS.

The currents IMs and ITs supplied from the AC power supply SUP1 are controlled to be sine waves in phase with the M-seat voltage VM and the T-seat voltage VT, respectively, and the input power Ps becomes Ps=IMs×VM+ITs×VT. In addition, the compensation currents IMc and ITc supplied from the first and second voltage source self-excited power converters CNV1 and CNV2 are respectively:

IMc=IML-IMs

ITc=ITL-ITs=-ITs

Active power Ped supplied from the energy storage device ESS is included in the M-seat compensation current IMc, and Ps=PL-Ped is supplied from the AC power supply SUP1.

The effective power PMs of the M seat is equal to the effective power PTs of the T seat, and half of Ps=PL-Ped is supplied from the M seat winding of the Scott connection transformer, and the remaining half is supplied from the T seat winding.

The power PTs=Ps/2 supplied from the T-block winding is regenerated by the second voltage source self-excited power converter CNV2 and supplied to the DC smoothing capacitor Cd. That is, ITc=-ITs.

Furthermore, its power and Ps/2 are supplied to the single-phase AC feeder Fa through the first voltage-type self-excited power converter CNV1. At this time, including the active power Ped supplied from the energy storage device ESS and the reactive power QL=VM×IML×sinθ of the load, it is supplied from the above-mentioned first voltage-type self-excited power converter CNV1, and is supplied from the Scott connection transformer M-seat windings of TR only supply effective power PMs=Ps/2.

The first and second voltage-type self-excited power converters CNV1 and CNV2 perform power interchange between the M block and the T block, and have the same power capacity. However, when the energy storage device ESS is installed, the power transferred between the energy storage device ESS and the single-phase feeder passes through the first AC output terminal connected to the single-phase AC feeder side (M block). The voltage-type self-excited power converter CNV1 is used for receiving and receiving.

By setting the output capacity of the first voltage-source self-excited power converter CNV1 to be larger than the output capacity of the second voltage-source self-excited power converter CNV2, it is possible to increase the exchange rate with the energy storage device ESS. Power, can increase the compensation amount for the peak load power, can reduce the load sharing of the three-phase AC power supply. In other words, it is possible to reduce the capacity of the substation equipment.

As described above, in the AC power feeding system of the present invention, the regenerative energy of the electric vehicle load is stored, and by releasing the energy when the load is powered, the energy can be effectively used, and the supply power Ps from the three-phase AC power supply SUP1 can be cut off. peak. That is, it is possible to realize a single-phase pull-through AC power feeding system, and seek to reduce the capacity of substation equipment and save electricity.

ninth embodiment

Fig. 27 is a block diagram showing an AC power feeding system for an electrified railway according to a ninth embodiment of the present invention. In the figure, SUP1 represents a three-phase AC power supply, TR represents a Scott connection transformer, Fa represents a single-phase AC feeder, Load represents a tram load, TRm and TRt represent a single-phase transformer, and CNV1 and CNV2 represent voltage-type self-excited power conversion Cd represents the DC smoothing capacitor, Lf and Lf represent the reactor and capacitor that constitute the LC filter, and ESS represents the energy storage device.

The difference from the embodiment shown in FIG. 21 is that an LC filter is connected in parallel to the DC smoothing capacitor Cd. The control mode is the same as that explained in the apparatus of Fig. 21 . For the M-seat voltage VM, the load current IML lags behind the phase angle θ. The single-phase load power PL fluctuates according to twice the frequency (120 Hz) of the AC feeder frequency f1 = 60 Hz.

That is, when the seat M voltage VM=Vsm×sinωt, and the load current IML=ILm×sin(ωt-θ), the power PL becomes

PL＝Vsm×sinωt×ILm×sin(ωt-θ)

＝(Vsm×ILm/2){cosθ-cos(2·ωt-θ)}

The first term is a constant value corresponding to the power Ps supplied from the aforementioned AC power supply SUP1 through the Scott connection transformer TR. The second term is the variation part ΔPL, which fluctuates according to twice the frequency of the power supply frequency.

When the DC smoothing capacitor Cd absorbs the power variation ΔPL accompanying the single-phase load, the variation ΔVd of the voltage Vd applied to the DC smoothing capacitor Cd is proportional to the load power PL, and is proportional to the DC smoothing capacitor Cd. Capacity is inversely proportional.

That is, the current Icap flowing through the DC smoothing capacitor Cd becomes: when the average value of the DC voltage is expressed as Vdo

IcapΔPL/Vdo

＝-{Vsm×ILm/(2·Vdo)}·cos(2·ωt-θ)

Therefore, the variation ΔVd of the DC voltage Vd becomes:

ΔVd=(1/Cd)∫Icap·dt

-{Vsm×ILm/(4·Vdo·ω·Cd)}

×sin(2·ωt-θ)

For example, when the power supply frequency f1=60Hz, the load power PL=20MW (power factor=0.95), the DC voltage Vdo=8kV, and Cd=10mF, the peak value of the current Icap flowing in the DC smoothing capacitor Cd becomes Icap(peak)= 20MW/0.95/8kV≈2632A, the voltage fluctuation ΔVd at this time is the peak value, and becomes ΔVd(peak)≈349V.

The fluctuation ΔVd of the DC voltage affects the compensation current control of the first and second self-excited power converters CNV1 , CNV2 , and causes distortion of the compensation current. In order to keep the DC voltage variation ΔVd small, it is necessary to increase the capacity of the capacitor Cd, which is an uneconomical system.

In the embodiment shown in FIG. 27 , an LC filter matching the resonant frequency around twice the frequency of the single-phase AC feeder line is connected in parallel to the DC smoothing capacitor Cd. For example, in the case of power supply frequency f1 = 60 Hz, prepare an LC filter that matches the resonance frequency at 2 · f1 = 120 Hz. That is, when Cf=4mF, Lf=0.44mH. The Icap is absorbed by the LC filter circuit, and the variation ΔVd of the DC voltage is suppressed. In addition, although the DC smoothing capacitor Cd cannot be omitted in order to absorb the high-order harmonic current accompanying the PWM control of the voltage source self-excited power converters CNV1 and CNV2, the capacity of Cd can be greatly reduced, and the volume of the device can be reduced. Small, lightweight and cost-effective.

As described above, the DC voltage fluctuation ΔVd is suppressed by absorbing the power fluctuation portion ΔPL accompanying the single-phase load fluctuation at twice the frequency of the AC feeder line by the LC filter. As a result, the capacity of the DC smoothing capacitor Cd can be reduced, and the variation ΔVd of the DC power supply can be significantly reduced. In the case of a sudden change in load, etc., there is concern about the electrical oscillation phenomenon caused by the LC filter, but since the DC voltage control is performed by the first and second voltage-source self-excited power converters CNV1 and CNV2, it is possible to use The damping effect of the electrical oscillation decay provides a stable system.

In addition, in the above-mentioned embodiment, the example in which the DC smoothing capacitor Cd is connected between the connection point of the first voltage-source self-excited power converter CNV1 and the second voltage-source self-excited power converter CNV2 has been described. And in order to achieve the purpose of the present invention, it is not necessarily required. In addition, in the above-mentioned embodiment, the case where the first voltage-type self-excited power converter CNV1 and the second voltage-type self-excited power converter CNV2 are used as the power converters constituting the two-phase power conversion device has been described. , but may not be limited to these devices.

Furthermore, in the above-mentioned embodiment, the electric double layer capacitor was cited as an example of the energy storage unit which is one of the structures of the energy storage device, but it is not limited to this, and a secondary battery etc. which can be quickly charged and discharged can also be used. .

In the above-described embodiments, a Scott connection transformer was given as an example of a transformer for converting three-phase power into two-phase power, but it is not necessarily limited to such a transformer.

Tenth embodiment

Fig. 28 is a block diagram of an AC power feeding system for electric railways according to a tenth embodiment of the present invention. In the figure, SUP1 is the first AC power supply, SUP0 and SUP2 are the second AC power supply, M/G is a rotary frequency converter (M/G device), TR1 is a Scott connection transformer, and Fa is a single-phase AC feeder , Load is the tram load, TRm and TRt are single-phase transformers, CNV1 and CNV2 are voltage-type self-excited power converters, Cd is a DC smoothing capacitor, Lf and Cf are reactors and capacitors that constitute LC filters, and TR2 is three Phase transformer, REC is a diode rectifier, CONT1 is a compensation current control unit that controls compensation currents IMc, ITc output from power converters CNV1, CNV2.

The compensation current control unit CONT1 is composed of a DC voltage control circuit Vd-CONT, a compensation current command generation circuit Ic-ref, a compensation current control circuit IMc-CONT, ITc-CONT and a pulse width modulation control circuit PWM1, PWM2.

The M/G device performs frequency conversion between the second AC power supply SUP0 (three-phase-50 Hz) and the first AC power supply SUP1 (three-phase-60 Hz). This M/G device is installed in a frequency conversion station, and transmits power from the first AC power supply SUP1 (three-phase-60 Hz) to the substation using an AC transmission line. That is, in general, a frequency conversion station is at a certain distance from a substation in which its subordinate equipment is installed.

The Scott connection transformer TR1 converts the three-phase AC voltages Vu, Vv, Vw of the first AC power supply SUP1 into two-phase AC voltages VM, VT having a phase difference of 90°. In addition, as a method of connecting a transformer for converting a three-phase voltage into a two-phase voltage, there is a modified Woodbridge connection transformer and the like.

The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-block terminal, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 is connected to the T-block terminal. Also, the AC terminal of the diode rectifier REC is connected to the second AC power supply SUP2 (three-phase-50 Hz) via the three-phase transformer TR2, and the DC output terminal of the diode rectifier REC is connected to the DC smoothing capacitor Cd. The LC filter (Lf, Cf) selects the values of the reactor Lf and the capacitor Cf such that they resonate at a frequency twice the frequency (60 Hz) of the AC feeder line.

A specific configuration example of the compensation current control unit CONT1 in the device of FIG. 28 is the same as that of FIG. 3 . However, in this embodiment, the voltage command generator Fd(x) provides the command value Vd ^* of the voltage Vd applied to the DC smoothing capacitor Cd according to the time average value PL(av) of the load power PL, and sets its characteristics. Graphics in order to adjust the DC output current Irec of the diode rectifier REC.

FIG. 29 shows an example of the characteristics of the voltage command generator Fd(x), showing the DC voltage command value Vd ^* with respect to the time average value PL(av) of the load power PL. In the case of PL(av)>0, that is, in the case of a power running load, the DC voltage command value Vd ^* decreases as the load power PL(av) increases. As a result, the voltage Vd=Vd ^* applied to the DC smoothing capacitor Cd also decreases, and the output current Irec from the diode rectifier REC increases in proportion to the load power PL(av). In other words, the above-mentioned DC voltage command value Vd ^* is provided in consideration of the voltage regulation of the diode rectifier REC including the transformer TR2.

When the power supplied from the diode rectifier REC is Prec=Irec×Vd, the power supplied through the Scott connection transformer TR1 becomes Ps1=PL(av)-Prec, and the Scott connection transformer TR1 or the first AC The capacity of the equipment of the power supply SUP1 and the M/G device is reduced by a portion corresponding to the supply power Prec from the diode rectifier. Of course, at this time, the currents IMs and ITs of the M-block winding and the T-block winding of the Scott connection transformer TR1 become sinusoidal currents with a power factor=1 that are balanced by two phases.

In the case of PL(av)<0, that is, in the case of regenerative operation of the electric car, the DC voltage command value Vd ^* =Vdo ^* becomes constant, and the set voltage Vdo ^* is set to be higher than the no-load voltage of the diode rectifier REC. The rectified voltage Vreco is high.

Since the diode rectifier REC does not have the function of regenerating power to the second AC power supply SUP2, all the power regenerated from the electric car is regenerated by the first AC power supply SUP1 through the Scott connection transformer TR1. At this time, Vdo ^* >Vreco is set so that useless current Irec is not supplied from the diode rectifier REC.

FIG. 30 shows another characteristic example of the voltage command generator Fd(x), showing the DC voltage command value Vd ^* with respect to the time average value PL(av) of the load power PL. In this case, the load power PL(av) includes the regenerative operation, PL(av)<PLo, and the DC voltage command value Vdo ^* =constant. However, Vdo ^* >Vreco is set so that the power setting value PLo is given as a positive value.

When PL(av)>PLo, as the load power PL(av) increases, the DC voltage command value Vd ^* decreases. As a result, the voltage VT=Vd ^* applied to the DC smoothing capacitor Cd also decreases, and the output current Irec from the diode rectifier REC increases in proportion to (PL(av)-PLo).

That is, in the region where the load power PL(av) is smaller than the set power PLo, the supply power from the diode rectifier REC becomes Prec=Irec×Vd=0, and all the load power PL(av) including the regenerative power passes through the Cote wiring voltage regulator TR1 grants and accepts. When PL(av)>PLo, the supply power Prec from the diode rectifier REC increases in proportion to (PL(av)-PLo). This part can reduce the load sharing of Scott connection transformer TR1 or M/G device etc.

For example, let PL(av)>PLo, in the case of providing the voltage command value Vd ^* such that Prec=(PL(av)-PLo), the power running power Ps1 supplied through the Scott connection transformer TR1 becomes Ps1=PLo, A Scott connection transformer TR1 of a capacity corresponding thereto may be prepared in advance.

Fig. 31 is the M seat, T seat voltage-current vector diagram when the electric car load Load that is connected to the AC feeder Fa on the device power operation of Fig. 28. The T-seat load current ITL=0, and the M-seat load current IML lags the voltage VM by a certain phase θ.

If it is assumed that the load power PL=VM×IML×cosθ, the loss is very small, then the load power PL is equal to the power Ps1=VM×IMs+VT×ITs supplied from the Scott connection transformer TR1 and the power Prec supplied through the diode rectifier REC =Sum of Irec×Vd.

The M winding current MS and the T winding current ITs of the Scott connection transformer TR1 are in phase with the M voltage VM and the T voltage VT respectively, forming a sine wave current with a power factor of 1. In addition, the amplitude values Ism are the same, and become two-phase balanced currents.

The compensation current IMc supplied from the first voltage source self-excited power converter CNV1 becomes a difference vector between the load current vector IML and the M-seat winding current vector IMS. When the effective current of the M-seat compensation current IMc is denoted as IMcp, the power supplied from the diode rectifier REC becomes Prec=Irec×Vd=(IMcp-IMs)×VM.

On the other hand, since the compensation current ITc supplied from the second voltage source self-excited power converter CNV2 becomes ITL=0, ITc=-ITs. Power PTs=ITs×VT supplied from the T-block winding is regenerated by the second voltage source self-excited power converter CNV2 and supplied to the DC smoothing capacitor Cd. This effective power PTs is supplied to the single-phase AC feeder line Fa via the first voltage-source self-excited power converter CNV1.

The voltage-current vector diagrams of the M-seat and the T-seat during the load regenerative operation of the electric car connected to the AC feeder Fa are also shown in FIG. 8 in this embodiment. Since the diode rectifier REC cannot regenerate power, the regenerative power is returned to the first AC power supply SUP1 through the Scott connection transformer TR1. Half of the regenerative power PL flows through the M-seat winding of the Scott connection transformer TR1, and the remaining half passes through the first voltage-type self-excited power converter CNV1→DC smoothing capacitor Cd→the second voltage-type self-excited power converter CNV2, flows through the T-seat winding. At this time, the reactive power QL of the load is compensated by the first voltage source self-excited power converter CNV1.

Generally, in electrified railways, the power during power operation is larger than that during regenerative operation. In particular, in the AC feed system of the present invention, since the power exchange is automatically performed between the power running electric car and the regenerative electric car, the power regenerated through the Scott connection transformer TR1 is not too large. Therefore, by supplying a part of the load power from the diode rectifier REC to the power running load PL, the effect of reducing the capacity of the Scott connection transformer TR1 or the M/G device can be increased.

The example waveforms of M-seat voltage VM, load current IML, load power PL, and voltage Vd applied to DC smoothing capacitor Cd during power running of electric car Load connected to single-phase AC feeder Fa are the same as those shown in FIG. 11 . But in the case of this embodiment, the formula for power PL

PL＝Vsm×sinωt×ILm×sin(ωt-θ)

＝(Vsm×ILm/2){cosθ-cos(2·ωt-θ)}

The first item in is a constant value, which is consistent with the sum of the power Ps1 supplied from the AC power supply SUP1 through the Scott connection transformer TR1 and the power Prec supplied through the diode rectifier REC. Furthermore, the second term is a power fluctuation portion ΔPL, which fluctuates at a frequency twice the power supply frequency.

In the present embodiment, the LC filter is also used to absorb the power fluctuation portion ΔPL of the single-phase load accompanying the fluctuation at twice the frequency of the AC feeder line, thereby suppressing the above-mentioned fluctuation ΔVd of the DC voltage. As a result, the capacity of the DC smoothing capacitor Cd can be reduced, and the variation ΔVd of the DC voltage can be significantly reduced. In addition, although there is concern about the electrical oscillation phenomenon caused by the LC filter at the time of transition such as a sudden change, since the DC voltage control is performed by the first and second voltage-source self-excited power converters CNV1 and CNV2 described above, it is possible to make the power The damping effect of oscillation damping provides a stable system.

In addition, by stabilizing the DC voltage Vd, the compensation current control performed by the first and second voltage-source self-excited power converters CNV1 and CNV2 is stabilized, thereby improving control performance. In addition, the withstand voltage range of the power converters CNV1 and CNV2 can be reduced by an amount corresponding to the reduction of the voltage fluctuation ΔVd, and an inexpensive device can be provided.

The first and second voltage-type self-excited power converters CNV1 and CNV2 perform power interchange between the M block and the T block, and have the same power capacity. And the power Prec supplied from the second AC power supply SUP2 through the diode rectifier REC passes through the first voltage-type self-excited power converter CNV1, and is supplied to the electric car load that pulls through the single-phase AC feeder. By setting the output capacity of the first voltage-source self-excited power converter CNV1 to be larger than the output capacity of the second voltage-source self-excited power converter CNV2, the power supplied from the diode rectifier REC can be increased and corresponding The burden on the already installed M/G device (frequency converter) is greatly reduced.

FIG. 32 is a block diagram showing a specific example different from FIG. 3 of the compensation current control unit CONT1 of the AC power feeding system according to the tenth embodiment. In the figure, Fs(x) represents the power command generator, Ks represents the proportional element, ASW represents the switch circuit, SH represents the level detector, C1~C3 represents the comparator, Gv(S) represents the voltage control compensation circuit, M1, M2 Represents a multiplier, AD1-AD4 represent an adder-subtractor, Gi1(S), Gi2(S) represent a current control compensation circuit, PWM1, PWM2 represent a pulse width modulation control circuit.

The power PL of the single-phase load Load fluctuates at a frequency twice the frequency f1 of the AC feeder line. The detection value of the load power PL is time-averaged to obtain the average value PL(av) of the load power.

The effective power command value Ps1 ^* corresponding to the load power average value PL(av) is generated by the input power command generator Fs(x), and converted into the effective current wave peak value command Isma ^* through the proportional element Ks, and input to the switch circuit A-side terminal of ASW.

On the other hand, the comparator C1 compares the DC voltage command value Vd ^* with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd, and uses the subsequent voltage control compensation circuit Gv(S) to convert its deviation εv=Vd ^* - Vd is proportionally or integrally amplified to generate another effective current peak value command Ismb ^* , which is input to the b-side terminal of the switch circuit ASW.

The level detector SH inputs the load power PL(av), judges whether it is higher or lower than the set level PLo, and transmits the switching signal LB to the switching circuit ASW.

When PL(av)>PLo, the output signal LB of the level detector SH=1, the switch circuit ASW is connected to a side, and the effective current peak value command Ism ^* =Isma ^* is input to the multipliers M1 and M2. That is, the compensation current is controlled based on the input power command value Ps1 ^* .

In addition, when PL(av)<PLo, the output signal LB of the level detector SH=0, the switch circuit ASW is connected to the b side, and the effective current peak value command Ism ^* =Ismb ^* is input to the multiplier M1, M2. That is, the compensation current is controlled based on the output signal Ismb ^* from the DC voltage control circuit Gv(S).

Fig. 33(a) is a diagram showing the characteristics of the input power command generator Fs(x), and Fig. 33(b) is a diagram showing the operation of the level detector SH. The input power command generator Fs(x) provides the command value Ps1 ^* of active power supplied from the first AC power supply SUP1 according to the load power PL(av), and generates the command value Ps1 ^* as follows, for example. That is, when a certain set power value is taken as PLo>0, it becomes

When PL(av)<PLo, Ps1 ^* ＝PL(av)

When PL(av)>PLo, Ps1 ^* =Pso ^* =constant

In Fig. 32, the proportional element Ks converts the above-mentioned effective power command value Ps1 ^* into the peak value Isma ^* of the two-phase effective current. When the peak value of the two-phase voltage of the Scott connection transformer TR1 is recorded as Vsm, the proportional constant Ks=1/Vsm. That is, Isma ^* =Ps1 ^* /Vsm.

Find the unit sine wave sinωt synchronous with the M-seat voltage VM of the Scott connection transformer TR1, multiply the input current wave peak command Ism ^* by the multiplier M1, and output the input current command IMs ^* =Ism ^* ×sinωt. In addition, find the unit sine wave cosωt synchronized with the T-seat voltage VT of the Scott connection transformer TR1, multiply the above-mentioned input current wave peak command Ism ^* by the multiplier M2, and output the input current command ITs ^* = Ism ^* ×cosωt .

The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* .

Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to obtain the T-seat compensation current command value ITc ^* =ITL-ITs ^* .

The comparator C2 compares the M-seat compensation current detection value IMc with the above-mentioned compensation current command value IMc ^* , and the current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction Magical device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, and the compensation current IMc is reduced so that IMc ^* =IMc.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the above-mentioned compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to Addition and subtraction device AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc.

As a result, the T-seat input current ITs supplied from the Scott connection transformer TR is controlled so that ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=-ITs ^* . This input current Its becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. Wherein, the T seat load current ITL=0.

The currents IMs and ITs of the M-seat and T-seat of the Scott connection transformer TR1 have the same amplitude value Ism ^* , and the two-phase balanced currents whose phases are shifted by 90°. As a result, the current supplied from the three-phase AC power supply SUP1 also becomes a sinusoidal current with power factor=1 balanced by the three phases.

In FIG. 33, when the load power PL(av)<PLo, the level detector SH outputs LB=0, and the switching circuit ASW is connected to the b side, and the DC voltage Vd ^* =Vdo ^* =is controlled to be constant. At this time, by setting the voltage command value Vdo ^* higher than the no-load rectified voltage Vreco of the diode rectifier REC, the DC output current Irec of the diode rectifier REC becomes 0, and all the load power is supplied or regenerated from the first AC power supply SUP1. PL(av).

When the voltage Vd applied to the DC smoothing capacitor Cd is lower than the command value Vd ^* , the effective current peak value command Ismb ^* increases, and the supply power Ps1 from the AC power supply SUP1 increases to become larger than the load power PL(av). DC smoothing capacitor Cd as part of energy storage Ps1-PL (av). As a result, Vd increases and is controlled so that Vd=Vd ^* . Conversely, when Vd>Vd ^* , the peak value command Ismb ^* decreases, and becomes Ps1<PL(av), and the stored energy of the DC smoothing capacitor Cd is reduced, so that Vd=Vd ^* is also controlled.

On the other hand, when PL(av)>PLo, the output signal from the level detector SH becomes LB=1, the switching circuit is connected to the a side, and the input power is controlled so that Ps1 ^* =PLo=constant. By limiting the input power from the first AC power supply SUP1 to Ps1 ^* =PLo=constant, as the load power PL(av) increases, the DC voltage Vd decreases and the DC current Irec of the diode rectifier REC increases. That is, the power of Prec=PL(av)-PLo is automatically supplied from the diode rectifier REC, and the DC voltage Vd is also stabilized in this state.

In the compensation current control unit of the present invention, there is no need to grasp the voltage adjustment rate of the diode rectifier REC, and the advantage of automatically adjusting the supply power Prec from the diode rectifier REC is by determining the input power Ps1 ^* from the first AC power supply SUP1. In addition, as a characteristic of the above-mentioned input power command generator Fs(x), for example, by setting Ps1 ^* =PLo+k·PL(av) under the condition of PL(av)>PLo, the power Prec from the diode rectifier REC can be adjusted. , which can optimize load sharing. Wherein, when the set value PLo>0, the proportionality constant k=0˜1.

Eleventh embodiment

Fig. 34 is a block diagram of an AC power feeding system for an electrified railway according to an eleventh embodiment of the present invention. In the figure, SUP1 represents the first AC power supply, SUP0 and SUP2 represent the second AC power supply, M/G represents the rotary frequency converter (M/G device), TR1 represents the Scott connection transformer, and Fa represents the single-phase AC feeder , Load means tram load, TRm, TRt means single-phase transformer, CNV1, CNV2 means voltage type self-excited power converter, Cd means DC smoothing capacitor, Lf, Cf means reactor and capacitor constituting LC filter, TR2 means three Phase transformer, REC denotes a diode rectifier, ESS denotes an energy storage device, CONT1 denotes a control unit for compensating current IMc, ITc outputted from the power converters CNV1, CNV2, and CONT2 denotes a control unit for the above energy storage device.

The compensation current control unit CONT1 is composed of a DC voltage control circuit Vd-CONT, a compensation current command generation circuit Ic-ref, a compensation current control circuit IMc-CONT, ITc-CONT and a pulse width modulation control circuit PWM1, PWM2.

The M/G device performs frequency conversion between the second AC power supply SUP0 (three-phase-50 Hz) and the first AC power supply SUP1 (three-phase-60 Hz). Generally, this M/G device is installed in a frequency conversion station, and power is transmitted from the first AC power supply SUP1 (three-phase-60Hz) to the substation using an AC transmission line. That is, the frequency conversion station is at a certain distance from the substation in which its subordinate equipment is installed.

The Scott connection transformer TR1 converts the three-phase AC voltages Vu, Vv, Vw of the first AC power supply SUP1 into two-phase AC voltages VM, VT having a phase difference of 90°.

The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-block terminal, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 is connected to the T-block terminal.

Also, to the second AC power supply SUP2 (three-phase -50 Hz), the AC terminal of the diode rectifier REC is connected via the three-phase transformer TR2, and the DC output terminal of the diode rectifier REC is connected to the DC smoothing capacitor Cd.

The power transfer between the energy storage device ESS and the above-mentioned DC smoothing capacitor Cd is composed of, for example, a bidirectional circuit breaker CHO, an electric double layer capacitor EDLC as a secondary battery, and the like. When there are many regenerative trains and the power is abundant, the power is usually returned (regenerated) to the first AC power supply SUP1, and part or all of the regenerated power is stored in the energy storage device ESS. The stored energy is released when the load of the power running train increases, so as to seek effective utilization of energy. Thereby, it is possible to save electricity costs, reduce the peak load power when viewed from an AC power source, and reduce the capacity of substation equipment.

The LC filter (Lf, Cf) determines the values of the reactor Lf and the capacitor Cf so as to resonate at a frequency twice the frequency (60 Hz) of the AC feeder line.

By absorbing the power fluctuation portion ΔPL of the single-phase load accompanying the frequency fluctuation at twice the frequency of the AC feeder line by the LC filter, the above-mentioned fluctuation ΔVd of the DC voltage is suppressed. As a result, the capacity of the DC smoothing capacitor Cd can be reduced, and the variation ΔVd of the DC voltage can be significantly reduced. Although there is concern about the electrical oscillation phenomenon caused by the LC filter during a transition such as a sudden load change, since the DC voltage control is performed by the first and second voltage-type self-excited power converters CNV1 and CNV2 described above, it is possible to make the electrical oscillation The damping effect of decay provides a stable system.

By stabilizing the DC voltage Vd, the compensation current control by the first and second voltage source self-excited power converters CNV1 and CNV2 or the control of the energy storage device ESS is stabilized, thereby improving the control performance. In addition, the withstand voltage range of the power converters CNV1 and CNV2 or the circuit breaker device CHO can be reduced by an amount corresponding to the reduction of the voltage fluctuation ΔVd, and a cheaper device can be provided.

In the present embodiment, the compensation current control circuit CONT1 for controlling the first and second voltage source self-excited power converters CNV1 and CNV2 is the same as that shown in FIG. 33 . Among them, in the case of this embodiment, the voltage command generator Fd(x) provides the command value Vd ^* of the voltage Vd applied to the DC smoothing capacitor Cd according to the time average value PL(av) of the load power PL, set Its characteristic graph is for adjusting the DC output current Irec of the diode rectifier REC.

FIG. 35 is a diagram showing an example of a characteristic graph of the voltage command generator Fd(x). When the load power is PL(av)<0 (regenerative operation), Vd ^* =Vdo ^* =constant. When PL(av) When >0 (power running), there is a voltage regulation rate as Vd ^* =Vdoo ^* -kv×PL(av). Wherein, Vdoo ^* is the no-load rectified voltage of the diode rectifier REC, which is taken as Vdo ^* >Vdoo ^* .

If the proportional constant kv is increased, the voltage regulation rate increases, and the ratio of the power Ps2=Prec supplied from the second AC power supply SUP2 through the diode rectifier REC increases. On the contrary, if the proportional constant kv is decreased, the voltage regulation rate is reduced, from The ratio of power Ps2=Prec supplied by the second AC power supply SUP2 via the diode rectifier REC is reduced.

Compensation currents IMc, ITc output from the first and second voltage source self-excited power converters CNV1, CNV2 are controlled in the same manner as above. As a result, the currents IMs and ITs of the M-seat and the T-seat of the Scott connection transformer TR1 have the same amplitude value Ism ^* and are two-phase balanced currents whose phases are shifted by 90°. As a result, the current supplied from the three-phase AC power supply SUP1 also becomes a sinusoidal current with power factor=1 balanced by the three phases.

The main circuit configuration of the energy storage device ESS is the same as that shown in FIG. 22 , the configuration of the control circuit is the same as that shown in FIG. 24 , and operates as follows.

The bidirectional circuit breaker CHO is composed of self-arc-extinguishing elements Sx, Sy, and high-speed diodes Dx, Dy, and controls the output voltage Vcho through pulse width modulation control (PWM control). That is, when the voltage applied to the DC smoothing capacitor Cd is denoted as Vd, it becomes:

When Sx is turned on (Sy is turned off), Vcho＝+Vd

When Sy is turned on (Sx is turned off), Vcho=0.

The average value Vcho(av) of the output voltage Vcho becomes Vcho(av) when the on-period of the self-arc-extinguishing element Sx is denoted as Ton (the on-period of Sy becomes T-Ton) for the switching period T of the circuit breaker. )=(Ton/T)×Vd.

The electric double layer capacitor EDLC is a large-capacity capacitor, which has the characteristics of fast charge and discharge and long life. The voltage Ved applied to the electric double layer capacitor EDLC is lower than the voltage Vd applied to the above-mentioned DC smoothing capacitor Cd.

A difference voltage between the output voltage Vcho of the bidirectional circuit breaker CHO and the voltage Ved of the electric double layer capacitor EDLC is applied to the DC reactor Ld, and the current Ied of the DC reactor Ld can be controlled by adjusting the difference voltage (Vcho-Ved).

The bidirectional circuit breaker CHO of the energy storage device ESS is controlled as follows. First, load power PL is detected, its time average value PL(av) is obtained, and input to the second voltage command generator Fe(x). The second voltage command generator Fe(x) provides the sum (=Ps1 ^* +Ps2 ^* ) of the effective power commands supplied from the first AC power supply SUP1 and the second AC power supply SUP2 according to the load power PL(av).

Also, the voltage Ved applied to the electric double layer capacitor EDLC is detected, and the comparator C4 calculates the deviation εd=Ved ^* −Ved from the command value Ved ^* . By integrating this deviation εed, the compensation power command ΔPs ^* is obtained and input to the adder-subtractor AD6.

In the adder-subtractor AD6, the output signal Pso ^* from the above-mentioned power command generator Fe(x) is added to the compensation power command ΔPs ^* to generate active power supplied from the first AC power supply SUP1 and the second AC power supply SUP2 The sum instruction Ps ^* =Pso ^* +ΔPs ^* .

Furthermore, the subsequent subtractor AD7 obtains the difference between the load power detection value PL(av) and the effective power command value Ps ^* , and generates the effective power command value Ped ^* =PL(av)-Ps output from the electric double layer capacitor EDLC ^* . The divider DV divides the effective power command value Ped ^* by the voltage Ved applied to the EDLC to obtain the command value Ied ^* of the current flowing through the DC reactor Ld.

The deviation between the current command value Ied ^* and the detected value of the current Ied flowing through the above-mentioned DC reactor Ld is generated by the comparator C5 εied=Ied ^* -Ied, and the current control compensation circuit Gi3(S) in the back converts the deviation εied is inverting and amplified, and its output signal e3 ^* is input to the adder-subtractor AD8. In the adder-subtractor AD8, a compensation signal Eed ^* equivalent to the voltage Ved of the electric double layer capacitor EDLC is added to the output signal e3 ^* , and the control signal echo ^* is input to the pulse width modulation control circuit PWM3 of the circuit breaker device CHO . The pulse width modulation control circuit PWM3 transmits a gate signal to the self-arcing elements Sx and Sy of the circuit breaker device, and a voltage Vcho proportional to the input signal echo ^* is generated from the circuit breaker device CHO.

When Ied ^* >Ied, the above-mentioned deviation εied becomes a positive value, and the signal e3 ^* obtained by inverting and amplifying it becomes a negative value, and the control signal echo ^* to the PWM control circuit PWM3 is reduced. As a result, the output voltage Vcho of the circuit breaker device CHO decreases, the voltage Ved-Vcho applied to the DC reactor Ld increases, and the current Ied increases in the direction of the arrow in FIG. 32 .

Conversely, when Ied ^* <Ied, the deviation εied becomes a negative value, and the inverted and amplified signal e3 ^* becomes a positive value to increase the control signal echo ^* to the PWM control circuit PWM3. As a result, the output voltage Vcho of the circuit breaker device CHO increases, the voltage Ved-Vcho applied to the DC reactor Ld decreases, and the current Ied decreases. In this way, the current Ied flowing through the DC reactor Ld is controlled so as to match its command value Ied ^* .

For example, when the output signal Pso ^* from the voltage command generator Fe(x) is kept constant, and the compensation signal ΔPs ^* =0, the sum of the effective power supplied from the first AC power supply SUP1 and the second AC power supply SUP2 The power command value Ps ^* = Pso ^* + ΔPs ^* becomes constant, and if the power running load power PL(av) increases, the power command value Ped ^* = PL(av)-Ps ^* supplied from the energy storage device ESS increases, so that The current Ied=Ied ^* of the DC reactor Ld increases in the arrow direction (discharging direction) of FIG. 32 . As a result, the energy stored in the electric double layer capacitor EDLC decreases, and the applied voltage Ved also decreases.

As a result, Ved ^* >Ved, the deviation εed becomes a positive value, the output signal ΔPs ^* of the voltage control compensation circuit H(S) is gradually increased, and the effective voltage supplied from the first AC power supply SUP1 and the second AC power supply SUP2 is increased. The power command value Ps ^* of the power sum. Therefore, the power command value Ped ^* =PL(av)-Ps ^* supplied from the energy storage device ESS decreases and becomes a negative value. Ped ^* <0 means that Ied=Ied ^* is also negative, and the direction of the current is opposite to that of the arrow in the figure. That is, a charging current is supplied to the electric double layer capacitor EDLC to gradually increase the voltage Ved. Finally, it is controlled so that Ved=Ved ^* . When the capacity of the electric double layer capacitor EDLC is large, the change in the applied voltage Ved accompanying charging and discharging is small, and approximately Ved≈Ved ^* .

FIG. 36 shows a characteristic example of the second power command generator Fe(x). Before the power running load power PL(av) reaches PLa, the power command value Pso ^* is provided as Pso ^* =k·PL(av). Wherein, k is a proportional constant, which can be selected within the range of k=0-1. If given as k=1, Pso ^* =PL(av), which becomes the total effective power of the load power PL(av) supplied from the first AC power supply SUP1 and the second AC power supply SUP2. The ratio of the power PLs1 supplied from the first AC power supply SUP1 to the power PLs2 supplied from the second AC power supply SUP2 is determined based on the characteristics of the voltage command generator Fd(x) described in FIG. 35 , that is, the adjustment characteristics of the DC voltage Vd. . For example, when Pso=Ps1+Ps2 is constant, if the DC voltage adjustment rate is increased, the ratio of power Ps2=Prec supplied from the second AC power supply SUP2 through the diode rectifier REC increases, and is supplied from the first AC power supply SUP1 The power of Ps1 is reduced.

When PL(av) exceeds the set value PLa, the effective power command value Pso ^* =PLa becomes constant. At this time, the difference power Ped=PL(av)-Pso=PL(av)- PLA.

In addition, during the regenerative operation, until the load power PL(av) reaches -PLb, the power command value Pso ^* is given as Pso ^* =k·PL(av). When k=1, Pso ^* =PL(av), since the diode rectifier REC cannot perform power regeneration, it is controlled to regenerate all the effective power of the regenerative load power PL(av) in the AC power supply SUP1.

When the regenerative power PL(av) exceeds the set value -PLb, the effective power command value Pso ^* =-Psb ^* becomes constant. At this time, electric power of Ped=PL(av)-Pso=PL(av)+PLb is regenerated to the energy storage device ESS.

Generally, in electrified railways, the time for powering the load is long and the time for regenerative load is short. In the characteristics of the power command generator Fe(x) in FIG. 36, the upper limit value PLa on the power running side is set larger, and the lower limit value PLb on the regenerative side is set smaller. Thereby, the charging and discharging energy W=Ped×t to the energy storage device ESS becomes 0 on average, and the applied voltage to the electric double layer capacitor EDLC can be Ved≈Ved ^* . When the applied voltage Ved to the EDLC deviates from the command value Ved ^* , as described above, the corrected power command ΔPs ^* acts to gradually control Ved=Ved ^* .

In addition, by setting the proportionality constant k=1, when the load power operates within the range of -PLb<PL(av)<PLa, Pso=PL(av) becomes Pso=PL(av), and power Ped does not need to be supplied from the energy storage device ESS. That is, energy transfer to and from the electric double layer capacitor EDLC occurs only when the above-mentioned set value is exceeded, and since the time is short, it is possible to suppress the capacity of the electric double layer capacitor EDLC to be small. advantage.

Fig. 37 shows the voltage-current vector diagrams of the M-seat and the T-seat during power running of the apparatus of Fig. 34 . The T-seat load current ITL=0, and the M-seat load current IML is delayed by a certain phase θ with respect to the voltage VM.

The load power PL=VM×IML×cosθ is equal to the sum of the power supply Pso=Ps1+Ps2 from the first AC power supply SUP1 and the second AC power supply SUP2 and the power supply Ped from the energy storage device ESS.

The currents IMs and ITs supplied from the first AC power supply SUP1 through the Scott connection transformer TR1 are controlled to be sine waves in phase with the M-seat voltage VM and the T-seat voltage VT respectively, and the input power Ps1 becomes Ps1=IMs×VM+ITs ×VT.

In addition, the compensation currents IMc and ITc supplied from the first and second voltage source self-excited power converters CNV1 and CNV2 are respectively:

IMc=IML-IMs

ITc=ITL-ITs=-ITs

In the M-seat compensation current IMc, it includes the power Ps2=Prec supplied from the second AC power supply SUP2 through the diode rectifier REC and the effective power Ped supplied from the energy storage device ESS, and the supply Ps1=PL-(Ped from the first AC power supply SUP1 +Ps2).

The effective power PMs of the M seat is equal to the effective power PTs of the T seat, half of Ps1=PL-(Ped+Ps2) is supplied from the M seat winding of the Scott connection transformer, and the remaining half is supplied from the T seat winding. The power PTs=Ps1/2 supplied from the T-block winding is regenerated by the second voltage source self-excited power converter CNV2, and supplied to the DC smoothing capacitor Cd. That is, ITc=-ITs.

Furthermore, its power Ps1/2 is supplied to the single-phase AC power feeder Fa through the first voltage-source self-excited power converter CNV1. At this time, including the power Ps2 supplied from the second AC power supply SUP2 through the diode rectifier REC, the active power Ped supplied from the energy storage device ESS, and the reactive power of the load QL=VM×ILM×sinθ, from the above-mentioned first voltage type Self-excited power converter CNV1 is supplied, and only effective power PMs=Ps1/2 is supplied from the M-seat winding of Scott connection transformer TR.

The first and second voltage-type self-excited power converters CNV1 and CNV2 perform power interchange between the M block and the T block, and have the same power capacity. However, when the energy storage device ESS is installed, the power transferred between the energy storage device ESS and the single-phase feeder passes through the first AC output terminal connected to the single-phase AC feeder side (M block). The voltage type self-excited power converter CNV1 accepts.

By setting the output capacity of the first voltage-source self-excited power converter CNV1 to be larger than the output capacity of the second voltage-source self-excited power converter CNV2, the power supplied from the diode rectifier REC can be increased, and The load on the already installed M/G device (frequency converter) is correspondingly relieved. In addition, the power transferred to and from the energy storage device ESS can be increased, the amount of compensation for peak load power can be increased, and the load sharing of the first three-phase AC power supply SUP1 can be reduced. In other words, it is possible to reduce the capacity of the substation equipment.

As described above, in the AC power feeding system of the present invention, the regenerative energy of the electric vehicle load is stored, and by releasing the energy when the load is powered, the energy can be effectively used, and the power supply from the first three-phase AC power supply SUP1 can be cut off. Peak of Ps1. That is, it is possible to realize a single-phase pull-through AC power feeding system, and seek to reduce the capacity of substation equipment and save electricity.

Fig. 38 is a block diagram showing a specific configuration of another example of the compensation current control circuit CONT1 of the device shown in Fig. 34 . In the figure, Fs(x) represents the first power command generator, Ks represents the proportional element, ASW represents the switch circuit, SH represents the level detector, C1~C3 represent the comparator, Gv(S) represents the voltage control compensation circuit, M1 , M2 represent a multiplier, AD1-AD4 represent an adder-subtractor, Gi1(S), Gi2(S) represent a current control compensation circuit, PWM1, PWM2 represent a pulse width modulation control circuit.

The power PL of the single-phase load Load fluctuates at a frequency twice the frequency f1 of the AC feeder line. The detected value of the load power PL is time-averaged to obtain the average value PL(av) of the load power. The first effective power command value Ps1 ^* corresponding to the load power average value PL(av) is generated by the first power command generator Fs(x), and converted into the effective current peak value command Isma ^* through the proportional element Ks, and input to A-side terminal of the switching circuit ASW.

On the other hand, the DC voltage command value Vd ^* is compared with the detected value Vd of the applied voltage on the DC smoothing capacitor Cd by the comparator C1, and the deviation εv=Vd ^* -Vd is calculated by the subsequent voltage control compensation circuit Gv(S). Proportional or integral amplification is performed to generate another peak value command Ismb ^* of the input current, which is input to the b-side terminal of the switch circuit ASW.

The level detector SH inputs the load power PL(av), judges whether it is higher or lower than the set level PLo, and transmits the switching signal LB to the switching circuit ASW.

In the case of PL(av)>PLo, the output signal LB of the level detector SH becomes LB=1, the switch circuit ASW is connected to the a side, and the effective current peak value command Ism ^* =Isma ^* is input to the multiplier M1 , M2. That is, the compensation current is controlled based on the first power command value Ps1 ^* .

At this time, as described above, the compensation currents IMc and ITc are controlled so as to match the respective command values IMc ^* and ITc ^* . As a result, the M-seat winding current IMs and the T-seat winding current ITs of the Scott connection transformer TR1 are controlled. as follows:

IMs=IML-IMc=IML-IMc*=IMs*=Isma*×sinωt

ITs=ITL-ITc=-ITc=ITc*=Isma*×cosωt

That is, the power Ps1 supplied from the first AC power supply SUP1 is controlled so as to match the above-mentioned power command value Ps1 ^* .

In addition, in the case of PL(av)<PLo, the output signal LB of the level detector SH=0, the switch circuit ASW is connected to the b side, and the effective current peak value command Ism ^* =Ismb ^* is input to the multiplier M1, M2. That is, the compensation current is controlled based on the signal Ismb ^* from the DC voltage control circuit.

By setting the voltage command value Vd ^* higher than the no-load rectified voltage Vreco of the diode rectifier REC, when PL(av)<PLo, the power Ps2 supplied from the second AC power supply SUP2 via the diode rectifier REC becomes zero.

On the other hand, the energy storage device ESS is controlled in the same manner as explained in FIG. 24 . That is, the load power PL is detected, its time average value PL(av) is calculated, and input to the second power command generator Fe(x). The second power command generator Fe(x) provides the sum Pso ^* (=Ps1 ^* +Ps2 ^* ).

Also, the voltage Ved applied to the electric double layer capacitor EDLC is detected, and the comparator C4 calculates the deviation εed=Ved ^* −Ved from the command value Ved ^* . By integrating this deviation εed, the compensation power command ΔPs ^* is obtained and input to the adder-subtractor AD6.

In the adder-subtractor AD6, the output signal Pso ^* from the above-mentioned power command generator Fe(x) is added to the above-mentioned compensation power command ΔPs ^* to generate the output signal supplied from the first AC power supply SUP1 and the second AC power supply SUP2. The command Ps ^* of the sum of active powers = Pso ^* +ΔPs ^* .

Furthermore, the difference between the load power detection value PL(av) and the effective power command value Ps ^* is obtained by the following adder-subtractor AD7, and the effective power command value Ped ^* =PL(av) output from the electric double layer capacitor EDLC is generated. -Ps ^* . The divider DV divides the effective power command value Ped ^* by the voltage Ved applied to the EDLC to obtain the command value Ied ^* of the current flowing through the DC reactor Ld.

The deviation between the current command value Ied ^* and the detected value of the current Ied flowing through the above-mentioned DC reactor Ld is generated by the comparator C5 εied=Ied ^* -Ied, and the current control compensation circuit Gi3(S) in the back converts the deviation εied is inverting and amplified, and its output signal e3 ^* is input to the adder-subtractor AD8. In the adder-subtractor AD8, the signal Eed ^* equivalent to the voltage Ved of the electric double layer capacitor EDLC is added to the above-mentioned signal e3 ^* , and the control signal echo ^* is input to the pulse width modulation control circuit PWM3 of the circuit breaker device CHO.

As described above, it is controlled to be Ied=Ied ^* , and the electric power Ped supplied from the energy storage device ESS is controlled to be Ped=Ped ^* =PL(av)-Ps ^* .

When Ved ^* >Ved, since ΔPs ^* increases, Ps ^* also increases, so Ped=Ped ^* decreases or becomes a negative value. As a result, the voltage Ved applied to the electric double layer capacitor EDLC rises and is controlled so that Ved ^* =Ved. Conversely, in the case of Ved ^* <Ved, ΔPs ^* decreases, and is similarly controlled so that Ved ^* =Ved.

As described above, the sum Pso ^* (=Ps1 ^* + Ps2 ^* ), in addition, since the power Ps1=Ps1 ^* supplied from the first AC power supply SUP1 is determined by the control circuit CONT1 of FIG. 38, the power supplied from the second AC power supply SUP2 through the diode rectifier REC automatically becomes Ps2=Pso ^* -Ps1 ^* . That is, regardless of the voltage regulation rate of the diode rectifier REC, the distribution of the power supply Ps1 from the first AC power supply SUP1 and the power supply Ps2=Prec from the second AC power supply SUP2 can be set, and optimal load sharing can be performed. run.

Fig. 39 shows other operating characteristic examples of the device of Fig. 34, for load power PL (av), the first power command generator Fs (x) of Fig. 38 makes:

In the range of 0≤PL(av)≤PLa1, Ps1 ^* =PL(av)

Under PL(av)>PLa1, Ps1 ^* =PLa1=constant

In addition, when PL(av)<0, the DC voltage command is controlled so that Vd ^* =constant.

Furthermore, the second voltage command generator Fe(x) in FIG. 24 makes:

Under PL(av)<-PLb, Pso ^* =PL(av)+PLb

In the range of -PLb≤PL(av)<0, Pso ^* =0

In the range of 0≤PL(av)≤PLa2, Pso ^* =PL(av)

Under PL(av)>PLa2, Pso ^* =PLa2=constant.

As a result, assuming ΔPs ^* =0, the command value Ped ^* =PL(av)-Ps ^* of the power supplied from or regenerated to the energy storage device ESS becomes:

Under PL(av)<-PLb, Ped ^* =-PLb=constant

In the range of -PLb≤PL(av)<0, Ped ^* =PL(av)

In the range of 0≤PL(av)≤PLa2, Ped ^* =0

Under PL(av)>PLa2, Ped ^* =PL(av)-PLa2

In addition, the power Ps1 supplied from the first AC power supply SUP1 or regenerated to the SUP1 becomes:

Under PL(av)<-PLb, Ps1=Pso ^* =PL(av)+PLb

In the range of -PLb≤PL(av)<0, Ps1=Pso ^* =0

In the range of 0≤PL(av)≤PLa1, Ps1=Ps1 ^* =PL(av)

Under PL(av)>PLa1, Ps1=Ps1 ^* =PLa1=constant

Furthermore, the power Prec=Ps2 supplied from the diode rectifier REC becomes:

Under PL(av)<PLa1, Prec=0

In the range of PLA1≤PL(av)≤PLa2, Prec=PL(av)-PLa1

Under PL(av)>PLa2, Prec=Pso ^* -Ps1 ^* =PLa2-PLa1=constant

That is, in the regenerative operation, within the range of -PLb<PL(av)<0, all the regenerative power is stored in the energy storage device ESS, and the regenerative power (PL(av)+PLb) outside the range returns to the first 1 AC power supply SUP1. In this way, energy storage is given priority, and the stored energy is supplied to the next power running load, thereby saving electricity costs. In addition, when the stored energy of the energy storage device ESS increases too much, the above-mentioned compensation power ΔPs ^* acts to keep the stored energy amount substantially constant.

In addition, under power running load, firstly, power Ps1=PL(av) is supplied from the first AC power supply SUP1, and then, power Ps2=Prec=PL(av)-PLa1 is supplied through the diode rectifier REC, and further, if the load increases , then Ped=PL(av)=PLa2 is supplied from the energy storage device ESS.

Thus, the distribution of the power supply Ps1 supplied from the first AC power supply SUP1, the power supply Ps2=Prec from the second AC power supply SUP2, and the power supply from the energy storage device ESS can be freely selected, and the already set Savings in electricity bills due to equipment (M/G units or Scott connection transformers, etc.) and efficient use of energy.

In the device of this embodiment, the power Ps1 supplied via the already installed rotary frequency converter (M/G device) and the Scott transformer TR1 can be reduced, reducing the capacity of conventional equipment and facilitating equipment renewal. In addition, the diode rectifier REC can be used as a low-cost, high-efficiency static 50Hz/60Hz frequency converter compared with the conventional rotary frequency converter (M/G device).

In addition, part or all of the regenerative power can be stored in the energy storage device ESS, and the stored energy can be released when the load of the power running train is high, so that efficient energy utilization can be achieved. Thereby, it is possible to save electricity costs, reduce peak load power when viewed from an AC power source, and reduce substation equipment capacity.

Twelfth Embodiment

Fig. 40 is a block diagram of an AC power feeding system for an electrified railway according to a twelfth embodiment of the present invention. In the figure, SUP1 represents the first AC power supply, SUP2 represents the second AC power supply, TR1 represents the Scott connection transformer, Fa represents the single-phase AC feeder, Load represents the tram load, TRm and TRt represent the single-phase transformer, CNV1 and CNV2 represent Voltage-type self-excited power converter, Cd represents DC smoothing capacitor, Lf, Cf represent the reactor and capacitor constituting the LC filter, TR2 represents the three-phase transformer, CNV3 represents the third voltage-type self-excited power converter (fixed frequency Pulse phase control converter), CONT1 represents the compensation current control unit that controls the compensation current IMc, ITc output from the power converters CNV1, CNV2, and CONT3 represents the control circuit of the third voltage-type self-excited electric converter CNV3.

The compensation current control unit CONT1 is composed of an active power command generation circuit Ps1-ref, a compensation current command generation circuit Ic-ref, compensation current control circuits IMc-Cont, ITc-Cont, and pulse width modulation control circuits PWM1, PWM2.

In addition, the control circuit CONT3 is composed of a DC voltage control circuit Vd-Cont, an effective current control circuit Iq-Cont, and a phase control circuit PHC.

The Scott connection transformer TR1 transforms the three-phase AC voltages Vu, Vv, and Vw of the first three-phase AC power supply SUP1 (three-phase-60Hz) into two-phase AC voltages VM, VT, and the two-phase voltage VM and VT have a 90° angle phase difference. In addition, as a connection method of a transformer for converting a three-phase voltage into a two-phase voltage, there is a modified Woodbridge connection transformer and the like. The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-block terminal, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 is connected to the T-block terminal. In addition, to the second AC power supply SUP2 (three-phase-50Hz), the AC terminal of the fixed pulse phase control converter CNV3 is connected via the three-phase transformer TR2, and the DC output terminal of the converter CNV3 is connected to the above-mentioned DC smoothing capacitor Cd. .

The values of the reactor Lf and the capacitor Cf are determined so that the LC filter (Lf, Cf) connected in parallel to the DC smoothing capacitor Cd resonates at twice the frequency (60 Hz) of the AC feeder line. By absorbing the power fluctuation portion ΔPL of the single-phase load accompanying the fluctuation at twice the frequency (60 Hz) of the AC feeder line by the LC filter, the fluctuation ΔVd of the DC voltage is suppressed. As a result, the capacity of the DC smoothing capacitor Cd can be reduced, and the variation ΔVd of the DC voltage can be significantly reduced. Although there is concern about the electrical oscillation phenomenon caused by the LC filter during a transition such as a sudden load change, since the DC voltage control is performed by the fixed pulse phase control converter CNV3, it acts as a damper to attenuate the electrical oscillation and can provide a stable system. .

In addition, by stabilizing the DC voltage Vd, the compensation current control performed by the first and second voltage-type self-excited power converters CNV1 and CNV2 or the control performed by the third voltage-type self-excited power converter CNV3 are stabilized to improve control performance. In addition, the breakdown voltage range of the power converters CNV1, CNV2, and CNV3 can be reduced by an amount corresponding to the portion where the voltage fluctuation ΔVd is reduced, and an inexpensive device can be provided.

The third voltage type self-excited power converter CNV3 is a fixed pulse phase control converter that operates with a certain pulse pattern (for example, 1 pulse, 3 pulses, 5 pulses, ..., etc.). When the DC voltage Vd is kept constant Below, the amplitude value of the AC side output voltage Vc is constant. That is, by adjusting the phase angle  of the AC-side output voltage Vc (Vcr, Vcs, Vct) of the converter CNV3 with respect to the voltage Vs (Vr, Vs, Vt) of the second AC power supply SUP2, the input current Is (Ir, Is , It).

In the device of this embodiment, through power running/regenerative running, the DC voltage Vd=Vd ^* can be controlled to be almost constant by the third voltage-type self-excited power converter CNV3. The voltage utilization ratios of the first and second self-excited power converters CNV1 and CNV2 are greatly improved.

FIG. 41 shows the specific main circuit configuration and its control circuit of the third voltage-source self-excited power converter CNV3 of the device shown in FIG. 40 . In the figure, SUP2 is a second AC power supply, TR2 is a three-phase transformer, and Cd is a DC smoothing capacitor, which is repeated with FIG. 40 . In addition, CNV3 is a voltage-type self-excited power converter with three-phase bridge connection, which is composed of self-arc-extinguishing elements S1-S6 and high-speed diodes D1-D6. In addition, as the control circuit CONT3, comparators C1 and C5, a voltage control compensation circuit Gv(S), a current control compensation circuit Giq(S), a coordinate converter Z, a synchronous phase signal generator PLL, and a phase control circuit PHC are prepared. The operation of this control circuit will be described below.

Fig. 42 is a diagram showing an AC-side equivalent circuit (one-phase part) for explaining the operation of the voltage-source self-excited power converter (fixed pulse phase control converter) CNV3 of Fig. 41, Vs showing the second AC power supply SUP2 Vc represents the AC side output voltage of the converter CNV3, Ls represents the leakage inductance (or AC reactor) of the transformer TR2, and Is represents the input current.

In addition, Fig. 43 is a voltage-current vector diagram in an equivalent circuit. When the phase angle of the AC voltage Vc of the converter CNV3 is shifted by  with respect to the power supply voltage Vs, a differential voltage Vs is applied to the AC reactor Ls. -Vc=jωLs·Is, the input current Is flows. Assuming that the amplitude value of the power supply voltage Vs is equal to the fundamental wave amplitude value of the AC voltage Vc of the converter CNV3, the phase of the input current Is to the power supply voltage Vs becomes [phi]/2. Therefore, the input power factor becomes cos(/2).

The amplitude value of the fundamental wave of the AC voltage Vc of the converter CNV3 is determined by the DC voltage Vd, because it is based on the request from the load side (in this case, the first and second voltage-type self-excited electrical converters CNV1, CNV2) Since the DC voltage Vd is determined, the transformer TR2 makes the power supply voltage (secondary voltage of the transformer TR2) Vs equal to the amplitude value of the AC voltage Vc of the converter CNV3.

If the phase angle  is increased, the applied voltage on the reactor Ls increases, and the input current Is also increases. When the phase angle  is made negative, the direction of the voltage jωLs·Is applied to the reactor Ls is reversed, the input current becomes a vector of Is', and power can be regenerated in the power supply SUP2. Of course, if the phase angle [phi]=0, the input current Is=0 can be made.

FIG. 44 shows an example of an operation waveform of the fixed-pulse (1-pulse) phase control converter CNV3 during power running, showing a portion of one phase. Converter CNV3 is a voltage-type self-excited power converter with a common three-phase bridge connection. As the R-phase part, the self-arc-extinguishing element S1 of the upper branch and the self-arc-extinguishing element S4 of the lower branch are considered. High-speed diodes D1, D4 connected in antiparallel, respectively.

With respect to the supply voltage Vr, the AC voltage Vcr of the converter CNV3 lags behind by a phase angle [phi]. When the fundamental wave amplitude values of the voltages Vr and Vcr are the same, the input current Ir becomes a current lagging behind the power supply voltage Vr by [phi]/2. Here, in order to simplify the description, the high-frequency component of the current Ir is omitted, and a sine wave current is used.

When the current Ir>0, if the self-arcing element S4 is turned on (the element S1 is turned off), the AC output voltage becomes Vcr=-Vd/2, and the current IS4 flows in the lower branch S4. When the current Ir>0, and ωt=0, if the self-arcing element S4 is turned off and the element S1 is turned on, the AC output voltage Vcr=+Vd/2 becomes IS4=0. The current Ir flows through the high-speed diode D1 of the upper branch until the current Ir reverses and flows through ID1=Ir. If current Ir<0, since element S1 is turned on, current IS1 flows through element S1. At the phase angle ωt=π, if the self-arcing element S1 is turned off and the element S4 is turned on, the voltage Vcr becomes Vcr=-Vd/2 again, and becomes IS1=0, and the current ID4 flows in the high-speed diode D4 of the lower branch. .

The cut-off current Imax of the self-arc-extinguishing elements S1 and S4, if the peak value of the input current Ir is recorded as Im, becomes Imax=Im×sin(/2), for example, if the control phase angle =20°, then Imax =0.174×Im. That is, the switching (on/off operation) of the self-arc-extinguishing elements S1 and S4 is performed near the zero crossing of the input current Ir, and the maximum cut-off current Imax of the self-arc-extinguishing elements S1 and S4 can be suppressed as sufficiently small. As a result, an element with a small current interruption capacity can be used, and an economical inverter can be provided. In addition, the switching loss can be reduced, and the capacity of cooling equipment can be reduced. Furthermore, the AC voltage Vcr of the converter CNV3 becomes a rectangular wave voltage, and the peak value Vcm of its fundamental wave component becomes Vcm=(4/π)×(Vd/2)=1.273×(Vd/2), and it is possible to obtain a voltage equal to or greater than DC The value of the voltage (Vd/2). That is, compared with a normal PWM control converter, it has a higher voltage utilization rate, and when it is configured with a self-arc-extinguishing element of the same withstand voltage, it has an advantage that a larger output can be generated.

Fig. 45 is a diagram showing the operation waveform of the fixed-pulse (1-pulse) phase control converter CNV3 in the regenerative operation, showing a part of one phase. With respect to the supply voltage Vr, the AC voltage Vcr of the converter CNV3 leads by a phase angle [phi]. The phase of the input current Ir is reversed, and the phase lags behind the power supply voltage Vr by π-/2. The input power Pr=Vr×Ir is negative, and power can be regenerated in the power supply SUP2. The input power factor at this time becomes cos(π-/2)=cos(/2), and it is also possible to operate with a high power factor.

When ωt=0, when the element S1 is turned on (S4 is turned off), Vcr=+Vd/2, Ir>0, and therefore a current ID1=Ir flows through the high-speed diode D1. When the phase of the current Ir is reversed, a current starts to flow through the self-arcing element S1, and IS1=Ir flows. When ωt=π, when the element S1 is turned off and the element S4 is turned on, the current ID4 flows through the high-speed diode D4, and the current Ir is reversed, and IS4=Ir starts to flow through the element S4.

In regenerative operation, most of the input current Ir flows through the self-arc-extinguishing element S1 or S4, and at this time, since the self-arc-extinguishing element S1, S4 is switched (on/off) near the zero crossing of the current Ir Therefore, the maximum cut-off current Imax of the elements S1 and S4 is Imax=Im×sin(/2) with respect to the peak value Im of the current Ir. For example, if =20°, it becomes Imax=0.174×Im.

In the above description, the number of control pulses of the converter CNV3 is set to 1 pulse for the sake of simplicity of description, but it is also possible to control the number of control pulses to 3 pulses, 5 pulses, . . . and so on. In order to reduce the higher harmonics of the input current Ir, it is effective to increase the number of control pulses, but in this case, the switching (on/off operation) can be performed near the zero crossing of the input current Ir, and the automatic The maximum cut-off current Imax of the arc suppression element is suppressed to be very small. Also, even when the number of pulses is increased, the voltage utilization factor of converter CNV3 can secure a value close to the voltage utilization factor during the one-pulse operation described above.

In FIG. 41, the voltage Vd applied to the DC smoothing capacitor Cd is controlled by the fixed pulse phase control converter CNV3 as follows. The description will be made assuming that the DC voltage command value Vd ^* =constant.

The comparator C1 compares the DC voltage command value Vd ^* with the DC voltage detection value Vd, and inputs the deviation εv to the voltage control compensation circuit Gv(S), and obtains the supply from the second AC power supply SUP2 through proportional or integral amplification. The effective current command value Iq ^* . In addition, the three-phase input currents Ir, IS, It from the second AC power supply SUP2 are detected, and three-phase/dq conversion is performed on them by the coordinate converter, and they are divided into effective current Iq and inactive current Id. The effective current command value Iq ^* is compared with the effective current detection value Iq by the comparator C5, and its deviation εq is input to the subsequent current control compensation circuit Giq(S), and the control phase signal  ^* is calculated by proportional or integral amplification .

In the phase control circuit PHC, the phase reference signals θr, θs, and θt synchronized with the three-phase voltages Vr, Vs, and Vt of the second AC power supply SUP2 are input, compared with the control phase signal  ^* , and the selection communication of the converter CNV3 is generated. Number. The AC voltages Vcr, Vcs, Vct of the converter are shifted by the phase angle = ^* with respect to the power supply voltages Vr, Vs, Vt, and the effective component Iq of the input currents Ir, Is, It can be controlled.

In the case of Iq ^* >Iq, the control phase signal  ^* (lag) increases to increase the effective component Iq of the input currents Ir, Is, It. Conversely, in the case of Iq ^* <Iq, the control phase signal [phi] ^* (lag) becomes a negative value, reducing the effective component Iq of the input currents Ir, Is, and It. Therefore, it is controlled so that Iq ^* =Iq.

In the case of Vd ^* >Vd, the deviation εv becomes a positive value, which is amplified by the voltage control compensation circuit Gv(S) to increase the effective current command value Iq ^* . As a result, the power Ps2 is supplied from the second AC power supply SUP2, the voltage Vd applied to the DC smoothing capacitor Cd is increased, and it is controlled to Vd ^* =Vd. Conversely, when Vd ^* <Vd, the deviation εv becomes a negative value, and the effective current command value Iq ^* becomes a negative value. As a result, the regenerative power Ps2 to the second AC power supply SUP2 reduces the voltage Vd applied to the DC smoothing capacitor Cd, and is also controlled to be Vd ^* =Vd.

FIG. 46 is another block diagram of the control circuit CONT3 of the third voltage-source self-excited power converter CNV3 in the apparatus of FIG. 40 . In the figure, C1 represents a comparator, Gv(S) represents a voltage control compensation circuit, and PHC represents a phase control circuit.

The voltage Vd applied to the DC smoothing capacitor Cd is detected. The voltage command value Vd ^* is compared with the voltage detection value Vd by the comparator C1, and the deviation εv=Vd ^* -Vd is obtained. The following voltage control compensation circuit Gv(S) amplifies the deviation εv proportionally or integrally, and inputs it to the phase control circuit PHC as a phase control command  ^* . That is, the phase control signal [phi] ^* is directly transmitted from the DC voltage control circuit Gv(S) to the phase control circuit PHC.

In the case of Vd ^* >Vd, the deviation εv is positive, and the control phase angle command  ^* is increased. This control phase angle command  ^* determines the delayed phase angle  of the AC voltage Vc of the converter CNV3 with respect to the voltage Vs of the second AC power supply SUP2. As described in FIG. 43, by increasing  ^* =, the input The current Is increases. As a result, the effective power Ps2 supplied from the power supply SUP2 increases to increase the voltage Vd applied to the DC smoothing capacitor Cd, and is controlled so that Vd ^* =Vd. Conversely, in the case of Vd ^* <Vd, the deviation εv is negative, and the control phase angle command  ^* decreases or takes a negative value (leading phase). If <0, in the vector diagram of FIG. 22 , the DC voltage of converter CNV3 becomes Vc', the direction of the vector of input current Is is reversed, and active power Ps2 is reproduced in AC power supply SUP2. As a result, the voltage Vd applied to the DC smoothing capacitor Cd decreases, and is also controlled so that Vd ^* =Vd. As described above, the input current control circuit (partial loop) can be omitted, and the control circuit can be simplified.

FIG. 47 shows a specific configuration of compensation current control unit CONT1 for controlling compensation currents IMc, ITc outputted from the first and second voltage source self-excited power converters CNV1, CNV2 of the apparatus shown in FIG. 40 . In the figure, Fs(x) represents the power command generator, Ks represents the proportional element, C2 and C3 represent the comparator, M1 and M2 represent the multiplier, AD1~AD4 represent the adder and subtractor, Gi1(S), Gi2(S) Represents the current control compensation circuit, PWM1, PWM2 represent the pulse width modulation control circuit.

The power Ps1 supplied from the first AC power supply SUP1 (three-phase-60 Hz) is controlled as follows. The power command generator Fs(x) provides the command value Ps1 ^* of the power Ps1 supplied from the first AC power supply SUP1 according to the time average value PL(av) of the load power PL, and multiplies it by the proportional constant Ks as a Scott connection The peak value command Ism ^* of the M-seat and T-seat winding currents IMs and ITs of the transformer TR1.

The multiplier M1 multiplies the input current peak command Ism ^* with the unit sine wave sinωt synchronized with the M-seat voltage VM of the Scott connection transformer TR1, and outputs the M-seat input current command IMs ^* =Ism ^* ×sinωt. The multiplier M2 multiplies the input current wave peak command Ism ^* with the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR1, and outputs the T-seat input current command ITs ^* =Ism ^* ×cosωt.

The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* . Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to obtain the T-seat compensation current command value ITc ^* =ITL-ITs ^* . Among them, ITL=0.

The comparator C2 compares the M-seat compensation current detection value IMc with the compensation current command value IMc ^* , and the subsequent current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction method device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width modulation of the converter CNV1 Control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* . The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows. Of course, when the leakage inductance of the single-phase transformer TRm is small, a reactor Lsmo may be inserted in series with the primary or secondary winding of the transformer TRm.

When IMc ^* >IMc, the deviation εm is positive, the signal em ^* is controlled to increase, and the compensation current IMc is increased so that IMc ^* =IMc. On the contrary, when IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also made. As a result, the M-seat input current IMs supplied from the Scott connection transformer TR1 is controlled so that IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to the accumulator Subtractor AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* . The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows.

When ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, when ITc ^* <ITc, the deviation εt is negative, and the control is performed to reduce the signal et ^* , reduce the compensation current ITc, and also make ITc ^* =ITc. As a result, the T-seat input current ITs supplied from the Scott connection transformer TR1 is controlled so that ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=ITs ^* . This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. However, the T-seat load current becomes ITL=0. The currents IMs and ITs of the M-seat and T-seat of the Scott connection transformer TR1 have the same amplitude value Ism ^* , and the two-phase balanced currents whose phases are staggered by 90°. As a result, the current supplied from the first three-phase AC power supply SUP1 also becomes a three-phase balanced sinusoidal current with a power factor=1. In this way, not only the capacity of the Scott connection transformer TR1 can be reduced, but also the capacity of the equipment of the first AC power supply SUP1 or the M/G device can be reduced.

FIG. 48 shows an example of the characteristics of the power command generator Fs(x) of the control circuit CONT1 of FIG. 47, and the power command value Ps1 ^* from the first AC power supply SUP1 is given to the load power PL(av) as follows. That is, when the set value PLo is adopted, within the range of -PLo<PL(av)<+PLo, Ps1 ^* =PL(av), and the entire load power PL is supplied or regenerated from the first AC power supply SUP1. In addition, under PL(av)<-PLo, Ps1 ^* =-PLo= is kept constant, and under PL(av)>+PLo, Ps1 ^* =+PLo= is kept constant. That is, during regenerative operation, in the case of PL(av)<-PLo, the power regenerated in the first AC power supply SUP1 becomes Ps1=-PLo=constant, and the regenerative power outside this range (PL(av)-PLo ) is regenerated in the second AC power supply SUP2 through the fixed pulse phase control converter CNV3.

In addition, in power running, when PL(av)>+PLo, the power supplied from the first AC power supply SUP1 becomes Ps1=+PLo=constant, and the power supplied outside this range (PL(av)-PLo) It is supplied from the second AC power supply SUP2 via the fixed pulse phase control converter CNV3.

Therefore, in this embodiment, the capacity of power supply or regenerative power to electric vehicle loads can be increased without increasing the capacity of existing equipment (M/G device or Scott connection transformer, etc.), and economical power supply can be provided. AC feed system for electrified railway.

Fig. 49 is a voltage-current vector diagram of the M seat and the T seat during power running of the device of Fig. 40 . The T-seat load current ITL=0, and the M-seat load current IML lags behind the voltage VM by a certain phase θ. The load power is PL=VM×IML×cosθ, which is equal to the sum Pso=Ps1+Ps2 of the power supply Ps1 from the first AC power supply SUP1 and the power supply Ps2 from the second AC power supply SUP2.

The current IMs and ITs supplied from the first AC power supply SUP1 through the stud connection transformer TR1 are controlled to be sine waves in phase with the M-seat voltage VM and the T-seat voltage VT respectively, and the input power Ps1 becomes Ps1=IMs×VM+ITs ×VT. In addition, the compensation currents IMc and ITc supplied from the first and second voltage source self-excited power converters CNV1 and CNV2 are respectively:

IMc=IML-Ims

ITc=ITL-ITs=-ITs

In the M-seat compensation current IMc, including the active power Ps2 supplied from the second AC power supply SUP2 through the fixed pulse phase control converter CNV3 and the reactive power QL of the load, the active power supplied from the first AC power supply SUP1 is Ps1=PL- PS2.

The effective power PMs=IMs×VM of the M seat is equal to the effective power PTs=ITs×VT of the T seat, half of the effective power Ps1 is supplied from the M seat winding of the Scott connection transformer TR1, and the remaining half is supplied from the T seat winding . The power PTs=Ps1/2 supplied from the T-block winding is regenerated by the second voltage source self-excited power converter CNV2 and supplied to the DC smoothing capacitor Cd. That is, Tc=-ITs. Furthermore, its power Ps1/2 is supplied to the single-phase AC power feeder Fa via the first voltage-source self-excited power converter CNV1. At this time, including the power Ps2 supplied from the second AC power supply SUP2 through the fixed pulse phase control converter CNV3 and the reactive power of the load QL=VM×ILM×sinθ, it is supplied from the first voltage-type self-excited power converter CNV1 , Only the effective power PMs=IMs×VM=Ps1/2 is supplied from the M-seat winding of the Scott connection transformer TR1.

By setting the output capacity of the first voltage-type self-excited power converter CNV1 to be larger than the output capacity of the second voltage-type self-excited power converter CNV2, the power Ps2 supplied from the second AC power supply SUP2 can be increased, and the power Ps2 can be increased. Accordingly, the power Ps1 supplied from the first AC power supply SUP1 is reduced, in other words, the capacity of existing M/G devices (frequency converters) or Scott connection transformers can be reduced.

thirteenth embodiment

Fig. 50 is a block diagram of an AC power feeding system for an electrified railway according to a thirteenth embodiment of the present invention. In the figure, SUP1 represents the first AC power source, SUP2 represents the second AC power source, TR1 represents the Scott connection transformer, Fa represents the single-phase AC feeder, Load represents the tram load, TRm, TRt represent the single-phase transformer, SWt, SWm, SW1 and SW2 represent switches, CNV1 and CNV2 represent voltage-type self-excited power converters, Cd represents DC smoothing capacitors, Lf and Cf represent reactors and capacitors constituting LC filters, TR2 represents three-phase transformers, and CNV3 represents fixed pulse phases Controlling the converter, CONT1 represents the compensation current control unit that controls the compensation current IMc, ITc output by the electrical converters CNV1, CNV2, and CONT3 represents the control circuit of the fixed pulse phase control converter CNV3.

The Scott connection transformer TR1 transforms the three-phase AC voltages Vu, Vv, and Vw of the first AC power supply SUP1 (three-phase-60Hz) into two-phase AC voltages VM, VT, and the two-phase voltage VM and VT have a phase of 90° Difference. The M-seat winding of the Scott connection transformer TR1 is connected to the AC feeder Fa through the switch SWm. In addition, the single-phase output terminal of the first voltage-type self-excited electric converter CNV1 is connected to the AC feeder line Fa via the single-phase transformer TRm and the switch SW1. Furthermore, the single-phase output terminal of the second voltage-source self-excited power converter CNV2 is connected to the T-block terminal of the Scott connection transformer TR1 via the single-phase transformer TRt and the switch SWt. In normal operation, the switches SWt, SWm, and SW1 are turned on, and the switch SW2 is turned off.

Also, to the second AC power supply SUP2 (three-phase -50 Hz), the AC terminal of the fixed pulse phase control converter CNV3 is connected via the three-phase transformer TR2, and the DC output terminal of the converter CNV3 is connected to the above-mentioned DC smoothing capacitor Cd. The LC filter (Lf, Cf) connected in parallel with the DC smoothing capacitor Cd determines the values of the reactor Lf and the capacitor Cf so as to resonate at twice the frequency (60 Hz) of the AC feeder line. By absorbing the power fluctuation portion ΔPL of the single-phase load accompanying the fluctuation at twice the frequency (60 Hz) of the AC feeder line by the LC filter, the fluctuation ΔVd of the DC voltage is suppressed. As a result, the capacity of the DC smoothing capacitor Cd can be reduced, and the fluctuation ΔVd of the DC voltage can be significantly reduced. Although there is concern about electrical oscillations caused by the LC filter during transitions such as sudden load changes, since the DC voltage control is performed by the fixed pulse phase control converter CNV3, it acts as a damper to attenuate electrical oscillations, providing a stable system. .

Converter CNV3 is a voltage-type self-excited power converter operating with a certain pulse pattern (for example, 1 pulse, 3 pulses, 5 pulses, etc.). When the DC voltage Vd is kept constant, the AC side output voltage Vc The amplitude value of becomes constant. Therefore, by adjusting the phase angle  of the AC-side output voltage Vc (Vcr, Vcs, Vct) of the converter CNV3 with respect to the voltage Vs (Vr, Vs, Vt) of the second AC power supply SUP2, the input current Is (Ir, Is , It). The operation is omitted since it has already been described in FIGS. 41 to 45 .

FIG. 51 is a diagram showing the compensation current control unit CONT1 of the device shown in FIG. 50, and FIG. 52 is a diagram showing a specific configuration of the control circuit CONT3 of the converter CNV3. In the figure, ASW1~ASW4 represent the signal switcher, Fs(x) represents the power command generator, Ks represents the proportional element, C1~C3, C6 represent the comparator, Gv(S) represents the voltage control compensation circuit, M1, M2 represent the multiplication AD1~AD4 represent adder and subtractor, Gi1(S), Gi2(S), Giq(S) represent current control compensation circuit, PWM1, PWM2 represent pulse width modulation control circuit, PHC represent phase control circuit.

The voltage Vd applied to the DC smoothing capacitor Cd is controlled by the fixed pulse phase control converter CNV3 as follows. The description will be made assuming that the DC voltage command value Vd ^* =constant. Comparator C1 compares the DC voltage command value Vd ^* with the DC voltage detection value Vd, and inputs its offset εv to the voltage control compensation circuit Gv(S), and obtains the output from the second AC power supply SUP2 through proportional or integral amplification. The supplied effective current command value Iq ^* . In addition, three-phase input currents Ir, Is, It from the second AC power supply SUP2 are detected, and are divided into effective current Iq and reactive current Id by performing coordinate transformation (three-phase/dq transformation) on them.

The effective current command value Iq ^* is compared with the effective current detection value Iq by the comparator C6, and its deviation εq is input to the subsequent current control compensation circuit Giq(S), and the control phase signal  ^* is obtained through proportional or integral amplification .

In the phase control circuit PHC, the phase reference signals θr, θs, and θt synchronized with the three-phase voltages Vr, Vs, and Vt of the second AC power supply SUP2 are input, compared with the control phase signal  ^* , and the selection communication of the converter CNV3 is generated. Number. The AC voltages Vcr, Vcs, Vct of the converter are shifted only by the control phase angle  ^* with respect to the power supply voltages Vr, Vs, Vt, and the effective components Iq of the input currents Ir, Is, It are controlled. In the case of Iq ^* >Iq, the control phase signal  ^* (lag) increases to increase the effective component Iq of the input currents Ir, Is, It. Conversely, in the case of Iq ^* <Iq, the control phase signal [phi] ^* (lag) becomes a negative value, reducing the effective component Iq of the input signals Ir, Is, It. Therefore, it is controlled so that Iq ^* =Iq. In the case of Vd ^* >Vd, the deviation εv becomes a positive value, which is amplified by the voltage control compensation circuit Gv(S) to increase the effective current command value Iq ^* . As a result, the power Ps2 is supplied from the second AC power supply SUP2, the voltage Vd applied to the DC smoothing capacitor Cd is increased, and it is controlled to Vd ^* =Vd. Conversely, in the case of Vd ^* <Vd, the deviation εv becomes a negative value, and the effective current command value Iq ^* becomes a negative value. As a result, the regenerative power Ps2 to the second AC power supply SUP2 reduces the voltage Vd applied to the DC smoothing capacitor Cd, and is also controlled to be Vd ^* =Vd.

On the other hand, when all the signal switchers ASW1 to ASW4 are connected to the b side, the power Ps1 supplied from the first AC power supply SUP1 (three-phase-60 Hz) is controlled as follows.

The power command generator Fs(x) provides the command value Ps1 ^* of the power Ps1 supplied from the first AC power supply SUP1 according to the time average value PL(av) of the load power PL, and multiplies it by the proportional constant Ks as a Scott connection The peak value command Ism ^* of the M-seat and T-seat winding currents IMs and ITs of the transformer TR1.

The multiplier M1 multiplies the input current peak command Ism ^* with the unit sine wave sinωt synchronized with the M-seat voltage VM of the Scott connection transformer TR1, and outputs the M-seat input current command IMs ^* =Ism ^* ×sinωt. The multiplier M2 multiplies the input current wave peak command Ism ^* with the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR1, and outputs the T-seat input current command ITs ^* =Ism ^* ×cosωt.

The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* . Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to obtain the T-seat compensation current command value ITc ^* =ITL-ITs ^* . Among them, ITL=0. The comparator C2 compares the M-seat compensation current detection value IMc with the compensation current command value IMc ^* , and the subsequent current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction method device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the above-mentioned current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width of the converter CNV1 Modulation control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows. Of course, when the leakage inductance of the single-phase transformer TRm is small, an AC reactor Lsmo may be inserted in series with the primary or secondary winding of the transformer TRm. In the case of IMc ^* >IMc, the deviation εm is positive, and control is performed to increase the signal em ^* to increase the compensation current IMc so that IMc ^* =IMc. Conversely, in the case of IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also set.

As a result, the M-seat input current IMs supplied from the Scott connection transformer TR1 is controlled to be:

IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to the accumulator Subtractor AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the above-mentioned current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width of the converter CNV2 Modulation control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows. In the case of ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, in the case of ITc ^* <ITc, the deviation εt is negative, the signal et ^* is controlled to decrease, the compensation current ITc is reduced, and ITc ^* =ITc is also set.

As a result, the T-seat input current ITs supplied from the Scott connection transformer TR1 is controlled to be:

ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=ITs ^* .

This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. However, the T-seat load current becomes ITL=0.

The currents IMs and ITs of the M-seat and T-seat of the Scott connection transformer TR1 have the same amplitude value Ism ^* , and the two-phase balanced currents whose phases are staggered by 90°. As a result, the current supplied from the first three-phase AC power supply SUP1 also becomes a three-phase balanced sinusoidal current with a power factor=1. As a result, not only the capacity of the Scott connection transformer TR1 but also the capacity of the equipment of the first AC power supply SUP1 or the M/G device can be reduced.

Consider the case where the first AC power supply SUP1 (three-phase -60 Hz) fails for some reason and cannot feed power in the apparatus of FIG. 50 . In this case, the operation of the device is temporarily stopped, and the switches SWm, SWt connected to the M-seat winding and the T-seat winding of the Scott connection transformer TR1 are released. Then, the switch SW2 is turned on to connect the single-phase output terminal of the second voltage-source self-excited power converter CNV2 to the single-phase AC feeder line Fa. That is, the first voltage-type self-excited power converter CNV1 and the second voltage-type self-excited power converter CNV2 are connected in parallel to the single-phase AC feeder line Fa.

Here, the signal switchers ASW1-ASW4 are all switched to the a side by the control circuit shown in FIG. 51, and the operation starts again. By connecting the signal switcher ASW1 to the a side, the command value Ps1 ^* of the active power fed from the first AC power supply SUP1=0. Therefore, IMs ^* =0, ITs ^* =0. Also, by connecting the signal switchers ASW2 and ASW3 to the a side, 1/2 of the load current IML is input to the adder-subtractors AD1 and AD3, respectively. Therefore, the compensation current command values IMc* and ITc ^* supplied from the first and second voltage-source self-excited power converters CNV1 and CNV2 are respectively expressed as ^{IMc* =} IML/2-IMs ^* =IML/2 and ITc ^* =IML /2-ITs ^* = provided by IML/2. Furthermore, by connecting the signal switcher ASW4 to the a side, the compensation signal EM ^* proportional to the M-seat voltage VM (the voltage of the single-phase AC feed Fa) is input to the adder-subtractor AD4. Accordingly, the first and second voltage source self-excited power converters CNV1 and CNV2 generate voltage VM proportional to the compensation signal EM ^* , and are controlled to be 1/2 of the shared load current IML.

In this way, according to the present embodiment, even if the power feeding from one substation is stopped, when the voltage VM of the single-phase AC power feeder Fa is established by the power feeding from the adjacent substation, by using the voltage VM based on the voltage VM The compensation signal EM ^* can realize a single-phase pull-through AC feed system. Also, the third voltage-type self-excited power converter (converter with fixed pulse phase control) CNV3 controls so that the voltage Vd applied to the DC smoothing capacitor Cd matches the command value Vd ^* =constantly. That is, the entire load power PL is supplied from the second AC power supply SUP2. At this time, in the single-phase AC feeder line Fa, since power is supplied through the first and second voltage-type self-excited power converters CNV1 and CNV2, the second voltage-type self-excited power converter CNV2 can be effectively utilized. , the load on the first voltage source self-excited power converter CNV1 can be reduced.

14th embodiment

Fig. 53 is a block diagram of an AC power feeding system for an electrified railway according to a fourteenth embodiment of the present invention. In the figure, SUP1 indicates the first AC power supply, SUP2 indicates the second AC power supply, TR1 indicates the Scott connection transformer, Fa indicates the single-phase AC feeder, Load indicates the Shinkansen tram load, TRm, TRt indicate the single-phase transformer, CNV1, CNV2 means voltage-type self-excited power converter, Cd means DC smoothing capacitor, Lf, Cf means reactor and capacitor forming LC filter, TR2 means three-phase transformer, HB-CNV means hybrid converter, CONT1 means control slave power The compensation current control unit of the compensation current IMc, ITc output by the converters CNV1, CNV2, CONT3 represents the control circuit of the hybrid converter HB-CNV.

The compensation current control unit CONT1 is composed of an active power command generation circuit Ps1-ref, a compensation current command generation circuit Ic-ref, compensation current control circuits IMc-Cont, ITc-Cont, and pulse width modulation control circuits PWM1, PWM2. In addition, the control circuit CONT3 is composed of a DC voltage control circuit Vd-Cont, an effective current control circuit Iq-Cont, and a phase control circuit PHC.

The Scott connection transformer TR1 transforms the three-phase AC voltages Vu, Vv, and Vw of the first three-phase AC power supply SUP1 (three-phase-60Hz) into two-phase AC voltages VM, VT, and the two-phase voltage VM and VT have a 90° angle phase difference.

The M block output is connected to the single-phase AC feeder Fa, and the T block becomes an open circuit (no load). The single-phase output terminal of the first voltage-type self-excited power converter CNV1 is connected to the M-seat terminal of the Scott connection transformer TR1 through the single-phase transformer TRm, and the single-phase output terminal of the second voltage-type self-excited power converter CNV2 Connect to the T-block terminal of the Scott connection transformer TR1 through the single-phase transformer TRt. In addition, to the second AC power supply SUP2 (three-phase-50Hz), the AC terminal of the hybrid converter HB-CNV is connected via the three-phase transformer TR2, and the DC output terminal of the hybrid converter HB-CNV is connected to the above-mentioned DC smoothing capacitor Cd.

The values of the reactor Lf and the capacitor Cf are determined so that the LC filter (Lf, Cf) connected in parallel to the DC smoothing capacitor Cd resonates at twice the frequency (60 Hz) of the AC feeder line. By absorbing the power fluctuation portion ΔPL of the single-phase load accompanying the fluctuation at twice the frequency (60 Hz) of the AC feeder line by the LC filter, the fluctuation ΔVd of the DC voltage is suppressed.

As a result, in the present embodiment, the capacity of the DC smoothing capacitor Cd can be reduced, and furthermore, the fluctuation ΔVd of the DC voltage can be significantly reduced. Although there is concern about the electrical oscillation phenomenon caused by the LC filter during a transition such as a sudden load change, since the DC voltage control is performed by the fixed pulse phase control converter CNV3, it acts as a damper to attenuate the electrical oscillation and can provide a stable system. . In addition, by stabilizing the DC voltage Vd, the compensation current control by the first and second voltage-source self-excited power converters CNV1 and CNV2 or the control by the hybrid converter is also stabilized, thereby improving control performance. In addition, the withstand voltage range of the power converters CNV1 and CNV2 or the hybrid converter HB-CNV can be reduced by an amount corresponding to the reduction of the voltage fluctuation ΔVd, and a cheaper device can be provided.

The hybrid converter HB-CNV is a component that combines the power diode rectifier REC and the voltage-type self-excited power converter CNV. By adjusting a certain pulse pattern (such as 1 pulse, 3 pulses, 5 pulses, ... ) operation of the converter HB-CNV AC side output voltage Vc (Vcr, Vcs, Vct) with respect to the phase angle  of the voltage Vs (Vr, Vs, Vt) of the second AC power supply SUP2, control the input current Is (Ir, Vt) The effective component Iq of Is, It) controls the voltage Vd applied to the DC smoothing capacitor Cd as a result.

In the device of this embodiment, the DC voltage Vd=Vd ^* can be controlled to be almost constant by the hybrid converter HB-CNV through power running and regenerative running. Voltage utilization of self-excited power converters CNV1 and CNV2.

FIG. 54 is a diagram showing a specific main circuit configuration of the hybrid converter HB-CNV of the device shown in FIG. 53 and its control circuit CONT3. The drawing of the second AC power supply SUP2, the three-phase transformer TR2, and the DC smoothing capacitor Cd overlaps with that in FIG. 53 . In the figure, REC is a diode rectifier for power, and is composed of diodes PD1 to PD6 for power. In addition, CNV is a voltage-type self-excited power converter, which is composed of self-arc-extinguishing elements S1-S6 and high-speed diodes D1-D6. La is an AC reactor, and plays a role of suppressing a recovery current flowing through power diodes PD1 to PD6 accompanying switching operations of self-arc extinguishing elements S1 to S6 . In addition, as the control circuit CONT3, comparators C1 and C6, a voltage control compensation circuit Gv(S), a current control compensation circuit Giq(S), a coordinate converter Z, a synchronous phase signal generator PLL, and a phase control circuit PHC are prepared.

The voltage Vd applied to the DC smoothing capacitor Cd is controlled by the hybrid converter HB-CNV as follows. The description will be made assuming that the DC voltage command value Vd ^* =constant. The comparator C1 compares the DC voltage command value Vd ^* with the DC voltage detection value Vd, and inputs the deviation εv to the voltage control compensation circuit Gv(S), and obtains the supply from the second AC power supply SUP2 through proportional or integral amplification. The effective current command value Iq ^* . In addition, three-phase input currents Ir, Is, It from the second AC power supply SUP2 are detected, and are divided into effective current Iq and reactive current Id by performing coordinate conversion (three-phase/dq conversion) on them. The effective current command value Iq ^* is compared with the effective current detection value Iq by the comparator C6, and its deviation εq is input to the subsequent current control compensation circuit Giq(S), and the control phase signal  ^* is obtained through proportional or integral amplification .

In the phase control circuit PHC, the phase reference signals θr, θs, θt synchronized with the three-phase voltages Vr, Vs, Vt of the second AC power supply SUP2 are input and compared with the control phase signal  ^* to generate the hybrid converter HB-CNV the strobe signal. The AC voltages Vcr, Vcs, Vct of the hybrid converter HB-CNV are shifted by the phase angle = ^* with respect to the power supply voltages Vr, Vs, Vt as described above, and the effective components Iq of the input currents Ir, Is, It can be controlled. In the case of Iq ^* >Iq, the control phase signal  ^* (lag) increases to increase the effective component Iq of the input currents Ir, Is, It. Conversely, in the case of Iq ^* <Iq, the control phase signal [phi] ^* (lag) becomes a negative value, and the effective component Iq of the input signals Ir, Is, and IT is reduced. Therefore, it is controlled so that Iq ^* =Iq. Also, when Vd ^* >Vd, the deviation εv becomes a positive value, which is amplified by the voltage control compensation circuit Gv(S) to increase the effective current command value Iq ^* . Thereby, the power Ps2 is supplied from the second AC power supply SUP2, the voltage Vd applied to the DC smoothing capacitor Cd is increased, and it is controlled so that Vd ^* =Vd. Conversely, when Vd ^* <Vd, the deviation εv becomes a negative value, and the effective current command value Iq ^* becomes a negative value. As a result, the regenerative power Ps2 to the second AC power supply SUP2 reduces the voltage Vd applied to the DC smoothing capacitor Cd, and is also controlled to be Vd ^* =Vd.

Hereinafter, the operation of the hybrid converter HB-CNV in the apparatus of this embodiment will be described. FIG. 55 is a diagram showing a main circuit configuration of a one-phase portion for explaining the operation of the hybrid converter HB-CNV. In the figure, PD1 and PD4 denote power diodes, S1 and S4 denote self-arc suppressors, D1 and D4 denote high-speed diodes, La denote AC reactors, and Vd denote DC applied voltage on DC smoothing capacitor Cd. The input current Is is an alternating current, which is described assuming that it flows in the direction of the arrow in the figure.

Before the mode (i), the current Is flows through the power diode PD1. In the mode (i), when the arc-extinguishing element S4 is turned on, the input current Is flows through the arc-extinguishing element S4 through the AC reactor La. At this time, a recovery current flows from the DC voltage Vd to the power diode PD1 along the path of PD1→AC reactor La→self arc-extinguishing element S4. What suppresses this recovery current (di/dt) is the AC reactor La. Without the AC reactor La, an excessive recovery current flows through the power diode PD1 or the self-arc-extinguishing element S4, resulting in increased loss or destruction of the element.

Next, in the mode (ii), when the self-arcing element S4 is turned off and S1 is turned on, the current Ia of the AC reactor La does not immediately become 0, but flows through the high-speed diode D1. In the following mode (iii), the current of the high-speed diode D1 gradually decays, and the current moves to the power diode PD1. That is, in general, since the forward voltage drop VFb of the high-speed diode D1 is larger than the forward voltage drop VFa of the power diode PD1, the current of the AC reactor La attenuates according to the difference voltage (Vb-Va), and the input The current Is flows from the high-speed diode D to the power diode PD1. Finally, as in the mode (iv), the input current Is flows through the power diode PD1. The AC reactor La also functions to suppress the recovery current of the power diode PD4 in the lower arm.

FIG. 56 shows an example of an operation waveform of the hybrid converter HB-CNV during power running, and shows a one-phase portion (R-phase). The hybrid converter HB-CNV is a component that combines the power diode rectifier REC of the three-phase bridge connection and the voltage-type self-excited power converter CNV. As the R-phase part, the power diodes PD1 and PD4 are considered to be self-extinguishing. Elements S1, S4 and high-speed diodes D1, D4 are connected in antiparallel to each of the self-arcing elements S1, S4.

The AC voltage Vcr of the converter HB-CNV lags the supply voltage Vr by a phase angle [phi]. When the fundamental wave amplitude values of the voltage Vr and Vcr are the same, the input current Ir is delayed from the power supply voltage Vr by a current of [phi]/2. Here, in order to simplify the description, the harmonic component of the current Ir is omitted, and a sine wave current is used. When current Ir>0, if self-arc suppression element S4 is turned on (element S1 is off), the AC output voltage becomes Vcr=-Vd/2, and current IS4 flows through self-arc suppression element S4 of the lower branch. When the current Ir>0, at ωt=0, if the self-arcing element S4 is turned off and the element S1 is turned on, the AC output voltage becomes Vcr=+Vd/2, and IS4=0. The current Ir first passes through the high-speed diode D1 of the upper branch, and flows through ID1. As mentioned above, since the forward voltage drop Vb of the high-speed diode D1 is greater than the forward voltage drop Va of the power diode PD1, the current Ia of the AC reactor La Gradually attenuates, and turns to flow into diode PD1 for power. That is, ID1 attenuates, IPD1 increases, and finally IPD1=Ir. IPD1=Ir flows until the current Ir is reversed. Then, when the current becomes Ir<0, the element S1 is turned on, so the current IS1 flows through the element S1.

At the phase angle ωt=π, if the self-arcing element S1 is turned off and the element S4 is turned on, the voltage Vcr becomes Vcr=-Vd/2 again, IS1=0, first, the current ID4 is in the high-speed diode D4 of the lower branch flow. Since the forward voltage drop Vb of the high-speed diode D4 is larger than the forward voltage drop Va of the power diode PD4, the current Ia of the AC reactor La gradually attenuates and turns to flow into the power diode PD4. That is, ID4 decays, IPD4 increases, and finally IPD4=Ir. Until the current Ir is reversed again, IPD4=Ir flows.

The cut-off current Imax of the self-arc-extinguishing elements S1 and S4, if the peak value of the input current Ir is recorded as Im, becomes Imax=Im×sin(/2), for example, if the control phase angle =20°, then becomes Imax=0.174×Im. That is, the switching (on/off operation) of the self-arc-extinguishing elements S1 and S4 is performed near the zero crossing of the input current Ir, and the maximum cut-off current Imax of the self-arc-extinguishing elements S1 and S4 can be suppressed relative to the current peak value Im for sufficiently small. As a result, in the present embodiment, an element with a small capacity of the converter can be used, and an economical converter can be provided. In addition, switching loss can be reduced, and cooling equipment capacity can be reduced. Furthermore, the AC voltage Vcr of the hybrid converter HB-CNV is a rectangular wave voltage, and the peak value Vcm of the fundamental wave component becomes:

Vcm=(4/π)×(Vd/2)=1.273×(Vd/2)

A value equal to or greater than DC voltage (Vd/2) can be obtained. That is, compared with a normal PWM control converter, the voltage utilization rate is high, and when it is comprised with the self-arc-extinguishing element of the same withstand voltage, it has an advantage that a larger output can be generated. The S phase and the T phase are also controlled in the same manner.

As described above, in the hybrid converter HB-CNV constituting the device of the present embodiment, most of the input currents Ir, Is, and It during power running flow through the power diodes PD1 to PD6, and the currents in the self-arc extinguishing element S1 are suppressed. ~S6 or the current flowing in high-speed diodes D1~D6. Therefore, in an electrified railway system in which power running load power is larger than regenerative power, high-efficiency operation can be performed, and there is an advantage of being able to provide a more economical system compared with conventional PWM converters.

FIG. 57 is another block diagram of the control circuit CONT3 of the hybrid converter HB-CNV of the apparatus of FIG. 53 . In the figure, C1 represents a comparator, Gv(S) represents a voltage control compensation circuit, and PHC represents a phase control circuit.

The voltage Vd applied to the DC smoothing capacitor Cd is detected. The voltage command value Vd ^* is compared with the voltage detection value Vd by the comparator C1, and the deviation εv=Vd ^* -Vd is obtained. The following voltage control compensation circuit Gv(S) amplifies the deviation εv proportionally or integrally, and inputs it to the phase control circuit PHC as a phase control command  ^* . That is, the phase control signal [phi] ^* is directly transmitted from the DC voltage control circuit Gv(S) to the phase control circuit PHC. When Vd ^* >Vd, the deviation εv becomes positive, and the control phase angle command  ^* increases. This control phase angle command  ^* determines the delayed phase angle  of the AC voltage Vc of the converter CNV3 with respect to the voltage Vs of the second AC power supply SUP2, and by increasing  ^* =, the input current Is increases. As a result, the effective power Ps2 supplied from the power supply SUP2 increases, the voltage Vd applied to the DC smoothing capacitor Cd increases, and is controlled to Vd ^* =Vd.

Conversely, in the case of Vd ^* <Vd, the deviation εv is negative, and the control phase angle command  ^* decreases or takes a negative value (leading phase). If <0, the direction of the vector of the input current Is is reversed, and the active power Ps2 is reproduced in the AC power supply SUP2. As a result, the voltage Vd applied to the DC smoothing capacitor Cd decreases, and is also controlled so that Vd ^* =Vd. As described above, the input current control circuit (partial loop) can be omitted, and the control circuit can be simplified.

FIG. 58 shows a specific configuration of the compensation current control unit CONT1 of the first and second voltage-source self-excited power converters CNV1 and CNV2 of the device shown in FIG. 53 . In the figure, Fs(x) represents the power command generator, Ks represents the proportional element, C2 and C3 represent the comparator, M1 and M2 represent the multiplier, AD1~AD4 represent the adder and subtractor, Gi1(S), Gi2(S) Represents the current control compensation circuit, PWM1, PWM2 represent the pulse width modulation control circuit.

The power Ps1 supplied from the first AC power supply SUP1 (three-phase-60 Hz) is controlled as follows. The power command generator Fs(x) provides the command value Ps1 ^* of the power Ps1 supplied from the first AC power supply SUP1 according to the time average value PL(av) of the load power PL, and multiplies it by the proportional constant Ks as a Scott connection The peak value command Ism ^* of the M-seat and T-seat winding currents IMs and ITs of the transformer TR1.

The multiplier M1 multiplies the input current peak command Ism ^* with the unit sine wave sinωt synchronized with the M-seat voltage VM of the Scott connection transformer TR1, and outputs the M-seat input current command IMs ^* =Ism ^* ×sinωt. The multiplier M2 multiplies the input current wave peak command Ism ^* with the unit sine wave cosωt synchronous with the T-seat voltage VT of the Scott connection transformer TR1, and outputs the T-seat input current command ITs ^* =Ism ^* ×cosωt. The M-seat input current command value IMs ^* is subtracted from the detected value of the M-seat load current IML by the adder-subtractor AD1 to obtain the M-seat compensation current command value IMc ^* =IML-IMs ^* . Similarly, the T-seat input current command value ITs ^* is subtracted from the detected value of the T-seat load current ITL by the adder-subtractor AD3 to obtain the T-seat compensation current command value ITc ^* =ITL-ITs ^* . Among them, ITL=0.

The comparator C2 compares the M-seat compensation current detection value IMc with the compensation current command value IMc ^* , and the subsequent current control compensation circuit Gi1(S) amplifies its deviation εm=IMc ^* -IMc, and inputs it to the addition and subtraction method device AD2. In the adder-subtractor AD2, the compensation signal EM ^* proportional to the M seat voltage VM is added to the output signal of the current control compensation circuit Gi1(S), and its signal em ^* is input to the pulse width modulation of the converter CNV1 Control circuit PWM1. The first voltage source self-excited power converter CNV1 generates a voltage VMc proportional to the input signal em ^* .

The difference (VMc-VM) between the output voltage VMc and the M-seat power supply voltage VM is applied to the leakage inductance Lsm of the single-phase transformer TRm, and a compensation current IMc flows. Of course, when the leakage inductance of the single-phase transformer TRm is small, a reactor Lsmo may be inserted in series with the primary or secondary winding of the transformer TRm.

In the case of IMc ^* >IMc, the deviation εm is positive, and control is performed to increase the signal em ^* to increase the compensation current IMc so that IMc ^* =IMc. Conversely, in the case of IMc ^* <IMc, the deviation εm is negative, the signal em ^* is controlled to decrease, the compensation current IMc is reduced, and IMc ^* =IMc is also set. As a result, the M-seat input current IMs supplied from the Scott connection transformer TR1 is controlled to be:

IMs=IML-IMc=IML-IMc ^* =IML-(IML-IMs ^* )=IMs ^* . This input current IMs becomes a sine wave current having the same phase (power factor=1) as the M-seat voltage VM.

Similarly, the comparator C3 compares the T-seat compensation current detection value ITc with the compensation current command value ITc ^* , and the subsequent current control compensation circuit Gi2(S) amplifies its deviation εt=ITc ^* -ITc, and inputs it to the accumulator Subtractor AD4. In the adder-subtractor AD4, the compensation signal ET ^* proportional to the T seat voltage VT is added to the output signal of the current control compensation circuit Gi2(S), and its signal et ^* is input to the pulse width modulation of the converter CNV2 Control circuit PWM2. The second voltage source self-excited power converter CNV2 generates a voltage VTc proportional to the input signal et ^* .

The difference (VTc-VT) between the output voltage VTc and the T-seat power supply voltage VT is applied to the leakage inductance Lst of the single-phase transformer TRt, and a compensation current ITc flows. In the case of ITc ^* >ITc, the deviation εt is positive, the signal et ^* is controlled to increase, and the compensation current ITc is increased so that ITc ^* =ITc. Conversely, in the case of ITc ^* <ITc, the deviation εt is negative, the signal et ^* is controlled to decrease, the compensation current ITc is reduced, and ITc ^* =ITc is also set. As a result, the T-seat input current ITs supplied from the Scott connection transformer TR1 is controlled to be:

ITs=ITL-ITc=ITL-ITc ^* =ITL-(ITL-ITs ^* )=ITs ^* .

This input current ITs becomes a sine wave current having the same phase (power factor=1) as the T-seat voltage VT. However, the T-seat load current becomes ITL=0.

The currents IMs and ITs of the M-seat and T-seat of the Scott connection transformer TR1 have the same amplitude value Ism ^* , and the two-phase balanced currents whose phases are staggered by 90°. As a result, the current supplied from the first three-phase AC power supply SUP1 also becomes a three-phase balanced sinusoidal current with a power factor=1. As a result, not only the capacity of the Scott connection transformer TR1 but also the capacity of the equipment of the first AC power supply SUP1 or the M/G device can be reduced.

FIG. 59 shows an example of the characteristics of the electric power command generator Fs(x) of FIG. 58, and the power command value Ps1 ^* from the first AC power supply SUP1 is given to the load power PL(av) as follows.

Ps1 ^* ＝k×PL(av) (where k＝0～1)

For example, when k=0.5, half of the load power PL(av) is supplied from the first AC power supply SUP1, and the remaining half is supplied from the second AC power supply SUP2. That is, the power Ps2 supplied (or regenerated) from the second AC power supply SUP2 through the hybrid converter HB-CNV becomes

Ps2=Phb=PL(av)-Ps1 ^* =(1-k)·Ps1 ^*

By changing the coefficient k, the distribution of the power Ps1 supplied (or regenerated) from the first AC power supply SUP1 and the power Ps2 supplied (or regenerated) from the second AC power supply SUP2 can be adjusted.

FIG. 60 is a diagram showing another characteristic example of the electric power command generator Fs(x) of FIG. 58, and the power command value Ps1 ^* from the first AC power supply SUP1 is given to the load power PL(av) as follows. That is, when the set value PLo is used, within the range of -PLo<PL(av)<+PLo, Ps1 ^* =PL(av), and the entire load power PL is supplied or regenerated from the first AC power supply SUP1. When PL(av)<-PLo, Ps1 ^* =-PLo=constant, and when PL(av)>+PLo, Ps ^* =+PLo=constant.

That is, in the regenerative operation, in the case of PL(av)<-PLo, the power regenerated in the first AC power supply SUP1 is made Ps1=-PLo=constant, and the regenerative power outside this range (PL(av)- PLo) is regenerated in the second AC power supply SUP2 through the hybrid converter HB-CNV. In addition, in power running, in the case of PL(av)>+PLo, the power supplied from the first AC power supply SUP1 is set to Ps1=+PLo=constant, and the power supplied outside this range (PL(av)-PLo ) is supplied from the second AC power supply SUP2 through the hybrid converter HB-CNV.

Therefore, according to this embodiment, it is possible to increase the capacity of power supply or regenerative power to electric vehicle loads without increasing the capacity of existing equipment (M/G device or Sterling transformer, etc.), thereby providing an economical AC feed system for electrified railway.

Fig. 61 is a voltage-current vector diagram of the M seat and the T seat during the power running of the device of Fig. 53 . The T-seat load current ITL=0, and the M-seat load current IML lags the voltage VM by a certain phase θ. The load power is PL=VM×IML×cosθ, which is equal to the sum Pso=Ps1+Ps2 of the power supply Ps1 from the first AC power supply SUP1 and the power supply Ps2 from the second AC power supply SUP2.

The currents IMs and ITs supplied from the first AC power supply SUP1 through the Scott connection transformer TR1 are controlled to be sine waves in phase with the M-seat voltage VM and the T-seat voltage VT respectively, and the input current Ps1 becomes:

Ps1=IMs×VM+ITs×VT

In addition, the compensation currents IMc and ITc supplied from the first and second voltage source self-excited power converters CNV1 and CNV2 are respectively:

IMc=IML-IMs

ITc=ITL-ITs=-ITs

The M-seat compensation current IMc includes the active power Ps2 supplied from the second AC power supply SUP2 through the hybrid converter HB-CNV and the reactive power QL of the load, and the active power Ps1=PL-Ps2 is supplied from the first AC power supply SUP1.

The effective power PMs=IMs×VM of the M-seat is equal to the effective power PTs=ITs×VT of the T-seat, which becomes half of the effective power Ps1 supplied from the M-seat winding of the Scott connection transformer TR1, and the rest is supplied from the T-seat winding half.

The power PTs=Ps1/2 supplied from the T-block winding is regenerated by the second voltage source self-excited power converter CNV2 and supplied to the DC smoothing capacitor Cd. That is, ITc=-ITs. Furthermore, its power Ps1/2 is supplied to the single-phase AC power feeder Fa via the first voltage-source self-excited power converter CNV1. At this time, the power Ps2 supplied from the second AC power supply SUP2 through the hybrid converter HB-CNV and the reactive power QL=VM×ILM×sinθ of the load are supplied from the first voltage-type self-excited power converter CNV1, Only effective power PMs=IMs×VM=Ps1/2 is supplied from the M-seat winding of the Scott connection transformer TR1.

In this embodiment, by making the output capacity of the first voltage-source self-excited power converter CNV1 larger than the output capacity of the second voltage-source self-excited power converter CNV2, the power supplied from the second AC power supply SUP2 can be increased. Ps2 can reduce the power Ps1 supplied from the first AC power supply SUP1 accordingly. In other words, it is possible to reduce the equipment capacity of existing M/G devices (frequency converters) or Scott connection transformers.

Claims (43)

1. electric railway alternating current feeding system, this electric railway alternating current feeding system uses the voltage transformer that three-phase power is for conversion into two-phase electric power, when obtaining the power supply of two single-phase flow wires, from above-mentioned each single-phase flow wire process power collector, to the electric car electric, this electric railway alternating current feeding system is characterised in that to possess:

From a side's of above-mentioned voltage transformer single phase A.C. to the opposing party's the single phase A.C., can exchange continuously or the device of supply capability continuously.

2. electric railway alternating current feeding system, this electric railway alternating current feeding system spreads all over a plurality of substations, draws logical single phase A.C. flow wire, it is characterized in that:

At least one substation possesses in above-mentioned a plurality of substation:

The three-phase alternating voltage transformation is become two-phase alternating current voltage, in this two phase output terminals, only a phase output terminals is connected to voltage transformer on the above-mentioned single phase A.C. flow wire;

Connect ac output end a side's of two phase output terminals of above-mentioned voltage transformer on mutually, alternating electromotive force is transformed into the 1st power converter of direct current power;

Connect ac output end the opposing party's of two phase output terminals of above-mentioned voltage transformer on mutually, alternating electromotive force is transformed into the 2nd power converter of direct current power;

With the unbalanced current of the above-mentioned the 1st and the 2nd power converter compensation by the generations such as load of above-mentioned voltage transformer, control is used for the compensating current control unit as the compensating current of equalizing current.

3. electric railway alternating current feeding system according to claim 2 is characterized in that possessing:

Common dc terminal bonded assembly direct current smooth condenser with the above-mentioned the 1st and the 2nd power converter.

4. electric railway alternating current feeding system according to claim 2 is characterized in that possessing:

Common dc terminal bonded assembly direct current smooth condenser with the above-mentioned the 1st and the 2nd power converter;

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

5. according to each described electric railway alternating current feeding system of claim 2～4, it is characterized in that:

Above-mentioned compensating current control unit possesses:

Control the dc voltage control circuit of the voltage and instruction value unanimity that makes on the common dc terminal that is applied to the above-mentioned the 1st and the 2nd power converter;

On output signal, multiply by synchronous with above-mentioned two-phase alternating current voltage respectively unit sine wave signal, the Watt current command circuit of output biphase-equilibrium Watt current command value from above-mentioned dc voltage control circuit;

By deducting above-mentioned biphase-equilibrium Watt current command value from the load current detected value respectively, the compensating current command circuit of output compensating current command value;

According to above-mentioned compensating current command value, control is from the compensating current control circuit of the compensating current of the above-mentioned the 1st and the 2nd power converter output.

6. according to each described electric railway alternating current feeding system of claim 2～4, it is characterized in that:

Above-mentioned compensating current control unit possesses:

Detect bearing power, ask the circuit of the aviation value of the bearing power in its time;

Ask the circuit of the Watt current crest value instruction that compensates with the proportional forward direction of the aviation value of above-mentioned bearing power;

Control the dc voltage control circuit of the voltage and instruction value unanimity that makes on the common dc terminal that is applied to the above-mentioned the 1st and the 2nd power converter;

The adder-subtractor that adds the Watt current crest value of above-mentioned forward direction compensation on from the output signal of above-mentioned dc voltage control circuit;

On output signal, multiply by synchronous with above-mentioned two-phase alternating current voltage respectively unit sine wave signal, the Watt current command circuit of output biphase-equilibrium Watt current command value from above-mentioned adder-subtractor;

By deducting above-mentioned biphase-equilibrium Watt current value respectively from the load current detected value, the compensating current command circuit of output compensation current;

According to above-mentioned compensating current command value, control is from the compensating current control circuit of the compensating current of the above-mentioned the 1st and the 2nd power converter output.

7. according to each described electric railway alternating current feeding system of claim 2～4, it is characterized in that:

Above-mentioned compensating current control unit detects load current, control so that from above-mentioned the 1st power converter output and the proportional compensating current of this load current, and, control makes the voltage and instruction value unanimity on the common dc terminal that is applied to the above-mentioned the 1st and the 2nd power converter from the compensating current of above-mentioned the 2nd power converter output.

8. according to each described electric railway alternating current feeding system of claim 2～4, it is characterized in that:

Above-mentioned compensating current control unit is obtained the time average of load effective power, according to each bearing power aviation value, control is from the compensating current of above-mentioned the 1st power converter output, and, control makes the voltage and instruction value unanimity on the common dc terminal that is applied to the above-mentioned the 1st and the 2nd power converter from the compensating current of above-mentioned the 2nd power converter output.

9. electric railway alternating current feeding system, this electric railway alternating current feeding system spreads all over a plurality of substations, draws logical single phase A.C. flow wire, it is characterized in that:

At least one substation possesses in above-mentioned a plurality of substation:

The three-phase alternating voltage transformation is become two-phase alternating current voltage, only the phase output terminals of a side in this two phase output terminals is connected to the voltage transformer of above-mentioned single phase A.C. flow wire;

Connect ac output end a side's of two phase output terminals of above-mentioned voltage transformer on mutually, alternating electromotive force is transformed into the 1st power converter of direct current power;

Connect ac output end the opposing party's of two phase output terminals of above-mentioned voltage transformer on mutually, alternating electromotive force is transformed into the 2nd power converter of direct current power;

The compensating current control unit of the compensating current of the unbalanced current that control is produced by the load of above-mentioned voltage transformer etc. from the compensation of the above-mentioned the 1st and the 2nd power converter output;

Between the common dc terminal of the above-mentioned the 1st and the 2nd power converter, the give and accept energy storage equipment of electric power.

10. electric railway alternating current feeding system according to claim 9 is characterized in that:

The output capacity of above-mentioned the 1st power converter that has connected ac output end in above-mentioned single phase A.C. flow wire one side is bigger than the output capacity of above-mentioned the 2nd power converter.

11. electric railway alternating current feeding system according to claim 9 is characterized in that:

Above-mentioned energy storage equipment possesses control circuit, direct current reactor, the energy-storage units of bidirectional, dc release unit, this release unit,

The control circuit of above-mentioned release unit is controlled above-mentioned two-way release unit according to the value that is applied to the voltage on the above-mentioned energy-storage units, so that be adjusted at the power of giving and accepting between the common dc terminal of the above-mentioned the 1st and the 2nd power converter.

12. electric railway alternating current feeding system according to claim 9 is characterized in that, possesses:

Common dc terminal bonded assembly direct current smooth condenser with the above-mentioned the 1st and the 2nd power converter.

13. electric railway alternating current feeding system according to claim 9 is characterized in that, possesses:

Common dc terminal bonded assembly direct current smooth condenser with the above-mentioned the 1st and the 2nd power converter;

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

14. each the described electric railway alternating current feeding system according to claim 9～13 is characterized in that:

Above-mentioned compensating current control unit possesses:

Control the dc voltage control circuit of the voltage and instruction value unanimity that makes between the common dc terminal that is applied to the above-mentioned the 1st and the 2nd power converter;

On output signal, multiply by synchronous with above-mentioned two-phase alternating current voltage respectively unit sine wave signal, the Watt current command circuit of output biphase-equilibrium Watt current command value from above-mentioned dc voltage control circuit;

By deducting above-mentioned biphase-equilibrium Watt current command value from the load current detected value respectively, the compensating current command circuit of output compensating current command value;

According to above-mentioned compensating current command value, control is from the compensating current control circuit of the compensating current of the above-mentioned the 1st and the 2nd power converter output.

15. an electric railway alternating current feeding system, this electric railway alternating current feeding system spreads all over a plurality of substations, draws logical single phase A.C. flow wire, it is characterized in that:

At least one substation in above-mentioned a plurality of substation possesses:

The 1st three-phase alternating-current supply with the 1st frequency;

The three-phase alternating voltage transformation of the 1st three-phase alternating-current supply is become two-phase alternating current voltage, in this two phase output terminals, only a phase output terminals is connected to voltage transformer on the above-mentioned single phase A.C. flow wire;

The 1st power converter that on a phase (Building M) of two phase output terminals of this voltage transformer, has connected ac output end;

The 2nd power converter that on another phase (Building T) of two phase output terminals of this voltage transformer, has connected ac output end;

Be connected to the direct current smooth condenser on the common dc terminal of the above-mentioned the 1st and the 2nd power converter;

Control is from the compensating current control unit of the compensating current of the above-mentioned the 1st and the 2nd power converter generation;

The 2nd three-phase alternating-current supply with the 2nd frequency;

The three-phase alternating current of the 2nd three-phase alternating-current supply is transformed to direct current, supplies with the diode rectifier of dc power to above-mentioned direct current smooth condenser.

16. electric railway alternating current feeding system according to claim 15 is characterized in that:

Above-mentioned compensating current control unit is when bearing power during less than the setting power value, decision biphase-equilibrium Watt current command value, make the voltage and instruction value unanimity that is applied on the above-mentioned direct current smooth condenser, when above-mentioned bearing power surpasses the setting power value, according to this bearing power, provide from the command value of the effective power of above-mentioned the 1st three-phase alternating-current supply supply, from this effective power value decision biphase-equilibrium Watt current command value, obtain the compensating current command value by deduct this Watt current command value respectively from the load current detected value, according to this compensating current command value, control is from the compensating current of the above-mentioned the 1st and the 2nd power converter output.

17. electric railway alternating current feeding system according to claim 15 is characterized in that:

Above-mentioned compensating current control unit possesses:

Control the dc voltage control circuit that makes the voltage and instruction value unanimity that is applied on the above-mentioned direct current smooth condenser;

On output signal, multiply by the unit sine wave signal synchronous, the Watt current command circuit of output biphase-equilibrium Watt current command value with above-mentioned two-phase alternating current voltage from above-mentioned dc voltage control circuit;

By deducting this biphase-equilibrium Watt current command value respectively from the load current detected value, the compensating current command circuit of output compensating current command value;

According to this compensating current command value, control is from the compensating current control circuit of the compensating current of the above-mentioned the 1st and the 2nd power converter output.

18., it is characterized in that according to the described electric railway alternating current feeding system of claim 17:

Above-mentioned compensating current control unit is applied to the command value of the voltage on the above-mentioned direct current smooth condenser and sets the non-loaded commutating voltage that is higher than above-mentioned diode rectifier for when regeneration is moved.

19. electric railway alternating current feeding system according to claim 17 is characterized in that:

Above-mentioned compensating current control unit when power moves, according to the size of the power consumption of above-mentioned AC mains bonded assembly electric car load, control is applied to the voltage on the above-mentioned direct current smooth condenser, so that adjust the dc power of supplying with from above-mentioned diode rectifier.

20. electric railway alternating current feeding system according to claim 17 is characterized in that:

Above-mentioned compensating current control unit is when power running load power during less than the setting power value, above-mentioned vdc command value is set at the non-loaded commutating voltage that is higher than above-mentioned diode rectifier, control the voltage that is applied on the above-mentioned direct current smooth condenser, when power running load power has surpassed the setting power value, size according to this bearing power, control is applied to the voltage on the above-mentioned direct current smooth condenser, so that adjust the dc power of supplying with from above-mentioned diode rectifier.

21. each the described electric railway alternating current feeding system according to claim 15～20 is characterized in that possessing:

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

22. each the described electric railway alternating current feeding system according to claim 15～20 is characterized in that:

Make in above-mentioned single phase A.C. flow wire one side and connected the output capacity of above-mentioned the 1st power converter of ac output end greater than the output capacity of above-mentioned the 2nd power converter.

23. an electric railway alternating current feeding system, this electric railway alternating current feeding system spreads all over a plurality of substations, draws logical single phase A.C. flow wire, it is characterized in that:

At least one substation in above-mentioned a plurality of substation possesses:

The 1st three-phase alternating-current supply with the 1st frequency;

The three-phase alternating voltage transformation of the 1st three-phase alternating-current supply is become two-phase alternating current voltage, in this two phase output terminals, only a phase output terminals is connected to voltage transformer on the above-mentioned single phase A.C. flow wire;

The 1st power converter that on a phase (Building M) of two phase output terminals of this voltage transformer, has connected ac output end;

The 2nd power converter that on another phase (Building T) of two phase output terminals of this voltage transformer, has connected ac output end;

Be connected to the direct current smooth condenser on the common dc terminal of the above-mentioned the 1st and the 2nd power converter;

Control is from the compensating current control unit of the compensating current of the above-mentioned the 1st and the 2nd power converter generation;

And the energy storage equipment of the power of giving and accepting between the above-mentioned direct current smooth condenser;

Control is to the charging and discharging currents control unit of the charging and discharging currents of above-mentioned energy storage equipment;

The 2nd three-phase alternating-current supply with the 2nd frequency;

The three-phase alternating current of the 2nd three-phase alternating-current supply is transformed to direct current, supplies with the diode rectifier of dc power to above-mentioned direct current smooth condenser.

24. electric railway alternating current feeding system according to claim 23 is characterized in that:

Above-mentioned compensating current control unit possesses:

The voltage instruction unit of vdc command value is provided according to bearing power;

Control make be applied on the above-mentioned direct current smooth condenser voltage with from the consistent dc voltage control circuit of the command value of this voltage instruction unit;

On output signal, multiply by the unit sine wave signal synchronous, the Watt current command circuit of output biphase-equilibrium Watt current command value with above-mentioned two-phase alternating current voltage from this dc voltage control circuit;

By deducting this biphase-equilibrium Watt current command value respectively from the load current detected value, the compensating current command circuit of output compensating current command value;

According to this compensating current command value, control is from the compensating current control circuit of the compensating current of the above-mentioned the 1st and the 2nd power converter output.

25. electric railway alternating current feeding system according to claim 24 is characterized in that:

Above-mentioned compensating current control unit when power moves, according to the size of the power consumption of above-mentioned AC mains bonded assembly electric car load, control is applied to the voltage on the above-mentioned direct current smooth condenser, so that adjust the dc power of supplying with from above-mentioned diode rectifier.

26. electric railway alternating current feeding system according to claim 23 is characterized in that:

Above-mentioned compensating current control unit is when bearing power during less than the setting power value, decision biphase-equilibrium Watt current command value, the feasible voltage and instruction value unanimity that is applied on the above-mentioned direct current smooth condenser, when bearing power has surpassed the setting power value, according to this bearing power, provide from the command value of the effective power of above-mentioned the 1st three-phase alternating-current supply supply, from this effective power value decision biphase-equilibrium Watt current command value, obtain the compensating current command value by deduct this Watt current command value respectively from the load current detected value, according to this compensating current command value, control is from the compensating current of the above-mentioned the 1st and the 2nd power converter output.

27. each the described electric railway alternating current feeding system according to claim 23～26 is characterized in that possessing:

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

28. each the described electric railway alternating current feeding system according to claim 23～26 is characterized in that:

The charging and discharging currents control unit of above-mentioned energy storage equipment is according to bearing power, the power that decision is supplied with from above-mentioned the 1st three-phase alternating-current supply and the power of supplying with through diode rectifier from above-mentioned the 2nd three-phase alternating-current supply and power command value, deduct this and power command value by detected value from above-mentioned bearing power, ask and above-mentioned energy storage equipment between the power command value of giving and accepting, based on this power command value, control is to the charging and discharging currents of above-mentioned energy storage equipment.

29. electric railway alternating current feeding system according to claim 28 is characterized in that:

Power command value is repaid in the charging and discharging currents control unit supplement of above-mentioned energy storage equipment, makes the accumulation of energy of this energy storage equipment become constant, this compensation power command value is added on the above-mentioned and power command value controls.

30. electric railway alternating current feeding system according to claim 28 is characterized in that, possesses:

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

31. an electric railway alternating current feeding system, this electric railway alternating current feeding system spreads all over a plurality of substations, draws logical single phase A.C. flow wire, it is characterized in that:

At least one substation in above-mentioned a plurality of substation possesses:

The 1st three-phase alternating-current supply with the 1st frequency;

The three-phase alternating voltage transformation of the 1st three-phase alternating-current supply is become two-phase alternating current voltage, in this two phase output terminals, only a phase output terminals is connected to voltage transformer on the above-mentioned single phase A.C. flow wire;

The 1st power converter that on a phase (Building M) of two phase output terminals of this voltage transformer, has connected ac output end;

The 2nd power converter that on another phase (Building T) of two phase output terminals of this voltage transformer, has connected ac output end;

Be connected to the direct current smooth condenser on the common dc terminal of the above-mentioned the 1st and the 2nd power converter;

Control is from the compensating current control unit of the compensating current of the above-mentioned the 1st and the 2nd power converter generation;

The 2nd three-phase alternating-current supply with the 2nd frequency;

The three-phase alternating current of the 2nd three-phase alternating-current supply is transformed to direct current, supplies with the 3rd power converter of dc power to above-mentioned direct current smooth condenser.

32. electric railway alternating current feeding system according to claim 31 is characterized in that:

Make in above-mentioned single phase A.C. flow wire one side and connected the output capacity of above-mentioned the 1st power converter of ac output end greater than the output capacity of above-mentioned the 2nd power converter.

33. electric railway alternating current feeding system according to claim 31 is characterized in that:

Above-mentioned compensating current control unit is according to bearing power, the effective power command value that decision is supplied with from above-mentioned the 1st source of AC, ask biphase-equilibrium Watt current command value from this effective power command value, by deduct this biphase-equilibrium Watt current command value respectively from the load current detected value, current instruction value is repaid in supplement, according to this compensating current command value, control is from the compensating current of the above-mentioned the 1st and the 2nd power converter output.

34. electric railway alternating current feeding system according to claim 31 is characterized in that:

Constitute of the single-phase lead-out terminal disengaging of ac output end of above-mentioned the 2nd power converter, make on its lead-out terminal that can be parallel to above-mentioned the 1st power converter from above-mentioned changer.

35. each the described electric railway alternating current feeding system according to claim 31～34 is characterized in that:

The received current that the control of above-mentioned the 3rd power converter is supplied with from above-mentioned the 2nd source of AC makes to apply voltage and instruction value unanimity on above-mentioned direct current smooth condenser.

36. each the described electric railway alternating current feeding system according to claim 31～34 is characterized in that:

Above-mentioned the 3rd power converter is according to certain pulse pattern action, by the phase angle of adjustment with respect to the AC side terminal voltage of the 3rd power converter of the power line voltage of above-mentioned the 2nd three-phase alternating-current supply, the received current that control is supplied with from above-mentioned the 2nd three-phase alternating-current supply.

37. each the described electric railway alternating current feeding system according to claim 31～34 is characterized in that:

Above-mentioned the 3rd power converter is according to certain pulse pattern action, by the phase angle of adjustment with respect to the AC side terminal voltage of above-mentioned the 3rd power converter of the power line voltage of above-mentioned the 2nd three-phase alternating-current supply, control is applied to the voltage on the above-mentioned direct current smooth condenser.

38. electric railway alternating current feeding system according to claim 31 is characterized in that, possesses:

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

39. an electric railway alternating current feeding system, this electric railway alternating current feeding system spreads all over a plurality of substations, draws logical single phase A.C. flow wire, it is characterized in that:

At least one substation in above-mentioned a plurality of substation possesses:

The 1st three-phase alternating-current supply with the 1st frequency;

The three-phase alternating voltage transformation of the 1st three-phase alternating-current supply is become two-phase alternating current voltage, in this two phase output terminals, only a phase output terminals is connected to voltage transformer on the above-mentioned single phase A.C. flow wire;

The 1st power converter that on a phase (Building M) of two phase output terminals of this voltage transformer, has connected ac output end;

The 2nd power converter that on another phase (Building T) of two phase output terminals of this voltage transformer, has connected ac output end;

Be connected to the direct current smooth condenser on the common dc terminal of the above-mentioned the 1st and the 2nd power converter;

Control is from the compensating current control unit of the compensating current of the above-mentioned the 1st and the 2nd power converter generation;

The 2nd three-phase alternating-current supply with the 2nd frequency;

The three-phase alternating current of the 2nd three-phase alternating-current supply is transformed to direct current, supplies with the diode rectifier of dc power to above-mentioned direct current smooth condenser;

The 3rd power converter that on the ac terminal of this diode rectifier, has connected ac terminal through reactor;

Make the 3rd power converter according to certain pulse pattern action, by the phase angle of adjustment with respect to the AC side terminal voltage of the 3rd power converter of the voltage of above-mentioned the 2nd three-phase alternating-current supply, received current or effective power that control is supplied with from above-mentioned the 2nd three-phase alternating-current supply, control is applied to the control unit of the 3rd power converter of the voltage on the above-mentioned direct current smooth condenser.

40., it is characterized in that according to the described electric railway alternating current feeding system of claim 39:

Above-mentioned compensating current control unit is according to bearing power, the effective power command value that decision is supplied with from above-mentioned the 1st three-phase alternating-current supply, ask biphase-equilibrium Watt current command value from this effective power command value, by deduct this biphase-equilibrium Watt current command value respectively from the load current detected value, current instruction value is repaid in supplement, according to this compensating current command value, control is from the compensating current of the above-mentioned the 1st and the 2nd power converter output.

41., it is characterized in that according to claim 39 or 40 described electric railway alternating current feeding systems:

The received current that the control of above-mentioned the 3rd power converter is supplied with from above-mentioned the 2nd three-phase alternating-current supply makes to apply voltage and instruction value unanimity on above-mentioned direct current smooth condenser.

42., it is characterized in that according to claim 39 or 40 described electric railway alternating current feeding systems:

The control of above-mentioned the 3rd power converter is with respect to the phase angle of the AC side terminal voltage of above-mentioned the 3rd power converter of the voltage of above-mentioned the 2nd three-phase alternating-current supply, makes to apply voltage and instruction value unanimity on above-mentioned direct current smooth condenser.

43., it is characterized in that possessing according to the described electric railway alternating current feeding system of claim 39:

Be connected in parallel near the LC filter consistent 2 times of the frequency of above-mentioned AC mains with resonant frequency with above-mentioned direct current smooth condenser.

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