CN1322408A - Transmission over bundled channels in CDMA mobile radio system - Google Patents

Abstract

A transmitter apparatus and transmitting method are provided for implementing a high data rate multi-channel link in a digital Code Division Multiple Access (CDMA) wireless communications system composed of three or more individual lower data rate traffic channels represented in a transmitted combined RF signal, which are distributed over a set of phase offsets of a sinusoid carrier in the range of 0 to pi radians, in which a combined complex spread spectrum signal, composed of an in-phase component and a quadrature-phase component, is formed prior to up-conversion in a manner that all the required phase offsets of the represented channels are introduced. The three or more individual lower data rate channels include two or more individual channels having phase offsets which are 0, pi /2, or pi radians, and one or more individual channels having phase offsets which are pi /4 or 3 pi /4 radians. Individual complex spread spectrum signals are formed for the respective individual channels, each composed of an in-phase component and a quadrature-phase component. Further complex spread spectrum signals are derived therefrom (or merely repeated) for the individual channels in a manner including the introduction of fractional scaling factors having an absolute value substantially equal to [square root of] 2ROOT 2/2 for the channels having phase offsets of (/4 or 3(/4 radians, and these derived signals are additively combined to form the complex combined spread spectrum signal.

Description

Transmission in clustered channels of CDMA mobile radio systems

Background of the invention

technical field of invention

The present invention relates to apparatus and methods for transmitting data of the same communication in multiple or clusters of traffic channels of a communication system so as to obtain multiples of the data rate or bandwidth equivalent to that provided for the communication by a single traffic channel. A unique aspect of the present invention is that it relates to a multi-channel reverse link transmitter architecture in a digital code division multiple access (CDMA) wireless communication system, wherein each individual channel represented in the combined RF signal transmitted is distribution over a set of phase shift values in order to reduce the peak-to-average ratio of the combined RF signal over the set of phase shift values, which can be done if the carrier phases of the individual channels are aligned arrive.

Related technical description

Such a system can be submitted to the TR45.5 committee of the Telecommunications Industry Association (TIA) at the meeting held in Philadelphia by E. Tiedemann et al. of Qualcomm (Qualcomm) Co., Ltd. on June 16-20, 1997, entitled A system of "Increased Reverse Link Peak-to-Average Ratio (Phase 1c)" is known from TIA/EIA's Interim Standard IS-95B, and from PCT Patent Application WO98/58457.

In 1992, the Direct Spread Sequence Code Division Multiple Access (DS-CDMA) system was adopted together by the TIA and the Electronics Industries Association (EIA) as an interim standard IS-95 for the 800 MHz cellular band. After successful field tests and experimental systems, the IS-95 system is now operational for tens of millions of users. CDMA is based on spread spectrum technology first proposed by Allies in World War II to combat enemy radio interference. The characteristic of the spread spectrum signal is that its signal occupies a bandwidth W much larger than the information rate R bits/second of the signal. Thus, the spread spectrum signal inherently contains a kind of redundant information that can be used to overcome various interferences, including signals from other users in the same band and self-interference due to multipath components of time delay. Another key property of spread spectrum signals is pseudorandomness. As a result, the signal appears similar to random noise, which makes it difficult for receivers other than pre-programmed receivers to demodulate it. In a CDMA system, each channel shares a common bandwidth, and the channels are distinguished by different code sequences. In the case of IS-95, each channel carries a coded information signal modulated by a specific sequence in a set of orthogonal sequences called Walsh codes (the Walsh codes are assigned to the channel (referred to as applying the Walsh cover)), then modulated or scrambled by a long pseudo-noise (PN) code, in order to directly perform the in-phase and quadrature-phase sinusoidal and quadrature-phase The up-conversion of sine wave multiplication, and then the in-phase modulation and quadrature phase PN short code modulation to form the in-phase (I) and quadrature phase (Q) spread spectrum signal components of the complex spread spectrum signal respectively, and the in-phase and quadrature phase The results of each multiplication are summed to form an RF signal, and then sent to the RF power amplifier for amplification and then provided to the antenna.

In 1995, IS-95 was expanded into Interim Standard IS-95A, which later became Interim Standard IS-95B. The latter extension provides for high-bandwidth data applications where a group or cluster of up to eight channels can be used to carry data in the same communication, effectively forming a high-data-rate channel from the group of low-data-rate channels.

IS-95B stipulates that when multiple channels are used to form a multi-channel link, if the in-phase sine wave and quadrature-phase sine wave pair of each channel carrier frequency have 0 phase shift, in order to reduce the combined RF signal in this case , then this pair of sinusoids of the carrier frequency of these channels in the reverse link (transmission from the mobile station to the base station) is distributed in a specific sub-optimal manner at a phase shift in the range 0 to π between radians. This allocation of phase shifts reduces the linearity and dynamic range requirements of the mobile station power amplifier. In particular, the phase shifts of the in-phase and quadrature-phase sinusoids applied to upconverted channels 0-3 and channels 4-7 are 0, π/2, π/4, and 3π/4 radians, respectively.

As is well known, the in-phase and quadrature phase sine waves at the carrier frequency are analog signals generated by an oscillator which are analog multiplied by analog signals converted from the in-phase and quadrature phase spread spectrum digital signal components of the channel. The use of separate D/A converters for the in-phase and quadrature-phase spread spectrum signal components for each of up to 8 channels, or the assumption that separate D/A conversions are provided, results in increased complexity and cost. Furthermore, when the in-phase and quadrature-phase sine waves are generated by the oscillator portion of the mobile station, it is problematic to stably generate π/4 and 3π/4 phase-shifted sine waves.

The aforementioned PCT patent application shows that in this patent application only two channels are used for combining by negating the in-phase spread spectrum signal component of the first channel and the quadrature phase spread spectrum signal component of the second channel Post-adding to form a combined in-phase spread spectrum signal component, and adding the quadrature phase spread spectrum signal component of the first channel and the in-phase spread spectrum signal component of the second channel to form a combined quadrature phase component, it is possible Introduces a π/2 phase shift for the second channel. The combined in-phase and quadrature-phase components are then directly up-converted by multiplying respectively with the in-phase and quadrature-phase sine waves at the carrier frequency, and these product terms are summed to form the combined RF signal. However, when three or more channels are used for combining, a π/4 phase shift, and possibly a 3π/4 phase shift, is bound to result, requiring a set of in-phase and quadrature-phase sinusoids of at least Includes in-phase and quadrature-phase sine waves with a π/4 phase shift.

Purpose and summary of the invention

It is an object of the present invention to provide a type of multi-channel link structure in a digital Code Division Multiple Access (CDMA) wireless communication system, in which a plurality of different traffic channels represented in a transmitted combined RF signal is distributed on a set of sinusoidal carriers with a phase shift ranging between 0 and π radians, which requires only D/ A-conversion, and for multiplication with the different ones of the paired signals derived from the combined in-phase and combined quadrature-phase spread spectrum signal components, only a pair of in-phase and quadrature-phase sine waves need be generated at the carrier frequency. In such mobile stations capable of using multi-channel links consisting of three or more individual channels, such an architecture would simplify and reduce the cost of the baseband and RF sections.

The satisfaction of this and other objects of the present invention is achieved by providing a transmitter apparatus and sending method in which all required phase shifts (including two or more phase shifts of 0, π/2, or π denoted phase shifts of individual channels, and one or more phase shifts of denoted independent channels other than 0, π/2, or π radians) in the in-phase and combined quadrature phase spread spectrum signal components are introduced. In order to realize this scheme, each different complex spread spectrum signal (each of which is composed of an in-phase component and a quadrature phase component) of each corresponding individual channel is first formed, and the various complex spread spectrum signals composed of each individual channel are additively combined The signals derived from the complex spread spectrum signal to form a combined complex spread spectrum signal. This additive combination is such that the signals derived from the individual complex spread spectrum signals of one or more channels with phase shifts other than 0, π/2 and π radians are scaled by applying the relevant fractional scaling factors from It is obtained by converting signals derived from each complex spread spectrum signal of two or more channels with phase shifts of 0, π/2 and π. More specifically, those phase shift values that are not 0, π/2, or π are π/4 or 3π/4, and the fractional scale factor has an absolute value

Furthermore, the means for additively combining or the means capable of acting in this way are arranged such that the in-phase and quadrature-phase components of the combined spread spectrum signal receive respective complex components of a signal derived from either, but not both, of the in-phase or quadrature-phase components of a spread-spectrum signal while receiving the respective complex spread from a channel with a phase shift other than 0, π/2, or π component of the signal derived from both the in-phase and quadrature-phase components of the frequency signal. According to a first embodiment of the present invention, the in-phase and quadrature-phase components of each complex spread spectrum signal of each different channel are respectively fed to the input of a respective finite impulse response (FIR) filter or filtering operation, and its output Or the result is sent to a device for additively combining or a device capable of doing so.

According to a second embodiment of the invention, the in-phase and quadrature-phase components of each complex spread spectrum signal of each individual channel are respectively fed to the input of a respective finite impulse response (FIR) filter or filtering operation, whose output Or the result is sent to a device for additively combining or a device capable of doing so. The advantage of this embodiment is that only two filters or filtering operations are required instead of two for each individual channel.

Other objects, features and advantages of the present invention will become apparent by studying the following detailed description while referring to the accompanying drawings, in which:

Brief description of the drawings

Figure 1 is a general schematic diagram of a mobile station or handset used in a digital cellular system (such as a CDMA system); and

Fig. 2 and Fig. 3 are functional schematic diagrams of the implementation part of the mobile station transmitter shown in Fig. 1 in the first and second embodiments according to the present invention, respectively.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

Referring first to FIG. 1, a mobile station transceiver 10 of a conventional digital cellular system, such as a CDMA system, is described in detail comprising a user interface unit 12 connected to a baseband unit 14 which in turn is connected to a radio frequency unit 16. User interface unit 12 is also connected to keypad 18 , LCD display 20 , microphone 22 and speaker 24 . Baseband unit 14 includes a digital signal processor (DSP) 26 which has access to random access memory and read only memory 30 containing firmware instructions. The baseband unit 14 also contains a clock 32, an analog-to-digital converter (A/D's) 34 specifically for receiving analog signals from the RF unit 16 in the forward (receive) link, and a circuit especially for feeding the reverse The RF unit 16 in the (transmit) chain provides digital-to-analog converters (A/D's) 34 for analog signals.

RF unit 16 includes an oscillator 38 for generating a sine wave at the carrier frequency and providing it to a zero IF frequency downconverter 40 for the forward link and a zero IF frequency upconverter 42 for the reverse link. The antenna 48 sends the signal to the input amplifier 44 through the antenna duplexer 46, and the down-converter 40 receives the RF signal from the input amplifier 44. On the contrary, the up-converter 42 provides the RF signal to the output or power amplifier 50, and then through the antenna duplexer Worker 46 is sent to antenna 48.

As will soon appear in the description of the first and second embodiments of the present invention, all carrier phase shifts required to form a multi-channel reverse link consisting of three or more channels are achieved as follows: Processing is carried out in the DSP 32 of the baseband unit 14, and the up-converter 42 in the RF unit 16 is activated to form a pair of mixers or multipliers for combining the in-phase and quadrature-phase components of the complex combined signal with the in-phase of the carrier frequency Multiplied by a quadrature phase sine wave.

Turning to Fig. 2 about the first embodiment, it shows a part 52 realized by the baseband section 14, which respectively receives the data signals DS ₀ -DS _N of the N channels forming the multi-channel reverse link, and forms a An analog complex combined spread spectrum signal composed of an analog combined spread spectrum in-phase component ACSS _I and an analog combined spread spectrum quadrature phase component ACSS _Q. Although there are as many as 8 channels forming a multi-channel link in IS-95B, only 4 channels are shown for ease of illustration. The baseband processing scheme for channels 0-3 is the same as for channels 4-7 because IS-95B provides a phase-shifted sequence of in-phase and quadrature-phase sinusoids for upconverting channels 0-3 and for channels 4-7 is the same, namely 0, π/2, π/4, 3π/4 radians. Clearly, these additional channels simply add more components to the pair of analog composite spread spectrum signal components ACSS _I and ACSS _Q. These components described above are fed into the zero IF up-converter 54 implemented in the RF unit 16 . In upconverter 54, the analog combined in-phase and quadrature-phase spread spectrum signal components ACSS _I and ACSS _Q are provided to low-pass filters 140 and 141, respectively, whose outputs are in-phase and quadrature-phase sine waves sinusoidal at the carrier frequency (w _c t) and cos(w _c t) are fed into mixers or multipliers 144 and 146, respectively. The outputs of mixers or multipliers 144 and 146 are fed into adder 150 to form a combined RF signal CRF which is provided to power amplifier 150 .

The baseband processing unit 52 includes a unit 58 and a unit 56 . Block 58 forms respective different pairs of in-phase and quadrature-phase spread spectrum digital signal components DSS _0I , DSS _0Q to DSS _3I , DSS _3Q from respective input _data signals DS 0 to DS ₃ in a conventional manner. Section 56 filters, scales, and combines each of the different in-phase and quadrature phase spread spectrum digital signal component pairs DISS _0I , DISS _0Q to DISS _3I , DISS _3Q to form a digitally combined in-phase and quadrature phase spread spectrum signal component pair DCSS _I and DCSS _Q , in this process, the influence of the carrier phase required by each channel is introduced or considered. Digital combined components DCSS _I and DCSS _Q are provided to digital-to-analog converters 140 and 142 respectively to form analog combined components ACSS _I and ACSS _Q .

In block 58, the respective input data signals DS ₀ , DS ₁ , DS ₂ and DS ₃ are provided to respective channel encoders 61, 71, 81 and 91, respectively, and the resulting frame-encoded data signals are respectively sent to into respective Walsh modulators 62, 72, 82, and 92 to modulate the encoded data signals using respective channel-specific Walsh codes. The resulting encoded data signals modulated by Walsh codes are then sent to respective multipliers (or modulo-2 adders) 63, 73, 83 and 93, where they are divided by respective pseudo-noise ( PN) sub-sequences, (these pseudo-noise (PN) sub-sequences have different start positions in the continuously generated PN long code sequence whose period is generally 2 ⁴² -1), thereby extending the Walsh code The spectrum of the modulated coded signal. Next, the long-code-modulated signals are supplied to respective multipliers or modulo-2 adders 64, 74, 84, and 94, where they are respectively modulated by the in-phase PN short-code sequence PN_I to generate respective spread spectrum In-phase signals DSS _0I , DSS _1I , DSS _2I and DSS _3I . They are also supplied to respective multipliers or modulo-2 adders 65, 75, 85 and 95, respectively, where they are respectively modulated by the quadrature phase PN short code sequence PN_Q. The results of quadrature phase PN short code sequence PN_Q modulation are respectively sent to respective 1/2 chip time delay devices 66, 76, 86 and 96 to generate respective spread spectrum quadrature phase signals DSS _0Q , DSS _1Q , DSS _2Q and DSS _3Q . The in-phase and quadrature-phase short code sequences PN_I and PN_Q generally have a period 2 ¹⁵ -1 and they are specific to the base station communicating with the mobile station.

In block 58, the PN long code subsequence applied in channel 0 is sent to the data burst randomizer 69b, which in turn generates a control signal after a time delay 70 so that the power can be turned on when required Amplifier 50.

Section 56 includes a section 56a in which signal DDSS is obtained by applying signals DSS _0I , DSS _{1I ,} DSS _2I and DSS _3I to respective finite impulse response shaping filters (FIR_I) 67, 77, 87 and 97 _0I , DDSS _1I , DDSS _2I , and DDSS _3I , while the signals DSS _0Q , DSS _1Q , DSS _2Q , and DSS _3Q are applied to respective finite impulse response shaping filters (FIR_Q) 68, 78, 88, and 98 in this block Then the signals DDSS _0Q , DDSS _1Q , DDSS _2Q and DDSS _3Q can be obtained. It should be pointed out that FIR_I77 introduces a scale factor -1, and FIR_I87, FIR_Q88 and FIR_Q98 introduce a real value A fractional scaling factor of , and FIR_I97 introduces a real-valued The fractional scale factor of .

Component 56 also includes a component 56b which additively combines the resulting signals DDSS _0I , DDSS _0Q , DDSS _1I , DDSS _1Q , DDSS _2I , DDSS _2Q , DDSS _3I and DDSS _3Q to generate digitally combined signal components DCSS ₁ and DCSS _Q. The term "additively combine" is meant to include subtraction and/or changing the signs of operands prior to addition. Component 56b includes an adder 89 and an adder 90 as shown in the figure. The adder 89 forms the intermediate spread spectrum in-phase signal IDSS _2I of channel 2 by adding the obtained DDSS _2I and DDSS _2Q . The resulting spread spectrum signal DDSS _2I is subtracted from the resulting spread spectrum signal DDSS _2Q to form an intermediate spread spectrum quadrature phase signal IDSS _2Q for channel 2 . Component 56b also includes an adder 99 and an adder 100. The adder 99 forms the intermediate spread spectrum in-phase signal IDSS _3I of channel 3 by adding the obtained signals DDSS _3I and DDSS _3Q . The spread spectrum signal obtained by subtracting the obtained frequency signal DDSS _2I from DDSS _2Q forms the intermediate spread spectrum quadrature phase signal IDSS _3Q of channel 3.

的小数比例因子的引入和由加法器89、90、99和100产生的和与差的引入相组合，就使得中间信号对IDSS_2I和IDSS_2Q的相位旋转了π/4弧度，使得中间信号对IDSS_3I和IDSS_3Q的相位旋转了3π/4弧度。It should be noted that the actual absolute value in component 56a is

The combination of the introduction of the fractional scale factor of , and the introduction of the sum and difference produced by the adders 89, 90, 99 and 100, rotates the phase of the intermediate signal pair IDSS _2I and IDSS _2Q by π/4 radians, so that the intermediate signal pair The phases of IDSS _3I and IDSS _3Q are rotated by 3π/4 radians.

Furthermore, for channel 1, the phase rotation of π/2 radians is accomplished only by exchanging the in-phase and quadrature phase components, since the required sign change of the in-phase component prior to the exchange is already incorporated in the scaling factor -1 of FIR_I77.

After accounting for all required phase offsets, the in-phase and quadrature-phase components from the respective channels are summed in respective adders 110 and 120, the outputs of which are the respective combined in-phase and quadrature-phase signal components DCSS _I and DCSS _Q. Since clustered channels of up to 8 are possible according to IS-95B, in practice summers 110 and 120 have additional inputs (not shown) for channels 4-7, which, as previously described, are connected to Channels 0-4 have the same format. As shown, adder 110 receives signals DDSS _0I , DDSS _1Q , IDSS _2I , and IDSS _2Q as its inputs, and adder 120 receives signals DDSS _0Q , DDSS _1I , IDSS _2Q , and IDSS _3Q as its inputs. In order to maintain a suitable dynamic range for the resulting signals DCSS _I and DCSS _Q , adders 110 and 120 also receive control signals of the same magnitude on lines 115 and 125, respectively.

It should be noted that member 56 may take a variety of forms. The required fractional scaling factors may be introduced at various locations in this unit 56 (for example, at the relevant inputs of adders 110 and 120), and the scaling factor -1 may be introduced as a sign change prior to addition. Additionally, the various additions and subtractions can be changed in order, further split, or combined. For example, adders 89 and 99 could be deleted and their functions combined in adder 110 , and adders 99 and 100 could be deleted and their functions combined in adder 120 . In this case, adders 110 and 120 would each have six inputs for receiving one component of channels 0 and 1 and two components of channels 2 and 3, respectively.

More preferably, the digital data (including the digital data that will be input to the FIR filter of part 56a) that is used for all operations in the part 58 is all bipolar single-bit, takes the value of +1, and part 56a's The output of the FIR filter, and all subsequent subsequence digital data including the outputs of adders 110 and 120 are bipolar 5-bit, with integer values ranging from -15 to +15.

Fig. 3 has provided second kind of embodiment, and it looks the same with the embodiment of Fig. 2, and difference is: the part 56 that is made up of parts 56a and 56b among Fig. 2 is changed into Fig. 3 by part 156a (it does not contain a FIR filter, only the required fractional scaling factor is introduced), 156b (in which adder 120 changes the sign of DDSS ₁₁ before adding to introduce a required factor of -1) 156, and Additional shaping filters FIR_I 167 and FIR_Q 168 receive the bipolar 5-bit outputs of the respective adders 110 and 120 respectively and form the bipolar 5-bit combined signals DCSS _I and DCSS _Q , respectively. Replace the digital FIR filters 67, 68, 77, 78, 87, 67, 68, 77, 78, 87, 88, 97 and 98, resulting in a further reduction in complexity, although filters 167 and 168 use 5-bit inputs.

It should now be noted that the objects of the invention have been met. While the invention has been described in great detail, it should be noted that many modifications are possible within the intended spirit and scope of the invention. When interpreting the appended claims, it should be understood that:

a) the word "comprising" does not exclude those other unit steps than those listed in the claims;

b) the presence of the sign "a" before a certain element does not exclude the presence of many elements of the same type;

c) any reference signs in the claims do not limit their scope; and

d) Some "means" may be represented by the same structure or function realized by hardware or software.

Claims (10)

1. A transmitter device for high data rate multi-channel links in a digital code division multiple access (CDMA) wireless communication system, wherein a plurality of different low Data rate traffic channels are distributed over a set of sinusoidal carriers with phase offsets in the range 0-π, consisting of: means for forming a combined complex spread spectrum signal consisting of an in-phase component (DCSSI or ACSSI) and a quadrature-phase component (DCSSQ or ACSSQ) in such a way as to introduce all the phase shifts required for the indicated channel; and Means (144, 146, 150) for upconverting a combined complex spread spectrum signal by combining a signal derived from an in-phase component (DCSSI or ACSSI) with an in-phase sine wave (sin( act)) multiplied (144), the signal derived from the quadrature phase component (DCSSQ or ACSSQ) with the quadrature phase sine wave (sin( act)) are multiplied (DCSSQ or ACSSQ), and the results of the multiplications (144, 146) are added (150) to form a combined RF signal (CRF); Wherein said plurality of different low data rate channels is three or more individual channels (channels 0, 1, 2, 3) including two or more phase shifts of 0, π/2 or π A single channel in radians, and one or more individual channels with a phase shift other than 0, π/2, or π radians. 2. The apparatus of claim 1, wherein the means for forming the combined complex spread spectrum signal comprises: Means (58) for forming respective complex spread spectrum signals of respective respective channels, wherein each complex spread spectrum signal consists of an in-phase component (DSS0I, DSS1I, DSS2I and DSS3I) and a quadrature phase component (DSS0Q, DSS1Q, DSS2Q and DSS3Q); and means (56b, 156b) for additively combining signals (56a, 156a) derived from respective complex spread spectrum signals of respective respective channels to form a combined complex spread spectrum signal; where the signals derived from the individual complex spread spectrum signals of one or more channels (channels 2 and 3) with phase shifts other than 0, π/2 and π radians are scaled from two (56a, 156a) are obtained by scaling the signals derived from the complex spread spectrum signals of channels (channels 0 and 1) with phase shifts of 0, π/2 and π or more. 3. The apparatus of claim 2, wherein the means (56b) for additively combining is configured such that the in-phase (DCSSI or ACSSI) and quadrature-phase components (DCSSQ or ACSSQ) of the combined spread spectrum signal are respectively received from the phase either, but not both, of the in-phase or quadrature-phase components of the respective complex spread-spectrum signals of the channels (channels 0 and 1) whose shift is 0, π/2, or π are components of both derived signals, while they receive Components of signals derived from both in-phase or quadrature-phase components of respective complex spread spectrum signals of channels (channels 2 and 3) with a phase shift other than 0, π/2, or π. 4. The device of claim 2, wherein said phase shift that is not 0, π/2 or π is π/4 or 3π/4, and the fractional scale factor has an absolute value 5. The device of claim 3, wherein said phase shift that is not 0, π/2 or π is π/4 or 3π/4, and the fractional scale factor has an absolute value 6. The device of claims 1-5, wherein the in-phase (DSSOI, DSS1I, DSS2I and DSS3I) and quadrature phase components (DSS0Q, DSS1Q, DSS2Q and DSS3Q) of each complex spread spectrum signal of each single channel are sent to respectively Inputs to respective finite impulse response (FIR) filters (67, 68, 77, 78, 87, 88, 97 and 98) whose outputs are fed to means for additively combining (56b). 7. The apparatus of claims 1-5, wherein the in-phase (DCSSI) and quadrature phase (DCSSQ) components of the complex spread spectrum signal of the combination are fed to the input terminals of the finite impulse response (FIR) filters (167 and 168) , the filter output is sent to means for upconversion (144, 146 and 150). 8. A transmission method for a high data rate multi-channel link in a digital code division multiple access (CDMA) wireless communication system, wherein a plurality of different low data rates represented in the transmitted combined RF signal (CRF) The rate traffic channel representation is distributed over a set of sinusoids with phase offsets in the range 0-π, consisting of: forming a combined complex spread-spectrum signal consisting of an in-phase component (DCSSI or ACSSI) and a quadrature-phase component (DCSSQ or ACSSQ) in such a way as to introduce all phase shifts required for the indicated channel; and Frequency upconversion (144, 146, 150) is performed on the combined complex spread spectrum signal by combining the signal derived from the in-phase component (DCSSI or ACSSI) with the in-phase sine wave (sin( act)) multiplied (144), the signal derived from the quadrature phase component (DCSSQ or ACSSQ) with the quadrature phase sine wave (sin( act)) are multiplied (DCSSQ or ACSSQ), and the results of the multiplications (144, 146) are added (150) to form a combined RF signal (CRF); Wherein said plurality of different low data rate channels is three or more individual channels (channels 0, 1, 2, 3) including two or more phase shifts of 0, π/2 or π A single channel in radians, and one or more individual channels with a phase shift other than 0, π/2, or π radians. 9. The method of claim 8, wherein said operation of forming a combined complex spread spectrum signal comprises: forming respective complex spread spectrum signals for respective channels, wherein each complex spread spectrum signal is composed of an in-phase component (DSS0I, DSS1I, DSS2I and DSS3I) and a quadrature phase component (DSS0Q, DSS1Q, DSS2Q and DSS3Q); and additively combining (56a, 156a) signals derived from respective complex spread spectrum signals of respective respective channels to form a combined complex spread spectrum signal; where the signals derived from the individual complex spread spectrum signals of one or more channels (channels 2 and 3) with phase shifts other than 0, π/2 and π radians are scaled from two or more channels (channels 0 and 1) with phase shifts of 0, π/2, and π are derived by scaling the signals derived from a single complex spread spectrum signal. 10. The method of claim 2, wherein said operation for additively combining (56b, 156b) is configured as such: the in-phase and quadrature-phase components of the combined spread spectrum signal are received from phase shifts of 0, π respectively Either, but not both, of the in-phase or quadrature-phase components of a single complex spread-spectrum signal for channels (channels 0 and 1) of /2 or π are part of both derived signals, while they are received from phase shifts other than 0 , π/2, or π channels (channels 2 and 3) of all derived signal components of the in-phase or quadrature-phase components of a single complex spread spectrum signal.

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