CN1292259C - Electronic watthour meter and power-associated quantity calculation circuit - Google Patents

Abstract

The present invention relates to an L-multiple sampling frequency compensation device used for tracing the sampling frequency of AD conversion to the power supply frequency and a voltage controlling oscillator (VCO). The present invention controls the sampling frequency of an AD converter with high precision and obtains a power correlation quantum with high precision as a result. The needed reactive power by the power supply efficiency can be measured and mastered with high precision especially in the mode containing higher-order harmonics, and a power factor of an index for effective utilization of the power can be obtained with high precision.

Description

Electronic watt-hour meter and power-related calculation circuit

technical field

The present invention relates to an electronic electric energy meter and a power correlation operation circuit for measuring electric energy, in particular to an electronic energy meter capable of calculating active power, active electric energy, reactive power, reactive electric energy, apparent power, distorted power, effective value of current, effective value of voltage, current An electronic watt-hour meter and a power-related calculation circuit for power-related quantities such as the phase difference of the voltage and the higher harmonic values of the above items.

Background technique

Existing Example 1

Existing electronic watt-hour meter, for example, with reference to Japanese Patent No. 3080207, is made wherein as the voltage sensor (PT) and the current sensor (CT) measure voltage and the analog quantity of electric current is converted into the digital value device, has the first And the second successive comparison type AD converter, and these digital outputs (voltage and current) are calculated by the multiplier, and the power W is obtained.

The first and second successive comparison AD converters usually quantize the output of the analog input signal to equal

The resolution discretely increases the digital value, so in order to obtain absolute digital conversion accuracy for low-level input signals, a high-resolution successive comparison AD converter is required.

On the other hand, as a method of improving the accuracy of digital conversion, it is known to increase the sampling frequency of the first and second successive comparison type AD converters ("up-sampling"). For example, when the sampling frequency is increased to 128 times the frequency determined by the Nyquist theorem, the quantization noise spreads over a wide frequency band, thereby reducing the vector level of each frequency component and improving the noise level of the signal frequency component. This is equivalent to increasing the resolution of the first and second successive comparison AD converters by several bits.

However, in order to obtain a high-precision electronic galvanometer using the above-mentioned prior art example 1, high-resolution first and second successive comparison AD converters and multi-bit input multipliers are required, so that the circuit structure is complicated and the cost increases. In particular, it becomes a very unfavorable condition when it is desired to realize mass production by monolithic IC.

Existing example 2

In the above-mentioned Japanese Patent No. 3080207, as an example for solving the above-mentioned problems, the method disclosed is: after integrating the current and the voltage respectively with an integrator, a digital value is output by a comparator, and the digital output is delayed at the same time, And the value obtained by D/A conversion is fed back to the input terminal of the above-mentioned integrator.

This case has 1st and 2nd delta-sigma AD modulators that quantize the current and voltage respectively with upsampling frequency, and 1st and 2nd moving average processing with digital filter for moving average the quantized current and voltage respectively A device, a multiplication device that multiplies the current and voltage after moving average processing to obtain a power value, and an accumulation device that accumulates the power values obtained by multiplication.

In this way, according to the conventional example 2, noise in the low frequency band can be significantly reduced. That is, compared with the successive AD converter, the effective output bits of the AD converter are substantially increased, so a high-precision electronic energy meter can be obtained with a simple circuit structure, and the circuit can be simplified especially when a single-chip IC is used. change.

However, as mentioned above, the existing electronic watt-hour meters require real-time measurement (or display) of electric energy, so each sampling timing (the sampling frequency of the AD converter is fixed) directly multiplies the current and voltage to calculate the electric energy. The high-frequency component input signal (current and voltage) above the Nyquist frequency is obtained through a low-pass filter to calculate the electric energy. The electric energy calculated in this way is the electric energy synthesized by the fundamental wave and the high-order harmonic, and it is not possible to measure only the fundamental wave or only the high-order harmonic. Can not measure reactive power and higher harmonic reactive power.

In addition, many power devices in recent years use thyristors and inverters, and the current often contains high-order harmonics, so the measurement of various power-related quantities containing high-order harmonics is required.

Here, if the well-known Fourier transform that detects the high-order harmonic component is used for calculation, then the existing energy meter can also be logically considered to be able to measure the high-order harmonic component. However, the sampling frequency of the AD converter is not a natural multiple of the power supply frequency, so the adjacent times (Fourier transform results, the K-order harmonic component becomes the K+1-order harmonic component) also include the Fourier transform of current and voltage transform value.

Therefore, in order to obtain high-order harmonic components and make the sampling frequency of the AD converter fixed, it is necessary to use high-order harmonic correction operations of the power supply frequency. For this correction calculation, not only the detection of the power supply frequency but also the calculation itself is required, and therefore an expensive processor (CPU) is required especially as an electronic energy meter requiring real-time performance.

At this time, as a component different from the application of the electronic watt-hour meter, the same component as a high-precision and expensive measuring device equipped with a high-speed sampling frequency AD converter and a high-speed processor is required, so it is commonly used as a civilian product. It cannot be used in electronic energy meters.

On the other hand, when the measurement is performed using the successive upsampling of the AD converter of the conventional example 1 (or when the delta-sigma AD converter of the conventional example 2 is used) as an example, the upsampling frequency is several hundred kHz ~ tens of MHz (>> several kHz), while the output clock resolution of the CPU is several MHz to tens of MHz, so the sampling frequency cannot be controlled with high precision from the COU.

In addition, as described in the patent application (PCT2002JP00045: unpublished) previously filed by the inventors, the active power and reactive power of the fundamental wave can be calculated by rotating the current using the Hilbert transform. However, the frequency range that can be rotated by 90 degrees with the same Hilbert transformer is narrow (the Hilbert transformer that can be rotated by 90 degrees under the fundamental wave is rotated by an angle that deviates from 90 degrees), so it is difficult to use one The Hilbert transformer obtains the reactive power other than the fundamental wave.

The existing electronic watt-hour meter and the power-related quantity calculation circuit are composed as above. In order to obtain high precision with the electronic galvanometer of the existing example 1, high-resolution first and second successive comparison type AD converters and Since a multiplier with multiple bits is input, there is a problem that the circuit configuration is complicated, resulting in an increase in cost.

The existing electronic energy meter directly multiplies the current and the voltage at each sampling timing to calculate the electric energy and calculate the electric energy synthesized by the fundamental wave and the higher harmonic, so it cannot only measure the fundamental wave or the higher harmonic, and it cannot Measure reactive power and higher harmonic reactive power. There is a problem.

When using the known Fourier transform to measure high-frequency components, the sampling frequency of the AD converter is not a natural multiple of the power supply frequency, and adjacent orders also include Fourier transform values, which need to be corrected by high-order harmonics of the power supply frequency Therefore, an expensive processor is required, and there is a problem that it cannot be used in a general-purpose electronic energy meter.

When measuring by up-sampling successive AD converters or using Δ-Σ AD converters, since the up-sampling frequency is very high (hundreds of kHz to several MHz), the resolution of the clock output by the CPU (several MHz to several Ten MHz) cannot control the sampling frequency with high precision, and it is necessary to track the sampling frequency of the AD converter according to the natural multiple of the power frequency, and perform correction operations. There is a problem.

When using the Hilbert transform to rotate the current to calculate the active power and reactive power of the fundamental wave, since the frequency range that can be rotated by 90 degrees with the same Hilbert transform is narrow, there is a possibility that one Hilbert transform cannot be used. The problem of calculating the reactive power other than the fundamental wave of the converter.

The present invention was made to solve the above-mentioned problems, and an object of the present invention is to provide an electronic watt-hour meter and a power-related quantity calculation circuit that can obtain electric energy with improved sampling frequency control precision.

Another object of the present invention is to provide an electronic energy meter and a power-related calculation circuit with simple structure and capable of measuring not only fundamental wave electric energy but also high-order harmonic electric energy.

In particular, the object of the present invention is to provide an electronic watt-hour meter and a power-related quantity calculation that can measure reactive power required to check power supply efficiency with high precision and include high-order harmonics, and obtain power factor, which is an index of effective power utilization, with high precision. circuit.

Contents of the invention

An electronic watt-hour meter according to the present invention includes: an AD converter for converting measurement signals representing current and voltage of a power supply line into digital values and taking them in; The microprocessor of the power calculation device used; the electronic energy meter also has: a power frequency detection device for detecting the power frequency of the power line, which is set separately from the clock circuit of the microprocessor and controlled according to the power frequency The sampling frequency control device of the sampling frequency of described AD converter is corresponding to the voltage-controlled oscillator that the sampling frequency of described AD converter is changed by the control voltage from described sampling frequency control device; Said sampling frequency control device controls all The sampling frequency is set to be a natural multiple of the power supply frequency, and the electric energy calculation device includes a high-order harmonic calculation unit for calculating high-order harmonic electric energy. Therefore, the sampling frequency control precision can be improved, and electric energy can be acquired with high precision.

The sampling frequency control device is structured to control the sampling frequency so that the sampling frequency is a natural multiple of the power frequency, and the power computing device includes a high-order harmonic computing unit for computing high-order harmonic electric energy. Therefore, the electric energy of the fundamental wave and higher harmonics can be measured with a simple structure.

The sampling frequency control device is structured to control the sampling frequency, so that the sampling frequency is the power of 2 of the power supply frequency, and n is a natural number. The power calculation device includes a high-speed Fourier transform device, and uses a high-speed Fourier transform (FFT) Higher harmonic power. Therefore, calculation speed can be increased, and real-time performance is excellent.

The AD converter is composed of a delta-sigma AD converter that needs upsampling, so it has high precision and is suitable for monolithic IC.

The sampling frequency control means includes a zero-cross point detection means for detecting a zero-cross point at which the power supply frequency rises or falls, and a power supply phase difference detection means for detecting a zero-cross point at the previous (last 1) rise or fall zero-cross point of the power supply frequency After one cycle has elapsed, it is detected from the AD conversion value of the power supply frequency whether the power supply frequency is late or advanced, and as the power supply phase difference, there is also a device for controlling the sampling frequency of the AD converter based on the power supply phase difference. Therefore, the CPU load can be reduced and the structure can be simplified.

The sampling frequency control device includes an absolute phase detection device that detects the absolute phase of the voltage of the power line, and a voltage phase difference detection device that detects that the absolute phase is late or advanced as a voltage phase difference at a time when one cycle has passed after the last absolute phase of the voltage , and a device for controlling the sampling frequency of the AD converter according to the voltage phase difference. Therefore, the influence of white noise can be suppressed.

The voltage phase difference detection device uses an absolute phase of 0 degrees, 90 degrees, 180 degrees or 270 degrees as a reference, and thus can widen the adjustment range.

Structurally, it is made into a sampling frequency control device, which has a sampling frequency compensating device for compensating the sampling frequency of the AD converter. The sampling frequency compensating device includes a control amount calculation device for obtaining the control amount of the sampling frequency, and the control amount of the sampling frequency is added to D/ D/A conversion means for A-converted output, and bias means for biasing the output voltage of the D/A conversion means. The bias means biases a predetermined frequency side of the power supply line by a predetermined voltage. Therefore, a simple D/A converter with a small number of bits can be used.

Since the bias voltage adjusting means for externally adjusting the predetermined voltage is provided, it can be used in various circuits, and a simple D/A converter with a small number of bits can be used.

The power calculation device includes an AD conversion data encapsulation device for encapsulating voltage and current measurement signals for one cycle of the power supply frequency, and a power calculation device for performing Fourier transform on the data encapsulated by the AD conversion data encapsulation device and calculating the first power value, The packaging time detection device that detects the packaging time required for the voltage and current required for one cycle of packaging, maintains the first power value at each packaging time, and outputs the first power value as the power output of the second power value every time a sampling command of a predetermined operation cycle is generated. device, and a power pulse output device that accumulates the second power value and outputs a power pulse every time the accumulated value of the second power value reaches a specified value. Therefore, real-time performance can be satisfied.

The electric energy calculation device includes an AD conversion data packaging device that encapsulates the voltage and current measurement signals for one cycle of the power supply frequency, performs Fourier transform on the data packaged by the AD conversion data packaging device, and calculates the power-related value of each higher harmonic (Current, voltage, active power, reactive power) and the power calculation unit of the phase difference of the power-related value, and the phase compensation device, which makes the phase difference of the power-related value become the real power of the measured party through the rotation calculation. The relative value phase difference compensates the current or voltage for one cycle. Therefore, measurement accuracy can be improved.

The power-related quantity calculation circuit of the present invention includes an AD converter for converting the measurement signal representing the current and voltage of the power supply line into a digital value and receiving it, and an AD conversion data packaging device for packaging the digital value per cycle. Data encapsulation device The Fourier transform device that performs Fourier transform on the data encapsulated, and the power-related quantity calculation device that calculates the power-related quantity based on the transformation result of the Fourier transform device, calculates the sampling of the AD converter based on the frequency of the current or voltage A sampling frequency compensation device for the frequency compensation amount, and a voltage-controlled oscillator for outputting the sampling frequency changed with the compensation amount to the AD converter. Therefore, the sampling frequency control precision can be improved, and electric energy can be acquired with high precision.

Like the circuit structure above, the present invention uses a sampling frequency compensator and a voltage-controlled oscillator (VCO) to track the sampling frequency of AD conversion to L times the power supply frequency (L is a natural number) to control the sampling frequency of the AD converter with high precision. , so the power-related quantity can be obtained with high precision. In particular, it is possible to accurately measure the reactive power required to grasp the power supply efficiency including harmonics, and to obtain the power factor, which is an index of effective power utilization, with high accuracy.

Description of drawings

Fig. 1 is an explanatory diagram of the calculation logic of power-related quantities according to Embodiment 1 of the present invention, in which vector diagrams of voltage and current are shown as examples.

Fig. 2 is a block diagram showing the circuit composition of key parts of the electronic energy meter according to the first embodiment of the present invention.

Fig. 3 is an explanatory diagram showing the fluctuation of the fundamental wave voltage when the power supply frequency changes in the electronic energy meter according to Embodiment 1 of the present invention.

Fig. 4 is an explanatory diagram showing the zero-cross point of the power supply voltage and the sampling start point in the electronic energy meter according to Embodiment 1 of the present invention.

Fig. 5 is a block diagram showing the circuit composition of key parts of the electronic watt-hour meter according to Embodiment 2 of the present invention.

Fig. 6 is an explanatory diagram of compensation processing for tracking the sampling frequency of the AD converter to a natural multiple of the power supply frequency in Embodiment 2 of the present invention.

Fig. 7 is an explanatory view showing a sampling point of an electronic energy meter according to Embodiment 3 of the present invention.

Fig. 8 is a block diagram showing the circuit composition of key parts of the electronic watt-hour meter according to Embodiment 4 of the present invention.

Fig. 9 is an explanatory diagram showing an operation example of biasing the VCO control voltage by the electronic watt-hour meter according to Embodiment 4 of the present invention.

Fig. 10 is a block diagram showing the composition of a voltage adding circuit that can further adjust the VCO control voltage by using the electronic watt-hour meter according to Embodiment 4 of the present invention.

Fig. 11 is a block diagram showing the circuit composition of key parts of the electronic watt-hour meter according to Embodiment 5 of the present invention.

Fig. 12 is a timing chart showing the energy pulse output of the electronic watt-hour meter according to Embodiment 5 of the present invention.

Fig. 13 is a block diagram showing the circuit composition of key parts of the electronic watt-hour meter according to Embodiment 6 of the present invention.

The best form for carrying out the invention

Embodiment 1

Embodiment 1 of the present invention will be described below.

The present invention is different from the calculation of the existing electronic energy meter, and uses Fourier transform (preferably FFT) of the data of one cycle to calculate electric energy and the like.

First, the operation logic used in Embodiment 1 of the present invention will be briefly described.

When the sampling frequency of the AD converter is a natural multiple of the power supply frequency, Fourier transform can be used to measure the following power-related quantities.

The sampling frequency of the AD converter is L times of the power supply frequency (L is a natural number), and the higher harmonic is n times. When n=0, it is a direct current (DC) component; when n=1, it is a fundamental wave (1st harmonic). At this time, the maximum harmonic H that can be obtained by Fourier transform is represented by the following equation (1).

H＝floor(L/2)        (1) Among them, in formula (1), floor() is a function to remove the decimal point, so H is a natural number.

Perform Fourier transform on the data of point L obtained by AD transformation, and then obtain complex values with amplitude and phase information from the direct current component to the H harmonic component. Let the nth order harmonic complex values obtained by Fourier transforming the AD conversion data L points of the voltage and current be Vn_cmp and In_cmp respectively, which can be represented by the following equations (2) and (3).

Vn_cmp＝Vn_re+j·Vn_im (2)

In_cmp＝In_re+j·In_im (3)

Wherein, in formula (2) and formula (3), j is an imaginary number unit, and "·" is a multiplication symbol. Vn_re, Vn_im, In_re, and In_im are real numbers. Below, the value after Fourier transform is the value normalized by the effective value.

Here, the active power Wn of the nth harmonic is obtained by the following equation (4).

Wn＝Vn_re·In_re+Vn_im·In_im (4)

The power W including all harmonics is obtained by the following equation (5).

$\mathrm{<span\; class="notranslate">\; W</span>}<span\; class="notranslate">\; =</span>\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}}{\overset{\mathrm{<span\; class="notranslate">\; h</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\mathrm{<span\; class="notranslate">\; W</span>}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 5</span>}<span\; class="notranslate">\; )</span>$

Here, the reactive power varn of the nth harmonic is obtained by the following equation (6).

Varn＝Vn_im·In_re-Vn_re·In_im (6)

At this time, if the reactive power is positive, the current will lag behind the voltage; if the reactive power is negative, the current will lead the voltage. In addition, the reactive power including all harmonics is obtained by the following equation (7).

$\mathrm{<span\; class="notranslate">\; Var</span>}<span\; class="notranslate">\; =</span>\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}}{\overset{\mathrm{<span\; class="notranslate">\; h</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\mathrm{<span\; class="notranslate">\; Varn</span>}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 7</span>}<span\; class="notranslate">\; )</span>$

Use the following formula (8) to find the voltage effective value Vrmsn of the nth harmonic.

$\mathrm{<span\; class="notranslate">\; Vrmsn</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; vn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; vn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; vn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; vn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 8</span>}<span\; class="notranslate">\; )</span>$

The voltage effective value Vrms including all harmonics is obtained by the following formula (9).

$\mathrm{<span\; class="notranslate">\; Vrms</span>}<span\; class="notranslate">\; =</span>\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}}{\overset{\mathrm{<span\; class="notranslate">\; h</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\mathrm{<span\; class="notranslate">\; Vrmsn</span>}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 9</span>}<span\; class="notranslate">\; )</span>$

Use the following formula (10) to find the current effective value Irmsn of the nth harmonic.

$\mathrm{<span\; class="notranslate">\; Irmsn</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 10</span>}<span\; class="notranslate">\; )</span>$

Use the following formula (11) to find the current effective value Irms including all harmonics.

$\mathrm{<span\; class="notranslate">\; Irms</span>}<span\; class="notranslate">\; =</span>\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}}{\overset{\mathrm{<span\; class="notranslate">\; h</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}}\mathrm{<span\; class="notranslate">\; Irmsn</span>}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 11</span>}<span\; class="notranslate">\; )</span>$

The apparent power VAn of the nth harmonic is obtained by the following equation (12).

VAn＝Vrmsn·Irmsn (12)

The apparent power VA including all harmonics is obtained by the following equation (13).

VA＝Vrms·Irms (13)

If only fundamental waves exist in the power supply voltage and current, the relationship between active power W, reactive power, and apparent power VA satisfies the following equation (14).

$\mathrm{<span\; class="notranslate">\; VA</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; W</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; W</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; Var</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; Var</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 14</span>}<span\; class="notranslate">\; )</span>$

However, when higher harmonics exist, the above formula (14) does not hold, and distortion power exists. The distortion power D is obtained by the following equation (15).

$\mathrm{<span\; class="notranslate">\; D.</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; VA</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; VA</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; W</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; W</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; Var</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; Var</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 15</span>}<span\; class="notranslate">\; )</span>$

Distortion power D is a positive real number, not a negative value.

The nth harmonic voltage and current complex values Vn_cmp and In_cmp obtained by Fourier transform are taken as vector diagrams, as shown in Figure 1. In FIG. 1 , the horizontal axis is the real axis, and the vertical axis is the imaginary axis.

In Fig. 1, θ is the absolute phase of the complex value Vn_cmp, and φ is the absolute phase of the complex value In_cmp. Here, the reference of the absolute phase is the positive direction of the horizontal axis, and the counterclockwise direction represents leading (positive), and the clockwise direction represents lagging (negative).

Using the complex value Vn_cmp as a reference, and the phase difference (φ-θ) between Vn_cmp and In_cmp is Phase_VnIn, the phase difference Phase_VnIn is obtained by the following equation (16).

if varn >= 0

Phase_VnIn＝-arccos(Wn/VAn)

else (16)

Phase_VnIn＝arccos(Wn/VAn)

The phase difference Phase_VnIn is "positive" when the current leads the voltage, and "negative" when it is late. The range of the phase difference Phase_VnIn is ±180 degrees.

In the above, only the single-phase two-wire system was described, but the single-phase three-wire system, three-phase three-wire system, and three-phase four-wire system can also be found. From the added value obtained from each phase and the added power, the active power, reactive power, apparent power and distortion power can be obtained respectively by using formula (16).

It is also possible to recalculate the phase between the voltages of each phase. That is, first, the nth order harmonic voltage Van_cmp of the A phase is expressed in the following formula (17).

Van_cmp＝Van_re+j·Van_im (17)

Also, the B-phase nth harmonic voltage Vbn_cmp is expressed in the following equation (18).

Vbn_cmp＝Vbn_re+j Vbn_im (18)

At this time, the phase difference Phase_VanVbn based on the A phase is obtained by the following equation (19).

Vabn=Van_re·Vbn_re+Van_im·Vbn_im

Vabn'=Van_im·Vbn_re-Van_re·Vbn_im

$\mathrm{<span\; class="notranslate">\; Varmsn</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; Van</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; Van</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; Van</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; Van</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}}$

$\mathrm{<span\; class="notranslate">\; Vbrmsn</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; Vbn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; Vbn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; Vbn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; Vbn</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}}$

if Vabn'>=0

$\mathrm{<span\; class="notranslate">\; Phase</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; Van\; Vbn</span>}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; arccos</span>}<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; Vabn</span>}}{\mathrm{<span\; class="notranslate">\; Varmsn</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; Vbrmsn</span>}}<span\; class="notranslate">\; )</span>$

else

$\mathrm{<span\; class="notranslate">\; Phase</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; Van\; Vbn</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; arccos</span>}<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; Vabn</span>}}{\mathrm{<span\; class="notranslate">\; Varmsn</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; Vbrmsn</span>}}<span\; class="notranslate">\; )</span>$

(19)

In formula (19), Vabn represents the virtual active power between ab phases, and Vabn' represents the virtual reactive power between ab phases.

Even for voltages and currents with different phases, the phase difference can be calculated by the same method.

In this way, if the sampling frequency of the AD converter is a natural multiple of the power supply frequency, then the calculation structure of the Fourier transform can be used to obtain all the power-related quantities including high-order harmonics or contain the required high-order harmonics. The amount of power correlation of the harmonics.

Especially for obtaining reactive power for all high-order harmonics, this should be highlighted. In addition, if the sampling frequency of the AD converter is N times the power frequency of 2 (N is a natural number), then the Fourier transform operation can use FFT. Therefore, the sampling frequency of AD conversion is usually a natural number multiple of the power supply frequency, and it is better to be 2 to the Nth power.

The specific operation of the electronic energy meter according to Embodiment 1 of the present invention will be described below.

Here, as the VCO control method according to Embodiment 1 of the present invention, a method of performing locking using the zero-cross point of the power supply voltage will be described.

Fig. 2 is a block diagram showing the specific circuit composition of the electronic watt-hour meter according to Embodiment 1 of the present invention.

In FIG. 2, an AD converter 1 converts a voltage V and a current I detected by a sensor (not shown) into digital values. The output terminal of the AD converter 1 is sequentially connected to the AD data encapsulation device 2 , the Fourier transform device 3 and the power-related quantity calculation device 4 .

The AD data encapsulation device 2 encapsulates voltage and current measurement signals for one cycle of the power supply frequency. This device 2 has the function of detecting the power frequency on the power line and means of detecting the zero crossing point of the rising or falling power frequency.

Fourier transform means 3 performs Fourier transform. The power-related quantity calculation device 4 (electric energy calculation device) functions as a harmonic calculation unit for calculating a power-related quantity including harmonics.

The data packaged by the AD conversion data packaging device 2 is input to a voltage controlled oscillator (hereinafter referred to as VCO) for controlling the AD converter 1 through the sampling frequency compensating device 5 .

The voltage V and current I input to the AD converter 1 are the output of the sensor, not the physical voltage and current itself on the power supply, but the values input by the AD converter 1 . An anti-aliasing filter, an operational amplifier for amplification, etc. (not shown in the figure) are sometimes provided at the input end of the AD converter 1 .

The AD conversion data encapsulation device 2 collects the AD conversion data for one cycle of the power supply for each L point.

The Fourier transform device 3 performs Fourier transform on the AD converted data (L points) for one power cycle input from the AD converted data encapsulating device 2 every L points (per cycle).

The power-related quantity calculating means 4 calculates the power-related quantity from complex values of the voltage and current after Fourier transform calculation processing.

The sampling frequency compensation device 5 controls the voltage output to the VCO6 based on the data input from the AD conversion data encapsulation device 2, so that the AD conversion sampling frequency is locked to a natural number L times of the power supply frequency.

VCO6 converts the voltage output of the sampling frequency compensation device into a clock output. The clock output from VCO6 to AD converter 1 is equal to the sampling frequency when the AD converter 1 is a successive AD converter 1, and equal to the up-sampling frequency in the case of a delta-sigma AD converter. But when the AD converter 1 has the clock signal frequency division function, VCO6 provides the clock output before frequency division.

Next, the operation of Embodiment 1 of the present invention shown in FIG. 2 will be described with reference to the explanatory diagrams of FIGS. 3 and 4. FIG.

First, the sampling frequency compensating device 5 will be described based on FIG. 3 . Like FIG. 1 , FIG. 3 shows a vector space composed of a real axis (horizontal axis) and an imaginary axis (vertical axis). As shown in Fig. 3, the phase of the fundamental wave is late when the power frequency is late, and advanced when it is leading, so the AD conversion sampling frequency of the example VCO6 is controlled so that the phase of the fundamental wave is advanced when the phase of the fundamental wave is leading, and it is late when it is late. The simplest method of performing this control is feedback control.

Here, let the feedback coefficient be ε, the VCO control voltage performed at the mth time is Vcntrl_m, and the phase difference between the mth voltage fundamental wave and the m+1th voltage fundamental wave is Ψ (leading is positive), then use the following formula (20) represents the VCO control voltage Vcntrl_m+1 performed for the m+1th time.

Vcntrl_m ₊₁ = _{Vcntrl_m} +ε·Ψ (20)

Here, the greater the value of the VCO control voltage, the higher the clock frequency. The feedback coefficient ε (ε > 0) is determined by VCO6, tracking speed, etc. using appropriate values for control.

When the phase difference is used as it is as the error amount for the feedback of the above formula (20), trigonometric function calculations for the phase are required, which increases the amount of calculation. Therefore, the phase difference Ψ has a one-to-one correspondence and monotonically increases or decreases. The magnitude of the relationship is used as the error magnitude. If the error amount Error is sampled, the above formula (20) can be expressed by the following formula (21).

Vcntrl_m ₊₁ ＝ _{Vcntrl_m} -ε·Error (21)

Here, the error amount Error has a positive value when the power supply frequency is late (phase late), and has a negative value when the power supply frequency is advanced (phase lead). Therefore, when the phase difference increases monotonically, the sign is reversed, and when it decreases monotonically, the original value can be used as the error amount Error.

In order to reduce the amount of calculation, the error amount Error is replaced by the amount other than the phase, and the following processing related to frequency tracking is performed.

Usually, a D/A converter is used to generate the voltage output of the VCO control voltage Vcntrl (sampling frequency compensation device 5).

This is because, for example, when using PWM (Pulse Width Modulation) output, a low-pass filter is required, and thus the tracking speed becomes slow. A low-pass filter can be inserted before the D/A conversion, so that the VCO control voltage Vcntrl does not oscillate, but the tracking speed also becomes slower at this time.

Here, the most basic feedback control is used as an example to explain, as long as the VCO control voltage can be determined according to the error amount Error, any feedback can be used.

FIG. 4 is an explanatory diagram showing power supply frequency, in which the horizontal axis represents time and the vertical axis represents power supply amplitude.

When the sampling frequency of the AD converter 1 is k times the power supply frequency (a natural number multiple of the power supply frequency and a multiple of the N power of 2), it is controlled so that one voltage (AD conversion value) obtained by sampling at L points is a rising zero-crossing point .

At this time, if the lock is complete (the voltage frequency does not fluctuate), the AD conversion value of the voltage is "0" every time. However, the AD conversion value is a positive value when the power supply frequency is advanced, and a negative value when it is late (refer to FIG. 4 ).

The error amount Error has a one-to-one correspondence with the phase difference in the range of 90 degrees and monotonously increases, so that the AD conversion value of each selected sampling point can be a value whose sign is inverted.

However, in order to lock at a frequency that is a natural multiple of the sampling frequency or a frequency that is a fraction of a natural number, it is necessary to limit the VCO control voltage Vcntrl to a value that is 1/2 to 2 times the sampling frequency.

In Embodiment 1 of the present invention, the sampling frequency is set as a natural number multiple of the voltage frequency and is 2 to the Nth power, so 1/2 times the sampling frequency is also locked. The feedback amount "ε Error" in the above formula (21) also needs to be limited.

Here, the sampling frequency is locked at the rising point of the power supply voltage zero-cross point, but it can also be locked at the falling point. In this case, the AD conversion value itself is used as the error amount Error.

As above, by tracking the sampling frequency of the AD converter to a natural number multiple of the power supply frequency, the Fourier transform device 3 can directly obtain high-order harmonic components from the Fourier transform result, so it can be measured with a simple structure Fundamental wave electric energy and higher harmonic electric energy.

It can also be made structurally so that the VCO6 is combined with a sampling frequency of several kHz and a CPU clock of several MHz. The AD converter 1 does not need to have a particularly high precision. Therefore, the chip area is not large when a single-chip IC is used, which is better.

Oscillation frequency is controlled by VCO6, so compared with using CPU clock to directly control the sampling frequency of AD converter 1, the control precision can be improved, and the electric energy of higher harmonics can be measured with high precision.

As the AD converter 1, even if the Δ-Σ type is used, the up-conversion sampling frequency can be fine-tuned by the VCO6, so the measurement accuracy of the power-related quantity including electric energy is high, and it is suitable for a single-chip IC.

As described above, the sampling frequency of the AD converter 1 is set to be a natural number multiple of the power supply frequency and to be the Nth power of 2. Therefore, from the viewpoint of calculation speed, it is better to use FFT for the Fourier transform device 3. .

Using the above structure, it is possible to measure and calculate active power, active energy, reactive power, reactive energy, apparent power, distortion power, current RMS value, voltage RMS value, phase difference between current and voltage, or higher orders of these items Harmonic values and other power-related quantities that cannot be measured and calculated in the prior art.

For example, reactive power necessary for calculating power supply efficiency can be measured with high precision including harmonics, and power factor, which is an index of effective power utilization, can be acquired with high precision.

The sampling frequency compensating device 5 can be realized by the computing function of the CPU, and the sampling frequency compensating device 5 with the above-mentioned structure can be realized only by combining the VCO 6, so that the structure can be simplified.

The sampling frequency compensating device 5 performs locking based on the output data of the AD data encapsulating device 2 on the basis of the zero-crossing point (rising or falling), so that no complicated calculation occurs in the CPU, and the structure can be simplified.

The error amount Error in the above formula (21) is tracked by feedback control, so the tracking performance is good and the real-time performance is excellent.

Implementation form 2

In the control of VCO6, the absolute phase of the fundamental wave of FFT can also be locked at the same position,

Fig. 5 is a block diagram showing the circuit composition of key parts in Embodiment 2 of the present invention. As a VCO control method, this figure shows the case where the absolute phase of the FFT fundamental wave is locked at the same position. In FIG. 5 , the same parts as those described above (refer to FIG. 2 ) are attached with “A” after the same symbols, and description thereof will be omitted.

In this case, the Fourier transform device is connected with the sampling frequency compensation device 5A and VCO6A to form a sampling frequency control device, which includes a power supply voltage absolute phase detection device and a voltage phase difference detection device for absolute phase delay or advance. device.

The sampling frequency compensation device 5A outputs a VCO control voltage to the VCO 6A based on the data from the Fourier transform device 3A.

FIG. 6 is an explanatory diagram showing compensation processing for tracking the sampling frequency of the AD converter 1A to a natural number L times the power supply frequency according to Embodiment 2 of the present invention. It is the same as the above-mentioned FIG. 1 and FIG. axis) and the imaginary axis (vertical axis) of the vector space.

As mentioned above, if the sampling frequency of AD converter 1A is locked to the natural number L times of the power supply frequency, the absolute phase of the fundamental wave of the power supply will be the same value every time, but the power supply frequency will rotate in the backward direction when it is late, and it will rotate in the backward direction when it is ahead. Rotate in forward direction.

In the case represented by the following equation (22), consider the previous value V1_cmp_pre and the current value V1_cmp of the complex value after FFT calculation of the fundamental wave.

Vl_cmp_pre＝Vl_pre_re+j Vl_pre_im (22)

V1_cmp=V1_re+j·V1_im

At this time, the phase difference Phase_V1_Error is obtained by the following equation (23).

$\mathrm{<span\; class="notranslate">\; Phase</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">V</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; error</span>}$

$<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; arctan</span>}<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">V</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; pre</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">V</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">V</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; pre</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">V</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="im\; V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">im</figure-callout></span><figure-callout\; id="1"\; label="im\; V"\; filenames="C0280548500251.png,C0280548500331.png"\; state="\{\{state\}\}">\; </figure-callout>}}{\mathrm{<span\; class="notranslate">\; </span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; twenty\; three</span>}<span\; class="notranslate">\; )</span>$

Here, the range of the phase difference Phase_V1_Error is ±90 degrees.

Assuming that the voltage amplitude is approximately constant each time, the denominator of the formula (23) can be regarded as a constant. The numerator of formula (23) corresponds to the phase difference one by one, and decreases monotonously. Therefore, if the error amount Error is the numerator of the formula (23), the error amount Error can be expressed by the following formula (24).

Error＝V1_pre_im V1_re-V1_pre_re V1_im (24)

At this time, if there is a phase difference of ±90 degrees or more, the sign is reversed. And because it is locked at a natural multiple or a fraction of the sampling frequency, it is necessary to limit the VCO control voltage Vcntrl to a value that forms 1/2 to 2 times the sampling frequency. The amount of feedback (ε Error) also needs to be limited.

With the above configuration, the embodiment of the present invention has the following effects in addition to the effect of the above-mentioned embodiment 1 (locking at the rise or fall of the zero point).

That is, the phase locking of the voltage (or current) after the Fourier transform (FFT is preferred) is performed by the Fourier transform device, so compared with the locking at the zero-cross point of the voltage waveform as in the first embodiment, the white noise It is superior in improving accuracy when superimposed with high-order harmonics.

Implementation form 3

The above-mentioned embodiment 2 does not specifically mention the reference of the absolute phase, but the absolute phase of the fundamental wave of the FFT can be locked at 0 degrees, 90 degrees, 180 degrees, and 270 degrees.

Next, a VCO control method according to Embodiment 3 of the present invention will be described. In this case, the position of the fixed voltage fundamental wave coordinates (absolute phase of the fundamental wave of FFT) is selected as 0 degrees and 90 degrees. 180 degrees or 270 degrees.

Fig. 7 is an explanatory diagram showing the VCO control operation according to the third embodiment, and shows a vector space composed of a real axis (horizontal axis) and an imaginary axis (vertical axis), similarly to Figs. 1, 3 and 6 .

First, consider the case of locking at 0 degrees.

When the power frequency is late, the phase is late, so the real value V1_im in the above formula (2) is negative, and when the power frequency is advanced, the real value is positive.

The real value V1_im has a one-to-one correspondence with the phase difference in the range of ±90 degrees and has a monotonically increasing relationship. Therefore, the error amount Error can be represented by the following equation (25).

Error＝-V1_im (25)

Here, as in the above-mentioned second embodiment, it is also necessary to limit the VCO control voltage Vcntrl and the amount of feedback.

With this method, the amount of calculation of the error amount Error can be reduced compared to that of the second embodiment, and there is no need to store the last (last time) coordinate.

When the phase difference exceeds +90 degrees, the error amount Error does not increase monotonically, but the sign does not change. Therefore, setting the error amount Error to the following formula (26) can make the error amount Error monotonously decrease with respect to the phase difference within the range of ±180 degrees.

if V1_re≥0

Error＝-V1_im

else if V1_im≥0

Error＝-(2·Vrms1-V1_im) (26)

else

Error＝-(-2·Vrms1-V1_im)

The effective value Vrms1 of the voltage fundamental wave is usually almost constant and can be treated as a constant. Using this method, the feedback range of ±90 degrees can be ±180 degrees.

When locking at 90 degrees, 180 degrees, or 270 degrees, the amount of error can also be fixed.

That is, the error amount Error at 90 degrees can be represented by the following equation (27) as in the above equation (25).

Error＝V1_re (27)

The error amount Error at 180 degrees can be represented by the following equation (28).

Error＝V1_im (28)

The error amount Error at 270 degrees can be represented by the following equation (29).

Error＝-V1_re (29)

In addition, the error amount Error at 90 degrees can be represented by the following equation (30) as in the above equation (26).

if V1 im≥0

Error＝V1_re

else if V1_re≥0

Error＝2·Vrms1-V1_re (30)

else

Error＝-2·Vrms1-V1_re

The error amount Error at 180 degrees can be represented by the following equation (31).

if V1_re≤0

Error＝V1_im

else if V1_im≥0

Error＝2·Vrms1-V1_im (31)

else

Error＝-2·Vrms1-V1_im

The error amount Error at 270 degrees can be represented by the following equation (32).

if V1_im≤0

Error＝-V1_re

else if V1_re≥0

Error＝-(2 Vrms1-V1_re (32)

else

Error＝-(-2·Vrms1-V1_re)

As above, the absolute phase of the FFT fundamental wave is locked at 0 degrees, 90 degrees, 180 degrees, and 270 degrees. In addition to the effect of the second embodiment, the feedback range of ±90 degrees can be expanded to ±180 degrees. The adjustment range can be enlarged.

Embodiment 4

Embodiments 1 to 3 above do not mention the number of bits of the D/A converter associated with the sampling frequency compensation device, but a D/A converter with a small number of bits can be used.

Fig. 8 is a block diagram showing the circuit composition of key parts of Embodiment 4 of the present invention, wherein the same parts as those described above (Fig. 2 and Fig. 5) are added with "B" after the same symbols, and detailed description is omitted.

In FIG. 8, only the peripheral part of the sampling frequency compensator 5B and VCO 6B related to the D/A converter with a small number of bits are shown.

In this case, the sampling frequency compensation device 5B has a D/A converter with a small number of bits.

An attenuator 51 and an adder 52 are inserted between the sampling frequency compensation device 5B and the VCO 6B. Adder 52 is associated with a low number of bits D/A to set the biased VCO control voltage.

The adder 52 outputs the VCO control voltage obtained by adding the offset voltage VOFF to the VCO 6B, so that the VCO control voltage becomes a predetermined frequency (for example, 60 Hz) of the power supply. This makes it possible to finely control the frequency per bit of the D/A converter.

When the clock frequency output by VCO6B is high, in order to finely control the sampling frequency, the D/A converter requires many bits. Therefore, when the output of the D/A converter is "0", the clock frequency of VCO6B is controlled to be the specified frequency (60 Hz) of the power supply.

The adder 52 adds the bias voltage VOFF so that the clock frequency of the VCO 6B becomes the predetermined frequency of the power supply.

Fig. 9 is an explanatory diagram showing the control operation of Embodiment 4 of the present invention, in which the output voltage of the D/A converter (sampling frequency compensating means 5B), the output voltage of the attenuator 51, the VCO control voltage and the bias voltage are shown VOFF relationship.

As shown in Figure 9, by controlling the output voltage of the D/A converter around the bias voltage VOFF, the number of bits required by the D/A converter can be suppressed, and a cheap and small D/A converter can be used with high precision. measure power-related quantities.

Next, the case where the VCO control voltage can be further adjusted structurally will be described. FIG. 10 is a block diagram specifically showing the composition of the attenuator 51 and the adder 52 in FIG. 8 , which shows the situation that the adder 52 can adjust the offset voltage VOFF in structure.

In FIG. 10 , the attenuator 51 is structured to have a resistor R1 and a variable resistor R2 for dividing the output voltage of the D/A converter. By adjusting the divided voltage, the control range of the VCO control voltage can be adjusted.

The adder 52 has a resistance R3 and a variable resistance R4 for dividing the bias voltage VOFF, and a divided voltage of the D/A output voltage (the output voltage of the attenuator 51) and a divided voltage of the bias voltage VOFF. A voltage addition circuit 52B that adds and outputs the VCO control voltage.

The variable adjustment parts of the variable resistors R2 and R4 are set outside the electronic energy meter, which can be adjusted arbitrarily from the outside.

With the structure of Fig. 10, not only the range of the VCO control voltage, but also the actual added bias voltage can be adjusted.

The range of the VCO control voltage corresponds to the length of the arrow of the VCO control voltage in FIG. 9 , and the voltage range at 45 Hz to 66 Hz is assigned to, for example, an 8-bit D/A converter.

The offset voltage VOFF is equivalent to the offset from 0V in FIG. 9 (refer to the arrow), and can be adjusted arbitrarily at this time.

That is, adjusting the variable resistor R2 in the attenuator 51 changes the range of the VCO control voltage, and adjusting the variable resistor R4 in the adder 52 changes the actual input bias voltage of the voltage adding circuit 52B.

In this way, the control range of the D/A converter (or the bias amount of the bias voltage) can be changed, so that the power-related quantity can be measured with high precision according to the applied circuit, and when the predetermined frequency is changed, it can also be used cheaply and Small D/A converter supports.

Embodiment 5

Although the above-mentioned Embodiments 2 to 4 are not described in detail, as shown in FIG. 11, an electric energy pulse output device 7 for pulse outputting electric energy may be provided at a subsequent stage of the power-related quantity calculation device 4C.

Fig. 11 is a block diagram showing the circuit composition of the key parts of Embodiment 5 of the present invention, and the same parts as those described above (Fig. 2 and Fig. 5) are added with "C" after the same symbols, and detailed description is omitted.

In this case, the power pulse output device 7 samples the calculated power according to a fixed clock, and performs pulse processing.

In an energy meter, it is generally required to output power supply pulses at intervals shorter than one cycle of the power supply.

The existing electric energy meter directly multiplies the current I and the voltage V to perform electric energy calculation, so it is easy to meet the above-mentioned electric energy pulse output requirements, but in the present invention, the current and voltage of each cycle are calculated by Fourier transform (FFT), The electric energy is calculated from the calculation result, so the electric energy pulse output in each cycle cannot meet the above requirements.

For example, in FIG. 11 , it is assumed that the power-related calculation device 4C is called to calculate the required power such as active power, reactive power, and apparent power each time the result of the FFT operation.

At this time, the length of one cycle of the power supply is kept for each cycle of the power supply frequency that changes every moment, such as a counter, and the count value (one cycle of the power supply) is multiplied by the power value and added to obtain the electric energy (power time integral value). However, with such calculation processing, the power pulse output device 7 can only output power pulses per power cycle.

The operation of Embodiment 5 of the present invention which solves this problem and improves the real-time performance when Fourier transform is used will be described below.

Fig. 12 is a sequence diagram showing the operation of CPU power calculation and power pulse output in Embodiment 5 of the present invention. In FIG. 12, t, t+1, . . . correspond to the data content at each execution timing of each processing operation.

At this time, the AD conversion data encapsulation means 2C has a function of detecting the encapsulation time required for encapsulation at the point L, and the power-related quantity calculation means 4C has a function of outputting the calculated power per sampling command.

First, at the timing shown in the uppermost part of FIG. 12, the AD conversion data encapsulation device 2C records the time required for encapsulation at point L (the time for one cycle).

Next, the CPU performs Fourier transform on the packaged data by using the Fourier transform device 3C, and calculates the power (first power value) by the power-related quantity computing device 4C (refer to the second column in FIG. 12 ). At this time, each content data at the power calculation timing is one cycle behind the content at the data packing processing timing (see the first column) (see t-1, t, . . . ).

Next, the time required for packaging at point L (recording time: refer to the first column) and the power calculated from the above package data (the calculated power after Fourier transform: refer to the second column) are obtained from the power-related quantity calculation device 4C Pass to the electric energy pulse output device 7 (with reference to the 3rd column in Fig. 12).

At this time, the time (unit width in the third column) transmitted to the power pulse output device 7 varies with the length of the recording time. The content data at each power output timing is 2 cycles later than the data encapsulation time (refer to t-2, t-1, . . . ).

The CPU samples the power transmitted to the power pulse output device 7 at a fixed cycle (fixed clock) shorter than one cycle of the power supply frequency, and uses the power sampled each time (the value in the third column, that is, the second power) as the electric energy Add (refer to column 4 in Figure 12).

That is, in the example shown in FIG. 12 , there are about six samples per cycle of each power supply frequency, and the electric energy of the same value is added for each sample of the power output corresponding to the same data package.

Finally, the electric energy pulse output device 7 outputs electric energy pulses each time the electric energy addition value reaches the desired electric energy (refer to column 5 in FIG. 12 ).

By making up the circuit as above, the real-time requirements can be met, and at the same time, the electric energy pulse can be output with high precision.

At this time, the shorter the sampling period, the higher the accuracy can be. Therefore, it is desirable to set the fixed period (fixed clock) as shorter as possible than the energy pulse output period.

Embodiment 6

Although not specifically mentioned in the above-mentioned fifth embodiment, it is also possible to compensate the phase difference between the voltage and the current of each higher harmonic.

Fig. 13 is a block diagram showing the composition of the key circuit of Embodiment 6 of the present invention, which is structurally made to compensate the phase difference of the voltage and current of each higher harmonic, and the same parts as those described above (Fig. 11) have the same symbols Add "D" after, omit detailed description.

In Fig. 13, a phase compensating device 8 is inserted between the Fourier transform device 3D and the power-related quantity computing device 4D, and the device 8 compensates the voltage of each higher harmonic by rotating the absolute value of the voltage and current and current phase difference.

Current sensors (CT, etc.) and voltage sensors (PT, etc.) generally affect the phase of the voltage and current of the unit line. The analog circuit provided at the input of the AD converter 1D also affects the phase of the voltage and current.

Therefore, in order to correctly measure the power-related quantity of the power supply line, it is necessary to compensate the phase distortion caused by the above-mentioned analog circuit system.

For example, when the phase of the voltage and current of the nth harmonic has the relationship shown in Figure 1, and the phase difference on the power line is "0", the calculation of the voltage rotation θ-φ or the current θ-φ can be performed. Operation, and in the case that the phase difference on the power line is not '0', the rotation operation can be performed according to the phase difference.

Here, the new current In_cmp_new obtained when the current is rotated can be represented by the following equation (33).

In_cmp_new＝In_new_re+j In_new_im (33)

At this time, the new current In_cmp_new can be obtained from the following equation (34).

$\left(\begin{array}{c}\mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; new</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}\\ \mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; new</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}\end{array}\right)<span\; class="notranslate">\; =</span>\left(\begin{array}{cc}\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; )</span>& <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; )</span>\\ \mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; )</span>& \mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&phi;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&theta;</span>}<span\; class="notranslate">\; )</span>\end{array}\right)<span\; class="notranslate">\; \&CenterDot;</span>\left(\begin{array}{c}\mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; re</span>}\\ \mathrm{<span\; class="notranslate">\; In</span>}<span\; class="notranslate">\; \_</span>\mathrm{<span\; class="notranslate">\; im</span>}\end{array}\right)<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 34</span>}<span\; class="notranslate">\; )</span>$

The new current In_cmp_new is used when calculating the power-related quantity. By performing this compensation for each harmonic, the power-related quantity can be calculated with high precision.

Here, voltage or current is selected as the power-related value, and its absolute phase is rotated to compensate for the phase difference of each higher harmonic. However, the active power and reactive power obtained by the power-related calculation device 4D can also be calculated. Rotation operation, of course, can also get the same effect at this time.

Industrial Applicability

As above, according to the present invention, it is possible to calculate active power, active electric energy, reactive power, reactive electric energy, apparent power, distortion power, current effective value, voltage effective value, phase difference between current and voltage, and the height of each of these items. It is useful for electronic watt-hour meters and power-related calculation circuits not only for general households but also for consumers who want to manage electric energy by time slots. In addition, since reactive power is also measured, it is useful for electronic energy meters and power-related quantity calculation circuits not only for those who need power factor management but also for those who use high-order harmonic generation equipment such as inverters.

Claims (11)

1. An electronic energy meter comprising: An AD converter for converting the measurement signals representing the current and voltage of the power supply line into digital values and taking them in, and A microprocessor including an electric energy calculation device for calculating the electric energy of the power supply line according to the digital value; it is characterized in that the electronic energy meter also has: power frequency detecting means for detecting the power frequency of said power line, a sampling frequency control device which is provided separately from the clock circuit of the microprocessor and controls the sampling frequency of the AD converter according to the power supply frequency, a voltage-controlled oscillator that changes the sampling frequency of the AD converter in response to a control voltage from the sampling frequency control device; The sampling frequency control device controls the sampling frequency so that the sampling frequency is a natural multiple of the power frequency, The electric energy computing device includes a high-order harmonic computing unit for computing high-order harmonic electric energy. 2. The electronic energy meter according to claim 1, characterized in that: The sampling frequency control device controls the sampling frequency so that the sampling frequency is the power of 2 of the power supply frequency, wherein n is a natural number, The electric energy computing device includes a high-speed Fourier transform device, which uses high-speed Fourier transform to compute high-order harmonic electric energy. 3. The electronic energy meter according to claim 1, characterized in that, The AD converter is a delta-sigma AD converter that needs up-sampling. 4. The electronic energy meter according to claim 1, characterized in that: Structurally, the sampling frequency control device comprises: zero-cross point detecting means for detecting a zero-cross point where said power supply frequency rises or falls, and When one cycle has elapsed since the last rising or falling zero-crossing point of the power supply frequency, it is detected from the AD conversion value of the power supply frequency that the power supply frequency lags behind or leads the power supply phase difference as a power supply phase difference detection device, The sampling frequency of the AD converter is controlled according to the phase difference of the power supply. 5. The electronic energy meter according to claim 1, characterized in that: Structurally, the sampling frequency control device comprises: absolute phase detection means for detecting the absolute phase of the voltage of said power line, and a voltage phase difference detecting device that detects that the absolute phase is behind or ahead of the last absolute phase of the voltage as a voltage phase difference at a time when one cycle has elapsed since the last absolute phase of the voltage, The sampling frequency of the AD converter is controlled according to the voltage phase difference. 6. The electronic energy meter according to claim 5, characterized in that: Structurally, the voltage phase difference detection device takes the absolute phase of 0 degree, 90 degree, 180 degree or 270 degree as a reference. 7. The electronic energy meter according to claim 1, characterized in that: Structurally, the sampling frequency control device has a sampling frequency compensating device for compensating the sampling frequency of the AD converter, The sampling frequency compensation device includes A control amount calculation device for obtaining a control amount of the sampling frequency, a D/A conversion device that outputs the control amount of the sampling frequency after D/A conversion, and biasing means for biasing the output voltage of said D/A converting means, The bias means biases a predetermined frequency side of the power supply line by a predetermined voltage. 8. The electronic energy meter according to claim 7, characterized in that, The bias device has a bias voltage adjusting device for externally adjusting the predetermined voltage. 9. The electronic energy meter according to claim 1, characterized in that: The electrical energy computing device includes: an AD conversion data encapsulation device encapsulating voltage and current measurement signals for one cycle of the power supply frequency, performing Fourier transform on the data encapsulated by the AD conversion data encapsulating means and calculating the first power value of the power calculation means, packaging time detecting means for detecting packaging time required for packaging said 1-period share voltage or current, A power output device that maintains the first power value at each packaging time and outputs the first power value as a second power value every time a sampling instruction of a predetermined computing cycle is generated, and A power pulse output device that accumulates the second power value and outputs a power pulse every time the accumulated value of the second power value reaches a predetermined value. 10. The electronic energy meter according to claim 1, characterized in that: The electrical energy computing device includes: an AD conversion data encapsulation device encapsulating voltage and current measurement signals for one cycle of the power supply frequency, performing Fourier transform on the data encapsulated by the AD conversion data encapsulation device, and calculating the power-related value of each higher harmonic and the phase difference of the power-related value, and a power calculation unit, and A phase compensating device that makes the phase difference of the power-related value become the real power-related value phase difference of the measured party through rotation calculation, and compensates the current or the voltage for the one-cycle share. 11. A power-related quantity calculation circuit, characterized in that the power-related quantity calculation circuit includes: The AD converter that converts the measurement signals representing the current and voltage of the power line into digital values and takes them in, An AD conversion data encapsulation device for encapsulating said digital value in 1 cycle share, a Fourier transform device for performing Fourier transform on the data encapsulated by the AD conversion data encapsulation device, According to the conversion result of the Fourier transform means, the power-related quantity calculating means for calculating the power-related quantity, A sampling frequency compensating device for computing a sampling frequency compensation amount of the AD converter according to the frequency of the current or the voltage, and Outputting the sampling frequency varying with the compensation amount to the voltage-controlled oscillator of the AD converter.

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