CN1120165A - Test method and equipment for microwave dielectric properties of non-metallic materials - Google Patents

Abstract

The invention is a method for testing the microwave dielectric properties of non-metallic materials. In order to implement the method, a cut-off waveguide is specially designed and coupled with a TE _01n mode resonant cavity through a coupling hole to form a composite resonant cavity. This method is to adjust the piston to make the composite resonant cavity reach resonance under the two conditions that the cut-off waveguide contains and does not contain the material sample to be tested, and record the readings of the piston position L _o , L _i and the Q factor Q of the composite resonant cavity respectively _o and Q _i are substituted into the corresponding formula to obtain the complex permittivity ε _r of the material. Applying this method can make the TE _01n resonant cavity, which can only measure low-dielectric and low-loss materials, also conveniently measure high-dielectric and low-loss materials or magnetic and non-magnetic high-loss materials.

Description

Test method and equipment for microwave dielectric properties of non-metallic materials

The invention belongs to a method for testing electrical performance parameters such as microwave complex permittivity and complex magnetic permeability of non-metallic materials and specially designed equipment for implementing the method.

With the development of microwave integration, mobile phones, satellite communications, radio, television, radar, electronic countermeasures, microwaves in industrial and agricultural production processes, daily microwaves and microwave biology, medicine and other fields, it involves everything from semiconductors to insulators. , from magnetic to non-magnetic materials, the electrical properties of these materials vary widely. At present, according to different material properties, three different measurement methods are used: the resonant cavity method of various wave modes is used for low dielectric constant and low loss materials; the dielectric resonant cavity method is used for high dielectric constant and low loss materials; For magnetic or non-magnetic materials with large loss, the method of standing wave measurement line and microwave vector network analyzer is used. Therefore, the required equipment and testing methods are relatively complicated, inconvenient to use, and expensive. Therefore, there are more and more calls for methods and equipment that can measure the electrical properties of materials with great differences.

The purpose of the present invention is to provide a test method and specially designed equipment for implementing the method, which can realize the measurement of the basic electromagnetic properties applicable to the above three types of materials.

Before describing the measurement method of the present invention, the structural scheme of the equipment required for implementing the method will be described first. The equipment includes the TE _o1n mode resonant cavity widely used at present, and the microwave frequency stabilization signal with a frequency of 2-40GHZ is fed through the microwave signal input port 1 on the cavity, and the resonance signal is fed to the resonance indicator from the detection signal output port 2 . The structural feature is that a cut-off waveguide 4 is installed on the cavity of the TE _o1n mode resonant cavity 3, and a sample 5 of the material to be tested (hereinafter referred to as the sample) is placed in the cut-off waveguide, and the cut-off waveguide is coupled to the TE _o1n mode resonant cavity by means of a coupling hole 6 to form a Composite resonator. The high sensitivity of the TE _o1n- mode high-Q resonator is exploited to measure the material properties in the cut-off waveguide.

The cut-off waveguides have the following size ranges:

Cut-off waveguide length = 25 ~ 50mm; ratio of cut-off waveguide diameter to TE _o1n mode resonant cavity diameter = 0.25 ~ 0.30; ratio of coupling hole equivalent area to TE _o1n mode resonant cavity circular section = (3 ~ 5) ×10 ^-3 .

Graphic description:

Fig. 1 is the schematic diagram of a kind of composite resonant cavity that the present invention designs;

Fig. 2 is the schematic diagram of another kind of composite resonant cavity that the present invention designs;

Fig. 3 is A-A sectional view of Fig. 1 and Fig. 2 .

Embodiments of the equipment solution of the present invention are described with reference to the accompanying drawings.

example 1:

As shown in Figure 1, when the microwave signal input port 1 and the detection signal output port 2 on the cavity of the TE _o1n mode resonant cavity 3 (hereinafter referred to as the cavity) are set on the side wall of the cavity, a section of copper cut-off waveguide 4 is It is fixed on the top of the cavity, the coupling hole 6 is set at the bottom of the cut-off waveguide, the measured material sample 5 is installed in the cut-off waveguide 4, and the tuning piston 7 of the TE _o1n cavity is installed at the bottom of the cavity.

Example 2:

As shown in Figure 2, when the above-mentioned input port 1 and output port 2 are set on the top of the cavity, the cut-off waveguide 4 is freely placed on the tuning piston 7 at the bottom of the cavity, and the coupling hole is set on the top of the cut-off waveguide.

The shape of the coupling hole consists of four symmetrical long holes (both ends of the hole are arc-shaped), as shown in Figure 3

The measurement method of the present invention not only maintains the measurement function of the best TE _o1n mode resonant cavity method for measuring low dielectric constant and low loss materials at present, but also carries out transformation and expansion, and establishes a new test method so that it can measure high dielectric constant Constants, electrical parameters or electromagnetic parameters of low-loss materials and absorbing materials. That is to say, several materials with very different electrical properties can be measured with one device.

The principle of the method of the present invention: in the cut-off waveguide 4 that does not contain samples, except for very small wall losses, no power is absorbed, so the impedance it introduces to the resonant cavity 3 through the coupling hole 6 is actually pure reactance; When the sample 5 enters the cut-off waveguide 4, part of the power is absorbed by the sample, and a reflection field is established, resulting in a resistance component and reactance change. At this time, the power passing through the coupling hole 6 to the cavity 3 decreases, and the resonant cavity produces a corresponding decrease in Q factor and a change in the resonance length. The distance from the sample to the coupling hole 6 of the cut-off waveguide and the size of the coupling hole 6 determine the electric field intensity vector in the sample 5, and thus also determine the power absorbed by the sample 5 and the magnitude of the reactance change. Therefore, the analysis of the resonant behavior of the resonant cavity 3 before and after the sample is placed in the cut-off waveguide 4 can obtain information about the electrical properties of the material sample 5 to be tested.

When there is no sample in the cut-off waveguide, under the condition of a certain angular frequency ω=2πf, the piston position reading L _o and Q factor Q _o at resonance are measured; when a sample with thickness d is placed at the distance from the coupling hole S, the above The corresponding values are L _f and Q _l , then the two components of the input impedance Z _e = R _e + jX _e are:

resistance $R_e={\left(\frac{{\mathrm{\λ}}_g}{{\mathrm{\λ}}_0}\right)}^2\frac{{\mathrm{\π}}^2{a}^2{\mathrm{f\μ}}_{0}l}{A}(\frac{1}{Q_i}-\frac{1}{Q_0}).....\left(1\right)$

Reactance $x_e=\frac{{\mathrm{\ω\μ}}_0}{{\mathrm{\α}}_0}-\frac{{\mathrm{\πa}}^2{\mathrm{\ω\μ}}_0}{A}(L_i-L_0)......\left(2\right)$

In the formula ${\mathrm{\α}}_0=\frac{2\mathrm{\π}}{{\mathrm{\λ}}_0}\sqrt{{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2-1},{\mathrm{\λ}}_c=2\mathrm{\πb}/3.832......\left(3\right)$ λ _c = cut-off wavelength; λ _g = waveguide wavelength; λ _o = free space wavelength; μ _o = vacuum permeability; A = equivalent area of coupling hole; waveguide radius; α _o =attenuation coefficient of the air-cut waveguide; angular frequency ω=2πf (f is the frequency).

According to the transmission line theory, the real and imaginary parts of the reflection coefficient ρ _e = ρ _e ′-jρ _e ″ of the air-medium interface can be obtained as: ${\mathrm{\ρ}}_e^{\′}=\frac{{{\mathrm{\α}}_0}^2({x_e}^2+{R_e}^2)-{\mathrm{\ω}}^2{{\mathrm{\μ}}_0}^2}{\exp (-2{\mathrm{\α}}_{0}S)[{R_e}^2{{\mathrm{\α}}_0}^2+{({\mathrm{\ω\μ}}_0+{\mathrm{\α}}_0{x}_e)}^2]}.....\left(4\right)$ ${\mathrm{\ρ}}_e^{\′\′}=\frac{2{\mathrm{\α}}_0{R}_e{\mathrm{\ω\μ}}_0}{\exp (-2{\mathrm{\α}}_{0}S)[{R_e}^2{{\mathrm{\α}}_0}^2+{({\mathrm{\ω\μ}}_0+{\mathrm{\α}}_0{x}_e)}^2]}.......\left(5\right)$

And the normalized impedance Z(0)/Z ₁ of the air-dielectric interface is: ${\left(\frac{Z\left(0\right)}{Z_1}\right)}_e=\frac{1+{\mathrm{\ρ}}_e}{1-{\mathrm{\ρ}}_e}=\frac{1+\frac{Z_1}{Z_1}\tanh 2\mathrm{\α}}{1+\frac{Z_1}{Z_2}\tanh 2\mathrm{\α}}......\left(6\right)$

In the formula, Z ₁ =jωμ ₀ /α ₀ , Z ₂ =jωμ ₀ /α; tanh is the tangent of the hyperbolic function; $\mathrm{\α}=j\frac{2\mathrm{\π}}{{\mathrm{\λ}}_0}\sqrt{{\mathrm{\ϵ}}_r-{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2}......\left(7\right)$ is the complex propagation factor of the sample section in the cut-off waveguide; ε _r = ε _r ′-jε _r ″=ε _r ′(1-jtanδ) is the relative complex permittivity of the material; here the dielectric loss tangent tanδ=ε _r ″ / _εr '.

In this way, using the measured complex reflection coefficient ρ _e , the ε _r ′ and tan δ of the non-magnetic material can be obtained by the computer by iterative method.

For magnetic materials, due to the $\mathrm{\α}=j\frac{2\mathrm{\π}}{{\mathrm{\λ}}_0}\sqrt{{\mathrm{\ϵ}}_r{\mathrm{\μ}}_r-{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2}......\left(8\right)$

Z ₂ ＝jωμ ₀ μ _r /α (9) ${\mathrm{\μ}}_r=\left(\frac{\mathrm{\α}}{{\mathrm{\α}}_0}\right)/\left(\frac{Z_1}{Z_2}\right)......\left(10\right)$ Where μ _r =μ _r ′-jμ _r ″ is the relative complex permeability of the material.

Therefore, in addition to the above measurement, it is necessary to short-circuit the terminal of the sample, and measure again with the same method as above to obtain the piston reading L _k and Q factor Q _k when the sample is contained, and calculate R _k from formulas (1) to (5) , X _k and ρ _k , get: ${\left(\frac{Z\left(0\right)}{Z_1}\right)}_k=\frac{1+{\mathrm{\ρ}}_k}{1-{\mathrm{\ρ}}_k}=\frac{Z_2}{Z_1}\tanh 2\mathrm{\α}.......\left(11\right)$

Solving the simultaneous equations composed of (6) and (11), we get ${\left(\frac{Z_1}{Z_2}\right)}^2=\frac{1+{\left(\frac{Z\left(0\right)}{Z_1}\right)}_k-{\left(\frac{Z\left(0\right)}{Z_1}\right)}_e}{{\left(\frac{Z\left(0\right)}{Z_1}\right)}_k{\left(\frac{Z\left(0\right)}{Z_1}\right)}_e}.....\left(12\right)$ Then get α from formula (11), substitute into formula (10) to get the complex number μ _r , and then calculate ε _r ′ from formula (8), that is ${\mathrm{\ϵ}}_r=\frac{1}{{\mathrm{\μ}}_r}[{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2-\frac{{\mathrm{\α}}^2{\mathrm{\λ}}_0}{4{\mathrm{\π}}^2}].....\left(13\right)$

Test method of the present invention:

1. Evaluate the equivalent area A of the coupling hole:

At a certain fixed frequency, adjust the resonant cavity to resonance under the condition that there is nothing in the cut-off waveguide, and read the piston reading L _o at resonance; then put a metal short circuit board in the cut-off waveguide so that it is close to the coupling hole 6, After retuning L _m , calculate A according to the following formula:

A＝πa ² α ₀ (L _m -L ₀ ) (14)

2. Measurement of non-magnetic materials with high dielectric constant (ε _r > 10):

Adjust the microwave signal generator to your desired frequency. When the cut-off waveguide does not contain samples, adjust the piston of the composite resonant cavity to resonance, record the reading L _o , and measure the Q factor Q _o ; put in the cut-off waveguide After the thickness of the sample is d, adjust the piston to restore resonance and measure L _i , Q _i , calculate R _e , X _e according to formulas (1) and (2), and obtain ρ _e = from formulas (4) and (5) ρ _e ′-jρ _e ″, and finally the relative complex permittivity ε _r =ε _r ′(1-jtanδ) of the material is obtained from equations (6) and (7) by the Newton iterative method.

When the sample is close to the coupling hole (that is, S=0), formulas (4)-(7) are combined and simplified to obtain $\frac{x_e{\mathrm{\α}}_0-j{R}_e{\mathrm{\α}}_0}{{\mathrm{\ω\μ}}_0}=\frac{1+\sqrt{\frac{{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2-1}{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}}\mathrm{the\; tan}\frac{2\mathrm{\πd}}{{\mathrm{\λ}}_0}\sqrt{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}}{1-\sqrt{\frac{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}{{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2-1}}\mathrm{the\; tan}\frac{2\mathrm{\πd}}{{\mathrm{\λ}}_0}\sqrt{{{\mathrm{\ϵ}}_r-({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}}....\left(15\right)$

In this way, the intermediate parameter ρ _e can be omitted, and the complex number ε _r can be obtained directly from the measured X _e and R _e by iterative method.

3. For the measurement of magnetic large loss absorbing materials:

Adjust the microwave signal generator to the required frequency. When there is no sample in the cut-off waveguide, adjust the composite resonant cavity to resonance, record the reading L _o and measure the Q factor Q _o . After putting a sample with a thickness d in the cut-off waveguide, adjust the piston to restore resonance, and measure L _i and Q _i ; then add a metal plate behind the sample to short circuit, adjust the piston, restore resonance again, and obtain L _k and Q _k . Substitute Q _i , Q _o and _Li , L _o into formulas (1), (2) to get R _e , X _e ; then replace L _i and Q _i with L _k and Q _k into formulas (1), (2) , get R _k and X _k . Substituting the two sets of R and X values into formulas (4) and (5), we get ρ _e = ρ _e ′-jρ _e ″ and ρ _k = ρ _k ′-jρ _k ″, and the corresponding normalized impedance is ${\left(\frac{Z\left(0\right)}{Z_1}\right)}_i=\frac{1+{\mathrm{\ρ}}_i}{1-{\mathrm{\ρ}}_i}(i=e,k).....\left(15\right)$ Z ₁ /Z ₂ is obtained from formula (12).

Then, the complex propagation factor α can be obtained from equation (11). In this way, the basic electromagnetic complex parameters ε _r μ _r , μ _r and ε _r of the material are finally obtained from formulas (8), (10) and (13).

The advantage of the method of the present invention is that in addition to measuring materials with greatly different properties, there is also a low requirement for the cooperation between the sample and the waveguide, that is, when the sample diameter is less than the cut-off waveguide diameter within 0.5mm, the measurement results will not be affected basically, unlike the waveguide or coaxial test systems, the sample must fit tightly with them.

The applicable frequency range of the present invention is 2-40 GHz.

In order to understand the inventive method use effect, carry out actual operation under laboratory condition, its result is as follows:

1. Under the frequency conditions of 8-12GHz and 35GHz, we install the cut-off waveguide on the TE _o1n single-mode resonant cavity with an inner diameter of 2a=50mm according to Figure 2. The inner diameter 2b of the cut-off waveguide is 15mm, the area of the coupling hole A is 6.03mm ² , ε _r ′ of the material BaTi ₄ O ₉ is measured at a frequency of 9.375GHz=39.05, and tanδ=0.69×10 ^-4 .

2. At a frequency of 2-4GHz, we install a cut-off waveguide on the TE _o1n cavity with an inner diameter of 2a=180mm according to Figure 1, with an inner diameter of 2b=50mm, and a coupling hole area A=120mm ² , and at a frequency of 4-8GHz, the inner diameter TE _o1n cavity with 2a=120mm is installed according to Figure 1. Cut-off waveguide inner diameter 2b=30mm, coupling hole A=42mm ² ; measuring epoxy resin containing carbonyl iron powder, the result is: frequency f(GHz) ε _r μ _r μ _r ε _r 2.2 35.66-j12.30 2.99-j0.94 12.06-j0.352.8 30.67-j14.19 2.37-j1.13 12.86-j0.143.8 30.36-j14.67 2.41-j1.05 12.85-j0.25.401 -j15.77 2.38-j1.29 11.02-j0.656.0 22.65-j16.57 2.02-j1.18 11.95-j1.257.1 21.13-j15.72 1.84-j1.23 11.92-j0.618.1 19.15-j15.22 1. -j1.20 12.07-j0.37

Claims (7)

1. The method for testing the microwave dielectric properties of non-metallic materials is characterized in that the method comprises the steps: 1) Determine the equivalent area A of the coupling hole [6] Feed in a microwave signal of a given frequency from the microwave signal port [1], adjust the tuning piston [7] of the resonant cavity [3] to make the resonant cavity reach resonance, and read the tuning piston position reading L _o when resonant; in the cut-off waveguide [ 4] Set a metal short-circuit plate close to the coupling hole [6], readjust the tuning piston, read L _m , and obtain the equivalent area A of the coupling hole according to the following formula: A＝πa ² α ₀ (L _m -L ₀ ) (14) 2) Do not put the material sample [5] in the cut-off waveguide [4], at a given frequency, adjust the tuning piston to make the composite resonant cavity reach resonance, record the reading L _{o of} the tuning piston position at this time, and measure the composite resonance Q factor value Q _o of the cavity; 3) Put a material sample [5] with a thickness of d in the cut-off waveguide, then adjust the tuning piston, restore the resonance and measure the corresponding L _i and Q _i ; 4) Using the L _o , Q _o and L _i , Q _i measured in steps 2) and 3), calculate the complex permittivity ε _r of the material through the formula. 2. according to the test method of the microwave dielectric property of nonmetallic material of claim 1, it is characterized in that for the nonmagnetic material of high dielectric constant (ε _r > 10), comprise the following method steps: 1) Substitute the measured L _o , Q _o and L _i , Q _i into the following two formulas to obtain the resistance R _e and reactance X _e : $R_e={\left(\frac{{\mathrm{\λ}}_g}{{\mathrm{\λ}}_0}\right)}^2\frac{{\mathrm{\π}}^2{a}^2{\mathrm{f\μ}}_{0}l}{A}(\frac{1}{Q_i}-\frac{1}{Q_0})......\left(1\right)$ $x_e=\frac{{\mathrm{\ω\μ}}_0}{{\mathrm{\α}}_0}-\frac{{\mathrm{\πa}}^2{\mathrm{\ω\μ}}_0}{A}(L_i-L_0).....\left(2\right)$ In the formula: ${\mathrm{\α}}_0=\frac{2\mathrm{\π}}{{\mathrm{\λ}}_0}\sqrt{{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2-1},{\mathrm{\λ}}_c=2b/3.832.....\left(3\right)$ 2) from the formula ${\mathrm{\ρ}}_e^{\′}=\frac{{{\mathrm{\α}}_0}^2({x_e}^2+{R_e}^2)-{\mathrm{\ω}}^2{{\mathrm{\μ}}_0}^2}{\exp (-2{\mathrm{\α}}_{0}S)[{R_e}^2{{\mathrm{\α}}_0}^2+{({\mathrm{\ω\μ}}_0+{\mathrm{\α}}_0{x}_e)}^2]}...\left(4\right)$ ${\mathrm{\ρ}}_e^{\′\′}=\frac{2{\mathrm{\α}}_0{R}_e{\mathrm{\ω\μ}}_0}{\exp (-2{\mathrm{\α}}_{0}S)[{R_e}^2{{\mathrm{\α}}_0}^2+{({\mathrm{\ω\μ}}_0+{\mathrm{\α}}_0{x}_e)}^2]}.....\left(5\right)$ Obtain the real part and imaginary part of the reflection coefficient ρ _e = ρ _e ′-jρ _e ″ of the air-medium interface; 3) from the formula ${\left(\frac{Z\left(0\right)}{Z_1}\right)}_e=\frac{1+{\mathrm{\ρ}}_e}{1-{\mathrm{\ρ}}_e}=\frac{1+\frac{Z_2}{Z_1}\tanh 2\mathrm{\α}}{1+\frac{Z_1}{Z_2}\tanh 2\mathrm{\α}}......\left(6\right)$ Japanese style $\mathrm{\α}=j\frac{2}{{\mathrm{\λ}}_0}\sqrt{{\mathrm{\ϵ}}_r-{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2}......\left(7\right)$ Use the measured complex reflection coefficient ρ _e to obtain the complex permittivity of non-magnetic materials according to formula (6) and iterative method with computer When the material sample is close to the coupling hole, that is, S=0, it can be directly obtained from the formula $\frac{x_e{\mathrm{\α}}_0-j{R}_e{\mathrm{\α}}_0}{{\mathrm{\ω\μ}}_0}=\frac{1+\sqrt{\frac{{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2-1}{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}}\mathrm{the\; tan}\frac{2\mathrm{\πd}}{{\mathrm{\λ}}_0}\sqrt{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}}{1-\sqrt{\frac{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}{{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2-1}\mathrm{the\; tan}\frac{2\mathrm{\πd}}{{\mathrm{\λ}}_0}\sqrt{{\mathrm{\ϵ}}_r-{({\mathrm{\λ}}_0/{\mathrm{\λ}}_c)}^2}}}...\left(15\right)$ The complex permittivity ε _r is obtained by an iterative method. 3. according to the test mode of the non-metallic material microwave dielectric property of claim 1, it is characterized in that for the large loss wave-absorbing material of magnetism, comprise the following method steps: 1) Use the same method as above to tune and obtain Q _o , L _o , Q _i and L _i ; 2) Put a short circuit on the back of the material sample and then tune to get L _k and Q _k values, first substitute Q _i , Q _o and L _i , L _o into the above (1) formula (2) to get R _e , X _e ; Then replace L _i and Q _i with L _k and Q _k into formulas (1) and (2) to get R _k and X _k , and substitute the above two sets of R and X values into the above formulas (4) and (5) , get ρ _e = ρ _e ′-jρ _e ″ and ρ _k = ρ _k ′-jρ _k ″, the normalized impedance is ${\left(\frac{Z\left(0\right)}{Z_1}\right)}_i=\frac{1+{\mathrm{\ρ}}_i}{1-{\mathrm{\ρ}}_i}(i=e,k).....\left(16\right)$ by formula ${\left(\frac{Z_1}{Z_2}\right)}^2=\frac{1+{\left(\frac{Z\left(0\right)}{Z_1}\right)}_k-{\left(\frac{Z\left(0\right)}{Z_1}\right)}_e}{{\left(\frac{Z\left(0\right)}{Z_1}\right)}_k{\left(\frac{Z\left(0\right)}{Z_1}\right)}_e}......\left(12\right)$ Calculate Z ₁ /Z ₂ ; From the formula ${\left(\frac{Z\left(0\right)}{Z_1}\right)}_k=\frac{1+{\mathrm{\ρ}}_k}{1-{\mathrm{\ρ}}_k}=\frac{Z_2}{Z_1}\tanh 2\mathrm{\α}.......\left(11\right)$ Find α; finally from the following two formulas ${\mathrm{\μ}}_r=\left(\frac{\mathrm{\α}}{{\mathrm{\α}}_0}\right)/\left(\frac{Z_1}{Z_2}\right)........\left(10\right)$ ${\mathrm{\ϵ}}_r=\frac{1}{{\mathrm{\μ}}_r}[{\left(\frac{{\mathrm{\λ}}_0}{{\mathrm{\λ}}_c}\right)}^2-\frac{{\mathrm{\α}}^2{\mathrm{\λ}}_0}{4{\mathrm{\π}}^2}].....\left(13\right)$ We get μ _r and ε _r respectively. 4. for implementing the equipment of the non-metallic material microwave dielectric property testing method described in claim 1, comprise TE _o1n mode resonant cavity, be the microwave frequency stabilization signal of _2-40GH by microwave signal input port [1] feed-in frequency, Its resonance signal is fed from the detection signal output port [2] to the resonance indicator. It is characterized in that a cut-off waveguide [4] is installed on the cavity of the TE _o1n mode resonant cavity [3], and a sample of the material to be tested is placed in the cut-off waveguide [5], the cut-off waveguide is coupled to the TE _o1n mode resonator by means of the coupling hole [6] to form a composite resonator. 5. The device according to claim 4, characterized in that the size range and shape of the cut-off waveguide [4] are: cut-off waveguide length = 25-50mm; cut-off waveguide diameter: cavity diameter = 0.25-0.30; coupling hole equivalent Area: the circular section of the cavity = (3~5)×10 ^-3 ; the shape of the coupling hole is composed of four symmetrical long holes (both ends of which are arc-shaped). 6. The device according to claim 4 or 5, characterized in that when the microwave signal input port [1] and the detection signal output port [2] of the TE _o1n mode resonant cavity [3] are arranged on the cavity side wall, the cut-off The waveguide [4] is fixed on the top of the cavity, and the coupling hole [6] is set at the bottom of the cut-off waveguide [4]. 7. The device according to claim 4 or 5, characterized in that when the microwave signal input port [1] and the detection signal output port [2] of the TE _o1n mode resonant cavity [3] are set on the top of the cavity [3] , the cut-off waveguide [4] is freely placed on the tuning piston [7] at the bottom of the cavity, and the coupling hole [6] is set on the top of the cut-off waveguide.

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