CN111987812A - Wireless charging system dynamic tuning method for string compensation topology - Google Patents

Abstract

The invention discloses a method for dynamic tuning of a wireless charging system oriented to a string compensation topology, and belongs to the technical field of dynamic tuning of wireless charging systems. Step 1, based on a hardware circuit with a soft-switching controllable capacitor with a symmetrical structure that can realize dynamic tuning, the zero-phase frequency stabilization criterion of the series compensation topology is obtained; step 2, the phase judgment method and the zero value search method are used to realize The zero-phase frequency stabilization criterion of the series compensation topology is described; in step 3, the equivalent capacitance value of the soft-switching controllable capacitor is adjusted according to the zero-phase frequency stabilization criterion, thereby performing dynamic tuning. The invention proposes a dynamic tuning method for a wireless charging system oriented to a string compensation topology, which realizes frequency stabilization control when a passive element in the string compensation topology has parameter drift. Meanwhile, compared with the existing tuning methods, the proposed method reduces the amount of passive components and reduces the control complexity.

Description

A method for dynamic tuning of wireless charging system for string compensation topology

technical field

The invention relates to a method for dynamic tuning of a wireless charging system based on a string compensation topology, and belongs to the technical field of dynamic tuning of wireless charging systems.

Background technique

Compensation topology, as a key link of wireless charging system, plays the role of reactive power compensation, system efficiency and output power improvement, and system stable operation. Existing research focuses on composite structure design and performance parameter optimization, so as to achieve constant current and constant voltage. output and reducing the offset impact on system performance. However, the existing literature on the frequency stability of compensation topology is still weak. On the one hand, affected by the cumulative temperature rise and device aging, the parameter drift of passive components in the compensation topology destroys the resonance state; on the other hand, affected by the processing technology and parameter tolerance, the compensation topology harmonic error leads to the natural frequency drift. The above two situations will cause the system performance to degrade, and in severe cases will cause the system to work abnormally.

SUMMARY OF THE INVENTION

The purpose of the present invention is to propose a method for dynamic tuning of a wireless charging system oriented to a serial (series-to-series) compensation topology, so as to solve the problems existing in the prior art and prevent the system performance from being degraded.

A method for dynamic tuning of a wireless charging system oriented to a string compensation topology, the dynamic tuning method comprising the following steps:

Step 1: Based on a hardware circuit with a soft-switched controllable capacitor with a symmetrical structure that can realize dynamic tuning, the zero-phase frequency stabilization criterion of the series compensation topology is obtained;

Step 2, realizing the zero-phase frequency stabilization criterion of the string compensation topology through a phase judgment method and a zero-value search method;

Step 3: Adjust the equivalent capacitance value of the soft-switching controllable capacitor according to the zero-phase frequency stabilization criterion, so as to perform dynamic tuning.

Further, in step 1, the hardware circuit that can realize dynamic tuning is equivalent to a mutual inductance model of a string compensation topology, and the mutual inductance model includes: a transmitting end and a receiving end, the transmitting end includes a DC power supply, a transmitting end The coil and the soft-switching controllable capacitor of the transmitting end are connected in sequence to form a closed loop, and the DC power supply, the coil of the transmitting end and the soft-switching controllable capacitance of the transmitting end are connected in turn to form a closed loop, and the receiving end includes an equivalent load, the coil of the receiving end and the soft switching controllable of the receiving end capacitor, the equivalent load, the receiving end coil and the receiving end soft switch controllable capacitor are connected in turn to form a closed loop,

Define L ₁ and L ₂ as the self-inductance values of the transmitter coil and the receiver coil respectively, C ₁ and C ₂ as the compensation capacitance values of the transmitter coil and the receiver coil respectively, M is the difference between the transmitter coil and the receiver coil Mutual inductance, i ₁ and i ₂ are the resonant currents in the transmitter coil and the receiver coil respectively, Ro and _Uo are the resistance value and voltage value of the equivalent load, respectively, and u _s is the output voltage value of the inverter.

Further, the zero-phase frequency stabilization criterion includes a receiving end frequency stabilization criterion and a transmitting end frequency stabilization criterion, wherein,

The frequency stabilization criterion _of the receiving end: when L2 and _C2 have parameter drift, the phasor expression between the resonant current _i2 in the receiving coil and the resonant current _i1 in the transmitting coil is:

It can be seen from equation (1) that: when the receiving end works in the resonance state, i ₂ leads i ₁ and the phase difference γ ₂ is 90°; when the receiving end works in the loss-of-vibration state, the γ ₂ shown in equation (2) is not 90° , the phase difference between i ₂ lag by 90° and i ₁ is defined as γ _d2 , that is, γ _d2 =0 when the receiving end works in the resonance state,

The frequency stabilization criterion of the transmitting end: assuming that the receiving end works in the resonant state, and when the transmitting end works in the resonant state, the phasor relationship between the output voltage u _s and i ₁ of the full-bridge inverter is expressed as formula (3),

It can be seen from equation (3): when the transmitter works in the resonant state, u _s and i ₁ are in the same phase, that is, γ ₁ is 0°; when the transmitter works in the detuned state, the γ ₁ shown in equation (4) is not 0. °, when considering the ZVS soft switching of the switch tube in the full-bridge inverter, correct γ ₁ to realize that i ₁ lags u _s appropriately, and use this as the criterion for the resonance state of the transmitter,

Further, the soft-switching controllable capacitor with a symmetrical structure includes: a switch S ₁ , a switch S ₂ , a diode D ₁ , a diode D ₂ , a capacitor C _a , a capacitor C _1s and a capacitor C _2s , the capacitors C _1s , switch tube S ₁ , capacitor C _2s and switch tube S ₂ are connected in series in sequence, the bridge arm formed by the capacitor C _a and the capacitor C _1s and the switch tube S ₁ , the capacitor C _2s and the switch tube S ₂ are formed The bridge arms are connected in parallel, the diode D ₁ and the switch S ₁ are in anti-parallel, the diode D ₂ and the switch S ₂ are in anti-parallel, the main circuit current of the soft-switched controllable capacitor with a symmetrical structure is is _s and the terminal voltage for _us .

Further, the control signal _{of S1} is synchronized with the zero _- crossing point of is and the control signal of S2 is delayed by _half a cycle, so as to realize the ZVS soft switching _of the switch S1 and the switch S2. At the same time, the symmetrical controllable capacitor adopts The two switches realize symmetrical conduction in the positive and negative half cycles, so that the waveform is symmetrical without distortion and there is no DC bias.

Assuming that C _1s =C _2s =C, according to the different value ranges of the duty cycle D of the control signals _of the switch S1 and the switch S2, the working modes of the soft _- switching controllable capacitor are divided into: 0≤D≤0.25, 0.25 <D≤0.5 and 0.5<D,

Case ₁ : 0≤D≤0.25, in _one cycle, switch S1 and switch S2 and diode D1 and diode D2 are each conducting for _2Dπ time, and the _six operating modes correspond to t ₀ ～ t ₆ 6 time periods,

Mode ₁ [t ₀ ～ t ₁ ]: the switch S1 is turned _on and the switch S2 is turned off, the diode D1 and the diode D2 do not work, the capacitor C _1s and the capacitor C _a are _{charged at} the same time, and the capacitor C _2s The terminal voltage u _2c remains unchanged;

Mode 2 [t ₁ ～ t ₂ ]: both switch tubes S ₁ and S ₂ are turned off, diode D ₁ and diode D ₂ do not work, capacitor C _1s and capacitor C _2s are not charged, and the terminal of capacitor C _1s The voltages u _1c and u _2c remain unchanged, and the capacitor C _a is charged;

Mode 3 [t ₂ ～ _{t 3} ]: switch S1 and switch S2 are _both turned off, diode D1 does not work, diode D2 works, capacitor C _a and capacitor C _2s are charged _at the same time and _{u 1c} remains unchanged ;

Mode 4 [t ₃ ~ _{t 4} ]: switch S1 is turned off and switch S2 is turned _on , diode D1 and diode D2 do not work, capacitor C _a is charged and capacitor C _{2s is} discharged, and _{u 1c} remains inactive. Change;

Mode ₅ [t ₄ ～ _{t 5} ]: switch S1 and switch S2 are _both turned off, diode D1 and diode D2 do not work, capacitor C _a discharges, and _{u 1c} and u _2c remain unchanged;

Mode ₆ [t ₅ ～ _{t 6} ]: switch S1 and switch S2 are _both turned off, diode D1 works, diode _{D2 does not work, capacitor C 1s and} capacitor C _a discharge, u _2c remains unchanged, When the diode D1 and the diode D2 are turned _on in Mode ₃ and Mode ₆ , the terminal voltage _of the switch S1 and the switch S2 is zero, so as to realize the _ZVS soft switching _of the switch S1 and the switch S2,

Based on the above analysis, the expression of the equivalent capacitance value C _eq when 0≤D≤0.25 is,

Case 2: 0.25<D≤0.5, the capacitor C _1s and the capacitor C _2s cross when charging, the conduction time of the diode D ₁ and the diode D ₂ changes, and the 6 working modes correspond to 6 in t ₀ ~ t ₆ period,

Mode ₁ [t ₀ ~ t ₁ ]: switch S1 is turned _on and switch S2 is turned off, diode D1 and diode _D2 do not work, capacitor C _1s and capacitor C _a are charged _at the same time, and u _2c remains inactive. Change;

Mode ₂ [t ₁ ～ _{t 2} ]: the switch S1 is turned on and the switch S2 is turned off, the diode D1 does not work, the diode D2 _{works, and the capacitor C 1s ,} the capacitor C _a and the capacitor C _2s are charged _at the same time;

Mode 3 [t ₂ ～ _{t 3} ]: switch S1 and switch S2 are _both turned off, diode D1 does not work, diode D2 works, capacitor C _a and capacitor C _2s are charged _at the same time, and _{u 1c} remains unchanged ;

Mode 4 [t ₃ ～ _{t 4} ]: switch S1 is turned off and switch S2 is turned _on , diode D1 and diode D2 do not work, capacitor C _a and capacitor C _2s are discharged _at the same time, and _{u 1c} remains inactive. Change;

Mode 5 [t ₄ ~ _{t 5} ]: the switch S1 is turned off and the switch S2 is turned _on , the diode D1 works, the diode _{D2 does not work, and the capacitor C 1s ,} the capacitor C _a and the capacitor C _{2s discharge} simultaneously;

Mode 6 [t ₅ ~ t ₆ ]: both switch tubes S ₁ and S ₂ are turned off, diode D ₁ works, diode D ₂ does not work, capacitor C _1s and capacitor C _a discharge at the same time, mode 3 and mode _In state ₆ , when the diode D1 and the diode D2 are turned _on , the terminal voltage _of the switch S1 and the switch S2 is zero, so as to realize the ZVS soft switching,

Based on the above analysis, the expression of the adjustable equivalent capacitance value C _eq when 0.25<D≤0.5 is:

Case 3: 0.5<D, the charge-discharge cycle of capacitor C _1s and capacitor C _2s is completely synchronized with the charge-discharge cycle of capacitor C _a . At this time, control D cannot adjust C _eq , that is, the soft-switching controllable capacitor fails. As mentioned above, the effective adjustment range of D is 0～0.5 and the adjustment range of C _eq is C _a ～(C _a +2C).

Further, in step 3, the working principle of the control circuit of the soft-switching controllable capacitor: the current sensor collects the resonant current flowing through the main circuit of the soft-switching controllable capacitor, and uses this as a reference to generate two PWM signals with a phase difference of 180°. , so as to realize the ZVS soft switching _of the switch S1 and the switch S2 in the soft _- switching controllable capacitor, and the output signal of the zero-phase difference search circuit controls the duty cycle D of the above two PWM signals, thereby adjusting the equivalent capacitance value.

Further, in step 3, the working principle of the dynamic tuning method: i ₁ and i ₂ collected by the current sensor, after passing through the signal conditioning circuit and high-speed A/D converter, the controller calculates i ₁ and i ₂ and i ₁ and i 2 . The phase difference between u _s and the variable-step disturbance observation method is used to search for the zero phase difference, so as to output the control signal of the soft-switching controllable capacitor.

The main advantages of the present invention are as follows: the present invention proposes a dynamic tuning method for a wireless charging system oriented to a series-series (series-to-series) compensation topology, and realizes frequency stabilization control (that is, the frequency stability control). Meanwhile, compared with the existing tuning methods, the proposed method reduces the amount of passive components and reduces the control complexity.

Description of drawings

Fig. 1 is the mutual inductance model diagram of the string compensation topology;

Figure 2 is a structural diagram of a soft-switching controllable capacitor based on a symmetrical structure;

Figure 3 is the working waveform of the soft-switching controllable capacitor, wherein Figure 3(a) is the working waveform of the soft-switching controllable capacitor when 0≤D≤0.25; Figure 3(b) is the soft-switching waveform when 0.25≤D≤0.5 The working waveform of the switch controllable capacitor;

Figure 4 is a phase relationship diagram between two sinusoidal signals;

Fig. 5 is the principle block diagram of the variable-step perturbation observation method;

Fig. 6 is the flow chart of the dynamic tuning method of the wireless charging system oriented to the string compensation topology, Fig. 6 (a) is the soft-switching controllable capacitor control circuit; Fig. 6 (b) is the principle block diagram of the dynamic tuning method;

Fig. 7 is an analysis diagram of the resonance state of the receiving coil, wherein Fig. 7(a) is a working waveform diagram; Fig. 7(b) is a diagram showing the relationship between the LC resonance frequency and the phase difference;

Fig. 8 is the resonance state analysis diagram of the transmitting coil;

Fig. 9 is the simulation result graph of soft-switching controllable capacitor, wherein Fig. 9(a) is the graph of duty cycle and equivalent capacitance value; Fig. 9(b) is the graph of duty cycle and operating frequency;

Fig. 10 is a simulation result diagram of the dynamic tuning method, wherein Fig. 10(a) is the working waveform of the tuning process; Fig. 10(b) is the disturbance observation method.

Detailed ways

The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, but not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without creative efforts shall fall within the protection scope of the present invention.

The mutual inductance model of the string compensation topology in the present invention is shown in FIG. 1 . L ₁ and L ₂ are the self-inductance values of the transmitting coil and the receiving coil respectively, C ₁ and C ₂ are the compensation capacitance values of the transmitting coil and the receiving coil, M is the mutual inductance value between the transmitting coil and the receiving coil, i ₁ and i ₂ are the resonant currents in the transmitting coil and the receiving coil, respectively, R _o and u _o are the equivalent load resistance value and voltage value, respectively, and u _s is the output voltage value of the inverter.

(1) Zero-phase frequency stabilization criterion for series compensation topology

(a) The frequency stabilization criterion of the receiver. When L ₂ and C ₂ have parameter drift, combined with Fig. 1 and the law of electromagnetic induction, the phasor expression between the resonant current i ₂ in the receiving coil and the resonant current i ₁ in the transmitting coil is:

It can be seen from equation (1) that: when the receiving end works in the resonance state, i ₂ leads i ₁ and the phase difference γ ₂ is 90°; when the receiving end works in the loss-of-vibration state, the γ ₂ shown in equation (2) is not 90° . For the convenience of subsequent analysis, the phase difference between i ₂ lag by 90° and i ₁ is defined as γ _d2 , that is, γ _d2 =0 when the receiving end works in a resonance state.

(b) The frequency stabilization criterion of the transmitter. Assuming that the receiving end works in the resonant state and the transmitting end works in the resonant state, the phasor relationship between the output voltage u _s and i ₁ of the full-bridge inverter is expressed as formula (3). Among them, "&" represents the "AND" operation between the two formulas.

It can be seen from equation (3): when the transmitter works in the resonant state, u _s and i ₁ are in the same phase, that is, γ ₁ is 0°; when the transmitter works in the detuned state, the γ ₁ shown in equation (4) is not 0. °. When considering the ZVS soft switching of the switch tube in the full-bridge inverter, correcting γ ₁ realizes that i ₁ lags u _s appropriately, and takes this as the criterion for the resonant state of the transmitter.

(2) Soft-switching controllable capacitor based on symmetrical structure

As shown in FIG. 2 , the controllable capacitor composition unit based on the symmetrical structure includes: switch tubes S ₁ and S ₂ , their anti-parallel diodes D ₁ and D ₂ , and capacitors C _1s , C _2s and C _a . The mains current flowing into the controllable _capacitor is is and the terminal voltage is _us . _The control signal of S1 is synchronized with the zero _- crossing point of is and the control signal of S2 is delayed by half _a cycle, thereby realizing the ZVS soft switching _of S1 and S1. At the same time, the symmetrical controllable capacitor uses two switching tubes to achieve symmetrical conduction in the positive and negative half cycles, so that the waveform is symmetrical and has no distortion and no DC bias.

In order to ensure the symmetry of the circuit and simplify the theoretical analysis, it is assumed that C _1s =C _2s =C and the symmetrical controllable capacitor with ZVS characteristics is referred to as a soft-switching controllable capacitor for short. According to the different value ranges of the duty ratio D _of the S1 and S2 control signals, the working modes of the soft _- switching controllable capacitor are divided into: 0≤D≤0.25, 0.25<D≤0.5 and 0.5<D.

Case 1: 0≤D≤0.25, the working waveform is shown in Figure 3(a). In one cycle, S ₁ and S ₂ and D ₁ and D ₂ are each conducting for 2Dπ time, and the 6 working modes correspond to 6 time periods in t ₀ to t ₆ .

[1] Mode 1 [t ₀ ~ t ₁ ]: S ₁ is turned on and S ₂ is turned off, D ₁ and D ₂ are not working, C _1s and C _a are charged at the same time, and the terminal voltage u _2c of C _2s remains unchanged Change;

[2] Mode 2 [t ₁ ~ t ₂ ]: Both S ₁ and S ₂ are off, D ₁ and D ₂ are not working, C _1s and C _2s are not charged and the terminal voltages u _1c and u _2c of C _1s remain unchanged, C _a is charged;

[3] Mode 3 [t ₂ ~ t ₃ ]: Both S ₁ and S ₂ are off, D ₁ does not work, D ₁ works, _Ca and C _2s are charged at the same time and u _1c remains unchanged;

[4] Mode 4 [t ₃ ~ t ₄ ]: S ₁ is turned off and S ₂ is turned on, D ₁ and D ₂ are not working, _Ca is charged and C _2s is discharged, and u _1c remains unchanged;

[5] Mode 5 [t ₄ ~ t ₅ ]: Both S ₁ and S ₂ are turned off, D ₁ and D ₂ are not working, _Ca is discharged, and u _1c and u _2c remain unchanged;

Mode 6 [t ₅ ~ t ₆ ]: Both S ₁ and S ₂ are off, D ₁ works, D ₂ does not work, C _1s and C _a discharge, and u _2c remains unchanged. When D1 and D2 are turned _on in modes ₃ and ₆ , the terminal voltages _of S1 and S2 are zero, thus realizing the _ZVS soft switching _of S1 and S2.

Based on the above analysis, the expression of the equivalent capacitance value C _eq when 0≤D≤0.25 is:

Case 2: 0.25<D≤0.5, the working waveform is shown in Figure 3(b). When C _1s and C _2s are charged, there is a crossover, and the conduction time of D ₁ and D ₂ varies. The six operating modes correspond to six time periods in t ₀ to t ₆ .

[1] Mode 1 [t ₀ ~ t ₁ ]: S ₁ is turned on and S ₂ is turned off, D ₁ and D ₂ are not working, C _1s and C _a are charged at the same time, and u _2c remains unchanged;

[2] Mode 2 [t ₁ ~ t ₂ ]: S ₁ is turned on and S ₂ is turned off, D ₁ does not work, D ₂ works, and C _1s , C _a and C _2s are charged at the same time;

[3] Mode 3 [t ₂ ~ t ₃ ]: Both S ₁ and S ₂ are off, D ₁ does not work, D ₂ works, _Ca and C _2s are charged at the same time, and u _1c remains unchanged;

[4] Mode 4 [t ₃ ~ t ₄ ]: S ₁ is turned off and S ₂ is turned on, D ₁ and D ₂ are not working, _Ca and C _2s are discharged at the same time, and u _1c remains unchanged;

[5] Mode 5 [t ₄ ~ t ₅ ]: S ₁ is turned off and S ₂ is turned on, D ₁ works, D ₂ does not work, C _1s , C _a and C _2s discharge simultaneously;

[6] Mode 6 [t ₅ ~ t ₆ ]: Both S ₁ and S ₂ are turned off, D ₁ works, D ₂ does not work, and C _1s and C _a discharge at the same time. When D ₁ and D ₂ are turned on in modes 3 and 6, the terminal voltages of S ₁ and S ₂ are zero, thus realizing ZVS soft switching.

Based on the above analysis, the expression of the adjustable equivalent capacitance value C _eq when 0.25<D≤0.5 is:

Case 3: 0.5<D. The charge-discharge cycle of C _1s and C _2s is completely synchronized with the charge-discharge cycle of C _a , at this time, the control D cannot adjust C _eq , that is, the soft-switching controllable capacitor fails. To sum up, the effective adjustment range of D is 0-0.5 and the adjustment range of C _eq is C _a ～(C _a +2C).

(3) Dynamic tuning method

(a) Phase judgment method. The phase relationship between the two signals is obtained by the digital zero-crossing phase detection method shown in Figure 4. The working principle is as follows: The two channels of the same frequency sine signal are processed and input to the A/D converter, and the controller performs data processing and phase difference. calculate.

In Fig. 4, the periods of the signals i ₁ and i _1p are both T and m and _n are the sample value serial _numbers . 1) Zero-crossing between samples and n samples. At this time, the method for determining the phase relationship between i ₁ and i _1p is expressed as follows: when i _{1p crosses} zero, i ₁ <0 corresponds to i _1p leading i ₁ , that is, γ _d >0; i ₁ >0 corresponds to i _1p lags i ₁ , that is, γ _d <0.

Assuming that i ₁ and i _1p are straight lines around the zero-crossing point and the sampling period is T _s , (i _{1p_m} -i _{1p_m-1} )/T _s is the slope of (m-1) point and m point, (i _{1p_m} T _s ) /(i _{1p_m} -i _{1p_m-1} ) is the interval between the zero-crossing point of the instantaneous value of i _1p and the m point, and the interval Δt between the two zero-crossing points is expressed as formula (7). Similarly, the analysis of i ₁ is similar. According to Fig. 4 and equation (7), the phase difference γ _d is deduced and expressed as equation (8).

(b) Zero value search method. As shown in Figure 5, referring to the maximum power tracking in photovoltaic power generation, the variable-step disturbance observation method is used to search for zero phase difference. The working principle is: the disturbance step is determined by the absolute value of the slope of each point on the curve. When the measured value is far from the maximum power point , use a larger step size to improve the tracking speed; when the measured value is close to the maximum power point, use a smaller step size to ensure the tracking speed, so as to achieve a fast and high-precision tracking effect.

(c) Schematic diagram of the dynamic tuning method

As shown in Figure 6, the hardware circuit for dynamic tuning mainly includes soft-switching controllable capacitors and a zero-phase difference search circuit.

The working principle of the control circuit of the soft-switching controllable capacitor shown in Figure 6(a): the current sensor collects the resonant current flowing through the main circuit of the soft-switching controllable capacitor, and generates two PWM signals with a phase difference of 180° based on this. Thus, the ZVS soft switching of the two switching tubes in the soft-switching controllable capacitor is realized. The output signal of the zero-phase-difference search circuit controls the duty cycle D of the above-mentioned two PWM signals, thereby adjusting the equivalent capacitance value.

The working principle of the dynamic tuning method shown in Fig. 6(b): i ₁ and i ₂ collected by the current sensor, after passing through the signal conditioning circuit and high-speed A/D converter, the controller calculates i ₁ and i ₂ and i ₁ and u The phase difference between _s and the variable-step perturbation observation method is used to search for the zero phase difference, so as to output the control signal of the soft-switching controllable capacitor.

Specific embodiments of the present invention are proposed below:

(1) Zero-phase frequency stabilization criterion

Assuming the parameter drift of the inductance or capacitance at the receiving end, the working waveform is shown in Figure 7(a) and the simulation results of γ ₂ and γ _2d varying with the LC resonant frequency f ₂ of the receiving end are shown in Fig. 7(b). Among them, waveform 1 is i ₁ , waveform 2 is i ₂ , waveform 3 is waveform i _2d with i ₂ lag by 90°, waveform 4 is square wave A after i ₁ zero-crossing comparison, and waveform 5 is square wave B after i _2d zero-crossing comparison , and waveform 6 is a square wave AB after A and B XOR.

As shown in Figure 7, the working states of the receiving end are divided into three types: i ₁ leads i ₂ , Z ₂ is capacitive and γ _d2 >0; i ₁ and i ₂ are in the same phase, Z ₂ is resistive and γ _d2 =0 ; i ₁ lags i ₂ , Z ₂ is inductive and γ _d2 <0. Obviously, the theoretical analysis and simulation results show that: Equation (1) can be used as the frequency stability criterion for the series compensation of the receiving coil.

When the receiving coil works in the resonance state, assuming that the self-inductance value of the transmitting coil or the compensation capacitor value has parameter drift, the simulation waveform between the output voltage and output current of the full-bridge inverter is shown in Figure 8. Obviously, there are three working states: u _s leads i ₁ , γ _II >0; u _s is in phase with i ₁ , γ _II = 0; u _s lags i ₁ , γ _II <0. Obviously, both theoretical analysis and simulation results show that: Equation (3) can be used as the frequency stability criterion for the series compensation of the transmitting coil.

(2) Specific examples of soft-switching controllable capacitors

Both C _1s and C _2s are 220nF, the proportional coefficient γ=C ₂ /C _1s ranges from 2.1 to 3.1, the inductance that forms series resonance with the soft-switching controllable capacitor is 53μH, and the normalized equivalent capacitance value C The simulation results between _eq /C ₂ and the operating frequencies f and D are shown in Figure 9. It can be seen that adjusting D achieves monotonic and continuous adjustment of C _eq and f, and different γ corresponds to C _eq and f in different value ranges.

(3) The specific embodiment of the dynamic tuning method

It can be seen from Fig. 10(a) that the variable-step disturbance observation method is used to control the duty cycle D of the PWM signal of C _{2_eq} , and γ _2a = 0 in the process of adjusting C _{2_eq} corresponds to the complete resonance of the receiving coil, and the amplitude of i ₂ at this time maximum. It can be seen from Figure 10(b) that the output voltage U _o is the largest when the receiving coil is fully resonated. Meanwhile, the dynamic tuning process of the transmitting coil is similar to that in Fig. 10.

Claims (7)

1. A wireless charging system dynamic tuning method facing a string compensation topology is characterized by comprising the following steps:

step one, obtaining a zero-phase frequency stabilization criterion of a series compensation topology based on a hardware circuit which is provided with a soft switch controllable capacitor with a symmetrical structure and can realize dynamic tuning;

step two, realizing a zero-phase frequency stabilization criterion of the series compensation topology through a phase judgment method and a zero value search method;

and step three, adjusting the equivalent capacitance value of the soft switch controllable capacitor according to a zero-phase frequency stabilization criterion, thereby carrying out dynamic tuning.

2. The method for dynamically tuning the wireless charging system oriented to the string compensation topology according to claim 1, wherein in step one, the hardware circuit capable of realizing dynamic tuning is equivalent to a mutual inductance model of the string compensation topology, and the mutual inductance model includes: the transmitting terminal comprises a direct current power supply, a transmitting terminal coil and a transmitting terminal soft switch controllable capacitor which are sequentially connected to form a closed loop, the receiving terminal comprises an equivalent load, a receiving terminal coil and a receiving terminal soft switch controllable capacitor which are sequentially connected to form a closed loop,

definition of L₁And L₂Self-inductance values, C, of the transmitting-side coil and the receiving-side coil, respectively₁And C₂Compensation capacitance values of the transmitting end coil and the receiving end coil respectively, M is a mutual inductance value between the transmitting end coil and the receiving end coil, i₁And i₂Resonant currents, R, in the transmitting-side coil and the receiving-side coil, respectively_oAnd Uo are the resistance and voltage values of the equivalent load, u, respectively_sIs the output voltage value of the inverter.

3. The dynamic tuning method of wireless charging system facing to string compensation topology as claimed in claim 1, wherein the zero-phase frequency stabilization criterion comprises a receiving-end frequency stabilization criterion and a transmitting-end frequency stabilization criterion, wherein,

the receiving end frequency stabilization criterion is as follows: l is₂And C₂In the presence of parameter drift, the resonant current i in the receiving coil₂With resonant current i in the transmitting coil₁The phasor expression between is:

the formula (1) shows that: when the receiver is operating in resonance, i₂Lead i₁And a phase difference of gamma₂Is 90 degrees; when the receiving end works in the state of vibration loss, gamma shown in formula (2)₂Not 90 degrees, will i₂Lag by 90 DEG and i₁The phase difference therebetween is defined as gamma_d2I.e. gamma when the receiver is operating in resonance_d2＝0，

The transmitting end frequency stabilization criterion is as follows: assuming that the receiving end works in a resonance state and the transmitting end works in the resonance state, the full-bridge inverter outputs a voltage u_sAnd i₁The phasor relationship between them is expressed by the formula (3),

as can be seen from the formula (3): when the transmitting end works in a resonance state, u_sAnd i₁In phase, i.e. gamma₁Is 0 degree; when the transmitting end works in a detuning state, gamma shown in formula (4)₁Not at 0 deg., correcting gamma when considering ZVS soft switch of switch tube in full bridge inverter₁Implementation of i₁With appropriate hysteresis u_sAnd using the obtained signal as the criterion of resonance state of transmitting terminal,

4. the dynamic tuning method for the wireless charging system oriented to the string compensation topology according to claim 1, wherein the soft-switched controllable capacitor with the symmetric structure comprises: switch tube S₁Switch tube S₂Diode D₁Diode D₂Capacitor C_aCapacitor C_1sAnd a capacitor C_2sSaid capacitor C_1sSwitch tube S₁Capacitor C_2sAnd a switching tube S₂Are sequentially connected in series, the capacitor C_aAnd said capacitor C_1sAnd a switching tube S₁Formed bridge arm and capacitor C_2sAnd a switching tube S₂The formed bridge arms are connected in parallel, and the diode D₁And a switching tube S₁Antiparallel, the diode D₂And a switching tube S₂The main circuit current of the soft switch controllable capacitor with a symmetrical structure is i_sAnd terminal voltage is u_s。

5. The method for dynamically tuning the wireless charging system of the string compensation topology according to claim 4, wherein S is₁Control signal of and i_sIs synchronized with the zero crossing point of S₂The control signal is delayed by half a cycle, thereby realizing the switch tube S₁And a switching tube S₂The ZVS soft switch, meanwhile, the symmetrical controllable capacitor adopts two switching tubes to realize symmetrical conduction in positive and negative half periods, thereby the waveform is symmetrical and has no distortion and no DC bias,

let C be_1s＝C_2sC, according to the switching tube S₁And a switching tube S₂The duty ratio D value intervals of the control signals are different, and the working modes of the soft switch controllable capacitor are as follows: d is more than or equal to 0 and less than or equal to 0.25, D is more than 0.25 and less than or equal to 0.5, and D is more than 0.5,

the first condition is as follows: d is more than or equal to 0 and less than or equal to 0.25, and in one period, the switch tube S is switched on and off₁And a switching tube S₂And a diode D₁And a diode D₂Each conducting for a time of 2D pi, and 6 operating modes corresponding to t₀～t₆In the 6 time periods, the number of the channels is equal to or less than the total number of the channels,

mode 1[ t ]₀～t₁]: switch tube S₁Conducting and switching tube S₂Turn-off, diode D₁And a diode D₂All are not working, capacitor C_1sAnd a capacitor C_aSimultaneous charging, capacitor C_2sTerminal voltage u of_2cKeeping the same;

mode 2[ t ]₁～t₂]: switch tube S₁And a switching tube S₂Are all turned off, diode D₁And a diode D₂All are not working, capacitor C_1sAnd a capacitor C_2sNot charged and capacitor C_1sTerminal voltage u of_1cAnd u_2cRemaining unchanged, capacitance C_aCharging;

mode 3[ t ]₂～t₃]: switch tube S₁And a switching tube S₂Are all turned off, diode D₁Non-operating, diode D₂Operation, capacitance C_aAnd a capacitor C_2sAre charged simultaneously and u_1cKeeping the same;

mode 4[ t ]₃～t₄]: switch tube S₁Switch off and switch tube S₂Conducting, diode D₁And a diode D₂All are not working, capacitor C_aCharging and capacitor C_2sDischarge, u_1cKeeping the same;

mode 5[ t ]₄～t₅]: switch tube S₁And a switching tube S₂Are all turned off, diode D₁And a diode D₂All are not working, capacitor C_aDischarge, u_1cAnd u_2cKeeping the same;

mode 6[ t ]₅～t₆]: switch tube S₁And a switching tube S₂Are all turned off, diode D₁Operating, diode D₂Off-working, capacitor C_1sAnd a capacitor C_aDischarge, u_2cDiode D in mode 3 and mode 6, which remains unchanged₁And a diode D₂When conducting, the switch tube S₁And a switching tube S₂End voltage of is zero, thereby realizing the switch tube S₁And a switching tube S₂The soft switching of the ZVS of (1),

based on the above analysis, the equivalent capacitance C is equal to or greater than 0 and equal to or less than 0.25_eqThe expression of (a) is as follows,

case two: d is more than 0.25 and less than or equal to 0.5, and the capacitance C_1sAnd a capacitor C_2sCross, diode D, during charging₁And a diode D₂The conduction time is changed, and 6 working modes correspond to t₀～t₆In the 6 time periods, the number of the channels is equal to or less than the total number of the channels,

mode 1[ t ]₀～t₁]: switch tube S₁Conducting and switching tube S₂Turn-off, diode D₁And a diode D₂All are not working, capacitor C_1sAnd a capacitor C_aSimultaneous charging, u_2cKeeping the same;

mode 2[ t ]₁～t₂]: switch tube S₁Conducting and switching tube S₂Turn-off, diode D₁Non-operating, diode D₂Operation, capacitance C_1sCapacitor C_aAnd a capacitor C_2sCharging at the same time;

mode 3[ t ]₂～t₃]: switch tube S₁And a switching tube S₂Are all turned off, diode D₁Non-operating, diode D₂Operation, capacitance C_aAnd a capacitor C_2sSimultaneous charging, u_1cKeeping the same;

mode 4[ t ]₃～t₄]: switch tube S₁Switch off and switch tube S₂Conducting, diode D₁And a diode D₂All are not working, capacitor C_aAnd a capacitor C_2sSimultaneous discharge of u_1cKeeping the same;

mode 5[ t ]₄～t₅]: switch tube S₁Switch off and switch tube S₂Conducting, diode D₁Operating, diode D₂Off-working, capacitor C_1sCapacitor C_aAnd a capacitor C_2sDischarging at the same time;

mode 6[ t ]₅～t₆]: switch tube S₁And a switching tube S₂Are all turned off, diode D₁Operating, diode D₂Off-working, capacitor C_1sAnd a capacitor C_aSimultaneous discharge, diode D in mode 3 and mode 6₁And a diode D₂When conducting, the switch tube S₁And a switching tube S₂The end voltage of the switching element is zero, so that ZVS soft switching is realized,

based on the above analysis, the adjustable equivalent capacitance C is more than 0.25 and less than or equal to 0.5_eqIs expressed as

Case three: d is more than 0.5 and the capacitance C_1sAnd a capacitor C_2sCharge-discharge period and capacitor C_aThe charging and discharging periods of (A) are completely synchronous, and at the moment, the control D can not adjust C_eqThat is, the controllable capacitance of the soft switch is failed, and in summary, the effective regulation interval of D is 0 to 0.5 and C_eqThe adjustment range is C_a～(C_a+2C)。

6. The dynamic tuning method of the wireless charging system oriented to the series compensation topology as claimed in claim 5, wherein in step three, the control circuit of the soft switch controllable capacitor operates according to the following principle: the current sensor collects the resonance current flowing through the soft switch controllable capacitor main circuit, and generates two paths of PWM signals with the phase difference of 180 degrees by taking the resonance current as a reference, thereby realizing the switching tube S in the soft switch controllable capacitor₁And a switching tube S₂The output signal of the ZVS soft switch and the zero phase difference search circuit controls the duty ratio D of the two paths of PWM signals, thereby adjusting the equivalent capacitance value.

7. The dynamic tuning method of the wireless charging system for the string compensation topology according to claim 5, wherein in step three, the working principle of the dynamic tuning method is as follows: i collected by current sensor₁And i₂After passing through the signal conditioning circuit and the high-speed A/D converter, the controller calculates i₁And i₂And i₁And u_sAnd searching for zero phase difference by adopting a variable step size disturbance observation method, thereby outputting a control signal of the soft switch controllable capacitor.

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