CN1115043C - Digital TV receiver circuitry for detecting and suppressing NTSC Co-channel interference - Google Patents

Abstract

Compare the symbol decoding results obtained by using the comb filter to suppress the artifacts of NTSC co-channel interference before symbol decoding and the symbol decoding results obtained without using the comb filter before symbol decoding, in order to determine which symbol to choose The decoding result is used as the final symbol decoding result. When NTSC co-channel interference occurs, the symbol decoding result obtained by using the comb filter to suppress the artifacts of NTSC co-channel interference is selected as the final symbol decoding result. Check the digital TV signal to determine if the 4.5 MHz NTSC intercarrier is acquired to confirm the presence of significant NTSC co-channel interference.

Description

Digital TV receiver circuit for detecting and suppressing NTSC co-channel interference

field of invention

The present invention relates to digital television systems, such as digital high definition television (HDTV) systems for US terrestrial broadcasting according to Advanced Television Subcommittee (ATSC) standards, particularly A digital television receiver with a circuit for detecting co-channel interference on television signals.

This application is a continuation-in-part of US Patent Application Serial No. 08/839,691 filed April 15,1997.

Background technique

The digital television standard promulgated by the Advanced Television Subcommittee (ATSC) on September 16, 1995 stipulates that the vestigial sideband (VSB) signal of a digital television (DTV) signal is transmitted in a 6MHz bandwidth television channel. (NTSC) A signal used in radio broadcasting of an analog television signal. The VSBDTV signal is designed so that its spectrum may interleave with the spectrum of the co-channel interfering NTSC analog TV signal. This is done by positioning the DTV signal's pilot carrier and main AM sideband frequencies between the quarter horizontal scans of the NTSC analog TV signals that fall between even multiples of the quarter horizontal scan line rate of the NTSC analog TV signals at odd multiples of the line rate, where the energy of most of the luminance and chrominance components of the co-channel interfering NTSC analog TV signal will fall at these even multiples. The video carrier of the NTSC analog TV signal is offset by 1.25 MHz from the lower frequency limit of the TV channel. The carrier of the DTV signal is offset from the video carrier by 59.75 times the horizontal scanning line rate of the NTSC analog TV signal so that the carrier of the DTV signal is located approximately 309,877.6 kHz from the lower frequency limit of the television channel. Therefore, the carrier of the DTV signal is approximately 2,690, 122.4 Hz from the intermediate frequency of the TV channel.

The exact symbol rate in the digital television standard is 684/286 times the 4.5MHz sound carrier offset from the video carrier in the NTSC analog TV signal. 684 is the number of symbols per horizontal scanning line in the NTSC analog TV signal, and 286 is multiplied by the horizontal scanning line rate in the NTSC analog TV signal to obtain the sound carrier with a 4.5MHz offset from the video carrier in the NTSC analog TV signal coefficient. The symbol rate is 10.762238 megasymbols per second, which can be contained in a VSB signal extending 5.381119 MHz from the DTV signal carrier. That is, the VSB signal can be limited to a frequency band extending 5.690997 MHz from the lower limit frequency of the television channel.

The ATSC standard for digital HDTV signals broadcast terrestrially in the United States is capable of carrying either of two high definition television (HDTV) formats with a 16:9 aspect ratio. One HDTV display format uses 1920 samples per scan line and 1080 effective horizontal scan lines per 30 Hz frame with 2:1 field interleaving. Another HDTV display format uses 1280 luminance samples per scan line and 720 scan lines of progressively scanned television pictures per 60 Hz frame. In addition to HDTV display formats, the ATSC standard also accommodates DTV display formats, eg, the parallel transmission of four television signals of normal definition compared to NTSC analog TV signals.

DTV transmitted by vestigial sideband (VSB) amplitude modulation (AM) during terrestrial broadcasting in the United States includes a series of time-continuous data fields, and each data field includes 313 time-continuous data segments. It can be considered that the digital fields are consecutively numbered modulo 2, forming a data frame from each odd data field and the subsequent even data field. The frame rate is 20.66 frames per second. The duration of each data segment is 77.3 microseconds. Thus, for a symbol rate of 10.76MHz, there are 832 symbols per data segment. Each segment of data begins with a row sync block of four symbols having consecutive values +S, -S, -S, and +S. The +S value is one level below the maximum positive data offset, and the -S value is one level above the maximum negative data offset. The initial row of each data field includes a field sync block that encodes the training signal used for the channel equalization and multipath suppression processes. The training signal is a 511 sample pseudonoise sequence (or PN sequence) followed by three 63 sample PN sequences. The middle part of the 63-sample PN sequence in the field synchronization code is transmitted according to the first logical transition in the first row of each odd-numbered data field and according to the second logical transition in the first row of each even-numbered data field, The first and second logic transitions are complementary to each other.

The data in the data row is trellis encoded using 12 interleaved trellis codes, with one uncoded bit per 2/3 rate trellis code. Reed-Solomon forward error correction coding is applied to the interleaved trellis code, which provides correction for burst errors arising from noise sources such as nearby unshielded automobile ignition systems. Reed-Solomon encoding results are transmitted as 8-level (3 bits/symbol) one-dimensional constellation symbol encoding for radio transmissions without symbol precoding separated from the trellis encoding process. Reed-Solomon encoding results are transmitted as 16-level (4 bits/symbol) one-dimensional constellation symbol encodings for cable broadcasting without precoding. A VSB signal suppresses its natural carrier, which can vary in amplitude depending on the modulation percentage.

The natural carrier is replaced by a pilot carrier of fixed amplitude, the amplitude of which corresponds to a predetermined modulation percentage. This fixed-amplitude pilot carrier is generated by introducing a DC component shift to the modulation voltage applied to the balanced modulator, which generates the AM sidebands that are supplied to the filter, which provides the VSB signal as its response. If the 8-level 4-bit symbol code has normalized values -7, -5, -3, -1, +1, +3, +5 and +7 in the carrier modulation signal, the pilot carrier has normalized values The normalized value is 1.25. The normalized value of +S is +5, and the normal value of -S is -5.

In the early development of DTV technology, it was considered that a DTV broadcast station could be requested to decide whether to use a symbol precoder at the transmitter, which follows the symbol generation circuit and provides precode filtering of the symbols. This decision by the broadcaster depends on whether interference from co-channeled NTSC broadcasters is expected. The symbol precoder supplements the symbol postcoding incidentally introduced in every DTV receiver by using a comb filter before the data-slicer in the symbol decoder circuit in order to eliminate NTSC Artifacts of co-channel interference signals. No symbol precoding is used for the data line synchronous coding group or during the data line during which the data field synchronous data is transmitted.

Co-channel interference is reduced at great distances from NTSC broadcasting stations. Co-channel interference is likely to occur when certain ionization conditions exist. The summer months of the year are known for their potential for co-channel interference. Of course, such interference would not occur without a co-channel NTSC broadcast station. If there is a possibility of NTSC interference in the broadcast coverage area, it is assumed that HDTV broadcasts will use a symbol precoder to make the HDTV signal easier to separate from NTSC interference; therefore, a comb filter is used as the symbol in the DTV receiver Post-encoder to perform matched filtering. If the possibility of NTSC interference does not exist or is not realistic, in order for the noise to have a flat spectrum and less likely to make wrong decisions about symbol values in the trellis decoder, it is assumed that the DTV broadcast station will use intermittently The symbol precoder, therefore, disables the symbol postcoder in every DTV receiver. When the broadcaster is unaware of the situation, due to anomalous skip conditions, due to cable broadcast leaks, due to insufficient IF picture suppression in NTSC receivers, due to tapes used for digital TV recordings with remnants of previous analog TV recordings, or due to In certain other exceptional cases, the actual co-channel interference may be considerable for parts of the broadcast reception area.

U.S. Patent No. 5,594,496, issued January 14, 1997 to Nielsen et al., entitled "Detection of Co-channel Interference in Digital Television Signals," describes an NTSC co-channel interference detector in which data segments containing field sync codes are Field comb filtering to obtain the data field comb filter response with periodic intervals without symbol encoding where received noise and co-channel interference can be estimated. The data field comb filter response during these periodic intervals is compared to the response subjected to further comb filtering which suppresses the artifacts of NTSC co-channel interference. If further comb filtering results in a significant reduction in signal level, the signal is assumed to contain artifacts of considerable NTSC co-channel interference and comb filtering to suppress the artifacts of NTSC co-channel interference is applied prior to symbol decoding. If further comb filtering did not result in a significant reduction in the detected noise, the noise was presumed to be mainly Johnson noise, and symbol decoding was performed without prior comb filtering to suppress artifacts of NTSC co-channel interference. This is due to the approximately 3dB increase in Johnson noise associated with the comb filtering of symbol encoding by linearly combining differential delays.

The current ATS CDTV standard does not allow the transmitter to use symbol precoding. After the data restriction process associated with symbol decoding, it is assumed that suppression of co-channel interference analog TV signals is to be performed in the trellis decoding process. This procedure avoids the problem of determining whether to precode at the transmitter. However, co-channel interfering with analog TV signals introduces undesired errors into the data limiting process, which imposes a burden on the error correction decoding process, trellis decoding, and Reed Solomon decoding. These errors would reduce the broadcast coverage area, which would result in lost revenue for commercial DTV broadcasters. Therefore, although the current ATS CDTV standard does not allow symbol precoding at the DTV transmitter, it is still desirable to provide suppression of co-channel interfering analog TV signals before data limitation.

The term "linear combination" generally refers to performing addition and subtraction, whether by conventional or modular arithmetic. The term "modular combination" refers to linear combinations performed according to modular arithmetic. Coding that recodes a stream of digital symbols by a linear combination of differential delays and differential delay terms, exemplified by symbol postcoding employed in prior art HDTV receivers, is defined in this specification as "Class I code element recoding". The encoding that re-encodes a stream of digital symbols by its modulo combination with the delayed result of the modulo combination, exemplified by the symbol precoding employed in prior art HDTV transmitters, is defined in this specification as "second class symbol recoding".

From the standpoint of being an interference problem that sometimes occurs with receivers, co-channel interference from analog television signals can be resolved by adaptive filtering circuitry in the receiver. As long as the dynamic range of the system channel is not exceeded such that co-channel interference can capture the system channel by destroying the signal transfer capability for DTV modulation, the performance of the system can be viewed as a signal superposition problem. The filter circuit in the receiver is adapted to select digital signals from co-channel interference produced by analog television signals based on their apparent correlation and anti-correlation properties in order to reduce the energy of analog television signals sufficiently to capture system channels from them .

In the case of co-channel interference from analog TV signals, it enters the system channel after the DTV transmitter and before the DTV receiver. The use or absence of symbol precoding at the DTV transmitter has no effect on co-channel interference from analog television signals. In a DTV receiver, as long as the co-channel interference is not so large that it overloads the receiver front-end and captures the system channel, it is best to add a comb filter before the data limiting circuit to reduce the energy of the high-energy spectral components of the co-channel interference, thereby An error occurred during data limit reduction. DTV broadcasters should adjust their carrier frequency, which is usually 310KHz above the lower frequency limit of television channel allocation, so that their carrier frequency is optimally offset in frequency from the video carrier of potentially interfering co-channel NTSC analog TV signals. The optimum deviation of the carrier frequency is exactly 59.75 times the horizontal scanning line frequency f _H of the NTSC analog TV signal. The artifacts of co-channel interference in the demodulated DTV signal will then include the extraneous noise between the digital HDTV carrier and the video carrier of the co-channel interfering analog TV signal at 59.75 times the horizontal scanning line frequency f _H of the NTSC analog TV signal. Beats due to difference; beats at 287.25 times f _H by heterodyning between the digital HDTV carrier and the chrominance subcarrier of the co-channel interfering analog TV signal, these beats have frequencies very close to those at f _H The fifth harmonic of the beat at 59.75 times of . These artifacts will also include beats at about 345.75 times f _H resulting from heterodyning between the digital HDTV carrier and the audio carrier of the co-channel interfering analog TV signal, which have frequencies very close to those at f _H The sixth harmonic of the beat at 59.75 times. The near harmonic relationship of these beats allows them to be rejected by a properly designed comb filter with a constant phase delay (epoch) of symbols containing only a few differential delays. The first type of symbol re-encoding is additionally performed using an NTSC rejection comb filter prior to data slicing in a DTV receiver to alter the symbols obtained by data slicing.

As far as data transmission is concerned, since the design data quantization level matches the symbol level, the data slicing operation after the re-encoding of the first type of symbol in the DTV receiver is a method that does not destroy the re-encoding of the first type of symbol. The quantization process of the coded symbols. However, the quantization identifies co-channel interfering analog TV signal residues that remain after filtering and are significantly smaller than the step size between symbol levels associated with the first type of symbol re-encoding. This is a trapping phenomenon in which a stronger signal is obtained at the expense of weakening one during quantization.

For data transfer, a stream of digital data symbols flows the entire length of the system channel. When the second type of symbol recoding is performed as symbol precoding at the DTV transmitter, the additive combination of differentially delayed data symbol streams is performed on a block-by-block basis without increasing transmitter power or increasing the average inter-symbol distance, Helps further overcome interference with analog TV signals. Alternatively, the primary mechanism for overcoming the interfering analog TV signal is its attenuation relative to the DTV signal provided by the comb filter of the DTV receiver such that the residual analog TV signal in the comb filter response is immediately filtered by the comb filter Quantization effects in the subsequent data slicer are suppressed.

Since neither encoding scheme destroys the transportability of the symbol stream, the order in which the first and second type of symbol re-encoding processes are carried out in this case has no appreciable effect on the signal transport over the system channels. The order in which the first and second type symbol recoding processes are performed suppresses co-channel interference for digital receivers as long as the second type symbol recoding does not overlap between the first type symbol recoding and subsequent data slicing The ability to simulate TV signals has no noticeable effect.

These understandings provide the inventor's U.S. Patent Application Serial No. 08/839,691, filed April 15, 1997, entitled "Digital Television Receiver With Adaptive Filter Circuit For Suppressing NTSC Co-Channel Interference" The general starting point for the device described in . The adaptive filter circuit receives a 2N-level symbol stream sensitive to artifacts associated with co-channel interference analog television signals for symbol decoding, where N is a positive integer. NTSC co-channel interference is detected in the adaptive filter circuit and it is determined whether the NTSC co-channel interference has sufficient energy to introduce uncorrectable errors into the data clipping process used to detect the VSB-AM DTV signal via synchronous detection The recovered baseband signal is directly decoded by symbols. If it is determined that the NTSC co-channel interference does not have sufficient energy to cause uncorrectable errors, the baseband signal is symbol decoded using the first data slicer to produce symbol decoded results. If it is determined that the NTSC co-channel interference has sufficient energy to cause uncorrectable errors, the baseband signal is filtered with the first comb filter to reduce the energy of the co-channel interference prior to symbol decoding with the second data slicer. The first comb filter additionally performs a first type of symbol re-encoding process to introduce errors into the decoded symbols produced by the second data slicer. As far as adaptive filtering for suppressing NTSC co-channel interference is concerned, the first type of symbol re-encoding process performed before the second data limiter performs data clipping is regarded as a pre-coding process. The second comb filter performs the second type of symbol re-encoding process after data clipping by the second data slicer, implements a post-encoding process to compensate for the first type of symbol re-encoding process and generates a symbol encoding result.

A first type of symbol re-encoding process re-encodes an input symbol stream by means of differential delays and first linear combinations of differential delay terms. A second type of symbol re-encoding process re-encodes the partially filtered symbol decoding results recovered by the second data slicer. This second type of symbol re-encoding process utilizes a second linear combination of the partially filtered symbol decoding result with the symbol decoding result previously fed back with the same delay as the differential delay introduced into the input symbol stream, according to the modulo 2N algorithm. A second linear combination is used to generate the decoded result of the postcoded symbols. Running errors in post-decoding element decoding results are reduced by forcing the symbol decoding results to coincide with ideal symbol decoding results retrieved from memory in the DTV receiver at the time the data field sync information and data segment sync information are present. One of the first and second linear combinations is negative and the other is positive.

The present invention relates to estimates based on selection from decoded results of post coded symbols obtained after comb filtering to suppress NTSC co-channel interference, rather than data clipping from encoding baseband symbols without suppressing NTSC co-channel interference The estimates selected from the intermediate symbol decoding results obtained by comb filtering are used to determine when it is more likely to correct the final symbol decoding results. This determination is made by comparing the decoding result of each post coded symbol with the corresponding intermediate symbol decoding result throughout each data segment. Assuming that due to the existence of NTSC artefacts in the baseband symbol encoding, the decoded results of the post-encoded symbols deviate greatly from the corresponding intermediate symbol decoding results, so the decoded results of the post-encoded symbols are selected to be included in the final symbol decoding results, and do not select intermediate decoding results, unless other information indicates that the selection is likely to be wrong.

Contents of the invention

A digital television signal receiver, comprising: a digital television signal detection device for providing a flow of 2N-level symbols each having a constant phase of symbols of a specified time length, N is a positive integer, and the flow of said 2N-level symbols is easy Subject to artifacts associated with co-channel interference analog television signals, said symbols are grouped into consecutive data segments with headers corresponding to the sync codes of the data segments grouped into initial successive data fields of the data segment, each data field comprising a data field sync code varying from data field to data field; circuitry for providing M unique comb filter responses to said 2N level symbol stream, Each of said unique comb filters is less affected by artifacts associated with co-channel interference analog television signals than said 2N-level symbol streams, where M is a positive integer; a plurality of symbol decoders, For generating corresponding estimated symbol decoding results, the first of the plurality of symbol decoders responds to the 2N level symbol stream to generate a first estimated symbol decoding result, and the plurality of symbol decoders Each of the other symbol decoders responds to a corresponding one of the M unique comb filter responses to generate a corresponding estimated symbol decoding result whose corresponding estimated symbol decoding result is post-coded so that for the M said corresponding one of the unique comb filter responses performs a corresponding matched filter, said corresponding estimated symbol decoding result is obtained from said M unique comb filter responses, and all of said plurality of symbol decoders The other symbol decoders include a second symbol decoder for generating a second estimated symbol decoding result;

According to the present invention, a digital television signal receiver comprises a circuit for detecting whether there is currently a deviation between the decoding results of said first and second estimated symbols; and a best estimate selection circuit for selecting from said corresponding estimated symbol The meta decoding result selects the best estimate to produce the final symbol decoding result at times between those times when the synchronization code occurs, the selection of the best estimate being dependent on the first estimated symbol decoding result in relation to the other estimated symbol decoding results The bias of the meta decoding result.

The present invention also proposes a digital television signal receiver, comprising: a digital television signal detection device, which is used to provide a stream of 2N level symbols each having a constant phase delay of a symbol of a specified time length, where N is a positive integer, so The 2N level symbol flow is susceptible to the impact of the pseudo-generated phenomenon of the analog television signal accompanied by co-channel interference, the symbols are grouped into continuous data segments with the headers of the corresponding data segment synchronization codes, and the data segments are grouped into bands Consecutive data fields having an initial data segment of each data field, each data field comprising a data field synchronization code varying from data field to data field; a first symbol decoder responsive to said 2N level symbols stream produces first estimated symbol decoding results; circuitry for providing a plurality of unique comb filter responses to said 2N level symbol stream, each said unique comb filter being subject to analog television with co-channel interference signal artifacts to a lesser extent than said 2N level symbol stream; a corresponding symbol decoder, responsive to a respective one of said plurality of unique comb filter responses, producing a respective estimated symbol decoding results, post-encoding said corresponding estimated symbol decoding result so that a corresponding matched filtering is performed for said respective one of said unique comb filter responses from which said corresponding estimated symbol decoding result is obtained ; a circuit for obtaining an internal carrier signal from the heterodyne between the audio and video carriers of said co-channel interfering analog television signal; detecting when the amplitude of this carrier signal exceeds a specified level for providing said co-channel interfering an indication that the level of the analog television signal is sufficient to cause an error in said first estimated symbol decoding result produced by said first symbol decoder, or that the level of said co-channel interfering analog television signal is insufficient A circuit that causes an indication of an error in the first estimated symbol decoding result; when detecting the occurrence of a synchronization code in the 2N level symbol stream, providing an indication of the occurrence of the synchronization code in the 2N level symbol stream and providing A circuit for indicating that no synchronization code appears in the 2N level symbol stream; in response to the detected synchronization code occurring in the 2N level symbol stream, for generating an ideal symbol decoding result for the synchronization code A circuit for determining which of said estimated symbol decoding results other than said first estimated symbol decoding result currently has the greatest absolute deviation from said first estimated symbol decoding result to generate which of said other estimated symbol decoding results A circuit for an indication that the meta-decoding result is least likely to be caused by said artifacts of co-channel interfering analog television signals; a multiplexer for reproducing a currently selected one of a plurality of input signals provided providing a final symbol decoding result, said plurality of input signals comprising said first estimated symbol decoding result, each of said other estimated symbol decoding results and said ideal symbol decoding result; selectively controlling said plurality of multiplexer to provide said final symbol decoded result by reproducing said ideal symbol decoded result in response to said indication of the occurrence of a synchronization code in said 2N level symbol stream; said indication that a sync code is not present in the 2N level symbol stream and said indication that said co-channel interfering analog television signal is not at a level sufficient to cause an error in said first estimated symbol decoding result, said multiple a multiplexer is selectively controlled to provide said final symbol decoded result by reproducing said first symbol decoded result; in response to no synchronization code being present in said simultaneously provided 2N level symbol stream , said indication that said level of co-channel interfering analog television signal is insufficient to cause an error in said first estimated symbol decoding result, and said indication that co-channel interfering analog television signal is least likely to be caused by co-channel interfering analog television signal said indication of said other estimated symbol decoding results of noise-induced errors, said multiplexer being selectively controlled to reproduce said noise-induced errors least likely to be caused by co-channel interfering analog television signals said other estimated symbol decoding results for providing said final symbol decoding results;

The present invention also proposes a digital television signal receiver, comprising: a digital television signal detection device, which is used to provide a stream of 2N level symbols each having a constant phase delay of a symbol of a specified time length, where N is a positive integer, so The 2N level symbol flow is susceptible to the impact of the pseudo-generated phenomenon of the analog television signal accompanied by co-channel interference, the symbols are grouped into continuous data segments with the headers of the corresponding data segment synchronization codes, and the data segments are grouped into bands Consecutive data fields having an initial data segment of each data field, each data field comprising a data field synchronization code varying from data field to data field; a first symbol decoder responsive to said 2N level code The element stream produces first estimated symbol decoding results; circuitry for providing first, second, and third unique comb filter responses to said 2N level symbol stream, each of said unique comb filters affected by The degree of influence of artifacts accompanying co-channel interference analog television signals is smaller than that of said 2N level symbol stream; a second comb filter response producing a second estimated symbol decoding result in response to said first comb filter response a symbol decoder, said second symbol decoder comprising a first post-encoding comb filter for providing said second estimate to said first comb filter response in a matched filter response symbol decoding results; a third symbol decoder responsive to said second comb filter to generate a third estimated symbol decoding result, said third symbol decoder comprising a second post-coded comb a filter for providing said third estimated symbol decoding result to said second comb filter response in a matched filter response; and generating a fourth estimated symbol response in response to said third comb filter response A fourth symbol decoder of the element decoding result, said fourth symbol decoder comprising a third post-coding comb filter for responding to said third comb filter in a matched filter response providing said fourth estimated symbol decoding result; a circuit for obtaining an inner carrier signal from a heterodyne between audio and video carriers of said co-channel interfering analog television signal; for detecting when the amplitude of the carrier signal exceeds providing an indication that said co-channel interfering analog television signal is at a level sufficient to cause errors in said first estimated symbol decoding results produced by said first symbol decoder, or providing said co-channel The level of channel interfering analog TV signal is not enough to cause an indication of error in the first estimated symbol decoding result; it is used to detect the occurrence of synchronous code in the 2N level symbol stream, and provide the 2N level An indication of a synchronous code occurring in the symbol stream, and a circuit providing an indication of no synchronous code occurring in the 2N level symbol stream; responding to the detected synchronous code occurring in the 2N level symbol stream, for all Said synchronous code produces the circuit of ideal symbol decoding result; For determining which one of said second, third and fourth estimated symbol decoding results has the largest absolute deviation from said first estimated symbol decoding result, to means for generating an indication of which of said second, third and fourth estimated symbol decoding results is least likely to be error-caused by said artifact of a co-channel interfered analog television signal; a multiplexer, for providing a final symbol decoding result by reproducing a currently selected one of a plurality of input signals provided comprising said first estimated symbol decoding result, said second estimated symbol decoding result , the third estimated symbol decoding result, the fourth estimated symbol decoding result, and the ideal symbol decoding result; selectively controlling the multiplexer so that by responding to the 2N level An indication of a synchronization code appearing in the symbol stream, reproducing the ideal symbol decoding result to provide the final symbol decoding result; in response to the simultaneously provided indication of no synchronization code appearing in the 2N level symbol stream and said indication that the level of said co-channel interfering analog television signal is insufficient to cause an error in said first estimated symbol decoding result, said multiplexer is selectively controlled to reproduce said said first symbol decoding result to provide said final symbol decoding result; in response to said indication that a synchronization code does not appear in said 2N level symbol stream provided at the same time, said co-channel interfering signal of an analog television signal said indication that the level is insufficient to cause an error in said first estimated symbol decoding result, and said second estimated symbol decoding result being least likely to cause an error by said noise of co-channel interfering analog television signals said multiplexer is selectively controlled to provide said final symbol decoding result by reproducing said second symbol decoding result; selectively controlling said multiplexer, so that by responding to said indication that a sync code is not present in said 2N level symbol stream, said co-channel interfering analog television signal is at a level sufficient to cause an error in said first estimated symbol decoding result. Simultaneously with providing an indication that said third estimated symbol decoding result is least likely to cause errors due to said artifacts of co-channel interfering analog television signals, reproducing said third symbol decoding result to provide said said final symbol decoding result; selectively controlling said multiplexer to pass said co-channel interference analog television signal in response to said indication that a sync code is not present in said 2N level symbol stream said indication of a level sufficient to cause an error in said first estimated symbol decode result, and said fourth estimated symbol decode result least likely to be caused by said artifact of a co-channel interfering analog television signal Concurrently with said indication, said fourth symbol decoding result is reproduced to provide said final symbol decoding result.

In one embodiment of the present invention, M is 2, and besides said first symbol decoder, the plurality of symbol decoders only includes a second symbol decoder for generating a second estimated symbol decoding result. In this embodiment of the invention, the best estimate selection circuit takes the following form. A multiplexer is connected to provide the ability to select between first and second estimated symbol decoding results for producing final symbol decoding results at times between those times when the synchronization code occurs. For deviations between first and second estimated symbol decoding results there is a squarer for finding the squared results as absolute values of those deviations, and an averager for producing an average of said squared results. a threshold detector responsive to the average of the squared results exceeding a predetermined threshold for controlling the multiplexer to select the second estimated symbol decoding result to produce the final symbol at times between those times at which said synchronization code occurs The decoded result, or otherwise controls the multiplexer to select the first estimated symbol decoded result to produce the final symbol decoded result at times between those times when the synchronization code occurs.

Description of drawings

Figure 1 is a block diagram of a digital television receiver using an NTSC band-stop comb filter before symbol decoding and a post-encoding comb filter after symbol decoding and using a co-channel jammer detector in accordance with the present invention, co-channel The interference detector compares the symbol decoding result obtained without the NTSC co-channel interference suppression measure with the symbol decoding result obtained with the NTSC co-channel interference suppression measure. One of the first and second linear combiners in Fig. 1 is a subtractor, and the other is an adder.

FIG. 2 is a block diagram showing details of the NTSC co-channel jammer detector used in the digital television signal receiver of FIG. 1. The NTSC co-channel jammer detector according to one aspect of the present invention provides a method for decoding As a result, the best estimate selection circuit selects the best estimate to produce final symbol decoding results at times between those times at which the synchronization code occurs.

Figure 3 is a block schematic diagram showing details of a portion of the digital television signal receiver of Figure 1 when the NTSC band-stop comb filter employs a 12 symbol delay.

Figure 4 is a block schematic diagram showing details of a portion of the digital television signal receiver of Figure 1 when the NTSC band-stop comb filter employs a 6 symbol delay.

Figure 5 is a block schematic diagram showing details of a portion of the digital television signal receiver of Figure 1 when the NTSC band-stop comb filter employs a 2 video line delay.

Figure 6 is a schematic block diagram showing details of a portion of the digital television signal receiver of Figure 1 when the NTSC band-stop comb filter employs a 262 video line delay.

Figure 7 is a schematic block diagram showing details of a portion of the digital television signal receiver of Figure 1 when the NTSC band-stop comb filter employs a 2 video frame delay delay.

FIG. 8 is a block schematic diagram showing details of a portion of the digital television signal receiver of FIG. 1 for generating predetermined symbol decoding results during a data synchronization interval.

Fig. 9 is a block schematic diagram showing a digital television signal receiver utilizing a plurality of NTSC band-stop comb filters for parallel symbol decoding. A, B and C in FIG. 9 are corresponding integers of different values selected from 1, 2, 3, 4 and 5.

Figure 10 is an assembly diagram of how Figures 10A and 10B fit together to form a single figure referred to as Figure 10 described in detail below, and Figure 10 shows a digital television signal receiver of the type shown in Figure 9. Details of the symbol encoding selection circuit.

FIG. 10A is a block schematic diagram showing details of circuitry in the digital television signal receiver of FIG. 9 for generating predetermined symbol decoding results during data synchronization intervals.

FIG. 10B is a diagram illustrating another method for selecting the best estimate from various symbol decoding results in accordance with another aspect of the present invention to produce the best estimate of the final symbol decoding result at times between those times when the synchronization code occurs. A block schematic diagram of details of the circuit in the digital television signal receiver of Fig. 9 for the evaluation selection circuit.

Detailed ways

At various points in the circuits shown in the figures, as will be understood by those skilled in the art of electronic design, shimming delays must be inserted to get the sequence of operations correct. Unless there are exceptional interstitial delay requirements, they are not covered in the description below.

Figure 1 shows a digital television signal receiver for recovering error-corrected data suitable for recording with a digital videocassette recorder or for MPEG (Motion Picture Experts Group)-2 decoding and display on a television. The DTV signal receiver of Figure 1 receives television broadcast signals from a receiving antenna, but may also receive signals from a cable network. The television broadcast signal is provided to the "head-end" electronics 10 as a received signal. "Front end" electronics 10 typically include a radio frequency amplifier and first detector for converting radio frequency television signals to intermediate frequency television signals as input to an intermediate frequency (IF) amplifier chain 12 for vestigial sideband DTV signals. The DTV receiver is preferably a multi-conversion type, and its IF amplifier chain 12 includes an IF amplifier for amplifying the DTV signal converted into the UHF band by the first detector, and an IF amplifier for converting the amplified DTV signal into a UHF band. A second detector for the high frequency band, and another IF amplifier for amplifying the DTV signal converted into the VHF band. If baseband is demodulated digitally, the IF amplifier chain 12 will also include a third detector for converting the amplified DTV signal to a final intermediate frequency band closer to baseband.

Preferably a surface acoustic wave (SAW) filter is used in the IF amplifier for the UHF band to shape the channel selection response and reject adjacent channels. The SAW filter cuts off quickly just above 5.38 MHz, removing it from the carrier frequency of the suppressed VSBDTV signal and the pilot frequency of similar frequency and fixed amplitude. Thus, the SAW filter removes a significant portion of the FM sound carrier of any co-channel interfering analog TV signal. Removing any co-channel interfering FM sound carrier of the analog TV signal in the IF amplifier chain 12 prevents carrier artefacts upon detection of the final IF signal to recover baseband symbols and prevents these artefacts during symbol decoding Data slicing that interferes with those baseband symbols. Data slicing to prevent these artifacts from interfering with those baseband symbols during symbol decoding is better than doing it with a comb filter before slicing.

The final IF output signal from IF amplifier chain 12 is provided to complex demodulator 14 which demodulates the vestigial sideband AM DTV signal in the final intermediate frequency band to recover real and imaginary baseband signals. Demodulation can be done digitally after analog-to-digital conversion in the final intermediate frequency band in the several MHz range, e.g., as described in C.B. Disclosed in US Patent No. 5,479,449 "Digital VSB Detector for Detectors". Alternatively, the demodulation can be performed in an analog fashion, in which case the result is usually converted from analog to digital for further processing. This complex demodulation is preferably performed by in-phase (I) synchronous demodulation and quadrature-phase (Q) synchronous demodulation. The digital result of the demodulation process described above typically has 8 bits or better precision and describes 2N levels of symbols encoding N bits of data. Now, 2N is 8 in case the DTV signal receiver of FIG. 1 receives by wireless broadcasting via the IF amplifier chain 12, and 2N is 16 in the case of the DTV signal receiver of FIG. 1 receiving via cable broadcasting. The present invention relates to terrestrial reception by radio broadcast, and Figure 1 does not show the part of the DTV receiver that provides symbol decoding and error correction decoding for received cable broadcast transmissions.

Symbol synchronizer and equalizer circuit 16 receives at least digitized real samples of the in-phase (I channel) baseband signal from complex demodulator 14; in the DTV receiver of FIG. channel) digitized dummy sampling of the baseband signal. Circuit 16 includes digital filters with adjustable weighting coefficients to compensate for ghosting and slanting in the received signal. Symbol synchronizer and equalizer circuit 16 provides symbol synchronization or "desrotation" as well as amplitude equalization and ghost cancellation. A symbol synchronizer and equalizer circuit in which symbol synchronization is achieved prior to amplitude equalization is known from US Patent No. 5,479,449. In this design, complex demodulator 14 will provide an oversampled demodulator response containing real and imaginary baseband signals to symbol synchronizer and equalizer circuit 16 . After symbol synchronization, the oversampled data is decimated to extract the baseband I-channel signal at the normal symbol rate for downsampling by digital filtering for amplitude equalization and ghost removal. Symbol synchronizer and equalizer circuits 16, in which amplitude equalization precedes symbol synchronization, "derotation" or "phase tracking" are also known to those skilled in the art of digital signal receiver design.

Each sample of the output signal of circuit 16 is decomposed into 10 or more bits and is effectively a digital representation of an analog symbol presenting one of 2N=8 levels. The output signal of circuit 16 is tightly gain controlled by any of several known methods, so that the ideal step levels of the symbols are known. Since the response speed of the gain control is very fast, one method of gain control is preferably to adjust the DC component of the real baseband signal supplied from the complex demodulator 14 to a normalized level of +1.25. This method of gain control is generally described in U.S. Patent No. 5,479,449 and C.B. Patel et al., issued June 3, 1997, entitled "Automatic Gain in a Wireless Receiver for Receiving Digital High Definition Television Signals Control" is more specifically described in U.S. Patent No. 5,573,454, incorporated herein by reference.

The signal output from the circuit 16 is provided as input data to the data synchronization detection circuit 18, and the data synchronization detection circuit recovers the data field synchronization information F and the data segment synchronization information S from the equalized baseband I channel signal. Alternatively, the input signal to the data sync detection circuit 18 may be obtained prior to equalization.

The equalized I-channel signal samples provided as an output signal from circuit 16 at a normal symbol rate are provided as an input signal to an NTSC band-stop comb filter 20 . Comb filter 20 includes first delay means 201 for generating a pair of differentially delayed 2N-level symbol streams, and first delay means 201 for linearly combining the differentially delayed symbol streams to generate the response of comb filter 20 Linear combiner 202 . As described in US Patent No. 5,260,793, the first delay means 201 can provide a delay equal to the period of 12 2N-level symbols, and the first linear combiner 202 can be a subtractor. Each sample of the comb filter 20 output signal is decomposed into 10 or more bits and is actually a digital representation of an analog symbol exhibiting one of (4N-1=15) levels.

Assume that the symbol synchronizer and equalizer circuit 16 is designed to suppress the DC offset component of its input signal (represented in data samples), which has a normalized level of +1.25 and is reduced due to pilot carrier detection. appears in the real baseband signal supplied from the complex demodulator 14. Thus, each sample of the output signal of circuit 16 is applied to comb filter 20 as an input signal, said each sample actually exhibiting the following normalized levels: -7, -5, -3, -1, + 1. A digital description of one of the analog symbols of +3, +5 and +7. These symbol levels are named "odd" symbol levels and are detected by the odd level data slicer 22 to produce intermediate symbol decoding results: 000, 001, 010, 011, 100, 101, 110, and 111, respectively .

Each sample of the comb filter 20 output signal actually exhibits the following normalized levels: -14, -12, -10, -8, -6, -4, -2, 0, +2, +4 , +6, +8, +10, +12 and +14 one of the digital description of the analog symbol. These symbol levels are named "even" symbol levels and are detected by the even level data slicer 24 to generate precoded symbol decoding results respectively: 001, 010, 011, 100, 101, 110, 111, 000, 001, 010, 011, 100, 101, 110 and 111.

As assumed in the description thus far, the data slicers 22 and 24 may be of the so-called "hard-decision" type, or of the so-called "soft-decision" type employed in implementing the Viterbi decoding scheme. Use a multiplexer connection to move its position in the circuit and provide bias to change its slicing range, replacing the odd-level data slicer 22 and the even-level data slicer with a single data slicer Configurations of the controller 24 are possible, but due to operational complexity, these configurations are not optimal.

It has been assumed in the above description that the symbol synchronizer and equalizer circuit 16 is designed to suppress the DC component of its input signal (represented in digital samples), which DC bias component has a normalized level of +1.25, and due to the induced The detection of the frequency carrier occurs in the real baseband signal provided from the complex demodulator 14. Alternatively, the symbol synchronizer and equalizer circuit 16 is designed to preserve the DC bias component of its input signal, which simplifies the design of the equalization filter in circuit 16 somewhat. In this case, the data slicing levels in the odd level data slicer 22 are biased to account for the DC offset component of its input signal accompanying the data pacing. Assuming that first linear combiner 202 is a subtractor, circuit 16 is designed to suppress or preserve the DC bias component of its input signal with no effect on the data slicing level in even level data slicer 24 . However, if the differential delay provided by the first delay means 201 is chosen such that the first linear combiner 202 is an adder, the data slicing level in the even level data slicer 24 should be biased to take into account its input The doubled DC bias component of the signal that accompanies the data pacing.

A comb filter 26 is used after the data slicers 22 and 24 to produce a post-encoding filter response to the pre-encoding filter response of the comb filter 20 . Comb filter 26 comprises a multiplexer 261 of three inputs, a second linear combiner 262, and a second delay device 263 with the same delay as first delay device 201 in comb filter 20 . If the first linear combiner 202 is a subtractor, the second linear combiner 262 is a modulo 8 adder, and if the first linear combiner 202 is an adder, the second linear combiner 262 is a modulo 8 subtraction device. The first linear combiner 202 and the second linear combiner 262 may constitute corresponding read-only memories (ROMs) to sufficiently speed up the linear combination operation to support the associated sampling rate. The output signal from multiplexer 261 provides the response from post-coding comb filter 26 and is delayed by second delay means 263 . The second linear combiner 262 combines the precoded symbol decoding result from the even-level data slicer 24 with the output signal from the second delay device 263 .

After being selected in response to the first, second and third states of the multiplexer control signal supplied to the multiplexer 261 from the controller 28, the output signal of the multiplexer 261 is reproduced and applied to the multiplexer 261. One of the three input signals to multiplexer 261. During the time period when the data synchronization detection circuit 18 restores the data field synchronization information F and the data segment synchronization information S from the equalized baseband I-channel signal, the first input port of the multiplexer 261 receives the data from the memory in the controller 28. Ideal symbol decoding results provided. Controller 28 provides a first state multiplexer control signal to multiplexer 261 during these times, which adjusts multiplexer 261 to provide the ideal symbol decoding supplied from memory within controller 28 The result serves as the final encoding result of its output signal. The odd-level data slicer 22 supplies the intermediate symbol decoding result to the second input port of the multiplexer 261 as its output signal. The multiplexer 261 is conditioned by the multiplexer control signal of the second state to reproduce the intermediate symbol decoded result as the final decoded result of its output signal. The second linear combiner 262 supplies the decoded result of the post coded symbols as its output signal to the third input port of the multiplexer 261 . The multiplexer 261 is conditioned by the multiplexer control signal of the third state to reproduce the decoded result of the post coded symbols as the final coded result of its output signal.

During the time when the time data sync detection circuit 18 restores the data field sync information F and the data segment sync information S, the comb filter from the post-coding is reduced by feeding back the ideal symbol decoding result provided from the memory in the controller 28. The running error in the decoding result of the post coded symbol of 26. This is an important aspect of the invention and will be described in further detail in this specification.

The output signal from the multiplexer 261 in the post-coding comb filter 26 includes the final symbol decoding results in 3 parallel bit groups assembled by the data assembler (assembler) 30 to the data interleaver 32 . Data interleaver 32 converts the packed data into parallel data streams which are applied to trellis decoder circuit 34 . Trellis decoder circuit 34 typically uses 12 trellis decoders. The trellis decoding result is provided from trellis decoder circuit 34 to data deinterleaver 36 for inverse conversion. Byte construction circuit 38 converts the output signal of data interleaver 36 into Reed-Solomon error-correcting encoded bytes that are applied to Reed-Solomon Solomon decoder circuit 40, which performs Reed-Solomon decoding to produce the provided Error corrected byte stream to data derandomizer 42. The data de-randomizer 42 provides the reproduced data to the remainder of the receiver (not shown). The remainder of the overall DTV receiver includes a packet sorter, an audio decoder, an MPEG-2 decoder, and so on. The remainder of the DTV receiver incorporated in the digital tape recorder/player includes circuitry for converting the data into recorded form.

NTSC co-channel jammer detector 44 provides controller 28 with an indication of whether NTSC co-channel jammers are of sufficient magnitude to cause uncorrectable errors in the data slicing performed by data slicer 22 . If the detector 44 indicates that the NTSC co-channel interference is not too strong, the controller multiplexes the second state to the second state at times other than those times when the data field sync information F and the data segment sync information S are recovered by the data sync detection circuit 18. The user control signal is provided to the multiplexer 261. This controls the multiplexer 261 to reproduce the intermediate symbol decoding result supplied from the odd-level data slicer 22 as its output signal. If the detector 44 indicates that the NTSC co-channel interference is strong enough to cause uncorrectable errors in the data slicing performed by the data slicer 22, the controller 28, in addition to recovering the data field sync information F by the data sync detection circuit 18 and The multiplexer control signal of the third state is supplied to the multiplexer 261 at times other than those times when the data segment sync information S is present. This controls the multiplexer 261 to reproduce, as its output signal, the post coded symbol decoding result supplied from the second linear combiner 262 as the second linear combination result.

Figure 2 shows a form of NTSC co-channel interference detector 44 that may be employed in one embodiment of the present invention. The subtractor 441 differentially combines the intermediate symbol decoding result supplied from the odd-level data slicer 22 with the post coded symbol decoding result supplied from the second linear combiner 262 as a second linear combination result. The amount of NTSC co-channel interference is negligible, and if the random noise in the baseband I-channel signal is negligible, these intermediate and post coded symbols should decode similarly, so the differential output signal from subtractor 441 should be reduced. However, if the amount of NTSC co-channel interference is significant, the differential output signal from subtractor 441 will not be low, but high.

The amount of energy in the differential output signal from subtractor 441 is found by squaring the difference output signal with squarer 442 and averaging the squarer response over predetermined short time intervals with averaging circuit 443. The squarer 442 can be implemented with a read only memory (ROM). Averaging circuit 443 may be implemented using a delay line memory for storing several consecutive digital samples and an adder for summing the digital samples currently stored in the delay line memory. A short-term average of the energy in the difference output signal from subtractor 441 determined by averaging circuit 443 is provided to a connected digital comparator for supply to threshold detector 444 . The threshold in threshold detector 444 is high enough, but does not exceed the short-term average of the difference in random noise applied to the decoded results of intermediate symbols and postcoded symbols applied to subtractor 441 . This threshold is exceeded if the NTSC co-channel interference is strong enough to cause uncorrectable errors in the data slicing performed by the data slicer 22 . Threshold detector 444 provides an indication to controller 28 of whether the threshold has been exceeded.

In an alternate embodiment, squarer 442 may be replaced by other circuitry for finding the absolute value of the deviation between said first and second estimated symbol decoding results, eg, a circuit such as an absolute value circuit. To speed up calculations, this absolute value circuit can be implemented using read-only memory.

FIG. 3 is a schematic block diagram showing the detailed construction of a portion of a digital television signal receiver using a type 120 of the NTSC band-stop comb filter 20 and a type 126 of the post-coding comb filter 26 of FIG. Subtractor 1202 acts as the first linear combiner in NTSC bandstop comb filter 120 and modulo-8 adder 1262 acts as the second linear combiner in post-coding comb filter 126 . The NTSC bandstop comb filter 120 uses a first delay device 1201 exhibiting a constant phase delay of 12 symbols, and the post-coding comb filter 126 uses a second delay device 1263 which also exhibits a constant phase delay of 12 symbols. The 12-symbol delay exhibited by each delay means 1201 and 1263 is approximately one cycle delay of the artifacts of the analog TV video carrier at 59.75 times the analog TV horizontal scan frequency _fH . The 12 symbol delay is approximately 5 cycles of the analog TV chrominance subcarrier artifact of 287.25 times _fH . The 12 symbol delay is approximately 6 cycles of the analog TV sound carrier artifact of 345.75 times _fH . This is due to the fact that the differential combined response of the subtractor 1202 to the audio carrier, to the video carrier and to the chrominance subcarrier frequency close to the differential delay by the first delay means 1201 tends to reduce co-channel interference. However, in the portion of the video signal where an edge intersects a horizontal scan line, the amount of correlation in an analog TV video signal at these distances in the horizontal spatial direction is very low.

One type 1261 of multiplexer 261 is controlled by a multiplexer control signal when it determines that there is not enough NTSC co-channel interference to cause an uncorrectable error in the output signal from data slicer 22 is controlled in its second state most of the time and in its third state most of the time when it determines that there is enough NTSC co-channel interference to cause an uncorrectable error in the output signal from the data slicer 22. Multiplexer 1261 is controlled by a control signal in its third state to feed back the modulo-8 addition result of adder 1262 to the adder as a 12-symbol constant phase delay delayed by delay means 1263 as a addend. This is a modulo-accumulation process, where a single error is propagated as a running error, with the error cycled every 12 symbols with a constant phase delay. During the four symbol constant phase delay at the beginning of each data segment and the entire portion of each data segment including the field sync, multiplexer 1261 in the first state reduces The running error in the decoding result of the post-coded symbol of the device 126. When the control signal is in its first state, multiplexer 1261 reproduces as its output signal the ideal symbol decoding result provided from memory in controller 28 . Introducing ideal symbol decoding results into the output signal of multiplexer 1261 prevents running errors. Since there are 4+69(12) symbols per data segment, the ideal symbol decoding result jumps back in phase by four symbols per data segment with a constant phase delay so that no running error persists beyond three data segments .

FIG. 4 is a block schematic diagram showing details of a portion of a digital television signal receiver of FIG. The NTSC bandstop comb filter 220 uses a first delay device 2201 exhibiting a six-symbol constant phase delay, and the post-code comb filter 226 uses a second delay device 2263 that also exhibits a six-symbol constant phase delay. The 6 symbols presented by delay means 2201 and 2263 are each delayed by a delay of 0.5 cycles of the artifact of the analog TV video carrier at approximately 59.75 times the analog TV horizontal scanning frequency _f , which is approximately 287.25 times f 2.5 cycles of the artifacts of the analog TV chrominance subcarrier for _H , and 3 cycles of any artifacts of the analog TV audio carrier which is close to 345.75 times _fH . Adder 2202 acts as a first linear combiner in NTSC bandstop comb filter 220 and modulo-8 subtractor 2262 acts as a second linear combiner in post-coding comb filter 226 . Since the delay presented by delay means 2201 and 2263 is shorter than the delay presented by delay means 1201 and 1263, although the frequency around zero converted from the analog TV carrier frequency is a narrower frequency band, it is more likely that adder 2202 adds the combined signal A good anticorrelation is less likely to be a good correlation in the signals subtractively combined by the subtractor 1202 . The rejection of the sound carrier is worse in the NTSC rejection comb filter 220 response than in the NTSC rejection comb filter 120 response. However, the poor sound carrier rejection of the comb filter 220 is not a problem if the sound carrier of the co-channel interfering analog TV signal has been suppressed by SAW filtering or by the sound limiter in the IF amplification chain 12. The response to the sync header is reduced during the use of the NTSC band-rejection comb filter 220 of FIG. 4 rather than the NTSC band-rejection comb filter 120 of FIG. the trend of.

One type 2261 of the multiplexer 261 is controlled by the multiplexer control signal, i.e. most of the time when it is determined that the NTSC co-channel interference is not sufficient to cause uncorrectable errors in the signal output from the data slicer 22 Time is in its second state, and most of the time when NTSC co-channel interference is determined to be sufficient to cause uncorrectable errors in the signal output from data slicer 22 is in its third state. Multiplexer 2261 is controlled by a control signal in a third state to feed back the modulo-8 addition result of adder modulo-8 subtractor 2262 to the adder as a 6-symbol constant phase delay delayed by delay means 2263 2262 as the summand. This is a modulo-accumulation process, where a single error is propagated as a running error, and the error cycles every 6 symbols with a constant phase delay. During the initial four symbol constant phase delay of each data segment and the entire portion of each data segment including the field sync, multiplexer 2261 in the first state reduces 226 running error in the decoding result of the post coded symbol. When the control signal is in its first state, multiplexer 2261 reproduces as its output signal the ideal symbol decoding result provided from memory in controller 28 . The ideal symbol decoding result is introduced into the output signal of the multiplexer 2261 to remove the running error. Since there are 4+138(6) symbols per data segment, the ideal symbol decoding result jumps back in phase to a constant phase of four symbols per data segment, so that no running error can persist beyond two data segments. Although running errors cycle more frequently and have twice the impact on 12 interleaved trellis codes, the likelihood of stalled cycles for running errors in post-code comb filter 226 is significantly less than in post-code comb filter 126 delay period.

FIG. 5 is a block schematic diagram showing details of a portion of the digital television signal receiver of FIG. The NTSC band-stop comb filter 320 uses a first delay device 3201 exhibiting a constant phase delay of 1368 symbols substantially equal to the constant phase of two horizontal scan lines of an analog TV signal, and the post-coding comb filter 326 uses A second delay means 3263 that also exhibits this delay. The first linear combiner in the NTSC bandstop comb filter 320 is a subtractor 3202, and the second linear combiner in the postcode comb filter 326 is a modulo-eight adder 3262.

One type 3261 of multiplexers 261 is controlled by a multiplexer control signal which is large enough when it is determined that NTSC co-channel interference is not sufficient to cause uncorrectable errors in the signal output from data slicer 22. It is in its second state part of the time and in its third state most of the time when NTSC co-channel interference is determined to be sufficient to cause uncorrectable errors in the signal output from data slicer 22 . The DTV receiver preferably includes circuitry for detecting changes between interlaced lines in NTSC co-channel interference so that controller 28 may not provide the third state multiplexer 3261 control signal under this condition.

The multiplexer 3261 is controlled by the control signal in the third state so that the modulo-8 addition result of the adder 3262 is fed back to the adder 3262 as a 1368-symbol constant phase delay delayed by the delay device 3263 as the added number. This is a modulo-accumulation process, where a single error is propagated as a running error, with the error cycled every 1368 symbols in constant phase. The span of the symbol encoding is longer than that of a set of Reed-Solomon encodings, so a single running error is easily corrected during Reed-Solomon decoding. Constant phase over the entire portion of each data segment containing the field sync, and the first four symbols of each data segment, with multiplexer 3261 in its first state reducing The running error in the decoding result of the post-coded symbol of the device 326. When the control signal is in its first state, multiplexer 3261 reproduces as its output signal the ideal symbol decoding results provided from memory in controller 28 . Introducing ideal symbol decoding results into the output signal of multiplexer 3261 prevents running errors from being generated. An NTSC video field of 16.67 millisecond duration exhibits a phase shift to a DTV data field of 24.19 millisecond duration, so the DTV data segment containing the field sync ends up scanning the entire NTSC frame raster. The 525 lines in the NTSC frame screen each contain 684 symbols of constant phase delay for a total of 359,100 symbols of constant phase delay. Since it is less than 432 times the constant phase delay of 832 symbols in the DTV data segment containing the field sync, it can be speculated with reasonable confidence that the multiplex that reproduces the ideal symbol decoding result during the DTV data segment containing the field sync Runtime errors longer than 432 data fields are removed by implementor 3261. There is also a phase shift between data segments for the beginning code group, for which ideal symbol decodable results are available, and for NTSC video scan lines. An estimated 359,100 symbol constant phase delay, which is 89,775 times the four symbol constant phase delay in the code start group, is scanned during 89,775 consecutive data segments. Since there are 313 data segments per DTV data field, it can be speculated with reasonable confidence that running errors of duration longer than 287 data fields will be removed by the multiplexer 3261 which reproduces the decoded result of the ideal symbol during the encoding start group . The two sources of suppressed running error are reasonably independent of each other, so running errors lasting longer than 200 or so data fields are quite unlikely. Furthermore, if NTSC co-channel interference is reduced when running error reproduction to suppress the multiplexer 3261 from reproducing the response of the data slicer 22 as its output signal, the error can be corrected earlier than it would otherwise be.

The NTSC bandstop comb filter 320 of FIG. 5 is effective in suppressing demodulation artifacts produced in response to the analog TV horizontal sync pulses, and many of the demodulation artifacts produced in response to the analog TV vertical sync pulses and equalization pulses very good. These artifacts are co-channel interference with the highest energy. Except for the line-to-line variation that exists in the video content of an analog TV signal over two scan line periods, the NTSC band-stop comb filter 320 provides suitably good rejection of that video content independent of its color. The FM audio carrier of the analog TV signal is fairly well suppressed in the absence of suppression by the tracking band-rejection filter in the symbol synchronization and equalization circuit 16 . Most analog TV burst artifacts are also suppressed in the response of the NTSC bandstop comb filter 320. In addition, the filtering provided by the NTSC band-stop comb filter 320 is "orthogonal" to the NTSC interference cancellation placed in the trellis decoding process.

FIG. 6 is a block schematic diagram showing details of a portion of the digital television signal receiver of FIG. NTSC band-stop comb filter 420 employs first delay means 4201 exhibiting a constant phase delay of 179,208 symbols, which is substantially equal to the period of 262 horizontal scan lines of an analog TV signal, and postcodes the comb filter 426 employs a second delay device 4263 that also exhibits this delay. Subtractor 4202 acts as the first linear combiner in NTSC bandstop comb filter 420 and modulo-8 adder 4262 acts as the second linear combiner in post-coding comb filter 426 .

One type 4261 of multiplexers 261 is controlled by a multiplexer control signal that is large enough to cause uncorrectable errors in the signal output from data slicer 22 when it is determined that NTSC co-channel interference is insufficient. It is in its second state part of the time and in its third state most of the time when NTSC co-channel interference is determined to be sufficient to cause uncorrectable errors in the signal output from data slicer 22 . The DTV receiver preferably includes circuitry for detecting field-to-field variations of NTSC co-channel interference so that the controller 28 can inhibit the multiplexer 4261 control signal from providing the third state under this condition.

The multiplexer 4261 is controlled by the control signal in the third state so that the modulo-8 addition result of the adder 4262 is fed back to the adder 4262 as a 179,208 symbol constant phase delay delayed by the delay device 4263 as the added number. This is a modulo-accumulation process where a single error is propagated as a running error, with the error recurring every 179,208 symbols with a constant phase delay. The symbol encoding interval is longer than that of a set of Reed-Solomon encodings, so it is easy to correct single run errors during Reed-Solomon decoding. During the entire portion of each data segment containing the field sync, and the four symbol constant phases at the beginning of each data segment, multiplexer 4261 in its first state reduces The running error in the decoding result of the post-coded symbol of the device 426. When the control signal is in its first state, the multiplexer 4261 reproduces as its output signal the ideal symbol decoding result provided from the memory in the controller 28 . Introducing ideal symbol decoding results into the output signal of multiplexer 4261 prevents running errors from occurring. It is assumed that the maximum number of data fields required to remove running errors from the multiplexer 4261 output signal is substantially the same as that required to remove running errors from the multiplexer 3261 output signal. However, the number of error recurrences decreased by a factor of 131 during this period.

The NTSC bandstop comb filter 420 of FIG. 6 suppresses most of the demodulation artifacts generated in response to the analog TV vertical sync pulses and equalization pulses, and all demodulation artifacts generated in response to the analog TV horizontal sync pulses . These artifacts are co-channel interference with the highest energy. In addition, the NTSC bandstop comb filter 420 suppresses artifacts produced by the video content of an analog TV signal that does not change from field to field or line to line, removing static patterns that are independent of their horizontal spatial frequency and color. Most of the artifacts of the analog TV color burst are also suppressed in the response of the NTSC bandstop comb filter 420.

FIG. 7 is a block schematic diagram showing details of a portion of the digital television signal receiver of FIG. NTSC bandstop comb filter 520 employs first delay means 5201 exhibiting a constant phase delay of 718,200 symbols, which is substantially equal to the period of two frames of an analog TV signal, and post-encoding comb filter 526 employs the same exhibiting this Delayed second delay means 5263. Subtractor 5202 acts as the first linear combiner in NTSC bandstop comb filter 520 and modulo-8 adder 5262 acts as the second linear combiner in post-coding comb filter 526 .

One type 5261 of multiplexers 261 is controlled by a multiplexer control signal that is large enough to cause uncorrectable errors in the signal output from data slicer 22 when it is determined that NTSC co-channel interference is insufficient. It is in its second state part of the time and in its third state most of the time when NTSC co-channel interference is determined to be sufficient to cause uncorrectable errors in the signal output from data slicer 22 . The DTV receiver preferably includes circuitry for detecting changes between alternate frames in NTSC co-channel interference so that the controller 28 can inhibit the multiplexer 5261 control signal from providing the third state under this condition.

The multiplexer 5261 is controlled by the control signal in the third state so that the modulo-8 addition result of the adder 5262 is fed back to the adder 5262 as a 718,200 symbol constant phase delay delayed by the delay device 5263 as the added value number. This is a modulo-accumulation process where a single error is propagated as a running error, with the error repeating at constant phase every 718,200 symbols. The symbol coding interval is longer than that of a set of Reed-Solomon codes, so it is easy to correct single running errors during Red-Solomon decoding. The multiplexer 5261 in its first state reduces the input from the post-encoding comb during the entire portion of each data segment including the field sync, and during the four-symbol constant phase delay at the beginning of each data segment. The running error in the decoded result of the postcoded symbols of filter 526. When the control signal is in its first state, the multiplexer 5261 reproduces as its output signal the ideal symbol decoding results provided from the memory in the controller 28 . Introducing ideal symbol decoding results into the output signal of multiplexer 5261 prevents running errors from occurring. It is assumed that the maximum number of data fields required to remove running errors from the multiplexer 5261 output signal is substantially the same as that required to remove running errors from the multiplexer 3261 output signal. However, the number of error recurrences during this period is reduced by a factor of 525.

The NTSC bandstop comb filter 520 of FIG. 7 suppresses all demodulation artifacts generated in response to the analog TV vertical sync pulses and equalization pulses, and all demodulation artifacts generated in response to the analog TV horizontal sync pulses. Phenomenon. These artifacts are co-channel interference with the highest energy. In addition, the NTSC bandstop comb filter 520 suppresses artifacts produced by the video content of an analog TV signal that does not change over two frames, removing static patterns that are independent of their horizontal spatial frequency and color. All analog TV burst artifacts are also suppressed in the response of the NTSC bandstop comb filter 520.

Those skilled in the art of television system design will appreciate that there are other types of correlation and anticorrelation characteristics in analog TV signals other than those shown in Figures 3-7 that may be employed in the design of NTSC bandstop filters. The 2N level of the baseband signal is raised to the 8N-1 data level by using an NTSC band-stop filter cascaded with two disclosed NTSC band-stop filter types. These filters are required to overcome the particularly unfavorable co-channel interference problem, although they have the disadvantage of degrading the signal-to-noise ratio of random noise interference from symbol decoding.

FIG. 8 shows in more detail the preferred structure of multiplexer 261 of FIG. Multiplexer 261 includes output buffer registers of read only memory (ROM) 46 , 48 , 50 for selectively reading from multiplexer 261 onto output bus 2610 which is 3 bits wide. Multiplexer 261 also includes a tri-state buffer 2611 for selectively delivering the 3-bit wide output of multiplexer 2612 to output bus 2610 during various times when ideal symbol decoding results are not produced . Multiplexer 261 is responsive to NTSC co-channel interference detector 44 at a "0" indicating that NTSC co-channel interference has no features that would render uncorrectable in the intermediate symbol decoding results provided by data slicer 22. The magnitude of the erroneous symbol to reproduce the intermediate symbol decoding result as an input signal to the tri-state data buffer 2611. Multiplexer 261 is responsive to NTSC co-channel jammer detector 44 being at a "1", a "1" indicating that NTSC co-channel jammer has caused uncorrectable errors in the intermediate symbol decoding results provided by data slicer 22 magnitude in order to reproduce the precoded symbol decoding result from the second linear combiner 262 as an input signal to the tri-state data buffer 2611.

Circuitry for generating ideal symbol decoded results applied to output bus 2610 includes ROMs 46, 48, 50; symbol clock generator 52; address counter 54 for addressing ROMs 46, 48, 50; Interference (jam) reset circuit 56; For generating the address decoder 60,62,64 of the enable signal of reading ROM 46,48,50; And for controlling the "NOR (NOR)" gate of tri-state buffer 2611 . Address counter 54 counts the input pulses received from symbol clock generator 52 at the symbol decode rate to generate successive addresses each describing a symbol in a frame of data. Appropriate portions of these addresses are applied to ROMs 46, 48, 50 as their input addresses. The disturbance reset circuit 56 resets the counter 54 to count properly in response to the data field sync information F and the data segment sync information S recovered by the data sync detection circuit 18 of FIG. 1 .

Counter 54 is preferably configured so that one set of most significant bits counts the number of data segments per data frame, and so that one set of less significant bits counts the number of symbols per data frame. This simplifies the design of the interference reset circuit 56; reduces the bit width of the input signal to the address decoder 60,62,64; is convenient to use the partial address from the counter 54 to address the ROM 46,48,50, reducing the ROM addressing time. bit width.

ROM 46 stores the ideal symbol decoding results for odd field sync segments and is selectively enabled for reading by receiving a "1" from address decoder 60 . The set of least significant bits output by counter 54, which counts the number of symbols per data segment group, addresses ROM 46; address decoder 60 responds with the set of most significant bits, which counts the number of data segments per data frame. Address decoder 60 produces a "1" if and only if the data segment portion of the address provided by address counter 54 corresponds to the address of an odd field sync segment.

ROM 48 stores the ideal symbol decoding results for even field sync segments and is selectively enabled for reading by receiving a "1" from address decoder 62 . The set of least significant bits output by counter 54, which counts the number of symbols per data segment group, addresses ROM 48; address decoder 62 responds with the set of most significant bits, which counts the number of data segments per data frame. Address decoder 62 generates "1" if and only if the data segment portion of the address provided by address counter 54 corresponds to the address of an even field sync segment.

ROM 50 stores the ideal symbol decoding results for the starting code group at the beginning of each sync segment and is selectively enabled for reading by receiving a "1" from address decoder 64 . ROM 50 responds to the two least significant bits output by counter 54; address decoder 64 responds to the set of least significant bits output by counter 54, which counts the number of symbols for each segment group. Address decoder 64 generates a "1" if and only if a data symbol of each segment count portion of the address provided by address counter 54 corresponds to the partial address starting the coded group.

The NOR gate receives the responses of address decoders 60, 62 and 64 at a respective one of its three input connections. One of address decoders 60, 62 and 64 provides a "1" as its output signal, which controls the NOR gate to provide a "0" response to tri-state data buffer 2611 when the ideal symbol decoding result is available. The tri-state data buffer 2611 is controlled so as to present a high source impedance to the bit lines of the data bus 2610, therefore, no multiplexer 2612 transfers will be transmitted on the 3-bit wide data bus 2610 from the multiplexer 2612 signal of. During those portions of the data segment where the ideal symbol decoding result cannot be predicted, none of the address decoders 60, 62, and 64 provides a "1" as its output signal, controlling the NOR gates to buffer the tri-state data 2611 provides a "1" response. The tri-state data buffer 2611 is controlled so as to present a low source impedance to the bit lines of the data bus 2610, therefore, the signal transferred from the multiplexer 2612 will be transferred on the 3 bit wide data bus 2610.

Figure 9 shows a modification of the digital television signal receiver thus far described, constructed in accordance with another aspect of the present invention, to operate a plurality of symbol decoders in parallel with corresponding even-level data slicers, each with its own Different types of NTSC band-stop comb filters are preceded and each followed by a corresponding post-coding comb filter to compensate for the precoding introduced by the preceding NTSC band-stop comb filters. An even-level data slicer A24 converts the response of the NTSC rejection filter A20 of the first type into the decoding result of the first precoded symbol applied to the post-coding comb filter A26 of the first type. An even level data slicer B24 converts the response of the second type of NTSC band-stop filter B20 into a second precoded symbol decoded result which is applied to the second type of post-coding comb filter B26. An even-level data slicer C24 converts the response of a third type NTSC band-stop filter C20 into a third precoded symbol decoded result which is applied to a third type of post-coding comb filter C26. When using the receiver part as shown in one of Figures 3-7, the initials A, B and C in the identification number of the part in Figure 9 correspond to one of the corresponding integers 1, 2, 3, 4 and 5 different integers.

The symbol decoding selection circuit 66 in FIG. The coded symbols are encoded as a result of selection to form the best estimate of the corrected symbol decoding which is applied via data assembler 30 to trellis decoding circuit 34 . The best estimate of the symbol decoding result is used to correct the addition process in the post comb filters A26, B26 and C26.

It is advantageous to select the circuitry of the NTSC band-stop comb filter A20 and the post-coding comb filter A26 to be the same as those of the NTSC band-stop comb filter 520 and post-code comb filter 526 of FIG. 7 . Since 718,200 symbols must be stored in each of the two video frame delays 5201 and 5263, this is done, although memory is a non-negligible cost. However, storage in 2 video frame delay 5201 can be used to achieve shorter delays 4201, 3201, 2201, 1201. Likewise, storage in 2 video frame delay 5203 can be used to achieve shorter delays 4263, 3263, 2263, 1263.

High energy demodulation artifacts generated in response to analog TV sync pulses, equalization pulses, and color burst signals are all suppressed when the NTSC bandstop comb filter A20 additively combines alternate video frames. In addition, artifacts produced by the video content of an analog TV signal that do not change over two frames are suppressed, removing static patterns that are not related to their spatial frequency or color.

The remaining problem of suppressing demodulation artifacts relates primarily to suppressing those demodulation artifacts arising from frame-to-frame differences at specific pixel locations within the analog TV signal raster. These demodulation artifacts can be suppressed by inter-frame filtering techniques. Optional NTSC band-stop comb filter B20 and post-comb filter B26 circuits to suppress residual demodulation artifacts depending on correlation in the horizontal direction, and optional NTSC band-stop comb filter C20 and post-comb filter C20 The comb filter C26 circuit is placed to suppress residual demodulation artifacts depending on the correlation in the vertical direction. Consider how to further implement this design decision.

If SAW filtering or sound wave limiting in the IF amplifier chain 12 does not suppress co-channel interference with the sound carrier of the analog TV signal, the type of NTSC band-stop comb filter B20 and post-comb filter B26 circuit selection is the same as for NTSC in Figure 3 Advantageously, bandstop comb filter 120 and post comb filter 126 are similar in circuitry. If SAW filtering or sound limiting in the IF amplifier chain 12 suppresses co-channel interference with the sound carrier of the analog TV signal, select the NTSC band-stop comb filter B20 and post-comb filter B26 circuits of the same type as the NTSC Advantageously, bandstop comb filter 220 and post comb filter 226 are similar in circuitry. This is due to the fact that the anticorrelation between video components with a constant phase delay of only six symbols from each other is generally better than the correlation between video components with a constant phase delay of 12 symbols from each other.

Since the choice must be made whether to select a temporally closer scan line in the same field or a spatially closer line in the previous field to be combined with the current scan line in the NTSC band-stop comb filter C20, the NTSC band Optimum selection of the rejection comb filter C20 and post comb filter C26 circuits is less straightforward. Since inter-field jumps are less likely to disrupt the NTSC bandstop of comb filter C20, it is usually better to select temporally closer scan lines in the same field. For this option, the NTSC band-stop comb filter C20 and post-comb filter C26 circuits are of a similar type to the NTSC band-stop comb filter 320 and post-comb filter 326 circuits of FIG. 5 . Alternatively, the NTSC band-stop comb filter C20 and post-comb filter C26 circuits are of a similar type to the NTSC band-stop comb filter 420 and post-comb filter 426 circuits of FIG. 6 .

Fig. 9 is an improved digital receiver device in another embodiment of the present invention, using additional parallel data clipping operations, respectively by connecting corresponding even-numbered stage data clippers after corresponding NTSC band-stop filters, This is followed by a cascade connection of the corresponding post-coded comb filters. Although FIG. 9 shows two additional parallel data slicing operations, improvements using further parallel data slicing operations can further provide the ability to improve the best estimate of the correct symbol decoding result.

In a preferred embodiment of the circuit of FIG. 9, when an ideal decoding result is available, the symbol decoding selection circuit 66 selects the ideal decoding result as the final decoding result. When it is determined that NTSC co-channel interference is actually obtained, the difference between the intermediate symbol decoding results and the various post-coded symbol decoding results may be due to NTSC co-channel interference. Therefore, when the ideal decoding result cannot be used, the final decoding result can be selected by the symbol decoding selection circuit 66 by comparing the decoding results of various post-coded symbols with each other and with the intermediate decoding results.

The initial wish to establish that NTSC co-channel interference is actually obtained is that during the noisy receive state when the level of "white" noise is sufficient to actually cause more errors in the decoded results of subsequent coded elements than in intermediate decoded results These differences can also arise in the various symbol decoding results. Using the technique described in US Patent No. 5,594,496 or a similar technique using a different NTSC co-channel rejection filter, the fact that NTSC co-channel interference is present during the data segment can be determined when field sync information is present. However, it is preferable to monitor the strength of NTSC co-channel interference on a continuous real-time basis so that changes in NTSC co-channel interference due to fading or due to changes in video content can be accounted for.

Figure 9 shows such monitoring provided by measuring the level between 4.5 MHz carriers in NTSC co-channel interference, as described by its inventors in a March 21, 1997 paper entitled "Using Intracarrier Signals to Detect NTSC co-channel interference" as taught in US Patent Application No. 08/821,945. The DTV signal converted to IF by the "front-end" electronics 10 is supplied to a quasi-parallel IF amplifier chain 68 for NTSC sound signals. The amplification stage in IF amplifier chain 68 for NTSC sound signals corresponds to a similar amplification stage in IF amplifier chain 12 for DTV signals, which has substantially linear gain and has the same automatic gain as the corresponding amplifier stage in IF amplifier chain 12 control. The frequency selectivity of the IF amplification chain 68 is set to emphasize the response within ±250 KHz of the NTSC audio carrier and within ±250 KHz of the NTSC video carrier. If a multi-conversion receiver circuit is used, the filtering process establishing the frequency selectivity of the IF amplification chain 68 may be performed by SAW filtering in the UHF IF amplifier. The response of the IF amplification chain 68 is provided to the intercarrier detector 70 which uses the modulated NTSC video carrier as the recovered carrier for the heterodyned NTSC audio carrier to generate an intercarrier audio IF with a carrier frequency of 4.5 MHz Signal. The intercarrier audio IF signal is amplified by an intercarrier audio IF amplifier 72. The 4.5 MHz IF amplifier 72 supplies the amplified intercarrier audio IF signal to an intercarrier amplitude detector 74. The response of amplitude detector 74 is provided to threshold detector 76 . The threshold of threshold detector 76 is exceeded if the NTSC co-channel interference is strong enough to possibly cause errors in the data slicing performed by data slicer 22 . Threshold detector 76 provides an indication to symbol detection selection circuit 66 whether the threshold is exceeded. If the indication is that NTSC co-channel interference is not of sufficient magnitude to possibly cause an error in the data slicing performed by data slicer 22, then the indication controls symbol decode selection circuit 66 to select an intermediate symbol from data slicer 22 The decoding result is used as the final symbol decoding result, as long as the ideal symbol decoding result cannot be provided for the current symbol constant phase delay. The time constant in intercarrier amplitude detector 74 should be carefully chosen if optimum performance is sought. Since isolated symbol decoding errors are correctable, eliminating short bursts of the output signal from intercarrier amplitude detector 74 with a fast time constant is critical for generating the choice for switching final symbol decoding results from intermediate symbol decoding results to A control signal post-coded symbol decoding result may be the best.

Various circuit configurations can be used to obtain the carrier signal by heterodyning between the audio and video carriers of the co-channel interfering analog television signal. A number of such structures are disclosed in US Patent Application No. 08/821,945.

10A and 10B show some detailed circuits included in the symbol decoding selection circuit 66 for selecting the final symbol decoding result. FIG. 10 is an assembly diagram showing how FIGS. 10A and 10B are assembled together to provide an overall block schematic diagram of the symbol decode selection circuit 66. As shown in FIG. The symbol decoding selection circuit 66 has a 3-bit wide output data bus 78, which goes from the bottom of Figure 10A to the bottom of Figure 10B, and thus to the data assembler 30, the data interleaver 32, the trellis decoding circuit 34, and the data deinterleaver 36. The beginning of the serial connection of the byte building circuit 38, the Reed-Solomon decoding circuit 40 and the data de-randomizer 42 is the same as that shown in FIG. 1 . FIG. 10A shows a circuit similar to that shown in FIG. 8 for selectively applying the ideal symbol decoded results read from ROMs 46, 48 and 50 to output data bus 78.

Figure 10B shows a best estimate selection circuit for selecting final symbol decoding results during time intervals between which data segments or data fields are provided in a DTV signal when ideal symbol decoding results are not available Interval between those decoded results of the synchronization code. Tri-state data buffer 2611 and multiplexer 2612 of FIG. 8 are replaced by tri-state data buffers 080, A80, B80 and C80 in the best estimate selection circuit of FIG. 10B. Tri-state data buffer 080 is controlled by the response of AND gate 082 at a logic "1" to transmit on output data bus 78 the intermediate symbol decoded results provided from odd level data slicer 22 of FIG. . The tri-state data buffer A80 is controlled by the response of the AND gate A82 at a logic "1" to pass on the output data bus 78 the post-encode symbols provided from the post-encode comb filter A26 of FIG. Decode the result. Tri-state data buffer B80 is controlled by the response of AND gate B82 at a logic "1" to transmit on output data bus 78 the post-encoded symbol decodes provided from post-comb filter B26 of FIG. result. Tri-state data buffer C80 is controlled by the response transfer of AND gate C82 at logic "1" to transfer the post-encoded code provided from post-encoded comb filter B26 of FIG. 9 on output data bus 78 Meta decoding result. NOR gate 58 of FIG. 10A provides its response to AND gates 082, A82, B82, and C82 as their respective inputs, thereby controlling tri-state buffers 080, A80, B80, and C80 so that only when None of the tri-state output buffers of ROMs 46, 48, and 50 present a low source impedance to the bit lines of output data bus 78 while the ideal symbol decoding results are transmitted on data bus 78.

The signal output from the threshold detector 76 of FIG. 9 is logic "1" when the NTSC co-channel interference is of sufficient strength to cause errors in the data slicing performed by the data slicer 22. The signal output from the threshold detector 76 of FIG. 9 is applied as a corresponding input signal to each of the AND gates A82, B82 and C82, thereby controlling the tri-state buffers A80, B80 and C80 so that only when NTSC co-channel The bit line to output data bus 78 exhibits a low source impedance when the disturbance is of sufficient magnitude to cause errors in the data slicing by data slicer 22 . The signal output from threshold detector 76 of FIG. The bit lines of the output data bus 78 present low source impedances at the magnitude of the error caused in the data clipping performed.

Considering now how to select from the post-encoding symbol decoding results provided by the post-encoding comb filters A26, B26 and C26 of FIG. 9 when the signal output by the threshold detector 76 of FIG. "1" indicates that the NTSC co-channel interference is strong enough to cause errors in the intermediate symbol decoding results provided by the data slicer 22 . Since the comb filtering used to achieve the decoded result of this post-coded symbol is more effective in suppressing the artifacts of NTSC co-channel interference than the comb filter used to achieve the decoded result of another post-coded symbol, it is assumed that in absolute The decoding result of the post-coded symbol with the largest deviation from the decoding result of the intermediate symbol in the item presents the absolute deviation. Therefore, the intermediate symbol decoding results provided by the data slicer 22 and the post-encoding symbol decoding results provided by the post-encoding comb filters A26, B26, and C26 are determined by digital subtractors A84, B84, and C84 of FIG. 10B. difference between the results. The absolute values of these differences are determined by absolute value circuits A86, B86 and C86 to determine the difference between the decoded results of the post-encoded symbols provided from the post-encoded comb filters A26, B26 and C26 and the intermediate values provided by the data slicer 22. Absolute deviation of symbol decoding result. Absolute value circuits A86, B86 and C86 can be implemented in read-only memory (ROM) to achieve faster calculation rates by selectively negating and adding "1". Faster calculation rates can be achieved by using ROM to simultaneously perform subtraction and absolute value processing.

The best estimate selection circuit of FIG. 10B includes digital comparators A88, B88 and C88 in addition to tri-state buffers 080, A80, B80 and C80 and AND gates 082, A82, B82 and C82. The digital comparator A88 determines whether the absolute deviation of the intermediate symbol decoding result from the post-encoding symbol decoding result provided by the post-encoding comb filter A26 is equal to or exceeds the difference between the intermediate symbol decoding result and the post-encoding comb filter provided by the post-encoding comb filter. The absolute deviation of the decoded result of the post-coded symbol provided by B26 provides a logic "1" if positive, and a logic "0" if negative. Digital comparator B88 determines whether the absolute deviation of the intermediate symbol decoding result from that provided by post-encoding comb filter B26 exceeds the difference between the intermediate symbol decoding result and that provided by post-encoding comb filter C26 The absolute deviation of the decoding result of the post-encoded symbol of , providing a logical "1" if positive, and a logical "0" if negative. Digital comparator C88 determines whether the absolute deviation of the intermediate symbol decoding result from that provided by post-encoding comb filter C26 exceeds the difference between the intermediate symbol decoding result and that provided by post-encoding comb filter A26 The absolute deviation of the decoding result of the post-encoded symbol of , providing a logical "1" if positive, and a logical "0" if negative.

To make the response of the AND gate A82 a logic "1" for controlling the tri-state buffer A80 to transmit on the output data bus 78 the decoded results of the post-encoded symbols provided by the post-encode comb filter A26 , the digital comparator A88 must determine that the absolute deviation between the intermediate symbol decoding result and the post-encoding symbol decoding result provided by the post-encoding comb filter A26 is equal to or exceeds the difference between the intermediate symbol decoding result and the post-encoding comb filter The absolute deviation of the decoding result of the post coded symbol provided by the device B26, at the same time, the digital comparator C88 will determine the difference between the decoded result of the intermediate symbol and the decoded result of the post coded symbol provided by the post coded comb filter C26 The deviation does not exceed the deviation of the decoded result of the intermediate symbols from the decoded result of the post-coded symbols provided by the post-coded comb filter A26. To make the response of the AND gate B82 a logic "1" for controlling the tri-state buffer B80 to transmit on the output data bus 78 the decoded results of the post-encoded symbols provided by the post-encode comb filter B26 , the digital comparator B88 must determine that the absolute deviation of the intermediate symbol decoding result from that provided by the post-encoding comb filter B26 exceeds the difference between the intermediate symbol decoding result and that provided by the post-encoding comb filter C26 The absolute deviation of the post code symbol decoding result provided, at the same time, the digital comparator A88 will determine that the absolute deviation of the post code symbol decoding result provided by the post code comb filter A26 is the same as that of the intermediate symbol decode result. Equal to or exceed the deviation of the intermediate symbol decoding result from the post-encoding symbol decoding result provided by the post-encoding comb filter B26. To make the response of the AND gate C82 a logic "1" for controlling the tri-state buffer C80 to transmit on the output data bus 78 the decoded results of the post-encoded symbols provided by the post-encode comb filter C26 , the digital comparator C88 must determine that the absolute deviation between the intermediate symbol decoding result and the post-encoding symbol decoding result provided by the post-encoding comb filter C26 is equal to or exceeds the difference between the intermediate symbol decoding result and the post-encoding comb filter The absolute deviation of the post coded symbol decoding result provided by the device A26, at the same time, the digital comparator B88 will determine the difference between the intermediate symbol decoded result and the post coded symbol decoded result provided by the post coded comb filter B26 The deviation does not exceed the deviation of the decoded result of the intermediate symbols from the decoded result of the post-coded symbols provided by the post-coded comb filter C26. One and only one of comparators A88, B88, and C88 (comparator A88 in FIG. , B86 and C86 provide all equal absolute deviations, and none of the tri-state data buffers A80, B80 and C80 can be controlled to drive the bit line of the output data bus 78 from a low source impedance.

Those skilled in the art of digital communication receiver design and familiar with the foregoing description and accompanying drawings will be able to devise many embodiments of the invention other than the preferred embodiment specifically described above. This should be kept in mind in defining the scope of the broader claims below. In the following claims, the word "said" is used whenever reference is made to a preceding example and the definite article "the" is used for grammatical purposes rather than referring to the preceding example.

Claims (15)

1. digital television signal receiver comprises:

The digital television signal checkout gear, be used to provide the stream of the 2N level code element of a code element constant phase that has stipulated time length separately, N is a positive integer, described 2N level code element stream is subject to be attended by the influence of common road interference simulation TV signal artifcates, described each code element packet becomes to have the continuous data segment of the stem of corresponding data section synchronous code, described data segment is grouped into the continuous data field of the primary data section that has each data fields, and each data fields comprises the data fields synchronous code of data fields to the variation between the data fields;

Be used for providing the circuit of M unique comb filter response to described 2N level code element stream, the artifcates effect that each described unique comb filter is attended by common road interference simulation TV signal is littler than described 2N level code element stream, and wherein M is a positive integer;

A plurality of symbol decoding devices, be used to produce corresponding estimation symbol decoding result, first of described a plurality of symbol decoding devices responds described 2N level code element stream and produces the first estimation symbol decoding result, other symbol decoding device of each of described a plurality of symbol decoding devices responds the corresponding corresponding estimation of a generation symbol decoding result in described M unique comb filter response, its corresponding estimation symbol decoding result is encoded by postposition, so that at described in the unique comb filter response of described M corresponding one finish corresponding matched filtering, obtain corresponding estimation symbol decoding result from described M unique comb filter response, described other symbol decoding device of described a plurality of symbol decoding devices comprises the second symbol decoding device that is used to produce the second estimation symbol decoding result;

Be used to detect the current circuit that whether has deviation between the described first and second estimation symbol decoding results; With

The best-estimated is selected circuit, be used for selecting the best-estimated from corresponding estimation symbol decoding result, so that the time between those times when synchronous code occurring produces final symbol decoding result, the described first estimation symbol decoding result and other estimation symbol decoding result's deviation is depended in the selection of described the best-estimated.

2. digital television signal receiver according to claim 1 further comprises:

Respond final symbol decoding result's trellis decoder circuit, be used to produce the built error correction decoded result; With

Respond the Reed-Solomon decoder circuit of described built error correction decoded result byte, be used to produce outside error correction decoding result.

3. digital television signal receiver according to claim 1, wherein M is 1.

4. digital television signal receiver according to claim 3, wherein said the best-estimated select circuit to comprise:

One multiplexer connects to be used to provide described first and second and estimates the ability of selecting between the symbol decoding results, and the time that is used between those times when described synchronous code occurring produces described final symbol decoding result;

Be used to obtain the circuit of the absolute figure of deviation between the described first and second estimation symbol decoding results;

Be used to produce the averager of the mean value of described absolute figure; With

A threshold dector, response surpasses described square of result's of predetermined threshold described mean value, be used to control described multiplexer and select the described second estimation symbol decoding result, produce described final symbol decoding result with each time between those times when described synchronous code occurring, perhaps control described multiplexer and select the described first estimation symbol decoding result, produce described final symbol decoding result with each time between those times when described synchronous code occurring.

5. digital television signal receiver according to claim 3, wherein said the best-estimated select circuit to comprise:

One multiplexer connects to be used to provide described first and second and estimates the ability of selecting between the symbol decoding results, and the time that is used between those times when described synchronous code occurring produces described final symbol decoding result;

The squarer of deviation is used to obtain square result as the absolute figure of those deviations between the one described first and second estimation symbol decoding results;

Be used to produce the averager of described square of result's mean value; With

One threshold dector, response surpasses described square of result's of predetermined threshold described mean value, be used to control described multiplexer and select the described second estimation symbol decoding result, produce described final symbol decoding result with each time between those times when described synchronous code occurring, perhaps control described multiplexer and select the described first estimation symbol decoding result, so that each time between those times when described synchronous code occurring produces described final symbol decoding result.

6. digital television signal receiver according to claim 5 further comprises:

Trellis decoder circuit in response to final symbol decoding result is used to produce the built error correction decoded result; With

In response to the Reed-Solomon decoder circuit of described built error correction decoded result byte, be used to produce outside error correction decoding result.

7. digital television signal receiver according to claim 1, wherein M is 2 at least, wherein M is a positive integer.

8. digital television signal receiver according to claim 7 further comprises:

Be used for determining whether the described current level of road interference simulation TV signal altogether is enough to cause error in the described first estimation symbol decoding result, be used for determining that the described level of road interference simulation TV signal altogether is not enough to provide first indication when the described first estimation symbol decoding result causes error and is determining that the described level of road interference simulation TV signal altogether provides the circuit of second indication when being enough to cause error in the described first estimation symbol decoding result; And this circuit is selected in the circuit at described the best-estimated; With

Respond described first indication and from the described first estimation symbol decoding result, only select described the best-estimated, and each time between those times when described synchronous code occurring produces described final symbol decoding result's circuit current.

9. digital television signal receiver according to claim 8, wherein said the best-estimated select circuit further to comprise:

Which of deviation that is used for determining other estimation symbol decoding result and the described first estimation symbol decoding result has maximum value, least may be caused the circuit of the indication of error by the described artifcates that is total to road interference simulation TV signal therein to produce among other estimation symbol decoding result which; With

And if only if provides described second when indication which among other estimation symbol decoding result least may be caused the indication of error by the described artifcates of road interference simulation TV signal altogether therein in response to simultaneously, from described other estimation symbol decoding result, select to be marked as the described the best-estimated that least may cause error, be used for the circuit that each time between those times when described synchronous code occurring produces described final symbol decoding result by the described artifcates of road interference simulation TV signal altogether.

10. digital television signal receiver according to claim 7 further comprises:

Be used for obtaining the circuit of intercarrier signal from the heterodyne between the Voice ﹠ Video carrier wave of described road interference simulation TV signal altogether;

When be used to detect the amplitude of this intercarrier signal above specified level, be used for providing the described level of road interference simulation TV signal altogether to be enough to cause the indication of error in the described first estimation symbol decoding result who produces by the described first symbol decoding device, perhaps, provide the described level of road interference simulation TV signal altogether to be not enough in the described first estimation symbol decoding result, cause the indicating circuit of error; And this circuit is selected in the circuit at described the best-estimated; With

Be not enough to estimate the described indication that causes error among the symbol decoding result in response to the described level of road interference simulation TV signal altogether, only select the circuit of current described the best-estimated from the described first estimation symbol decoding result described first.

11. digital television signal receiver according to claim 10, wherein said the best-estimated select circuit further to comprise:

Be used for determining other estimation symbol decoding result which have maximum value with the described first estimation symbol decoding result's deviation, least may be caused the circuit of the indication of error by the described artifcates that is total to road interference simulation TV signal therein to produce among other estimation symbol decoding result which; With

And if only if provides described second when indication which among other estimation symbol decoding result least may be caused the indication of error by the described artifcates of road interference simulation TV signal altogether therein in response to simultaneously, be marked as the described the best-estimated that least may cause error from the institute that described other estimation symbol decoding result selects, be used for the circuit that time between those times when the described synchronous code of appearance produces described final symbol decoding result by the described artifcates of road interference simulation TV signal altogether.

12. a digital television signal receiver comprises:

The digital television signal checkout gear, be used to provide one each have the stream of 2N level code element of the code element epoch of stipulated time length, N is a positive integer, described 2N level code element stream is subject to be attended by the influence of common road interference simulation TV signal artifcates, described code element packet becomes to have the continuous data segment of the stem of corresponding data section synchronous code, described data segment is grouped into the continuous data field of the primary data section that has each data fields, and each data fields is included in the data fields synchronous code that changes between data fields and the data fields;

One first symbol decoding device responds described 2N level code element stream and produces the first estimation symbol decoding result;

Be used for providing to described 2N level code element stream the circuit of a plurality of unique comb filter responses, the artifcates effect that each described unique comb filter is attended by common road interference simulation TV signal is littler than described 2N level code element stream;

Corresponding symbol decoding device, respond described a plurality of unique comb filter responses corresponding each, produce corresponding estimation symbol decoding result, the corresponding estimation symbol decoding result of rearmounted coding is so that finish corresponding matched filtering at corresponding of the described unique comb filter response that is obtained corresponding estimation symbol decoding result by it;

Be used for obtaining the circuit of intercarrier signal from the heterodyne between the Voice ﹠ Video carrier wave of described road interference simulation TV signal altogether;

When the amplitude that detects this carrier signal surpasses specified level, be used for providing the described level of road interference simulation TV signal altogether to be enough to cause the indication of error in the described first estimation symbol decoding result who produces by the described first symbol decoding device, perhaps, provide the described level of road interference simulation TV signal altogether to be not enough in the described first estimation symbol decoding result, cause the circuit of the indication of error;

Detect when synchronous code occurs in the described 2N level code element stream, provide the indication of synchronous code to occur in the described 2N level code element stream and the circuit that does not occur the indication of synchronous code in the described 2N level code element stream is provided;

Synchronous code in response to occurring in the described 2N level code element stream that is detected is used to described synchronous code to produce the circuit of desirable code element decoded result;

Be used for determining that the current and described first estimation symbol decoding result of which described estimation symbol decoding result except that the described first estimation symbol decoding result has maximum absolute deviation, to produce the circuit which described other estimation symbol decoding result least may be caused the indication of error by the described artifcates that is total to road interference simulation TV signal;

One multiplexer, one of a plurality of input signals that provided that are used for by reproducing current selection provide final symbol decoding result, described a plurality of input signals to comprise the described first estimation symbol decoding result, each described other estimation symbol decoding result and described desirable code element decoded result; Optionally control described multiplexer, so that provide described final symbol decoding result by reproducing described desirable code element decoded result in response to the described indication that occurs synchronous code in the described 2N level code element stream; Be not enough to estimate the described indication that causes error among the symbol decoding result in response to described indication that synchronous code is not provided in the described 2N level code element stream that provides simultaneously and the described level of road interference simulation TV signal altogether described first, described multiplexer is Be Controlled optionally, in order to provide described final symbol decoding result by reproducing the described first symbol decoding result; In response to the described indication that synchronous code is not provided in the described 2N level code element stream that provides simultaneously, the described level of road interference simulation TV signal altogether is not enough to cause the described indication of error in the described first estimation symbol decoding result, and the described indication that least may cause described other estimation symbol decoding result of error by the described noise that is total to road interference simulation TV signal, described multiplexer is Be Controlled optionally, in order to may cause that described other estimation symbol decoding result of error provides described final symbol decoding result by the described noise that is total to road interference simulation TV signal by reproducing least;

13. digital television signal receiver according to claim 12 further comprises:

Trellis decoder circuit in response to final symbol decoding result is used to produce the built error correction decoded result; With

In response to the Reed-Solomon decoder circuit of described built error correction decoded result byte, be used to produce outside error correction decoding result.

14. a digital television signal receiver comprises:

The digital television signal checkout gear, be used to provide one each have the stream of 2N level code element of the code element epoch of stipulated time length, N is a positive integer, described 2N level code element stream is subject to be attended by the influence of common road interference simulation TV signal artifcates, described code element packet becomes to have the continuous data segment of the stem of corresponding data section synchronous code, described data segment is grouped into the continuous data field of the primary data section that has each data fields, and each data fields is included in the data fields synchronous code that changes between data fields and the data fields;

One first symbol decoding device produces the first estimation symbol decoding result in response to described 2N level code element stream;

Be used for providing to described 2N level code element stream the circuit of first, second and the response of the 3rd unique comb filter, the artifcates effect that each described unique comb filter is attended by common road interference simulation TV signal is littler than described 2N level code element stream;

One the second symbol decoding device in response to described first comb filter response generation, the second estimation symbol decoding result, the described second symbol decoding device comprises one first rearmounted coding comb filter, and being used for provides the described second estimation symbol decoding result at the filter response of coupling to described first comb filter response;

One the 3rd code element decoder in response to described second comb filter response generation the 3rd estimation symbol decoding result, described the 3rd code element decoder comprises one second rearmounted coding comb filter, and being used for provides described the 3rd estimation symbol decoding result at the filter response of coupling to described second comb filter response;

One the 4th symbol decoding device in response to described the 3rd comb filter response generation the 4th estimation symbol decoding result, described the 4th symbol decoding device comprises one the 3rd rearmounted coding comb filter, and being used for provides described the 4th estimation symbol decoding result at the filter response of coupling to described the 3rd comb filter response;

Be used for obtaining the circuit of intercarrier signal from the heterodyne between the Voice ﹠ Video carrier wave of described road interference simulation TV signal altogether;

When be used to detect the amplitude of this carrier signal above specified level, provide the described level of road interference simulation TV signal altogether to be enough in the described first estimation symbol decoding result who produces by the described first symbol decoding device, cause the indication of error, perhaps, provide the described level of road interference simulation TV signal altogether to be not enough in the described first estimation symbol decoding result, cause the circuit of the indication of error;

Be used for detecting described 2N level code element stream synchronous code and occur, provide the indication of synchronous code to occur in the described 2N level code element stream and the circuit that does not occur the indication of synchronous code in the described 2N level code element stream is provided;

The synchronous code that occurs in the described 2N level code element stream that response is detected thinks that described synchronous code produces the circuit of desirable code element decoded result;

Be used for determining described second, third and the 4th estimation symbol decoding result which have maximum absolute deviation with the described first estimation symbol decoding result, to produce the device that described artifcates that among described second, third and the 4th estimation symbol decoding result which least may be total to road interference simulation TV signal causes the indication of error;

One multiplexer, one of a plurality of input signals that provided that are used for by reproducing current selection provide final symbol decoding result, described a plurality of input signals to comprise the described first estimation symbol decoding result, the described second estimation symbol decoding result, described the 3rd estimation symbol decoding result, described the 4th estimation symbol decoding result and described desirable code element decoded result; Optionally control described multiplexer, so that by in response to the indication that occurs synchronous code in the described 2N level code element stream, reproducing described desirable code element decoded result provides described final symbol decoding result; Be not enough to estimate the described indication that causes error among the symbol decoding result in response to described indication that synchronous code is not provided in the described 2N level code element stream that provides simultaneously and the described level of road interference simulation TV signal altogether described first, described multiplexer is Be Controlled optionally, in order to provide described final symbol decoding result by reproducing the described first symbol decoding result; Be not enough in the described first estimation symbol decoding result, cause the described indication of error and the described indication that least may cause the described second estimation symbol decoding result of error in response to the described indication that synchronous code is not provided in the described 2N level code element stream that provides simultaneously, the described level of road interference simulation TV signal altogether by the described noise that is total to road interference simulation TV signal, described multiplexer is Be Controlled optionally, in order to provide described final symbol decoding result by reproducing the described second symbol decoding result; Optionally control described multiplexer, so that being enough to cause in the described first estimation symbol decoding result by the level in response to the described indication that does not occur synchronous code in the described 2N level code element stream, described altogether road interference simulation TV signal provides when the described indication of error and described the 3rd estimation symbol decoding result least may be caused the described indication of error by the described artifcates of road interference simulation TV signal altogether, reproducing described the 3rd code element decoded result provides described final symbol decoding result; Optionally control described multiplexer, so that being enough to cause in the described first estimation symbol decoding result by the level in response to the described indication that does not occur synchronous code in the described 2N level code element stream, described altogether road interference simulation TV signal provides when the described indication of error and described the 4th estimation symbol decoding result least may be caused the described indication of error by the described artifcates of road interference simulation TV signal altogether, reproducing described the 4th symbol decoding result provides described final symbol decoding result.

15. digital television signal receiver according to claim 14 further comprises:

Trellis decoder circuit in response to final symbol decoding result is used to produce the built error correction decoded result; With

In response to the Reed-Solomon decoder circuit of described built error correction decoded result byte, be used to produce outside error correction decoding result.

Images