CN102546512B - OFDM transmitting device and OFDM receiving device - Google Patents

Abstract

The present invention provides an OFDM sending device and an OFDM receiving device. The OFDM sending device includes: a mapping unit for generating a signal for estimating transmission whose amplitude is switched according to an insertion interval. The symbols of the path change and the signal sequence configured with the data symbol corresponding to the amplitude of each modulation method; the inverse Fourier transform unit performs inverse transformation on the signal sequence generated by the mapping unit to generate an OFDM symbol sequence; sending The unit is configured to send the OFDM symbol sequence generated by the inverse Fourier transform unit.

Description

Orthogonal frequency division multiplexing transmitting device and orthogonal frequency division multiplexing receiving device

The application date is August 24, 2006, the application number is 200680030825.1, and the title of the invention is "MIMO-OFDM transmission device and MIMO-OFDM transmission method" The divisional application of the invention patent application.

technical field

The present invention relates to a MIMO-OFDM (Multiple-Input Multiple-Output-Orthogonal Frequency Division Multiplexing) sending device and a MIMO-OFDM sending method. In particular, it relates to a technique for realizing a symbol structure suitable for frequency offset estimation, channel variation (channel variation) estimation, synchronization, and signal detection in MIMO-OFDM communication.

Background technique

FIG. 1 shows the structure and frame structure of a wireless LAN (Local Area Network, Local Area Network) transmitting and receiving device as an example of a conventionally realized wireless communication system using OFDM (Orthogonal Frequency Division Multiplexing).

(a) of FIG. 1 shows an example of a configuration of a transmission device. A frame structure signal generator 10 receives control information 9 such as a modulation method as input, determines a frame structure, and outputs a frame structure signal 11 . The serial-to-parallel conversion unit (S/P) 2 takes the frame structure signal 11 and the digitally modulated baseband signal 1 as input, performs serial-to-parallel conversion, and outputs a parallel signal 3 conforming to the frame structure. An inverse Fourier transform unit (ifft) 4 takes the parallel signal 3 as input, performs inverse Fourier transform, and outputs a signal 5 after inverse Fourier transform. The wireless unit 6 takes the inverse Fourier-transformed signal 5 as input, performs frequency conversion, etc., and outputs a transmission signal 7 . The transmission signal 7 is transmitted through an antenna 8 as radio waves.

(b) of FIG. 1 shows a configuration example of a receiving device. The wireless unit 14 receives a received signal 13 received via the antenna 12 as input, performs processing such as frequency conversion, and outputs a baseband signal 15 . Synchronization unit 16 takes baseband signal 15 as input, establishes time synchronization with the transmitting device, and outputs timing signal 17 . Fourier transform unit (fft) 18 receives baseband signal 15 and timing signal 17 as input, performs Fourier transform on baseband signal 15 based on timing signal 17 , and outputs Fourier transformed signal 19 .

Channel variation estimation section 20 receives Fourier-transformed signal 19 and timing signal 17 as input, detects a preamble in the Fourier-transformed signal, estimates channel variation, and outputs channel variation estimation signal 21 . The frequency offset estimating unit 22 takes the signal 19 and the timing signal 17 after the Fourier transform as input, detects pilot symbols and preambles in the signal after the Fourier transform, estimates the frequency offset based on these symbols, and outputs the frequency offset Shift estimated signal 23.

The demodulation unit 24 receives the Fourier-transformed signal 19, the timing signal 17, the channel variation estimation signal 21, and the frequency offset estimation signal 23 as inputs, and compensates for the channel variation and frequency offset in the Fourier-transformed signal 19. and demodulation, and output the received digital signal 25.

(c) of FIG. 1 is an image of a frame structure of IEEE (Institute of Electrical and Electronics Engineers) 802.11a (not a correct frame structure). The horizontal axis represents frequency, and the vertical axis represents time. A preamble is inserted at the beginning in order to estimate propagation channel fluctuations and frequency offsets (and to perform signal detection in some cases). Also, pilot symbols are inserted into specific carriers such as carrier 2 and carrier 5, and are used for estimating frequency offset and phase noise at the receiving device. Regarding the preamble and the pilot symbol, the signal point configuration on the in-phase I-orthogonal Q plane is a known signal point configuration. Also, data is transmitted by data symbols.

In addition, non-patent literature 1 describes a wireless LAN system.

[Non-Patent Document 1] High speed physical layer (PHY) in 5GHz band "IEEE802.11a, 1999

Contents of the invention

The problem to be solved by the invention

Non-Patent Document 1 shows the configuration of symbols used for frequency offset estimation, channel variation (channel variation) estimation, and synchronization and signal detection when OFDM is used.

However, in a wireless LAN, if the method shown in Non-Patent Document 1 is combined with a MIMO system using Spatial Multiplexing or SDM: Spatial Division Multiplexing, further improvement in transmission speed can be expected. Thus, a wide range of services can be provided to users.

In order to obtain high reception quality in this MIMO-OFDM system, it is necessary to perform high-precision frequency offset estimation, high-precision transmission path variation estimation, and high-precision synchronization and signal detection.

However, the present situation is that symbols for channel estimation and frequency offsets for realizing the above-described high-precision frequency offset estimation, high-precision channel fluctuation estimation, and high-precision synchronization and signal detection are not sufficiently considered. Estimates the method of transmission in symbols.

An object of the present invention is to provide a MIMO-OFDM transmission device and a MIMO-OFDM transmission method capable of performing high-precision frequency offset estimation, high-precision channel variation estimation, and high-precision synchronization and signal detection.

The OFDM transmission device of the present invention includes: a mapping unit that generates symbols for estimating transmission path fluctuations whose amplitudes are switched according to insertion intervals, and a signal sequence in which data symbols having amplitudes corresponding to each modulation method are arranged; The inverse Fourier transform unit performs inverse transform on the signal sequence generated by the mapping unit to generate an OFDM symbol sequence; the sending unit sends the OFDM symbol sequence generated by the inverse Fourier transform unit.

The OFDM receiving apparatus of the present invention includes: a receiving unit that receives symbols for estimating transmission path fluctuations whose amplitudes are switched according to insertion intervals, and a signal sequence in which data symbols having amplitudes corresponding to each modulation method are arranged; The Fourier transform unit performs Fourier transform on the received OFDM symbol sequence, and outputs the signal sequence; the transmission path variation estimation unit uses the symbols used for estimating the transmission path variation included in the signal sequence to perform transmission estimate the path distortion, and output the obtained distortion value; the compensation unit uses the estimated value to compensate the distortion of the signal sequence; the demodulation unit demodulates the distortion-compensated signal sequence and converts the data recovery.

The MIMO-OFDM transmission device of the present invention transmits OFDM-modulated data symbols through multiple antennas during data transmission, and transmits pilot symbols through specific carriers of the multiple antennas during the data transmission, which The structure adopted includes: an OFDM signal forming unit, which forms OFDM signals transmitted through each antenna; and a pilot symbol mapping unit, which assigns the sequences in an orthogonal relationship to each other along the time axis direction to the OFDM signals transmitted through each antenna at the same time The same carrier between OFDM signals, thus forming a pilot carrier.

According to this configuration, since sequences in a mutually orthogonal relationship are allocated to corresponding subcarriers between OFDM signals transmitted at the same time through each antenna along the time axis direction, thereby forming pilot carriers, even if the pilot symbols are in A plurality of channels (antennas) are multiplexed, and high-precision frequency offset and phase noise estimation can also be performed. Furthermore, since the pilot symbols of each channel can be extracted without using channel estimation values (transmission path estimation values), the structure for compensating for frequency offset and phase noise can be simplified.

In addition, the structure adopted by the MIMO-OFDM transmission device of the present invention is that, in the case of transmitting OFDM signals through two antennas, the pilot symbol mapping unit forms the pilot carrier so that the first between the first and second antennas, using pilot signals of mutually orthogonal sequences, and in each antenna of the first and second antennas, using different sequences of pilot signals on different carriers, and The same sequence of pilot signals is used by the first and second antennas.

According to this configuration, when MIMO-OFDM transmission is performed using two transmission antennas, an increase in transmission peak power can be suppressed without deteriorating frequency offset and phase noise estimation accuracy, and a transmission device with a simple configuration can be realized.

In addition, the structure adopted by the MIMO-OFDM transmitting device of the present invention is that, in the case of transmitting OFDM signals through three antennas, the pilot symbol mapping unit forms the pilot carrier so that the first 1. Between the second and third antennas, pilot signals of mutually orthogonal sequences are used, and there are antennas using pilot signals of different sequences in different carriers configured with the pilot signals , and there are two or more antennas using pilot signals of the same sequence.

According to this configuration, when MIMO-OFDM transmission is performed using three transmission antennas, an increase in transmission peak power can be suppressed without deteriorating frequency offset and phase noise estimation accuracy, and a transmission device with a simple configuration can be realized.

The pilot symbol mapping unit also includes: a storage unit that stores basic sequences that are in an orthogonal relationship with each other; and a shift register that shifts the basic sequences to form a number that is greater than the basic sequence and that is in the A sequence of orthogonal relationships.

The MIMO-OFDM transmission method of the present invention is used for transmitting OFDM-modulated data symbols through multiple antennas during data transmission, and transmitting pilot symbols through specific carriers of the multiple antennas during the data transmission , the MIMO-OFDM transmission method comprises: an OFDM signal forming step, forming an OFDM signal transmitted through each antenna; and a pilot carrier forming step, distributing sequences in an orthogonal relationship to each other along the time axis The same carrier between the OFDM signals sent at the same time, thus forming the pilot carrier.

In the case of transmitting an OFDM signal through two antennas, the pilot carrier forming step forms the pilot carrier so that between the first and second antennas of the same carrier, sequences of sequences in an orthogonal relationship to each other are used. pilot signals, and in the respective antennas of the first and second antennas, pilot signals of different sequences are used on different carriers, and pilot signals of the same sequence are used by the first and second antennas .

In the case of transmitting an OFDM signal through three antennas, the pilot carrier forming step forms the pilot carrier so that among the first, second and third antennas of the same carrier, using The pilot signal of the sequence; and in the different carriers configured with the pilot signal, there are antennas using the pilot signal of the different sequence; and there are two or more antennas using the pilot signal of the same sequence antenna.

The effect of the invention

According to the present invention, it is possible to realize a MIMO-OFDM transmission device and a MIMO-OFDM transmission method capable of high-precision frequency offset estimation, high-precision channel variation estimation, and high-precision synchronization and signal detection.

Description of drawings

FIG. 1 is a diagram for explaining a conventional wireless communication system. FIG. 1(a) is a diagram showing an example of a configuration of a transmitting device, and FIG. 1(b) is a diagram showing an example of a configuration of a receiving device. FIG. 1 (c) is a diagram showing an example of a transmission frame structure.

FIG. 2 is a block diagram showing the configuration of a MIMO-OFDM transmission device according to Embodiment 1 of the present invention.

FIG. 3A is a block diagram showing the configuration of a MIMO-OFDM receiving apparatus according to Embodiment 1 of the present invention.

FIG. 3B is a diagram showing the relationship between transmission and reception antennas in Embodiment 1. FIG.

4 is a diagram showing the frame structure of signals transmitted by each antenna in Embodiment 1. FIG. 4(a) is a diagram showing the frame structure of channel A, and FIG. 4(b) is a diagram showing the frame structure of channel B. picture.

5 is a diagram showing a signal point configuration of a data symbol, FIG. 5A is a diagram showing a signal point configuration of BPSK, FIG. 5B is a diagram showing a signal point configuration of QPSK, and FIG. 5C is a diagram showing a signal point configuration of 16QAM, FIG. 5D is a diagram showing a signal point arrangement of 64QAM, and FIG. 5E is a diagram showing normalization coefficients multiplied by signals of respective modulation schemes.

FIG.6 is a diagram for explaining signal point arrangement of pilot symbols according to Embodiment 1. FIG.

Fig. 7 is a block diagram showing the configuration of a frequency offset and phase noise compensation unit.

FIG. 8 is a diagram for explaining a signal point arrangement of a preamble according to Embodiment 1. FIG.

FIG. 9 is a block diagram showing the configuration of a channel variation estimation unit.

FIG. 10 is a diagram for explaining a signal point arrangement of a preamble according to Embodiment 1. FIG.

FIG. 11 is a graph showing temporal changes in reception strengths of preambles and data symbols.

FIG. 12 is a diagram for explaining a signal point arrangement of a preamble according to Embodiment 1. FIG.

FIG. 13 is a block diagram showing the configuration of a mapping unit according to the embodiment of the present invention.

14 is a block diagram showing the configuration of a MIMO-OFDM transmission device according to Embodiment 2 of the present invention.

FIG.15 is a block diagram showing the configuration of a MIMO-OFDM receiving apparatus according to Embodiment 2. FIG.

FIG. 16 is a diagram showing the relationship between transmission and reception antennas according to Embodiment 2. FIG.

FIG. 17 is a diagram showing the frame structure of signals transmitted by each antenna in Embodiment 2, FIG. 17( a ) is a diagram showing the frame structure of channel A, and FIG. 17( b ) is a diagram showing the frame structure of channel B. (c) of FIG. 17 is a diagram showing a channel C frame structure.

FIG. 18 is a diagram illustrating a signal point arrangement of a preamble according to Embodiment 2. FIG.

FIG. 19 is a diagram for explaining a signal point arrangement of a preamble according to Embodiment 2. FIG.

FIG. 20 is a block diagram illustrating the configuration of Embodiment 3, FIG. 20( a ) is a block diagram showing the configuration of a terminal, and FIG. 20( b ) is a block diagram showing the configuration of an access point.

FIG. 21 is a diagram for explaining the relationship between the communication method and the normalization coefficient according to the third embodiment.

FIG. 22 is a diagram for explaining a signal point arrangement of a preamble according to Embodiment 3. FIG.

FIG. 23 is a diagram for explaining a signal point arrangement of a preamble according to Embodiment 3. FIG.

24 is a diagram showing the frame structure of Embodiment 4, FIG. 24 (a) is a diagram showing the frame structure of channel A, FIG. 24 (b) is a diagram showing the frame structure of channel B, and FIG. 24 (c ) is a diagram showing the frame structure of channel C.

FIG. 25 is a diagram illustrating an example of the structure of a preamble according to Embodiment 4. FIG.

Fig. 26 is a diagram showing another expression of the relationship between the communication method and the normalization coefficient.

27 is a diagram showing the frame structure of Embodiment 5, FIG. 27 (a) is a diagram showing the frame structure of channel A, FIG. 27 (b) is a diagram showing the frame structure of channel B, and FIG. 27 (c ) is a diagram showing the frame structure of channel C.

FIG.28 is a diagram for explaining signal point arrangement of pilot symbols according to Embodiment 5. FIG.

FIG.29 is a block diagram showing the configuration of a MIMO-OFDM transmission device according to Embodiment 5. FIG.

FIG. 30 is a block diagram showing the configuration of a mapping unit according to Embodiment 5. FIG.

31 is a block diagram showing an example of another configuration of the mapping unit according to the fifth embodiment.

32 is a block diagram showing the configuration of a frequency offset and phase noise estimating unit according to Embodiment 5.

33 is a block diagram showing another configuration of the frequency offset and phase noise estimating section according to the fifth embodiment.

34 is a diagram showing an example of another frame structure according to Embodiment 5. (a) of FIG. 34 is a diagram showing a frame structure of channel A, and (b) of FIG. 34 is a diagram showing a frame structure of channel B. (c) of 34 is a diagram showing the frame structure of channel C.

35 is a diagram showing an example of another frame structure according to Embodiment 5. (a) of FIG. 35 is a diagram showing the frame structure of channel A. FIG. (c) is a diagram showing the frame structure of channel C.

36 is a diagram showing an example of another frame structure according to Embodiment 5. (a) of FIG. 36 is a diagram showing a frame structure of channel A, and (b) of FIG. 36 is a diagram showing a frame structure of channel B. (c) of 36 is a figure which shows the frame structure of channel C.

37 is a block diagram showing the configuration of a frequency offset and phase noise estimating unit according to the sixth embodiment.

FIG. 38 is a block diagram showing the configuration of a mapping unit according to Embodiment 6. FIG.

39 is a diagram showing the frame structure of Embodiment 7, FIG. 39( a ) is a diagram showing the frame structure of channel A, and FIG. 39( b ) is a diagram showing the frame structure of channel B.

FIG.40 is a diagram for explaining signal point arrangement of pilot symbols according to Embodiment 7. FIG.

FIG.41 is a block diagram showing the configuration of a MIMO-OFDM transmission device according to Embodiment 7. FIG.

FIG. 42 is a block diagram showing the configuration of a mapping unit according to Embodiment 7. FIG.

43 is a block diagram showing the configuration of a frequency offset and phase noise estimating unit according to Embodiment 7. FIG.

FIG.44 is a block diagram showing the configuration of a MIMO-OFDM transmission device according to Embodiment 8. FIG.

45 is a diagram showing the frame structure of Embodiment 8, FIG. 45 (a) is a diagram showing the frame structure of channel A, FIG. 45 (b) is a diagram showing the frame structure of channel B, and FIG. 45 (c ) is a diagram showing the frame structure of channel C, and (d) of FIG. 45 is a diagram showing the frame structure of channel D.

FIG.46 is a diagram showing the relationship between the modulation scheme of the reference symbol and the normalization coefficient when transmission is performed by the 4-transmission spatial multiplexing MIMO scheme.

FIG.47 is a diagram showing an example of mapping of reference symbols when transmission is performed in the 4-transmission spatial multiplexing MIMO scheme.

FIG.48 is a diagram showing an example of mapping of reference symbols when transmission is performed in the 4-transmission spatial multiplexing MIMO scheme.

49 is a diagram showing an example of another frame structure according to Embodiment 8. (a) of FIG. 49 is a diagram showing a frame structure of channel A, and (b) of FIG. 49 is a diagram showing a frame structure of channel B. 49(c) is a diagram showing the frame structure of channel C, and FIG. 49(d) is a diagram showing the frame structure of channel D.

detailed description

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

(Embodiment 1)

In this embodiment, the configuration of a MIMO system using spatial multiplexing (Spatial Multiplexing) and the configurations of a transmitting device and a receiving device in this configuration will be described, and the estimation accuracy of frequency offset, channel variation, and synchronization can be improved. , and the structure of the detection probability of the signal, the pilot symbol, the preamble and the reference symbol.

FIG. 2 is a block diagram showing the configuration of the MIMO-OFDM transmission device 100 according to this embodiment. However, FIG. 2 shows the case where the number of transmission antennas m=2 as an example.

The frame structure signal generating section 112 receives control information 111 such as a modulation method as input, generates a frame structure signal 113 including information on the frame structure, and outputs it.

Mapping section 102A receives transmission digital signal 101A of channel A and frame structure signal 113 as input, generates frame structure based baseband signal 103A, and outputs it.

Serial-to-parallel conversion unit 104A takes baseband signal 103A and frame structure signal 113 as input, performs serial-to-parallel conversion based on frame structure signal 113 , and outputs parallel signal 105A.

Inverse Fourier transform unit 106A takes parallel signal 105A as input, performs inverse Fourier transform, and outputs inverse Fourier transformed signal 107A.

Wireless unit 108A takes inverse Fourier-transformed signal 107A as input, performs processing such as frequency conversion, and outputs transmission signal 109A. Transmission signal 109A is output as radio waves through antenna 110A.

MIMO-OFDM transmission apparatus 100 also performs the same processing as channel A on channel B, thereby generating channel B transmission signal 109B. In addition, the element indicated by adding "B" at the end of the reference numerals is the part related to channel B, but the signal as the object is not channel A but channel B, which is basically the same as that of adding "B" at the end of the above reference numerals. A" is the same process as the part related to channel A.

FIG. 3A shows an example of the configuration of a receiving device according to this embodiment. However, FIG. 3A shows a case where the number of receiving antennas is n=2 as an example.

In receiving device 200 , wireless unit 203X takes received signal 202X received via receiving antenna 201X as input, performs processing such as frequency conversion, and outputs baseband signal 204X.

The Fourier transform unit 205X takes the baseband signal 204X as input, performs Fourier transform, and outputs a Fourier transformed signal 206X.

The same operation is performed at the receiving antenna 201Y. The synchronization unit 211 takes the baseband signals 204X and 204Y as input, for example, by detecting reference symbols, establishes time synchronization with the transmitting device, and outputs a timing signal 212 . The structure of the reference symbol and the like will be described later in detail using FIG. 4 and the like.

Frequency offset and phase noise estimation unit 213 takes Fourier transformed signals 206X and 206Y as input, extracts pilot symbols, estimates frequency offset and phase noise according to pilot symbols, and outputs phase distortion estimation signal 214 (including phase distortion with frequency offset). The structure of the pilot symbol and the like will be described in detail later using FIG. 4 and the like.

The channel A transmission path variation estimation unit 207A takes the Fourier transformed signal 206X as input, extracts the reference symbol of channel A, for example, estimates the channel A transmission path variation according to the reference symbol, and outputs the channel A transmission path Signal 208A is estimated.

The transmission path variation estimation unit 207B of the channel B takes the Fourier transformed signal 206X as input, extracts the reference symbol of the channel B, for example, estimates the variation of the transmission path of the channel B according to the reference symbol, and outputs the transmission path of the channel B Signal 208B is estimated.

For the channel A channel variation estimation unit 209A and the channel B channel variation estimation unit 209B, only the target signal is not the signal received by the antenna 201X but the signal received by the antenna 201Y. The above channel A channel variation estimation unit 207A and channel B channel variation estimation unit 207B perform the same processing.

Frequency offset and phase noise compensation unit 215 takes channel A transmission path estimation signals 208A and 210A, channel B transmission path estimation signals 208B and 210B, Fourier-transformed signals 206X and 206Y, and phase distortion estimation signal 214 as input, The phase distortion of each signal is removed, and the phase-compensated transmission path estimation signals 220A and 222A of channel A, the phase-compensated transmission path estimation signals 220B and 222B of channel B, and the phase-compensated Fourier-transformed signal 221X are output and 221Y.

For example, the signal processing unit 223 performs an inverse matrix operation, and outputs a baseband signal 224A of channel A and a baseband signal 224B of channel B. Specifically, as shown in FIG. 3B , for example, in a certain subcarrier, let the transmitted signal from the antenna AN1 be Txa(t), the transmitted signal from the antenna AN2 be Txb(t), and the received signal from the antenna AN3 be The received signal of R1(t) and antenna AN4 is R2(t), and the transmission path variation is respectively set to h11(t), h12(t), h21(t) and h22(t), then the following relation holds true .

Among them, t is time, n1(t) and n2(t) are noises. The signal processing unit 223 obtains the signal of the channel A and the signal of the channel B by using the formula (1), for example, by performing an inverse matrix operation. The signal processing unit 223 performs this calculation for all subcarriers. Also, estimation of h11(t), h12(t), h21(t), and h22(t) is performed by channel variation estimation sections 207A, 209A, 207B, and 209B.

The frequency offset estimation and compensation unit 225A takes the baseband signal 224A of channel A as an input, extracts pilot symbols, estimates and compensates the frequency offset of the baseband signal 224A based on the pilot symbols, and outputs the frequency offset compensated baseband Signal 226A.

Channel A demodulation unit 227A takes frequency offset compensated baseband signal 226A as input, demodulates the data symbols, and outputs received data 228A.

The MIMO-OFDM receiving apparatus 200 also performs the same processing on the baseband signal 224B of the channel B, thereby obtaining received data 228B.

FIG. 4 shows the frame configurations of channel A ((a) in FIG. 4 ) and channel B ((b) in FIG. 4 ) in time-frequency according to this embodiment. In (a) of FIG. 4 and (b) of FIG. 4 , signals of the same carrier at the same time are spatially multiplexed.

From time 1 to time 8, symbols for estimating transmission path variations equivalent to h11(t), h12(t), h21(t) and h22(t) of equation (1) are sent, such as called the preamble. The preamble is composed of a guard symbol 301 and a reference symbol 302 . Let the guard symbol 301 be (0, 0) on the in-phase I-orthogonal Q plane. The reference symbol 302 is, for example, a symbol with known coordinates other than (0, 0) on the in-phase I-orthogonal Q plane. In addition, channel A and channel B are configured so as not to interfere with each other. That is to say, for example, as carrier 1, time 1, when guard symbol 301 is configured in channel A, reference symbol 302 is configured in channel B; as carrier 2, time 1, reference symbol 302 When channel A is placed, guard symbol 301 is placed on channel B, and different symbols are placed on channel A and channel B in this way. By configuring in this way, for example, when focusing on channel A at time 1, it is possible to estimate the channel variation of carrier 3 from the reference symbols 302 of carrier 2 and carrier 4 . Since the carrier 2 and the carrier 4 are the reference symbols 302, it is possible to estimate channel variation. Therefore, in time 1, it is possible to accurately estimate propagation channel fluctuations of all carriers of channel A. Similarly, it is also possible to estimate propagation path fluctuations of all carriers on channel B with high accuracy. For time 2 to time 8, it is also possible to estimate channel fluctuations of all the carriers of channel A and channel B. FIG. Therefore, with regard to the frame structure of FIG. 4 , since it is possible to estimate propagation path fluctuations of all carriers in all the times from time 1 to time 8, it can be said that it is possible to realize extremely accurate estimation of propagation path fluctuations. code structure.

In FIG. 4, an information symbol (data symbol) 303 is a symbol for data transmission. Here, it is assumed that the modulation schemes are BPSK, QPSK, 16QAM, and 64QAM. The signal point arrangement and the like on the in-phase I-quadrature Q plane at this time will be described in detail using FIG. 5 .

The control symbol 304 is a symbol for transmitting control information such as a modulation method, an error correction coding method, and a coding rate.

Pilot symbols 305 are symbols for estimating phase fluctuations due to frequency offset and phase noise. As the pilot symbol 305, for example, a symbol with known coordinates on the in-phase I-orthogonal Q plane is used. Pilot symbols 305 are configured in carrier 4 and carrier 9 on both channel A and channel B. In this way, especially in a wireless LAN, when a system of the same frequency and the same frequency band is constructed by IEEE802.11a, IEEE802.11g and a spatially multiplexed MIMO system, the frame structure can be shared, and thus the reception device can be simplified. .

FIG. 5 shows the signal point arrangement on the in-phase I-quadrature Q plane of BPSK, QPSK, 16QAM, and 64QAM, which are the modulation schemes of the information symbol 303 in FIG. 4, and their normalization coefficients.

FIG. 5A is the signal point configuration of BPSK on the in-phase I-quadrature Q plane, and its coordinates are shown in FIG. 5A . FIG. 5B is a signal point configuration of QPSK on the in-phase I-orthogonal Q plane, and its coordinates are shown in FIG. 5B . FIG. 5C is a signal point configuration of 16QAM on the in-phase I-quadrature Q plane, and its coordinates are shown in FIG. 5C . FIG. 5D is a signal point configuration of 64QAM on the in-phase I-quadrature Q plane, and its coordinates are shown in FIG. 5D . FIG. 5E is a diagram showing the relationship between the modulation method and the multiplication coefficient (that is, the normalization coefficient) for correcting the signal point arrangement in FIGS. 5A to 5D so as to keep the average transmission power constant among the modulation methods. For example, in the case of transmitting with the QPSK modulation scheme shown in FIG. 5B , as can be seen from FIG. 5E , it is necessary to multiply the coordinates in FIG. 5B by the value of 1/sqrt(2). Among them, sqrt(x) is the square root of x (square root of x).

FIG.6 shows the arrangement on the in-phase I-orthogonal Q plane of the pilot symbol 305 in FIG.4 in this embodiment. (a) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 4 of channel A shown in FIG. 4( a ). (b) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 4 of channel B shown in FIG. 4( b ). (c) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 9 of channel A shown in (a) of FIG. 4 . (d) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 9 of channel B shown in (b) of FIG. 4 . Here, these configurations use BPSK modulation, but are not limited thereto.

The feature of the signal point configuration of the pilot symbol 305 in FIG. 6 is that the signal point configurations of channel A and channel B of the same carrier are orthogonal (the cross-correlation is zero).

For example, the signal point configuration of carrier 4 of channel A from time 11 to time 14 is orthogonal to the signal point configuration of carrier 4 of channel B from time 11 to time 14. Furthermore, the same applies to time 15 to time 18 . In addition, the signal point configuration of carrier 9 of channel A from time 11 to time 14 is also orthogonal to the signal point configuration of carrier 9 of channel B from time 11 to time 14. Furthermore, the same applies to time 15 to time 18 . At this time, for the orthogonality of signals, Walsh-Hadamard transform, orthogonal codes, and the like are suitably used. In addition, although the case of BPSK is shown in FIG. 6 , as long as it is orthogonal, QPSK modulation may be used, and the regulation of the modulation method may not be followed.

Moreover, in the case of this embodiment, in order to simplify the receiving device, it is assumed that carrier 4 of channel A and carrier 9 of channel B, and carrier 9 of channel A and carrier 4 of channel B are configured with the same signal point (the same pattern) (here, as shown in FIG. 6, the respective patterns are named pattern #1 and pattern #2). The reason for this will be described in detail using FIG. 7 . However, the same pattern does not use exactly the same signal point configuration. For example, on the in-phase I-quadrature Q plane, the same pattern can be considered only when the phase relationship is different.

In addition, the signal point arrangement of the pilot symbol 305 is made different between the carriers 4 and 9 of the channel A (or channel B), because if they are set to be the same, the transmission peak power may increase. However, the same can be said for the patterns defined above. That said, it is important that the signal point configurations are different.

Here, the advantage of orthogonality will first be described in detail using FIG. 3A and FIG. 7 .

FIG. 7 shows an example of the configuration of frequency offset and phase noise estimating section 213 in FIG. 3A . Pilot carrier extraction section 602 receives Fourier-transformed signal 206X (or 206Y) as input, and extracts the subcarrier of pilot symbol 305 . Specifically, the signals of carrier 4 and carrier 9 are extracted. Therefore, the pilot carrier extraction unit 602 outputs the baseband signal 603 of the carrier 4 and the baseband signal 604 of the carrier 9 .

For example, the code storage unit 605 stores the pattern #1 in FIG. 6 , and outputs the signal 606 of the pattern #1 according to the timing signal 212 .

For example, the code storage unit 607 stores the pattern #2 in FIG. 6 , and outputs the signal 608 of the pattern #2 according to the timing signal 212 .

The selection unit 609 takes the timing signal 212, the signal 606 of the pattern #1, and the signal 608 of the pattern #2 as inputs, outputs the signal of the pattern #2 as the selection signal 610 (X), and outputs the signal of the pattern #1 as the selection signal 611 (Y) signal of.

The code multiplication unit 612A takes the baseband signal 603 of the carrier 4 and the selection signal 611 (Y) as input, and multiplies the baseband signal 603 of the carrier 4 and the selection signal 611 (Y), thereby generating the baseband signal 613A of the channel A of the carrier 4, and output it. The reason for this is as follows.

The baseband signal 603 of carrier 4 is a signal in which the baseband signal of channel A and the baseband signal of channel B are multiplexed. In contrast, by multiplying the signal of pattern #1 which is the selection signal 611 (Y), the baseband signal component of channel B with zero cross-correlation is removed, so that only the baseband signal component of channel A can be extracted.

Similarly, the code multiplication unit 614A takes the baseband signal 604 of carrier 9 and the selection signal 610 (X) as input, and multiplies the baseband signal 604 of carrier 9 and the selection signal 610 (X), thereby generating the baseband of channel A of carrier 9 signal 615A and output it.

The code multiplication unit 612B takes the baseband signal 603 of the carrier 4 and the selection signal 610(X) as input, and multiplies the baseband signal 603 of the carrier 4 and the selection signal 610(X), thereby generating the baseband signal 613B of the channel B of the carrier 4, and output it.

The code multiplication unit 614B takes the baseband signal 604 of the carrier 9 and the selection signal 611 (Y) as input, and multiplies the baseband signal 604 of the carrier 9 and the selection signal 611 (Y), thereby generating the baseband signal 615B of the channel B of the carrier 9, and output it.

As described above, by making the signal point arrangement of channel A and channel B of the same carrier orthogonal, even if the pilot symbol 305 is multiplexed on channel A and channel B, it is possible to perform frequency offset and phase noise estimation with high accuracy. estimate. As another important advantage, since channel estimation values (transmission path variation estimation values) are not required, the configuration of a part that compensates for frequency offset and phase noise can be simplified. If the signal point configurations of the pilot symbols 305 of channel A and channel B are not orthogonal to each other, then it becomes the following structure: perform signal processing for MIMO separation, such as ZF (zero forcing, ZeroForcing), MMSE ( Minimum mean square error, Minimum Mean Square Error) and MLD (Maximum Likelihood Detection, Maximum Likelihood Detection), followed by estimation of frequency offset and phase noise. On the other hand, according to the configuration of the present embodiment, as shown in FIG. 3A , it is possible to compensate for frequency offset and phase noise at a stage preceding signal separation (signal processing section 223 ). Furthermore, in the signal processing unit 223, even after the signal of the channel A and the signal of the channel B are separated, the frequency offset and the phase noise can be removed by the pilot symbol 305, so it is possible to perform frequency offset and phase noise with higher precision. Compensation for phase noise.

However, when the signal point configurations of channel A and channel B of the same carrier are not orthogonal, the frequency offset and phase noise estimation accuracy of the frequency offset and phase noise estimation section 213 in FIG. interference component), it is difficult to add the frequency offset and phase noise compensation unit 215 in FIG. 3A , so that high-precision frequency offset and phase noise compensation cannot be performed.

Furthermore, according to the present embodiment, by setting carrier 4 of channel A and carrier 9 of channel B, and carrier 9 of channel A and carrier 4 of channel B, in the same signal point configuration (same pattern), it is possible to realize 7 code storage units 605 and 607 share, resulting in simplification of the receiving device.

However, in the present embodiment, it is necessary to make the signal point arrangements of channel A and channel B of the same carrier orthogonal, but they do not necessarily have to be set in the same pattern.

In the present embodiment, the pilot symbols 305 orthogonal to each other in units of four symbols have been described as examples from time 11 to time 14, but the present invention is not limited to units of four symbols. However, considering the effect on orthogonality due to fading fluctuations in the time direction, it is expected that forming an orthogonal pattern with about 2 to 8 symbols can ensure the estimation accuracy of frequency offset and phase noise . If the period of the orthogonal pattern is too long, there is a high possibility that the orthogonality cannot be ensured, and the estimation accuracy of the frequency offset and the phase noise deteriorates. In addition, although the case where the number of transmitting antennas is 2 and two modulated signals are transmitted is described, it is not limited thereto. For the case where the number of transmitting antennas is 3 or more and three or more modulated signals are transmitted, it is also possible to use The pilot symbols 305 are orthogonalized in units of several symbols, and the same effects as above are obtained.

Next, the structure of the reference symbol 302 in the preamble of FIG. 4 will be described in detail, which can simplify the receiving device and suppress deterioration of channel estimation accuracy due to quantization errors occurring in the receiving device.

Fig. 8 shows the signal point arrangement on the in-phase I-orthogonal Q plane of the preamble of this embodiment, especially shows that the reference symbol 302 is arranged at times 1 and 3 of carriers 2, 4, 6, 8, 10 and 12 , 5 and 7 signal point configurations.

Here, the signal formed at times 1, 3, 5, and 7 of carrier 2, the signal formed at times 1, 3, 5, and 7 of carrier 4, the signal formed at times 1, 3, 5, and 7 of carrier 6, Signals formed at times 1, 3, 5, and 7 of carrier 8; signals formed at times 1, 3, 5, and 7 of carrier 10, and signals formed at times 1, 3, 5, and 7 of carrier 12, in-phase On the I-quadrature Q plane, the phase relationship is different but the same pattern. This enables simplification of the receiving device.

Similarly, carriers 1, 3, 5, 7, 9, and 11 have different phase relationships but have the same pattern, thereby enabling simplification of the receiving device. Here, the receiving apparatus can be further simplified by making the pattern of the even-numbered carrier and the pattern of the odd-numbered carrier the same. Even in these different cases, however, there are certain advantages with regard to the simplification of the receiving device. This is because the required pattern signal is increased by only one. Similarly, if the same pattern is used for channel A and channel B, the receiving device can be further simplified, but even if different patterns are used, there is a certain advantage.

Hereinafter, an aspect to realize the simplification of the configuration of the receiving device will be described. However, in the following, a case where the pattern of the even-numbered carrier is the same as that of the odd-numbered carrier will be described as an example.

FIG. 9 shows a detailed configuration of channel fluctuation estimating sections 207 and 209 of the receiving apparatus in FIG. 3A. Here, the estimation of channel A channel variation will be described as an example.

The signal extraction unit 802_1 of carrier 1 takes the signal 206X (206Y) after Fourier transform as input, and extracts the reference symbol 302 (time 2, 4, 6 and 8), and output the reference symbol 803_1 of carrier 1.

The signal extraction unit 802_2 of the carrier 2 takes the Fourier transformed signal 206X (206Y) as input, and extracts the reference symbol 302 (time 1, 3, 5 and 7), and output the reference symbol 803_2 of carrier 2.

The same operation is performed in the signal extraction units of carrier 3 to carrier 12 .

The pattern signal generation unit 804 outputs a pattern signal 805 of (1,0), (-1,0), (-1,0) and (1,0) on the in-phase I-orthogonal Q plane (refer to the pattern of FIG. 8 ).

The multiplication unit 806_1 takes the reference symbol 803_1 of the carrier 1 and the pattern signal 805 as input, performs signal processing such as averaging while multiplying the reference symbol 803_1 of the carrier 1 and the pattern signal 805, and outputs the transmission path of the carrier 1 Variation estimation signal 807_1.

The multiplication unit 806_2 to the multiplication unit 806_12 also perform the same operation, thereby outputting the channel variation estimation signal 807_2 of the carrier 2 to the channel variation estimation signal 807_12 of the carrier 12 .

Parallel-to-serial conversion unit 810 takes channel variation estimation signal 807_1 of carrier 1 to channel variation estimation signal 807_12 of carrier 12 as input, performs parallel-to-serial conversion, and outputs channel variation estimation signal 208A (208B, 210A, and 210B).

In this way, since the pattern signal generation section 804 can be shared by the carrier 1 to the carrier 12, the storage capacity of the pattern signal in the pattern signal generation section 804 can be reduced and the signal processing can be shared, thereby simplifying the receiving device accordingly.

However, shown in FIG. 8 is the signal point configuration on the in-phase I-orthogonal Q plane when the reference symbol 302 has been BPSK modulated, and it is the same as that of the data symbol 303 that has been BPSK modulated. The signal point arrangement in this case is the same, and the normalization coefficient used for multiplication is also the same as that in the case where the data symbol 303 is BPSK-modulated. However, in this case, when an analog-to-digital converter is provided for digital signal processing in the receiving apparatus, the influence of the quantization error becomes large. An example of signal point arrangement on the in-phase I-quadrature Q plane to alleviate this problem will be described below.

FIG. 10 shows an example of signal point arrangement on the in-phase I-quadrature Q plane to alleviate this problem. As an example, BPSK modulation is used. At this point, the normalization coefficient is set to 1.0, and the signal point configuration of the reference symbol 302 is or That is, a signal point arrangement obtained by multiplying the signal point arrangement when BPSK modulation is performed on the data symbol 303 by a coefficient of 1.414 is employed.

The temporal change in received signal strength in such a case will be described with reference to FIG. 11 .

In FIG. 11, (a) of FIG. 11 shows the time-varying waveform of the received signal when the signal point arrangement of the preamble is performed as shown in FIG. 8; (b) of FIG. A waveform showing a time variation of a received signal when the signal point arrangement of the preamble is performed. When the signal points are arranged as shown in FIG. 8 , the average received power of the preamble is smaller than the average received power of the data symbol 303 . This phenomenon occurs because the guard symbol 301 exists in the reference symbol 302 of the preamble if the same signal point arrangement as that of the data symbol 303 is performed. As a result, especially when the received signal is converted into a digital signal by an analog-to-digital converter, the quality of the received signal of the preamble deteriorates due to the influence of the quantization error.

On the other hand, when the signal point arrangement of the preamble is performed as shown in FIG. 10, as shown in FIG. 11(b), the average received power of the preamble is equal to the average received power of the data symbol 303. level. Therefore, even if the received signal is converted into a digital signal by the analog-to-digital converter, the influence of the quantization error on the received signal of the preamble is reduced, and the quality is ensured.

FIG. 12 shows a method of signal point arrangement when the signal point arrangement of the reference symbol 302 is set to QPSK based on the same idea as above.

FIG. 12 shows an example of signal point arrangement on the in-phase I-quadrature Q plane when the normalization coefficient is set to 1 and the reference symbol 302 is subjected to QPSK modulation. Thus, as shown in FIG. 11(b), the average received power of the preamble is at the same level as the average received power of the data symbol 303. Even if it is converted into a digital signal by an analog-to-digital converter, the For the received signal, the influence due to the quantization error is mitigated, thereby ensuring the quality.

From the above, it is important that when the modulation scheme #X of the data symbol 303 is used for the reference symbol 302, the in-phase I-quadrature Q signal point configuration on the plane of The multiplied signal point configuration is set as the signal point configuration of the reference symbol 302 on the in-phase I-orthogonal Q plane after multiplication by the normalization coefficient. The coefficient of (2) times is a value determined by arranging the reference symbol 302 every other symbol on the frequency axis.

For example, when #X is QPSK, the signal point configuration on the in-phase I-orthogonal Q plane after the normalization coefficient multiplication operation is (±1/sqrt(2), ±1/sqrt(2)), According to the above rules, the signal points on the in-phase I-quadrature Q plane after the normalization coefficient multiplication of the reference symbol 302 are configured as (±1, ±1) (see FIG. 12 ).

FIG. 13 shows data as an example of the configuration of mapping section 102A ( 102B) of the transmission device in FIG. 2 according to this embodiment. The modulation unit 1103 takes the transmission digital signal 101A (101B) and the frame structure signal 1102 as input, and modulates the transmission digital signal 101A (101B) based on the information and timing of the modulation method contained in the frame structure signal 1102, and outputs the data Modulated signal 1104 of symbol 303.

The preamble mapping unit 1105 takes the frame structure signal 1102 as input, and outputs a preamble modulation signal 1106 based on the frame structure.

Code storage unit #1 (1107) outputs signal 1108 of pattern #1. Likewise, code storage unit #2 (1109) outputs signal 1110 of pattern #2.

Pilot symbol mapping section 1111 receives pattern #1 signal 1108, pattern #2 signal 1110, and frame structure signal 1102 as input, generates modulated signal 1112 of pilot symbol 305, and outputs it.

The signal generating unit 1113 takes the modulated signal 1104 of the data symbol 303, the modulated signal 1106 of the preamble and the modulated signal 1112 of the pilot symbol 305 as input, generates a baseband signal 103A (103B) conforming to the frame structure, and outputs it .

Although in the above description, it has been explained that if the structure of the pilot symbol 305 as shown in FIG. 4 and FIG. In the meta-structure, the code storage units 1107 and 1109 can be shared as shown in FIG. 13 in the transmitting device, so that the transmitting device can be simplified.

In the above, the method of generating the preamble and the pilot signal (pilot symbol) of the present embodiment and the transmitting device for generating them have been described, and the detailed configuration and action. According to the present embodiment, since it is possible to improve the estimation accuracy of frequency offset, channel variation, and synchronization, the probability of signal detection can be improved, and the transmission device and the reception device can be simplified.

The above-mentioned important feature of the present embodiment is, in other words, a MIMO-OFDM transmission device that transmits OFDM-modulated data symbols 303 through a plurality of antennas during a data transmission period, and in a period different from the data transmission period The OFDM-modulated transmission path estimation symbols are transmitted through the plurality of antennas, and the MIMO-OFDM transmission device includes: a data mapping unit (data modulation unit 1103) to form a data symbol 303; a transmission path estimation symbol mapping unit (preamble mapping unit 1105), when the number of subcarriers is assumed to be m, when the signal point amplitude of n subcarriers is set to 0, and α=m/(m-n), the following transmission path is formed The symbol used for estimation, that is, the signal point amplitude of the remaining m-n subcarriers is √/α times the signal point amplitude of the same modulation mode in the modulation mode of the data symbol 303; and the OFDM modulation unit, for the data The symbols 303 and the symbols for channel estimation are OFDM-modulated. Accordingly, since the quantization error of the channel estimation symbols at the receiving end can be reduced, it is possible to perform highly accurate channel variation estimation.

In the present embodiment, an example using the OFDM method was described, but it is not limited to this, and it can be implemented similarly even when using the single-carrier method, other multi-carrier methods, and spread spectrum communication methods. In addition, in this embodiment, although the case where there are two antennas for transmission and reception has been described as an example, it is not limited to this. Even if the number of receiving antennas is three or more, this embodiment will not be affected, and the same can be achieved. implement. Moreover, the frame structure is not limited to this embodiment. In particular, for the pilot symbols 305 used for estimating distortions such as frequency offset and phase noise, as long as it is configured in a specific subcarrier and passed through multiple antennas The configuration to be transmitted is sufficient, and the number of subcarriers for transmitting the pilot symbol 305 is not limited to two as in this embodiment. In addition, the embodiment in the case of other numbers of antennas and other transmission methods will be described in detail later. In addition, although the pilot symbol 305, the reference symbol 302, the guard symbol 301, and the preamble are named and described here, using other calling methods has no influence on this embodiment. This is also the same in other embodiments.

(Embodiment 2)

In this embodiment, a case where the number of transmitting and receiving antennas is set to three in Embodiment 1 will be described in detail.

FIG. 14 shows an example of the configuration of a transmission device according to this embodiment. In FIG. 14 , the same reference numerals as in FIG. 2 are assigned to parts that perform the same operations as those in FIG. 2 . The MIMO-OFDM sending device 1200 in FIG. 14 is different from that in FIG. 2 in that a sending unit of channel C is added.

FIG. 15 shows an example of the configuration of a receiving device according to this embodiment. In FIG. 15 , parts that perform the same operations as those in FIG. 3 are assigned the same reference numerals. In FIG. 15 , since modulation signals of three channels are transmitted from the transmitting device, compared with the structure of FIG. 3A , channel C channel fluctuation estimation units 207C and 209C are added, and the number of antennas is increased by one, so it becomes an additional corresponding to the desired structure.

FIG. 16 shows the relationship between the transmitting and receiving antennas of this embodiment. For example, in a certain subcarrier, let the transmission signal from antenna 1401 be Txa(t), the transmission signal from antenna 1402 be Txb(t), and the transmission signal from antenna 1403 be Txc(t); The reception signal is R1(t), the reception signal of the antenna 1405 is R2(t), the reception signal of the antenna 1406 is R3(t), and the transmission path variation is set to h11(t), h12(t), h13( t), h21(t), h22(t), h23(t), h31(t), h32(t) and h33(t), the following relational expressions hold.

Among them, t is time, n1(t), n2(t) and n3(t) are noises. The signal processing unit 223 in FIG. 15 obtains the signal of the channel A, the signal of the channel B, and the signal of the channel C by using the formula (2), for example, by performing an inverse matrix operation. The signal processing unit 223 performs this calculation for all subcarriers. In addition, the estimates of h11(t), h12(t), h13(t), h21(t), h22(t), h23(t), h31(t), h32(t) and h33(t) are estimated by The channel variation estimation sections 207A, 209A, 1301A, 207B, 209B, 1301B, 207C, 209C, and 1301C perform.

FIG. 17 shows an example of the frame structure of this embodiment, and the parts corresponding to those in FIG. 4 are denoted by the same reference numerals. (a) of FIG. 17 shows an example of the frame structure of the time-frequency channel A; FIG. 17 (b) shows an example of the frame structure of the time-frequency channel B; FIG. 17 (c) shows an example of the time-frequency channel An example of the frame structure of C. In (a) of FIG. 17, (b) of FIG. 17, and (c) of FIG. 17, signals of the same time and the same carrier of channels A, B, and C are multiplexed in space.

From time 1 to time 8, it is used to estimate h11(t), h12(t), h13(t), h21(t), h22(t), h23(t), h31(t ), h32(t) and h33(t) are transmitted. These symbols are composed of guard symbols 301 and reference symbols 302. Let the guard symbol 301 be (0, 0) on the in-phase I-orthogonal Q plane. The reference symbol 302 is, for example, a symbol with known coordinates other than (0, 0) on the in-phase I-orthogonal Q plane. In addition, channel A, channel B, and channel C are configured so that they do not interfere with each other. That is to say, for example, as carrier 1, time 1, when the reference symbol 302 is configured on channel A, guard symbols 301 are configured on channel B and channel C; When the symbol 302 is arranged on channel B, the guard symbol 301 is arranged on channel A and channel C; as in carrier 3 and time 1, when the reference symbol 302 is arranged on channel C, the protection symbol 301 is arranged on channel A and channel C. Channel B configures guard symbols 301 . In this way, at a certain carrier and time, the reference symbol 302 is allocated to only one channel, and the guard symbol 301 is allocated to the remaining channels. By configuring in this way, for example, when focusing on channel A at time 1, it is possible to estimate channel fluctuations of carriers 2 and 3 from the reference symbols 302 of carriers 1 and 4 . In addition, since the carrier 1 and the carrier 4 are the reference symbols 302, it is possible to estimate channel variation. Therefore, in time 1, it is possible to accurately estimate propagation channel fluctuations of all carriers of channel A. Similarly, it is also possible to estimate channel fluctuations of all carriers of channel B and channel C with high accuracy. For time 2 to time 8, it is also possible to estimate channel fluctuations of all the carriers of channel A, channel B, and channel C. Therefore, with regard to the frame structure of FIG. 17 , since it is possible to estimate propagation path fluctuations of all carriers in all the times from time 1 to time 8, it can be said that it is possible to estimate propagation path fluctuations with very high accuracy. The structure of the preamble.

Next, the preamble on the in-phase I-orthogonal Q plane (in particular, the reference symbol 302 ) signal point configuration. As a precondition, it is assumed that the signal point arrangement of the data symbol 303 in FIG. 17 is adopted, and the normalization coefficient based on FIG. 5 is adopted.

FIG. 18 shows an example of the signal point arrangement of the preamble on the in-phase I-orthogonal Q plane (the arrangement example of time 1, 2 and 3 of channel A). Here, it is assumed that the modulation scheme of the reference symbol 302 is BPSK. At this point, consider the case where the normalization coefficient is 1 and the signal point configuration of FIG. 5A is adopted. As shown in (a) of FIG. Since the analog-to-digital converter of the receiving device is equipped, the influence of the quantization error increases only on the preamble, thereby incurring a reduction in the reception quality. In contrast, in FIG. 18 , the signal points on the in-phase I-orthogonal Q plane of the reference symbol 302 are configured as or for That is, a signal point arrangement obtained by multiplying a signal point arrangement of BPSK-modulated data symbols by a coefficient of 1.732 is employed.

(b) of FIG. 11 is an image (image) of the received signal strength on the time axis of the preamble and the data symbol 303 at this time. In this way, the influence of the quantization error on the preamble caused by the analog-to-digital converter provided in the receiving device can be reduced, thereby improving the quality.

FIG. 19 shows a method of signal point arrangement when the signal point arrangement of the reference symbol 302 is set to QPSK based on the same idea as above.

FIG. 19 shows an example of signal point arrangement on the in-phase I-quadrature Q plane when the normalization coefficient is set to 1 and the reference symbol 302 is subjected to QPSK modulation. Thus, as shown in (b) of FIG. 11 , the average received power of the preamble becomes the same level as the average received power of the data symbol 303, and even if it is converted into a digital signal by an analog-to-digital converter, the preamble For the received signal, the influence due to the quantization error is also reduced, so that the quality is ensured. Here, 1.225 in Figure 19 is based on Find the value.

From the above, it is important that when the modulation scheme #X of the data symbol 303 is used for the reference symbol 302, the in-phase I-quadrature The signal point configuration on the Q plane is The multiplied signal point configuration is set as the signal point configuration of the reference symbol 302 on the in-phase I-orthogonal Q plane after multiplication by the normalization coefficient. The coefficient of multiplying is a value determined by arranging the reference symbol 302 every two symbols on the frequency axis.

For example, when #X is QPSK, the signal point configuration on the in-phase I-orthogonal Q plane after the normalization coefficient multiplication operation is (±1/sqrt(2), ±1/sqrt(2)), And according to above-mentioned rules, the signal point configuration on the in-phase I-orthogonal Q plane after the normalization coefficient multiplication operation of reference symbol 302 is (±sqrt(3)/sqrt(2),±sqrt(3)/sqrt (2)) (refer to FIG. 19).

As above, the method for generating the preamble and the pilot signal and the transmitting device for generating them according to the present embodiment have been described, and the detailed configuration and operation of the receiving device for receiving the modulated signal according to the present embodiment have been described. The case where the number of receiving antennas is 3, respectively. According to the present embodiment, since it is possible to improve the estimation accuracy of frequency offset, channel variation, and synchronization, the probability of signal detection can be improved, and the transmission device and the reception device can be simplified.

In the present embodiment, an example using the OFDM method was described, but it is not limited to this, and it can be implemented similarly even when using the single-carrier method, other multi-carrier methods, and spread spectrum communication methods. Furthermore, even if the number of receiving antennas is four or more, it does not affect this embodiment and can be implemented in the same way.

(Embodiment 3)

In this embodiment, the structure of the preamble in the communication method that switches between two-transmission spatial multiplexing MIMO and three-transmission spatial multiplexing MIMO will be described in detail according to the communication environment (for example, reception quality, etc.). Transmit spatial multiplexing MIMO is a spatial multiplexing MIMO system in which the number of transmitting antennas is 2 and the number of transmitting modulated signals is 2, and three-transmitting spatial multiplexing MIMO is a spatial multiplexing MIMO system in which the number of transmitting antennas is 3 and the number of transmitting modulated signals is 3 .

FIG. 20 is a diagram showing a communication form of the present embodiment. (a) of FIG. 20 shows a terminal, and (b) of FIG. 20 shows an access point (AP). At this point, a case where the AP switches between the MIMO scheme and the modulation scheme is taken as an example for description.

In the terminal shown in (a) of FIG. 20 , transmitting device 1902 receives transmission digital signal 1901 as input, and outputs modulated signal 1903 , and modulated signal 1903 is output through antenna 1904 as radio waves. At this time, it is assumed that the transmitted digital signal 1901 includes communication status information for the AP to switch the communication mode, such as bit error rate, packet error rate, and received electric field strength.

In the AP of FIG. 20( b ), receiving device 1907 receives received signal 1906 received via antenna 1905 as input, and outputs received digital signal 1908 .

Transmission method determination unit 1909 receives received digital signal 1908 as input, determines a communication method (that is, MIMO method and modulation method) based on communication status information included in received digital signal 1908, and outputs control information 1910 including the information.

Transmitter 1912 receives control information 1910 and digital transmission signal 1911 as input, modulates digital transmission signal 1911 based on the determined communication method, and outputs modulated signal 1913 which is transmitted through an antenna.

FIG. 14 is an example of a detailed configuration of the transmission device 1912 in FIG. 20(b). Frame signal generator 112 in FIG. 14 receives control information 111 , that is, control information 1910 in FIG. 20 , determines the modulation method and MIMO method based on the information, and outputs frame signal 113 including the information. For example, when the frame configuration signal 113 indicates a 2-transmission spatial multiplexing MIMO scheme, the transmission unit of channel C does not operate. This enables switching between the 2-transmission spatial multiplexing MIMO scheme and the 3-transmission spatial multiplexing MIMO scheme.

Here, the frame structure when the 2-transmission spatial multiplexing MIMO method is selected is as shown in FIG. 4 described in Embodiment 1, and the frame structure when the 3-transmission spatial multiplexing MIMO method is selected is as shown in FIG. like that.

The signal point configuration on the in-phase I-quadrature Q plane of the data symbol 303 adopts the signal point configuration shown in FIG. 5A , FIG. 5B , FIG. 5C and FIG. 5D . Wherein, the normalization coefficient adopts the normalization coefficient as shown in FIG. 21 .

FIG. 21 shows normalization coefficients employed for each MIMO scheme and each modulation scheme when the normalization coefficient for two-transmission spatial multiplexing MIMO and BPSK is set to 1. The reason for setting the normalization coefficient in this way is to keep the total transmission power of the modulated signals transmitted by the AP constant regardless of the modulation scheme and the number of modulated signals to be transmitted. Therefore, in the same modulation mode, if the normalization coefficient of the two-transmission spatial multiplexing MIMO scheme is set to X, the normalization coefficient of the three-transmission spatial multiplexing MIMO scheme is sqrt(2)/sqrt(3 ) times.

Next, the signal point arrangement on the in-phase I-quadrature Q plane of the preamble at this time will be described.

In the case of the two-transmission spatial multiplexing MIMO scheme, in order to reduce the influence of the quantization error that occurs in the analog-to-digital converter of the receiving device, the signal on the in-phase I-orthogonal Q plane of the reference symbol 302 in the preamble The dot arrangement is set as the arrangement shown in FIG. 10 or FIG. 12 described in the first embodiment.

At this time, considering the normalization coefficient in FIG. 21 and the description of Embodiment 2, in order to reduce the influence of the quantization error that occurs in the analog-to-digital converter of the receiving device, it is necessary to set the signal points as shown in FIG. 22 and FIG. 23 configuration. With this setting, even if the AP switches between the 2-transmission spatial multiplexing MIMO scheme and the 3-transmission spatial multiplexing MIMO scheme, the influence of the quantization error in the preamble can be reduced in the receiving apparatus.

In the above description, it is important that, as shown in FIG. 21 , when the total transmission power of modulated signals to be transmitted is constant regardless of the modulation method and the number of modulated signals to be transmitted, data symbols 303 When the modulation scheme #X is used in the reference symbol 302, in the reference symbol 302, the two-transmission spatial multiplexing MIMO scheme and the three-transmission spatial multiplexing MIMO scheme make the in-phase I-orthogonal after the normalized multiplication The signal point arrangement on the Q plane is the same signal point arrangement.

By adopting the above configuration, it is possible to reduce the influence of the quantization error in the preamble, and therefore it is possible to suppress a reduction in reception quality.

In the above description, the case where the average received power of the data symbol 303 and the preamble are equal is described as an example, but when the average received power of the preamble is made larger than the average received power of the data symbol 303, the reception There are many situations where quality can be assured. The idea described above can also be applied to this case. That is to say, when the modulation method #X of the data symbol 303 is used for the reference symbol 302, it only needs to comply with the following rule, that is, the MIMO method is multiplexed in the second transmission space and the MIMO method in the third transmission space, so that in In the reference symbol 302, the signal point configurations on the in-phase I-quadrature Q plane after normalization and multiplication are the same signal point configurations.

That is, in the n-transmission spatial multiplexing MIMO scheme in which the number of transmission antennas is n and the number of transmission modulated signals is n, when the reference symbol 302 is inserted into the preamble every n-1 symbols , when the modulation scheme #X of the data symbol 303 is used in the reference symbol 302, in the reference symbol 302, as long as the signal points on the in-phase I-orthogonal Q plane after the normalized multiplication operation are configured so as not to be based on The same operation as above can be performed by changing n to the same signal point arrangement.

In the present embodiment, an example using the OFDM method was described, but it is not limited to this, and it can be implemented similarly even when using the single-carrier method, other multi-carrier methods, and spread spectrum communication methods. Furthermore, even if the number of receiving antennas is three or more, it does not affect this embodiment and can be implemented in the same way.

(Embodiment 4)

In Embodiment 2, the structure of the preamble of the three-transmission spatial multiplexing MIMO system is described. In the case of using the preamble structure shown in Embodiments 1 and 2, as the number of antennas increases, the interval between the existence of the reference symbol 302 becomes longer, so the estimation accuracy of the channel variation in the receiving device deteriorates. The odds will increase. In the present embodiment, a method of constructing a preamble for alleviating this problem is proposed.

FIG. 24 shows an example of the frame structure of this embodiment. The characteristic part in Fig. 24 is the structure of the preamble. The basic operation in the case of transmitting the frame signal in FIG. 24 is the same as that in the case of transmitting the frame signal in FIG. The antenna transmits and is multiplexed in space.

In FIG. 24 , at time 1, all of carrier 1 to carrier 12 of channel A are reference symbols 302 . Furthermore, at time 2, all of carrier 1 to carrier 12 of channel B are reference symbols 302, and at time 3, all of carrier 1 to carrier 12 of channel C are reference symbols 302.

In particular, FIG. 25 shows the detailed structure of the preambles of carrier 1 and carrier 2, time 1 and time 2. At time 1, for channel A in which all of carriers 1 to 12 are reference symbols 302, a normal OFDM modulation signal is generated. Also, for channel B and channel C where reference symbol 302 and guard symbol 301 exist in carrier 1 to carrier 12, special signal processing is performed after inverse Fourier transformer 106 in FIG. 14 . Similarly, for channel B at time 2, for channel C at time 3, and for channel A... at time 4, a normal OFDM modulation signal is generated. In addition, at time 2 for channel A and channel C, at time 3 for channel A and channel B, at time 4 for channel B and channel C, . . . after the inverse Fourier transform 106 of FIG.

In FIG. 25 , it is assumed that at time 1 and channel A of carrier 1, a signal point configuration of (I, Q)=(1, 0) is performed on the in-phase I-quadrature Q plane. At this time, the signal points of channel C are arranged such that (I, Q)=(0, 0) so as not to interfere with other channels. In addition, it is assumed that on channel B, the same signal point arrangement as channel A is performed. Then, the above-mentioned special signal processing is performed. The special signal processing is the processing of shifting (shifting) the phase of the modulated signal by, for example, 0.5 symbols with respect to the time axis for channel B. However, the signals that exceed the channel A without overlapping are arranged one by one from the beginning (folded back). For this operation, it is disclosed in the literature "Channel estimation for OFDM systems with transmission diversity in mobile wireless channels," IEEE Journal on SelectedAreas in Communications, vol.17, no.3, pp-461-471, March 1999, and "Simplified channel estimation for OFDM systems with multiple transmit antennas," IEEE Transaction on wireless communications, vol.1, no.1, pp.67-75, 2002, at time 1 and carrier 1, perform cyclic delay diversity on channel A and channel B (Cyclic Delay Diversity ).

Similarly, assuming that at time 1 and channel A of carrier 2, a signal point configuration of (I, Q)=(-1, 0) is performed on the in-phase I-orthogonal Q plane. At this time, the signal points of channel B are arranged such that (I, Q)=(0, 0) so as not to interfere with other channels. In addition, it is assumed that on channel C, the same signal point arrangement as channel A is performed. Then, the above-mentioned special signal processing is performed. This special signal processing is the processing of shifting the phase of the modulated signal obtained by Fourier transforming the channel C, for example, by 0.5 symbols with respect to the time axis. However, the signals that do not overlap with the channel A but exceed it are sequentially arranged from the beginning (folded back). Thus, at time 1 and carrier 2, cyclic delay diversity is performed for channel A and channel B.

Therefore, at time 1, the channel A, channel B, and channel C are alternately subjected to cyclic delay diversity with respect to the time axis. In this case, the channel estimation section of the receiving device can perform channel estimation for the channel on which cyclic delay diversity is performed by performing equalization processing. Therefore, in carrier 1 at time 1, the propagation path fluctuations of channel A and channel B can be estimated simultaneously, and in carrier 2 at time 1, the propagation path fluctuations of channel A and channel C can be estimated simultaneously. In this way, unlike the case of Embodiment 2, channel fluctuations corresponding to two channels can be estimated simultaneously, and thus the number of symbols of the preamble can be reduced. As a result, an improvement in the data transmission speed is achieved.

However, in principle, cyclic delay diversity capable of simultaneously estimating transmission path fluctuations corresponding to three channels is possible, but there are disadvantages such as a decrease in diversity gain and an increase in the circuit scale of the receiving device, so as in this embodiment, As shown in , it is more preferable to estimate the propagation path variation corresponding to two channels at the same time. For this, it is important that a guard symbol ((I, Q)=(0, 0)) must be inserted for one of the channels.

In addition, when focusing on time 1, although the phases of channel B and channel C are shifted, it is very important to match the amount of shifted phase (which can also be represented by the amount of symbols or time), for example, 0.5 symbol. This is because, in the receiving apparatus, a circuit for simultaneously estimating propagation path variations of channel A and channel B and a circuit for simultaneously estimating propagation path variations of channel A and channel C can be shared.

As described above, by arranging the preamble of the three-transmit spatial multiplexing MIMO system so that two channels of cyclic delay diversity are performed on different channels for each carrier, it is possible to estimate propagation path variation with high accuracy and to By reducing the number of preambles, the data transmission speed can be increased.

In this embodiment, although the case of three transmissions is described, it can also be applied to four or more antennas. In addition, in this embodiment, although an example using the OFDM method was described, it is not limited to this, and it can be implemented similarly even when using a single-carrier method, other multi-carrier methods, and a spread spectrum communication method.

In addition, although the above-mentioned Embodiment 3 described switching between two-transmission spatial multiplexing MIMO and three-transmission spatial multiplexing MIMO, it is not limited to this. For example, when switching one-system transmission (when MIMO is not performed), can also be implemented. The relationship between the communication method and the normalization coefficient at this time is shown in FIG. 26 .

(Embodiment 5)

In Embodiment 1, the case where the number of transmission antennas is two is described, but in this embodiment, the configuration of the pilot carrier when the number of transmission antennas is three is described.

FIG. 27 shows an example of a frame structure of a transmission signal formed by the transmission device according to this embodiment, and parts corresponding to those in FIG. 4 are denoted by the same reference numerals. (a) of FIG. 27 shows the frame structure of channel A, (b) of FIG. 27 shows the frame structure of channel B, and (c) of FIG. 27 shows the frame structure of channel C. As can be seen from FIG. 27 , pilot symbols (pilot carriers) 305 are arranged on carriers 3, 5, 8, and 10, except for the timing when reference symbols 302 and control symbols 304 are transmitted.

FIG. 28 shows the signal point arrangement of pilot symbols 305 of channel A, channel B and channel C and their characteristics. The feature in this case is that, similarly to Embodiment 1, signal points of channel A, channel B, and channel C of the same carrier are arranged to be orthogonal (cross-correlation is zero).

For example, the signal point configuration of carrier 3 of channel A from time 11 to time 14 (Figure 28(a)), the signal point configuration of carrier 3 of channel B from time 11 to time 14 (Figure 28(b)), and channel The signal point configuration from time 11 to time 14 of carrier 3 of C ( FIG. 28( c )) is orthogonal. The signal point arrangement is performed so that such orthogonality holds true even after time 15 . In this case, for orthogonality of signals, Walsh-Hadamard transform, orthogonal codes, and the like are suitably used. In addition, although the case of BPSK is shown in FIG. 28, as long as it is orthogonal, QPSK modulation may be used, and the regulation of the modulation method may not be followed.

In addition, in the case of this embodiment, in order to simplify the transmitting device and the receiving device, the same signal point configuration (same sequence) is used for the following carriers: carrier 3 of channel A ((a) in FIG. 28 ) and carrier 5 ((f) of Figure 28), carrier 3 of channel B ((b) of Figure 28 ), carrier 8 of channel B (28(h) of Figure 28), carrier 3 of channel C ((c) of Figure 28 ) and carrier 10 of channel A ((j) in Figure 28), carrier 5 of channel A ((d) in Figure 28 ), carrier 8 of channel C ((i) in Figure 28 ), carrier 5 of channel B ( (e) of FIG. 28 and carrier 10 of channel B ((k) of FIG. 28 ), and carrier 8 of channel A ((g) of FIG. 28 ) and carrier 10 of channel C ((l) of FIG. 28 ) . The reason for this will be described in detail using FIGS. 30 , 31 , 32 and 33 . Wherein, the same sequence refers to exactly the same signal point configuration. Here, as shown in FIG. 28, the respective sequences are named sequence #1, sequence #2, sequence #3, sequence #4, sequence #5, and sequence #6.

In addition, in the carriers 3, 5, 8, and 10 of channel A (or channel C), the signal point configurations of the pilot symbols 305 are different (different sequences), because if they are the same (the same sequence is used) , it may lead to an increase in the peak transmission power. However, in this embodiment, channel B is an example that does not satisfy this condition.

Here, the simplification of the transmission device and the reception device and the necessity of orthogonality will be described using FIGS. 29 , 30 , 31 , 32 , and 33 .

FIG. 29 shows an example of the configuration of the MIMO-OFDM transmission device according to this embodiment. In FIG. 29 , the same reference numerals as those in FIG. 14 are assigned to parts that perform the same operations as those in FIG. 14 . In MIMO-OFDM transmission apparatus 2800, mapping unit 2802 takes transmission data 2801 and frame structure signal 113 as input, and outputs channel A baseband signal 103A, channel B baseband signal 103B, and channel C baseband signal 103C. Since the same operation as that described in Embodiment 1 or Embodiment 2 is performed thereafter, description thereof will be omitted.

FIG. 30 shows an example of a detailed configuration of mapping section 2802 in FIG. 29 . Data modulation section 2902 receives transmission data 2801 and frame signal 113 as input, outputs modulated signal 2903 of data symbol 303 based on frame signal 113 , and outputs it.

The preamble mapping unit 2904 takes the frame structure signal 113 as input, generates a preamble modulated signal 2905 according to the frame structure signal 113, and outputs it.

Sequence #1 storage unit 2906 outputs signal 2907 of sequence #1 in FIG. 28 . Sequence #2 storage unit 2908 outputs signal 2909 of sequence #2 in FIG. 28 . Sequence #3 storage unit 2910 outputs signal 2911 of sequence #3 in FIG. 28 . Sequence #4 storage unit 2912 outputs signal 2913 of sequence #4 in FIG. 28 . Sequence #5 storage unit 2914 outputs signal 2915 of sequence #5 in FIG. 28 . Sequence #6 storage unit 2916 outputs signal 2917 of sequence #6 in FIG. 28 .

Pilot symbol mapping section 2918 maps sequence #1 signal 2907, sequence #2 signal 2909, sequence #3 signal 2911, sequence #4 signal 2913, sequence #5 signal 2915, sequence #6 signal 2917 and The frame structure signal 113 is input, and the modulated signal 2919 based on the pilot symbol 305 of the frame structure signal 113 is generated and output.

The signal generation unit 2920 takes the modulated signal 2903 of the data symbol 303, the modulated signal 2905 of the preamble, and the modulated signal 2919 of the pilot symbol 305 as input, and outputs the modulated signal 103A of the channel A, the modulated signal 103B of the channel B and the channel Modulation signal 103C of C.

In the structure of Figure 30, only six sequential memory cells are required. This is because, in the present invention, a certain sequence is used for two or more subcarriers as shown in FIG. 28 (use of two subcarriers in FIG. 28 ). Accordingly, the circuit scale of the transmission device can be reduced. On the other hand, unlike FIG. 28 , when mutually different sequences are used, twelve sequence memory cells are required, and the circuit scale increases.

In addition, mapping section 2802 in FIG. 29 may also be configured as shown in FIG. 31 . In FIG. 31 , parts that perform the same operations as those in FIG. 30 are given the same reference numerals as in FIG. 30 . The code #1 storage unit 3001 stores "1, 1, -1, -1", and the code #2 storage unit 3003 stores "1, -1, 1, -1". Pilot symbol mapping section 2918 takes signal 3002 of pattern #1, signal 3004 of pattern #2 output from code #1 storage section 3001 and code #2 storage section 3003, and frame structure signal 113 as input, and outputs a pilot symbol. Modulated signal 2920 of symbol 305.

At this time, it can be seen from FIG. 28 that there are only two basic patterns of signals. Pilot symbol mapping section 2918 generates six kinds of sequences #1 to #6 from two basic patterns by shifting the codes using a shift register. Therefore, as shown in FIG. 31, only two memory cells can be used.

As mentioned above, from FIG. 29, FIG. 30, and FIG. 31, it can be seen that by configuring the pilot carrier as shown in FIG. 28, the configuration of the transmission device can be simplified.

Next, the sending device will be described. FIG. 15 is an example of the configuration of a receiving device. Hereinafter, the configuration of frequency offset and phase noise estimating section 213 in FIG. 15 will be described in detail using FIG. 32 and FIG. 33 .

FIG. 32 shows an example of the configuration of frequency offset and phase noise estimating section 213 in FIG. 15 according to this embodiment. The frequency offset and phase noise estimation unit 213 in FIG. 32 includes: a pilot subcarrier extraction unit 3101, sequence storage units 3108_1~3108_6, sequence selection unit 3110, frequency offset and phase noise estimation unit 3123_#3 of carrier 3, carrier 5 frequency offset and phase noise estimation unit 3123_#5, carrier 8 frequency offset and phase noise estimation unit 3123_#8, and carrier 10 frequency offset and phase noise estimation unit 3123_#10.

Pilot subcarrier extraction section 3101 receives Fourier-transformed signal 206X (or 206Y, 206Z) as input, and extracts the subcarrier of pilot symbol 305 . Specifically, the signals of carriers 3, 5, 8 and 10 are extracted. Therefore, pilot subcarrier extracting section 3101 outputs baseband signal 3102_#3 of carrier 3, baseband signal 3102_#5 of carrier 5, baseband signal 3102_#8 of carrier 8, and baseband signal 3102_#10 of carrier 10.

Sequence #1 storage unit 3108_1 stores sequence #1 in FIG. 28 , and outputs signal 3109_1 of sequence #1 based on timing signal 212 . Sequence #2 storage unit 3108_2 stores sequence #2 in FIG. 28 , and outputs signal 3109_2 of sequence #2 based on timing signal 212 . Sequence #3 storage unit 3108_3 stores sequence #3 in FIG. 28 , and outputs signal 3109_3 of sequence #3 based on timing signal 212 . Sequence #4 storage unit 3108_4 stores sequence #4 in FIG. 28 , and outputs signal 3109_4 of sequence #4 based on timing signal 212 . Sequence #5 storage unit 3108_5 stores sequence #5 in FIG. 28 , and outputs signal 3109_5 of sequence #5 based on timing signal 212 . Sequence #6 storage unit 3108_6 stores sequence #6 in FIG. 28 , and outputs signal 3109_6 of sequence #6 based on timing signal 212 .

The sequence selection unit 3110 converts the signal 3109_1 of the sequence #1, the signal 3109_2 of the sequence #2, the signal 3109_3 of the sequence #3, the signal 3109_4 of the sequence #4, the signal 3109_5 of the sequence #5, the signal 3109_6 of the sequence #6, and the timing signal 212 as input and assign sequence #5 to signal 3111, sequence #1 to signal 3112, sequence #4 to signal 3113, sequence #3 to signal 3114, sequence #6 to signal 3115, Assign sequence #5 to signal 3116, sequence #2 to signal 3117, sequence #1 to signal 3118, sequence #3 to signal 3119, sequence #4 to signal 3120, sequence #6 to to signal 3121, and sequence #2 is assigned to signal 3122, and output.

The frequency offset and phase noise estimation unit 3123_#3 of carrier 3 includes: code multiplication units 3103A, 3103B and 3103C, and phase variation estimation units 3105A, 3105B and 3105C, which perform frequency offset and phase noise of each channel of carrier 3 estimate.

The code multiplication unit 3103A takes the baseband signal 3102_#3 of the carrier 3 and the signal 3111 of the sequence #5 as input, and multiplies the baseband signal 3102_#3 of the carrier 3 and the signal 3111 of the sequence #5 to generate the signal 3111 of the channel A of the carrier 3. Baseband signal 3104A_#3, and output it. The reason for this is as follows.

The baseband signal 3102_#3 of carrier 3 is a multiplexed signal of the baseband signal of channel A, the baseband signal of channel B and the baseband signal of channel C. When the multiplexed signal is multiplied by the signal 3111 of sequence #5, the baseband signal components of channel B and channel C with zero cross-correlation are removed, so that only the baseband signal component of channel A can be extracted.

Phase fluctuation estimation section 3105A receives baseband signal 3104A_#3 of channel A of carrier 3 as input, estimates phase fluctuation based on the signal, and outputs phase fluctuation estimation signal 3106A_#3 of channel A.

Similarly, the code multiplication unit 3103B takes the baseband signal 3102_#3 of the carrier 3 and the signal 3112 of the sequence #1 as input, and multiplies the baseband signal 3102_#3 of the carrier 3 and the signal 3112 of the sequence #1 to generate the signal 3112 of the carrier 3 The baseband signal 3104B_#3 of channel B is output. Furthermore, the code multiplication unit 3103C takes the baseband signal 3102_#3 of the carrier 3 and the signal 3113 of the sequence #4 as input, and multiplies the baseband signal 3102_#3 of the carrier 3 and the signal 3113 of the sequence #4 to generate the channel of the carrier 3 C's baseband signal 3104C_#3, and output it.

The phase variation estimating units 3105B and 3105C respectively take the baseband signal 3104B_#3 of channel B of carrier 3 and the baseband signal 3104C_#3 of channel C of carrier 3 as input, estimate the phase variation based on these signals, and output the phase of channel B, respectively. Fluctuation estimation signal 3106B_#3 and channel C phase fluctuation estimation signal 3106C_#3.

For the frequency offset and phase noise estimation unit 3123_#5 of carrier 5, only the signal to be processed is different. The phase variation estimation signal 3106A_#5 of channel A, the phase variation estimation signal 3106B_#5 of channel B, and the phase variation estimation signal 3106C_#5 of channel C of carrier 5. For the frequency offset and phase noise estimation unit 3123_#8 of carrier 8, only the signal to be processed is different, and it performs the same operation as the above-mentioned frequency offset and phase noise estimation unit 3123_#3 of carrier 3, and outputs The phase variation estimation signal 3106A_#8 of channel A, the phase variation estimation signal 3106B_#8 of channel B, and the phase variation estimation signal 3106C_#8 of channel C of carrier 8. Furthermore, the frequency offset and phase noise estimating unit 3123_#10 of the carrier 10 is only different in the signal to be processed, and performs the same operation as the frequency offset and phase noise estimating unit 3123_#3 of the carrier 3 described above. , and output phase variation estimation signal 3106A_#10 of channel A, phase variation estimation signal 3106B_#10 of channel B, and phase variation estimation signal 3106C_#10 of channel C of carrier 10.

As described above, by making the signals of channel A, channel B, and channel C of the same carrier orthogonal, even if the pilot symbol 305 is multiplexed on channel A, channel B, and channel C, high-precision frequency offset can be performed and phase noise estimates. As another important advantage, since channel estimation values (transmission path variation estimation values) are not required, the configuration of a part that compensates for frequency offset and phase noise can be simplified. If the signal point configurations of the pilot symbols 305 of channel A, channel B, and channel C are not orthogonal, then it becomes the following structure: perform signal processing for MIMO separation, such as ZF, MMSE, and MLD, and then Estimate frequency offset and phase noise. On the other hand, according to the present embodiment, as shown in FIG. 15 , it is possible to compensate for frequency offset and phase noise at a stage preceding signal separation (signal processing section 223 ). Furthermore, in the signal processing unit 223, even after separation into the signal of channel A, the signal of channel B, and the signal of channel C, the frequency offset and phase noise can be removed by using the pilot symbol 305, so that higher precision Compensation of frequency offset and phase noise.

However, in the case where the signal point configurations of channel A, channel B, and channel C of the same carrier are not orthogonal, the frequency offset and phase noise estimation accuracy of the frequency offset and phase noise estimation unit 213 of FIG. 15 decreases ( become interference components of others), it is difficult to add the frequency offset and phase noise compensation unit 215 shown in FIG. 15, and high-precision frequency offset and phase noise compensation cannot be performed.

In addition, as a structure different from the structure of FIG. 32, the structure of FIG. 33 is conceivable. The difference between FIG. 33 and FIG. 32 is that the sequence storage units 3108_1 to 3108_6 are replaced by code storage units 3201_1 and 3201_2. The code #1 storage unit 3201_1 stores "1, 1, -1, -1"; the code #2 storage unit 3201_2 stores "1, -1, 1, -1". Code selection section 3203 shifts two basic codes input from code storage sections 3201_1 and 3201_2 using a shift register to generate six types of sequences #1 to #6. Thereby, only two memory cells can be configured, and the structure corresponding to this part can be simplified compared with the structure of FIG. 32 . Also, this is possible because pilot symbols 305 of the same sequence are allocated to multiple channels and/or multiple carriers.

As described above, by using the same sequence of pilot symbols 305 on multiple channels and/or multiple carriers, the sequence storage units 3108_1 to 3108_6 in FIG. 32 can be implemented among multiple channels and/or multiple carriers. Sharing, or sharing of the sequence storage units 3201_1 and 3201_2 in FIG. 33 , brings about simplification of the receiving device.

In addition, in the above, as shown in FIG. 27, it has been explained that the pilot symbol (pilot carrier) 305 used to estimate the distortion caused by frequency offset or phase noise, etc., is allocated to a specific subcarrier. In the following, The frame structure of pilot symbols 305 that differs from that of FIG. 27 is discussed.

34, 35 and 36 show examples of frame structures of transmission signals different from those in FIG. 27 .

In FIG. 34, the pilot symbol 305 is arranged at a specific time of a specific carrier.

In addition, (a) of FIG. 34 shows the frame structure of channel A, (b) of FIG. 34 shows the frame structure of channel B, and (c) of FIG. 34 shows the frame structure of channel C. In the example of FIG. 34 , where channel A, channel B, and channel C are multiplexed with pilot symbols 305, pilot symbol sequences having an orthogonal relationship between channels of the same carrier are used, and repeatedly used A sequence of pilot symbols of the same sequence. Also, on channel A, pilot symbols 305 of different sequences are used for different subcarriers. That is to say, in the example of FIG. 34, from time 6 to time 9, the same mapping as the pilot from time 11 to time 14 in FIG. 28 is performed, and next, from time 12 to time 15 in FIG. Mapping is performed by the same rules as time 11 to time 14 in FIG. 28 . Therefore, if the frame structure of FIG. 34 is used under the same conditions as above, the same effect as above can be obtained.

In FIG. 35 , pilot symbols 305 are arranged on a plurality of consecutive subcarriers at a specific time. At this time, for example, time 6 of channel A, pilot symbol sequence of carrier 2 to carrier 5, time 6 of channel B, pilot symbol sequence of carrier 2 to carrier 5, and time 6 of channel C, carrier 2 The pilot symbol sequence to carrier 5 is orthogonal. Similarly, time 6 of channel A, pilot signal (pilot symbol) 305 of carrier 8 to carrier 11, time 6 of channel B, pilot signal of carrier 8 to carrier 11, and time 6 of channel C, carrier The pilot signal 305 of 8 to carrier 11 is orthogonal.

Also, time 12 for channel A, pilot signal from carrier 2 to carrier 5, time 12 for channel B, pilot signal from carrier 2 to carrier 5, and time 12 for channel C, pilot signal from carrier 2 to carrier 5 Orthogonal. Similarly, time 12 of channel A, pilot signal of carrier 8 to carrier 11, time 12 of channel B, pilot signal of carrier 8 to carrier 11, and time 12 of channel C, pilot signal of carrier 8 to carrier 11 The signals are in quadrature.

Also, for example, if the same sequence is used for the pilot signals of carrier 2 to carrier 5 at time 6 of channel A, and carrier 8 to carrier 11 of time 6 of channel C, the same sequence is used for others as shown above in the same way. , the circuit scale can be reduced, and the same effect as that in the case of FIG. 27 can be obtained. However, although a plurality of consecutive subcarriers have been described as an example, the present invention is not limited to this, and the same effect can be obtained even if the pilot symbols 305 are allocated discretely to the subcarriers without losing the orthogonality. Furthermore, as shown in FIG. 36, the same effect can be obtained even if allocation is performed on both the time axis and the frequency axis. In any case, as long as the pilot symbols 305 are allocated in the direction of the frequency axis or the time axis without losing the orthogonality, the same effect as above can be obtained.

In this embodiment, the pilot symbols 305 orthogonal to each other in units of four symbols have been described as an example, but it is not limited to units of four symbols. However, considering the impact on orthogonality due to fading fluctuations in the time direction and/or frequency direction, it can be considered that if an orthogonal pattern is formed with about 2 to 8 symbols, it is possible to ensure Estimation accuracy of frequency offset and phase noise. If the period of the orthogonal pattern is too long, the possibility that the orthogonality cannot be ensured increases, and the estimation accuracy of frequency offset and phase noise deteriorates.

Important aspects of the method of constructing the pilot symbol 305 in this embodiment are as follows.

●The pilot signals of channel A, channel B and channel C of the same carrier are orthogonal

● There are channels using different sequences in different carriers on which pilot signals are arranged

● There are more than two channels using the same sequence (for example, a certain sequence is used by both antenna A and antenna B, that is, a certain sequence is shared by different multiple antennas)

As a result, when MIMO-OFDM transmission is performed using three transmission antennas, it is possible to realize a transmission device with a simple configuration that suppresses an increase in transmission peak power without deteriorating frequency offset and phase noise estimation accuracy. In addition, although selecting the pilot signal to form the pilot carrier is the best condition to satisfy all of the above three conditions, for example, if you only want to obtain a part of the above effects, for example, you can also select The pilot signal is used to form the pilot carrier and only two of the above three conditions are satisfied.

In the present embodiment, an example using the OFDM method was described, but it is not limited to this, and it can be implemented similarly even when using the single-carrier method, other multi-carrier methods, and spread spectrum communication methods. In addition, in this embodiment, although the case where three antennas are provided for transmission and reception has been described as an example, it is not limited thereto. In addition, the implementation in the case of other numbers of antennas and other transmission methods will be described in detail later. In addition, although the pilot symbols, reference symbols, guard symbols, and preambles are named and described here, using other calling methods will not have any influence on this embodiment. This is also the same in other embodiments. In addition, in the embodiment, terms such as channel A, channel B, and channel C are used for description, but these are for ease of description, and even if other calling methods are used, the embodiment will not be affected at all.

(Embodiment 6)

In Embodiment 1, when describing the structure of a pilot symbol, words such as a pattern are used for description, but in this embodiment, as in Embodiment 5, the description of Embodiment 1 is described using a word like a sequence. That is, the present embodiment is the same as that of the first embodiment in terms of the basic idea and the basic structure.

FIG. 2 shows the configuration of the MIMO-OFDM transmission device 100 according to this embodiment. However, FIG. 2 shows the case where the number of transmission antennas m=2 as an example.

The frame structure signal generating section 112 receives control information 111 such as a modulation method as input, generates a frame structure signal 113 including information on the frame structure, and outputs it.

Mapping section 102A receives transmission digital signal 101A of channel A and frame structure signal 113 as input, generates frame structure based baseband signal 103A, and outputs it.

Serial-to-parallel conversion unit 104A takes baseband signal 103A and frame structure signal 113 as input, performs serial-to-parallel conversion based on frame structure signal 113 , and outputs parallel signal 105A.

Inverse Fourier transform unit 106A takes parallel signal 105A as input, performs inverse Fourier transform, and outputs inverse Fourier transformed signal 107A.

Wireless unit 108A receives inverse Fourier-transformed signal 107A as input, performs processing such as frequency conversion, and outputs transmission signal 109A. Transmission signal 109A is output as radio waves through antenna 110A.

MIMO-OFDM transmission apparatus 100 also performs the same processing as channel A on channel B, thereby generating channel B transmission signal 109B. In addition, the element indicated by adding "B" at the end of the reference numerals is the part related to channel B, but the signal as the object is not channel A but channel B, which is basically the same as that of adding "B" at the end of the above reference numerals. A" is the same process as the part related to channel A.

FIG. 3A shows an example of the configuration of a receiving device according to this embodiment. However, FIG. 3A shows a case where the number of receiving antennas is n=2 as an example.

In receiving device 200 , wireless unit 203X takes received signal 202X received via receiving antenna 201X as input, performs processing such as frequency conversion, and outputs baseband signal 204X.

The Fourier transform unit 205X takes the baseband signal 204X as input, performs Fourier transform, and outputs a Fourier transformed signal 206X.

The same operation is performed at the receiving antenna 201Y. The synchronization unit 211 takes the baseband signals 204X and 204Y as input, for example, by detecting reference symbols, establishes time synchronization with the transmitting device, and outputs a timing signal 212 . The structure of the reference symbol and the like will be described later in detail using FIG. 4 and the like.

Frequency offset and phase noise estimation unit 213 takes the signals 206X and 206Y after Fourier transform as input, extracts pilot symbols, estimates frequency offset and phase noise according to pilot symbols, and outputs phase distortion estimation signal 214 (including phase distortion of the frequency offset). The structure of the pilot symbol and the like will be described in detail later using FIG. 4 and the like.

The transmission path variation estimation unit 207A of channel A takes the Fourier transformed signal 206X as input, extracts the reference symbols of channel A, for example, estimates the transmission path variation of channel A according to the reference symbols, and outputs the transmission path estimation signal 208A of channel A .

The transmission path variation estimation unit 207B of channel B takes the signal 206X after Fourier transform as input, extracts the reference symbol of channel B, for example, estimates the transmission path variation of channel B according to the reference symbol, and outputs the transmission path estimation signal of channel B 208B.

For the channel A channel variation estimation unit 209A and the channel B channel variation estimation unit 209B, only the target signal is not the signal received by the antenna 201X but the signal received by the antenna 201Y. The above channel A channel variation estimation unit 207A and channel B channel variation estimation unit 207B perform the same processing.

Frequency offset and phase noise compensation unit 215 takes channel A transmission path estimation signals 208A and 210A, channel B transmission path estimation signals 208B and 210B, Fourier-transformed signals 206X and 206Y, and phase distortion estimation signal 214 as input, The phase distortion of each signal is removed, and the phase-compensated transmission path estimation signals 220A and 222A of channel A, the phase-compensated transmission path estimation signals 220B and 222B of channel B, and the phase-compensated Fourier-transformed signal 221X are output and 221Y.

For example, the signal processing unit 223 performs an inverse matrix operation, and outputs a baseband signal 224A of channel A and a baseband signal 224B of channel B. Specifically, as shown in FIG. 3B , if, for example, in a certain subcarrier, let the transmitted signal from the antenna AN1 be Txa(t), the transmitted signal from the antenna AN2 be Txb(t), and the received signal from the antenna AN3 be The received signal of R1(t) and antenna AN4 is R2(t), and the transmission path changes are respectively set to h11(t), h12(t), h21(t) and h22(t), then the equation (1) relationship is established.

Among them, t is time, n1(t) and n2(t) are noises. The signal processing unit 223 obtains the signal of channel A and the signal of channel B by using formula (1), for example, by performing an inverse matrix operation. The signal processing unit 223 performs this calculation for all subcarriers. Also, estimation of h11(t), h12(t), h21(t), and h22(t) is performed by channel variation estimation sections 207A, 209A, 207B, and 209B.

The frequency offset estimation and compensation unit 225A takes the baseband signal 224A of channel A as an input, extracts pilot symbols, estimates and compensates the frequency offset of the baseband signal 224A based on the pilot symbols, and outputs the frequency offset compensated baseband Signal 226A.

Channel A demodulation unit 227A takes frequency offset compensated baseband signal 226A as input, demodulates the data symbols, and outputs received data 228A.

The MIMO-OFDM receiving apparatus 200 also performs the same processing on the baseband signal 224B of the channel B, thereby obtaining received data 228B.

FIG. 4 shows the frame configurations of channel A ((a) in FIG. 4 ) and channel B ((b) in FIG. 4 ) in time-frequency according to this embodiment. Signals of the same time and the same carrier in (a) of FIG. 4 and (b) of FIG. 4 are spatially multiplexed.

From time 1 to time 8, symbols for estimating transmission path variations equivalent to h11(t), h12(t), h21(t) and h22(t) of equation (1) are sent, such as called the preamble. The preamble is composed of a guard symbol 301 and a reference symbol 302 . Let the guard symbol 301 be (0, 0) on the in-phase I-orthogonal Q plane. The reference symbol 302 is, for example, a symbol with known coordinates other than (0, 0) on the in-phase I-orthogonal Q plane. In addition, channel A and channel B are configured so as not to interfere with each other. That is to say, for example, like carrier 1, time 1, when guard symbol 301 is configured in channel A, reference symbol 302 is configured in channel B; like carrier 2, time 1, in reference symbol 302 When channel A is placed, guard symbol 301 is placed on channel B, and different symbols are placed on channel A and channel B in this way. By configuring in this way, for example, when focusing on channel A at time 1, it is possible to estimate the channel variation of carrier 3 from the reference symbols 302 of carrier 2 and carrier 4 . Since the carrier 2 and the carrier 4 are the reference symbols 302, it is possible to estimate channel variation. Therefore, in time 1, it is possible to accurately estimate propagation channel fluctuations of all carriers of channel A. Similarly, it is also possible to estimate propagation path fluctuations of all carriers on channel B with high accuracy. For time 2 to time 8, it is also possible to estimate channel fluctuations of all the carriers of channel A and channel B. FIG. Therefore, with regard to the frame structure of FIG. 4 , since it is possible to estimate propagation path fluctuations of all carriers in all the times from time 1 to time 8, it can be said that it is possible to realize extremely accurate estimation of propagation path fluctuations. code structure.

In FIG. 4, an information symbol (data symbol) 303 is a symbol for data transmission. Here, it is assumed that the modulation schemes are BPSK, QPSK, 16QAM, and 64QAM. The signal point arrangement and the like on the in-phase I-quadrature Q plane at this time will be described in detail using FIG. 5 .

The control symbol 304 is a symbol for transmitting control information such as a modulation method, an error correction coding method, and a coding rate.

Pilot symbols 305 are symbols for estimating phase fluctuations due to frequency offset and phase noise. As the pilot symbol 305, for example, a symbol with known coordinates on the in-phase I-orthogonal Q plane is used. Pilot symbols 305 are configured on carrier 4 and carrier 9 on both channel A and channel B. In this way, especially in a wireless LAN, when a system of the same frequency and the same frequency band is constructed by IEEE802.11a, IEEE802.11g and a spatially multiplexed MIMO system, the frame structure can be shared, and thus the reception device can be simplified. .

FIG. 5 shows the signal point arrangement on the in-phase I-quadrature Q plane of BPSK, QPSK, 16QAM, and 64QAM, which are the modulation schemes of the information symbol 303 in FIG. 4, and their normalization coefficients.

FIG. 5A shows the signal point configuration of BPSK on the in-phase I-quadrature Q plane, and its coordinates are shown in FIG. 5A. FIG. 5B shows the signal point configuration of QPSK on the in-phase I-orthogonal Q plane, and its coordinates are shown in FIG. 5B. FIG. 5C shows the signal point configuration of 16QAM on the in-phase I-quadrature Q plane, and its coordinates are shown in FIG. 5C. FIG. 5D shows the signal point configuration of 64QAM on the in-phase I-quadrature Q plane, and its coordinates are shown in FIG. 5D. FIG. 5E shows the relationship between modulation schemes and multiplication coefficients (that is, normalization coefficients) for correcting the signal point arrangement of FIGS. 5A to 5D so as to keep the average transmission power constant among modulation schemes. For example, in the case of transmitting with the QPSK modulation scheme shown in FIG. 5B , as can be seen from FIG. 5E , it is necessary to multiply the coordinates in FIG. 5B by the value of 1/sqrt(2). Among them, sqrt(x) is the square root of x (square root of x).

FIG.6 shows the arrangement on the in-phase I-orthogonal Q plane of the pilot symbol 305 in FIG.4 in this embodiment. (a) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 4 of channel A shown in FIG. 4( a ). (b) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 4 of channel B shown in FIG. 4( b ). (c) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 9 of channel A shown in (a) of FIG. 4 . (d) of FIG. 6 shows an example of signal point arrangement of pilot symbols 305 from time 11 to time 18 of carrier 9 of channel B shown in (b) of FIG. 4 . Here, these configurations use BPSK modulation, but are not limited thereto.

The feature of the signal point configuration of the pilot symbol 305 in FIG. 6 is that the signal point configurations of channel A and channel B of the same carrier are orthogonal (the cross-correlation is zero).

For example, the signal point configuration of carrier 4 of channel A from time 11 to time 14 is orthogonal to the signal point configuration of carrier 4 of channel B from time 11 to time 14. Furthermore, the same applies to time 15 to time 18 . In addition, the signal point configuration of carrier 9 of channel A from time 11 to time 14 is also orthogonal to the signal point configuration of carrier 9 of channel B from time 11 to time 14. Furthermore, the same applies to time 15 to time 18 . In this case, for orthogonality of signals, Walsh-Hadamard transform, orthogonal codes, and the like are suitably used. In addition, although the case of BPSK is shown in FIG. 6 , as long as it is orthogonal, QPSK modulation may be used, and the regulation of the modulation method may not be followed. Moreover, in the case of this embodiment, in order to simplify the receiving device, it is assumed that carrier 4 of channel A and carrier 9 of channel B, and carrier 9 of channel A and carrier 4 of channel B are configured with the same signal point (the same pattern). However, the same pattern does not use exactly the same signal point configuration. For example, on the in-phase I-quadrature Q plane, only in the case of a different phase relationship, they can also be regarded as the same pattern.

Here, as defined in Embodiment 5, assuming that "the same sequence refers to exactly the same signal point configuration", in order to simplify the receiving device, between carrier 4 of channel A and carrier 9 of channel B, and carrier 9 of channel A It is only necessary to set the same signal point configuration as carrier 4 of channel B, that is, set the same sequence.

In addition, the signal point arrangement of the pilot symbol 305 is made different between the carriers 4 and 9 of the channel A (or channel B), because if they are set to be the same, the transmission peak power may increase.

Here, first, the advantage of orthogonality will be described in detail using FIG. 3A and FIG. 37 .

FIG. 37 shows an example of the configuration of frequency offset and phase noise estimating section 213 in FIG. 3A . Pilot carrier extraction section 602 receives Fourier-transformed signal 206X (or 206Y) as input, and extracts the subcarrier of pilot symbol 305 . Specifically, the signals of carrier 4 and carrier 9 are extracted. Therefore, the pilot carrier extraction unit 602 outputs the baseband signal 603 of the carrier 4 and the baseband signal 604 of the carrier 9 .

For example, sequence #1 storage unit 3601 stores sequence #1 of “1, -1, 1, -1” in FIG. 6 , and outputs signal 3602 of sequence #1 according to timing signal 212 .

The sequence #2 storage unit 3603 stores, for example, the sequence #2 of “1, 1, -1, -1” in FIG. 6 , and outputs the signal 3604 of the sequence #2 according to the timing signal 212 .

The selection unit 609 takes the timing signal 212 , the signal 3602 of the sequence #1 and the signal 3604 of the sequence #2 as input, outputs the signal of the sequence #2 as the selection signal 610 , and outputs the signal of the sequence #1 as the selection signal 611 .

Code multiplying unit 612A takes carrier 4 baseband signal 603 and selection signal 611 as input, multiplies carrier 4 baseband signal 603 and selection signal 611 to generate carrier 4 channel A baseband signal 613A, and outputs it. The reason for this is as follows.

The baseband signal 603 of carrier 4 is a signal in which the baseband signal of channel A and the baseband signal of channel B are multiplexed. In contrast, by multiplying the signal of sequence #1 which is the selection signal 611, the baseband signal component of channel B with zero cross-correlation is removed, so that only the baseband signal component of channel A can be extracted.

Similarly, the code multiplication unit 614A takes the baseband signal 604 of the carrier 9 and the selection signal 610 as input, and multiplies the baseband signal 604 of the carrier 9 and the selection signal 610 to generate the baseband signal 615A of the channel A of the carrier 9, and converts it to output.

The code multiplication unit 612B takes the carrier 4 baseband signal 603 and the selection signal 610 as input, multiplies the carrier 4 baseband signal 603 and the selection signal 610 to generate a carrier 4 channel B baseband signal 613B, and outputs it.

Code multiplication unit 614B takes carrier 9 baseband signal 604 and selection signal 611 as input, multiplies carrier 9 baseband signal 604 and selection signal 611 to generate carrier 9 channel B baseband signal 615B, and outputs it.

As described above, by making the signal point arrangement of channel A and channel B of the same carrier orthogonal, even if the pilot symbol 305 is multiplexed on channel A and channel B, it is possible to perform frequency offset and phase noise estimation with high precision. estimate. As another important advantage, since channel estimation values (transmission path variation estimation values) are not required, the configuration of a part that compensates for frequency offset and phase noise can be simplified. If the signal point configurations of the pilot symbols 305 of channel A and channel B are not orthogonal to each other, the following configuration is performed: performing signal processing for MIMO separation (for example, ZF, MMSE, and MLD), and then estimating frequency offset and phase noise. On the other hand, according to the configuration of the present embodiment, as shown in FIG. 3A , it is possible to compensate for frequency offset and phase noise at a stage preceding signal separation (signal processing section 223 ). Furthermore, in the signal processing unit 223, even after the signal of the channel A and the signal of the channel B are separated, the frequency offset and the phase noise can be removed by the pilot symbol 305, so it is possible to perform frequency offset and phase noise with higher precision. Compensation for phase noise.

However, when the signal point configurations of channel A and channel B of the same carrier are not orthogonal, the frequency offset and phase noise estimation accuracy of the frequency offset and phase noise estimation section 213 in FIG. interference component), it is difficult to add the frequency offset and phase noise compensation unit 215 in FIG. 3A , so that high-precision frequency offset and phase noise compensation cannot be performed.

Furthermore, according to this embodiment, by setting carrier 4 of channel A and carrier 9 of channel B, and carrier 9 of channel A and carrier 4 of channel B, to the same signal point configuration (same sequence), the The shared use of the sequence storage units 3601 and 3603 of 37 simplifies the receiving device.

However, in this embodiment, it is necessary to make the signal point arrangements of channel A and channel B of the same carrier orthogonal, but it is not necessarily necessary to set them to the same sequence.

In the present embodiment, the pilot symbols 305 orthogonal to each other in four-symbol units, such as time 11 to time 14, have been described as an example, but it is not limited to four-symbol units. However, considering the influence on orthogonality caused by fading fluctuations in the time direction, it is considered that if an orthogonal pattern is formed with about 2 to 8 symbols, the frequency offset and phase Noise estimation accuracy. If the period of the orthogonal pattern is too long, the possibility that the orthogonality cannot be ensured increases, and the estimation accuracy of frequency offset and phase noise deteriorates.

FIG. 38 shows an example of the configuration of mapping section 102A ( 102B) of the transmission device in FIG. 2 according to this embodiment. The data modulation unit 1103 takes the transmission digital signal 101A (101B) and the frame structure signal 1102 as input, and modulates the transmission digital signal 101A (101B) based on the information and timing of the modulation method contained in the frame structure signal 1102, and outputs Modulated signal 1104 of data symbol 303 .

The preamble mapping unit 1105 takes the frame structure signal 1102 as input, and outputs a preamble modulation signal 1106 based on the frame structure.

The sequence #1 storage unit 3701 outputs a signal 3702 of the sequence #1. Likewise, sequence #2 storage unit #3703 outputs sequence #2 signal 3704.

Pilot symbol mapping section 1111 receives sequence #1 signal 3702, sequence #2 signal 3704, and frame structure signal 1102 as input, generates modulated signal 1112 of pilot symbol 305, and outputs it.

The signal generating unit 1113 takes the modulated signal 1104 of the data symbol 303, the modulated signal 1106 of the preamble and the modulated signal 1112 of the pilot symbol 305 as input, generates a baseband signal 103A (103B) conforming to the frame structure, and outputs it .

Although in the above description, it has been explained that if the structure of the pilot symbol 305 as shown in FIG. 4 and FIG. As for the meta structure, as shown in FIG. 38, the sequence storage units 3701 and 3703 can be shared in the transmission device, so that the transmission device can be simplified.

The method for generating the preamble and the pilot signal according to the present embodiment and the transmitting device for generating them have been described above, as well as the detailed configuration and operation of the receiving device for receiving the modulated signal according to the present embodiment. According to the present embodiment, since it is possible to improve the estimation accuracy of frequency offset, channel variation, and synchronization, the probability of signal detection can be improved, and the transmission device and the reception device can be simplified.

As in Embodiment 1, important aspects of the method of configuring pilot symbols 305 in this embodiment are as follows.

●The pilot signals of channel A and channel B of the same carrier are orthogonal

●In the same channel, use different sequences on different carriers configured with pilot signals

● Use the same sequence on each channel (channel A and channel B)

Thus, when MIMO-OFDM transmission is performed using two transmission antennas, it is possible to realize a transmission device with a simple configuration that suppresses an increase in transmission peak power without deteriorating frequency offset and phase noise estimation accuracy. In addition, although selecting the pilot signal to form the pilot carrier is the best condition to satisfy all of the above three conditions, for example, if you only want to obtain a part of the above effects, for example, you can also select The pilot signal is used to form the pilot carrier and only two of the above three conditions are satisfied.

In the present embodiment, an example using the OFDM method was described, but it is not limited to this, and it can be implemented similarly even when using the single-carrier method, other multi-carrier methods, and spread spectrum communication methods. In addition, in this embodiment, although the case where there are two antennas for transmission and reception has been described as an example, it is not limited to this. Even if the number of receiving antennas is three or more, this embodiment will not be affected, and the same can be achieved. implement. In addition, the frame structure is not limited to this embodiment. In particular, for the pilot symbol 305 used for estimating distortions such as frequency offset and phase noise, as long as the pilot symbol 305 is configured on a specific subcarrier and is transmitted through multiple antennas, The transmission configuration is sufficient, and the number of subcarriers for transmitting the pilot symbol 305 is not limited to two as in this embodiment. In addition, the implementation in the case of other numbers of antennas and other transmission methods will be described in detail later. Furthermore, although the pilot symbol 305, the reference symbol 302, the guard symbol 301, and the preamble are named and described here, using other calling methods has no influence on this embodiment. This is also the same in other embodiments. In addition, in the embodiment, terms such as channel A and channel B are used for description, but these are for ease of description, and even if other calling methods are used, the embodiment will not be affected at all.

Furthermore, although the frame structure has been described using the frame structure of FIG. 4 as an example, it is not limited thereto. In particular, although the case in which the pilot symbol 305 is arranged on a specific subcarrier has been described as an example, it is not limited thereto. ground configuration can also be implemented. However, the point is to adopt a configuration that ensures the orthogonality of the pilot signals.

(Embodiment 7)

In Embodiment 1 and Embodiment 6, when the number of transmission signals is 2 and the number of antennas is 2, the method of arranging pilot symbols 305 on two subcarriers is described. A method of arranging and transmitting pilot symbols 305 on four subcarriers will be described.

FIG. 39 shows an example of a frame structure of a transmission signal according to this embodiment, and parts corresponding to those in FIG. 4 are denoted by the same reference numerals. (a) of FIG. 39 shows the frame structure of channel A, and (b) of FIG. 39 shows the frame structure of channel B. As can be seen from FIG. 39 , pilot symbols (pilot carriers) 305 are arranged on carrier 3, carrier 5, carrier 8, and carrier 10, except when reference symbol 302 and control symbols are transmitted.

FIG.40 shows the signal point arrangement of pilot symbols 305 of channel A and channel B and their characteristics. The feature in this case is that, similarly to Embodiment 1, the signal point arrangements of channel A and channel B of the same carrier are orthogonal (cross-correlation is zero).

For example, the signal point configuration of carrier 3 of channel A from time 11 to time 14 ((a) in Figure 40), and the signal point configuration of carrier 3 of channel B from time 11 to time 14 ((b) of Figure 40) Orthogonal. The signal point arrangement is performed so that such orthogonality holds true even after time 15 . In this case, for orthogonality of signals, Walsh-Hadamard transform, orthogonal codes, and the like are suitably used. In addition, although the case of BPSK is shown in FIG. 40, as long as it is orthogonal, QPSK modulation may be used, and the regulation of the modulation method may not be followed.

In addition, in order to simplify the transmitting device and the receiving device, the same signal point configuration (same sequence) is used in the following carriers: carrier 3 of channel A ((a) in FIG. 40 ) and carrier 10 of channel B ((h) in FIG. 40 ), carrier 3 of channel B ((b) in Figure 40) and carrier 5 of channel A ((c) in Figure 40), carrier 5 of channel B ((d) in Figure 40) and carrier 8 of channel A ( (e) of FIG. 40 ), and carrier 8 of channel B ((f) of FIG. 40 ) and carrier 10 of channel A ((g) of FIG. 40 ). Wherein, the same sequence refers to exactly the same signal point configuration. Here, as shown in FIG. 40, the respective sequences are named sequence #1, sequence #2, sequence #3, and sequence #4.

In addition, in the carriers 3, 5, 8, and 10 of the channel A and the channel B, the signal point arrangement of the pilot symbol 305 is made different (different sequences), because if they are made the same (the same sequence is used), then It may lead to an increase in the transmission peak power.

Here, the simplification of the transmission device and the reception device and the necessity of orthogonality will be described using FIG. 41 , FIG. 42 , and FIG. 43 .

FIG. 41 shows an example of the configuration of the MIMO-OFDM transmission device according to this embodiment. In FIG. 41 , the parts that perform the same operations as those in FIGS. 2 and 29 are denoted by the same reference numerals. The MIMO-OFDM transmitting apparatus 400 in FIG. 41 differs from that in FIG. 29 in that there is no transmitting unit for channel C, and the other parts operate in the same manner as in FIG. 29 .

FIG. 42 shows an example of a detailed configuration of mapping section 2802 in FIG. 41 .

Data modulation section 2902 receives transmission data 2801 and frame signal 113 as input, generates modulation signal 2903 of data symbol 303 based on frame signal 113, and outputs it.

The preamble mapping unit 2904 takes the frame structure signal 113 as input, generates a preamble modulated signal 2905 according to the frame structure signal 113, and outputs it.

Sequence #1 storage unit 2906 outputs signal 2907 of sequence #1 in FIG. 40 . Sequence #2 storage unit 2908 outputs signal 2909 of sequence #2 in FIG. 40 . Sequence #3 storage unit 2910 outputs signal 2911 of sequence #3 in FIG. 40 . Sequence #4 storage unit 2912 outputs signal 2913 of sequence #4 in FIG. 40 .

The pilot symbol mapping section 2918 takes the signal 2907 of sequence #1, the signal 2909 of sequence #2, the signal 2911 of sequence #3, the signal 2913 of sequence #4, and the frame structure signal 113 as input, and generates a signal based on the frame structure signal 113 The modulated signal 2919 of the pilot symbol 305 is output.

Signal generator 2920 receives modulation signal 2903 of data symbol 303, modulation signal 2905 of preamble, and modulation signal 2919 of pilot symbol 305, and outputs modulation signal 103A of channel A and modulation signal 103B of channel B.

In the structure of Figure 42, only four sequence memory cells are required. This is because, in the present invention, as shown in FIG. 42 , a certain sequence is used for two or more subcarriers (use of two subcarriers in FIG. 42 ). Accordingly, the circuit scale of the transmission device can be reduced. On the other hand, unlike FIG. 42 , when mutually different sequences are used, eight sequence memory cells are required, and the circuit scale increases.

Next, the receiving device will be described. FIG. 3 is an example of the configuration of a receiving device. Hereinafter, the configuration of frequency offset and phase noise estimating section 213 in FIG. 3 will be described in detail using FIG. 43 .

FIG. 43 is an example of the configuration of frequency offset and phase noise estimating section 213 in FIG. 3A according to this embodiment. The frequency offset and phase noise estimation unit 213 in FIG. 43 includes: a pilot subcarrier extraction unit 3101, sequence storage units 3108_1~3108_4, sequence selection unit 3110, frequency offset and phase noise estimation unit 3123_#3 of carrier 3, carrier 5 frequency offset and phase noise estimation unit 3123_#5, carrier 8 frequency offset and phase noise estimation unit 3123_#8, and carrier 10 frequency offset and phase noise estimation unit 3123_#10.

Pilot subcarrier extraction section 3101 receives Fourier-transformed signal 206X (or 206Y) as input, and extracts the subcarrier of pilot symbol 305 . Specifically, the signals of carriers 3, 5, 8 and 10 are extracted. Therefore, pilot subcarrier extracting section 3101 outputs baseband signal 3102_#3 of carrier 3, baseband signal 3102_#5 of carrier 5, baseband signal 3102_#8 of carrier 8, and baseband signal 3102_#10 of carrier 10.

Sequence #1 storage unit 3108_1 stores sequence #1 in FIG. 40 , and outputs sequence #1 signal 3109_1 based on timing signal 212 . Sequence #2 storage unit 3108_2 stores sequence #2 in FIG. 40 , and outputs signal 3109_2 of sequence #2 based on timing signal 212 . Sequence #3 storage unit 3108_3 stores sequence #3 in FIG. 40 , and outputs signal 3109_3 of sequence #3 based on timing signal 212 . Sequence #4 storage unit 3108_4 stores sequence #4 in FIG. 40 , and outputs signal 3109_4 of sequence #4 based on timing signal 212 .

Sequence selection unit 3110 takes signal 3109_1 of sequence #1, signal 3109_2 of sequence #2, signal 3109_3 of sequence #3, signal 3109_4 of sequence #4 and timing signal 212 as inputs, assigns sequence #1 to signal 3111, assigns sequence #1 to signal 3111, and Assign #2 to signal 3112, assign sequence #2 to signal 3114, assign sequence #3 to signal 3115, assign sequence #3 to signal 3117, assign sequence #4 to signal 3118, assign sequence #4 to signal 3120, and assign sequence #1 to signal 3121, and output.

The frequency offset and phase noise estimation unit 3123_#3 of carrier 3 includes: code multiplication units 3103A and 3103B, and phase variation estimation units 3105A and 3105B, which estimate the frequency offset and phase noise of each channel of carrier 3.

The code multiplication unit 3103A takes the baseband signal 3102_#3 of the carrier 3 and the signal 3111 of the sequence #1 as input, and multiplies the baseband signal 3102_#3 of the carrier 3 and the signal 3111 of the sequence #1 to generate the signal 3111 of the channel A of the carrier 3. Baseband signal 3104A_#3, and output it. The reason for this is as follows.

The baseband signal 3102_#3 of carrier 3 is a signal in which the baseband signal of channel A and the baseband signal of channel B are multiplexed. When the multiplexed signal is multiplied by the signal 3111 of sequence #1, the baseband signal component of channel B with zero cross-correlation is removed, so that only the baseband signal component of channel A can be extracted.

Phase fluctuation estimation section 3105A receives baseband signal 3104A_#3 of channel A of carrier 3 as input, estimates phase fluctuation based on the signal, and outputs phase fluctuation estimation signal 3106A_#3 of channel A.

Similarly, the code multiplication unit 3103B takes the baseband signal 3102_#3 of the carrier 3 and the signal 3112 of the sequence #2 as input, and multiplies the baseband signal 3102_#3 of the carrier 3 and the signal 3112 of the sequence #2 to generate the signal 3112 of the carrier 3 The baseband signal 3104B_#3 of channel B is output.

Phase fluctuation estimation section 3105B receives baseband signal 3104B_#3 of channel B of carrier 3 as input, estimates phase fluctuation based on the signal, and outputs phase fluctuation estimation signal 3106B_#3 of channel B.

For the frequency offset and phase noise estimation unit 3123_#5 of carrier 5, only the signal to be processed is different. The phase variation estimation signal 3106A_#5 of channel A and the phase variation estimation signal 3106B_#5 of channel B of carrier 5. For the frequency offset and phase noise estimation unit 3123_#8 of carrier 8, only the signal to be processed is different, and it performs the same operation as the above-mentioned frequency offset and phase noise estimation unit 3123_#3 of carrier 3, and outputs The phase variation estimation signal 3106A_#8 of channel A and the phase variation estimation signal 3106B_#8 of channel B related to carrier 8. Furthermore, the frequency offset and phase noise estimating unit 3123_#10 of the carrier 10 is only different in the signal to be processed, and performs the same operation as the frequency offset and phase noise estimating unit 3123_#3 of the carrier 3 described above. , and output phase fluctuation estimation signal 3106A_#10 of channel A and phase fluctuation estimation signal 3106B_#10 of channel B of carrier 10.

As described above, by making the signals of channel A and channel B of the same carrier orthogonal, even if pilot symbols 305 are multiplexed on channel A and channel B, it is possible to estimate frequency offset and phase noise with high accuracy. As another important advantage, since channel estimation values (transmission path variation estimation values) are not required, the configuration of a part that compensates for frequency offset and phase noise can be simplified. If the signal point configurations of channel A and channel B are not orthogonal, the configuration is such that signal processing for MIMO separation such as ZF, MMSE, and MLD is performed, and then frequency offset and phase noise are estimated. On the other hand, according to the configuration of this embodiment, as shown in FIG. 3 , it is possible to compensate for frequency offset and phase noise at a stage preceding signal separation (signal processing section 223). Furthermore, in the signal processing unit 223, even after the signal of the channel A and the signal of the channel B are separated, the frequency offset and the phase noise can be removed by the pilot symbol 305, so it is possible to perform frequency offset and phase noise with higher precision. Compensation for phase noise.

However, when the signal point configurations of channel A and channel B of the same carrier are not orthogonal, the frequency offset and phase noise estimation accuracy of the frequency offset and phase noise estimation section 213 in FIG. interference component), it is difficult to add the frequency offset and phase noise compensation unit 215 in FIG. 3 , so that high-precision frequency offset and phase noise compensation cannot be performed.

As described above, by using the same sequence of pilot symbols 305 on multiple channels and/or multiple carriers, the sequence storage units 3108_1 to 3108_4 in FIG. 43 can be implemented among multiple channels and/or multiple carriers. Sharing brings simplification of the receiving device.

In this embodiment, the pilot symbols 305 orthogonal to each other in units of four symbols have been described as an example, but it is not limited to units of four symbols. However, considering the effect on orthogonality caused by fading fluctuations in the time direction, it is considered that if an orthogonal pattern is formed with about 2 to 8 symbols, the frequency offset and Estimation accuracy of phase noise. If the period of the orthogonal pattern is too long, the possibility that the orthogonality cannot be ensured increases, and the estimation accuracy of frequency offset and phase noise deteriorates.

As in Embodiment 1, important aspects of the method of configuring pilot symbols 305 in this embodiment are as follows.

●The pilot signals of channel A and channel B of the same carrier are orthogonal

●In the same channel, use different sequences on different carriers configured with pilot signals

● Use the same sequence on each channel (channel A and channel B)

Thus, when MIMO-OFDM transmission is performed using two transmission antennas, it is possible to realize a transmission device with a simple configuration that suppresses an increase in transmission peak power without deteriorating frequency offset and phase noise estimation accuracy. In addition, although selecting the pilot signal to form the pilot carrier is the best condition to satisfy all of the above three conditions, for example, if you only want to obtain a part of the above effects, for example, you can also select The pilot signal is used to form the pilot carrier and only two of the above three conditions are satisfied.

In the present embodiment, an example using the OFDM method was described, but it is not limited to this, and it can be implemented similarly even when using the single-carrier method, other multi-carrier methods, and spread spectrum communication methods. In addition, in this embodiment, although the case where there are two antennas for transmission and reception has been described as an example, it is not limited to this. Even if the number of receiving antennas is three or more, this embodiment will not be affected, and the same can be achieved. implement. In addition, the frame structure is not limited to this embodiment. In particular, for the pilot symbol 305 used for estimating distortions such as frequency offset and phase noise, as long as the pilot symbol 305 is configured on a specific subcarrier and is transmitted through multiple antennas, The transmission configuration is sufficient, and the number of subcarriers for transmitting the pilot symbol 305 is not limited to four as in the present embodiment. In addition, the implementation in the case of other numbers of antennas and other transmission methods will be described in detail later. In addition, although the pilot symbol 305, the reference symbol 302, the guard symbol 301, and the preamble are named and described here, using other calling methods has no influence on this embodiment. These are also the same in other embodiments. In addition, in the embodiment, terms such as channel A and channel B are used for description, but these are for ease of description, and even if other calling methods are used, the embodiment will not be affected at all.

In addition, although the frame structure has been described using the frame structure of FIG. 39 as an example, it is not limited thereto. In particular, although the case in which the pilot symbol 305 is arranged on a specific subcarrier has been described as an example, it is not limited thereto. ground configuration can also be implemented. However, it is important to adopt an arrangement that ensures the orthogonality of the pilot signals.

(Embodiment 8)

In the embodiment, a method of configuring a preamble in a spatial multiplexing MIMO system in which the number of transmission antennas is 4 and the number of transmission modulation signals is 4 will be described in detail.

An example of the configuration of the MIMO-OFDM transmission device according to this embodiment is shown in FIG. 44 . In FIG. 44 , the same reference numerals as those in FIG. 2 are assigned to parts that perform the same operations as those in FIG. 2 . In the MIMO-OFDM transmission device 4300, the mapping unit 4302 takes the transmission digital data 4301 and the frame structure signal 113 as input, and outputs the digital signal 103A of the channel A, the digital signal 103B of the channel B, the digital signal 103C of the channel C and the digital signal of the channel D Digital signal 103D.

In addition, the elements indicated by adding "A" at the end of the reference signs are the parts related to channel A; the elements indicated by adding "B" at the end of the reference signs are the parts related to channel B; The elements indicated by "C" are related to channel C; the elements indicated by adding "D" at the end of the reference numerals are related to channel D, but the target signal is different, basically the same as in the embodiment 1. The same processing is performed for the part related to the channel A described above with "A" appended to the end of the reference numerals.

FIG. 45 shows an example of a frame structure of a transmission signal formed by the transmission device according to this embodiment, and the same reference numerals are assigned to parts corresponding to those in FIG. 4 . (a) of Figure 45 shows the frame structure of channel A; (b) of Figure 45 shows the frame structure of channel B; (c) of Figure 45 shows the frame structure of channel C; (d) of Figure 45 shows the frame of channel D structure.

In Fig. 45, the most characteristic feature is the structure of the preamble, that is to say, at a certain time, in two channels, the reference symbol 302 and the protection (empty, null) symbol 301 are alternately arranged in frequency axis (e.g., in time 1 and time 2, channel A and channel C in Fig. 45, and time 3 and time 4, channel B and channel D), while in the remaining two channels (e.g., in Fig. Time 1 and time 2, channel B and channel D, and time 3 and time 4, channel A and channel C) do not configure the reference symbol 302, and only use the protection (empty, null) symbol 301 to form.

In the case of transmitting modulated signals of four channels, as the simplest structure, a method of inserting reference symbols 302 every three symbols can be considered, and the following wireless communication system can also be considered, that is, transmission on the frequency axis Correlation of path variations decreases as the distance on the frequency axis increases due to the influence of multipath. In such a wireless communication system, it is preferable not to insert the reference symbol 302 every third symbol. Plus, it's not necessarily bad to do so. Depending on the radio communication system, even if the reference symbol 302 is inserted every third symbol, reception quality may not be affected.

Therefore, in this embodiment, reference symbols 302 are alternately inserted into OFDM symbols at a certain time (a general term for symbols of all subcarriers present at a certain time. Refer to FIG. 45( d )). and guard (empty) symbols 301. However, it is not possible to adopt a configuration in which reference symbols 302 and guard (empty) symbols 301 are alternately arranged in OFDM symbols of all channels. This is because the reference symbols 302 collide between channels if handled in this way. By adopting the frame structure as shown in FIG. 45, even if the number of channels increases, reference symbols 302 are not spread too far on the frequency axis, and collision of reference symbols 302 between channels can be prevented from occurring.

FIG.46 shows the relationship between the modulation scheme of the reference symbol 302 and the normalization coefficient when four modulation signals are transmitted.

FIG. 47 and FIG. 48 show an example of mapping on the in-phase I-orthogonal Q plane in OFDM symbol units when the relationship between the modulation scheme of the reference symbol 302 and the normalization coefficient when four modulation signals are transmitted is as in FIG. 46 . Among them, the examples in FIG. 47 and FIG. 48 are the cases where the normalization coefficient is 1. At this time, according to the rule of Embodiment 3, it is preferable to set the power of the reference symbol 302 to 2×2+0×0=4 as shown in FIG.47 and FIG.48. Accordingly, in the receiving device, since the influence of the quantization error can be reduced, it is possible to suppress a decrease in reception quality.

In addition, although the case where the average received power of the data symbol 303 and the preamble is equal to the example has been described here, when the average received power of the preamble is made larger than the average received power of the data symbol 303, the reception There are many situations where quality can be assured. The idea described above can also be applied to this case.

In addition, in the present embodiment, even if the number of channels increases, the reference symbols 302 are not spread too much on the frequency axis, and the collision of the reference symbols 302 between channels can be prevented from happening beforehand. Although FIG. 45 was taken as an example to describe, the present invention is not limited thereto, and the same effect can be obtained even if a frame structure as shown in FIG. 49 is used.

(other embodiments)

In the above-mentioned embodiment, the symbol (preamble) for estimating the channel variation is arranged at the head of the frame, but it is not limited to this, and if the data symbol 303 can be separated, it can be arranged at any position. For example, in order to improve estimation accuracy, a method of inserting between data symbols 303 and 303 may be considered.

In addition, although in FIG. 4 and FIG. 16 , the preamble is configured on all carriers, that is, on carriers 1 to 12, but it may also be partially configured, such as on carriers 3 to 10. In addition, although it is called a preamble in the above-mentioned embodiment, this term itself does not have any meaning. Therefore, the name is not limited to this, and it may also be called a pilot symbol control symbol, for example.

Also, the present invention is similarly applicable even when the one antenna shown in the above-mentioned embodiment is configured by a plurality of antennas, and the above-mentioned one-channel signal is transmitted by the plurality of antennas.

Moreover, in the above-mentioned embodiments, words such as channels are used, which is an expression used to distinguish signals transmitted through each antenna, even if words such as channels are replaced by streams, modulated signals, or even transmitting antennas, etc. The words and phrases also do not have any impact on the above-mentioned implementation.

This specification is based on Japanese Patent Application No. 2005-243494 filed on August 24, 2005 and Japanese Patent Application No. 2006-228337 filed on August 24, 2006. Its contents are included here in its entirety.

Industrial Applicability

The MIMO-OFDM transmission device and MIMO-OFDM transmission method of the present invention can be widely applied to wireless communication systems such as wireless LAN and cellular systems.

Claims (6)

1.OFDM dispensing devices, including：

Map unit, based on frame structure signal sequence is generated, and the frame structure includes that being each assigned to time period and frequency carries Multiple element positions of ripple, the frame structure includes code element and the data symbols changed for transmission path estimation, wherein described The code element changed for transmission path estimation and the amplitude of the data symbols depend on one or more modulation systems and Individual or multiple MIMO methods；

Fourier inverse transformation unit, the signal sequence to being generated by the map unit carries out inverse fourier transform and generates OFDM codes Metasequence；

Transmitting element, the OFDM symbol sequence to being generated by the Fourier inverse transformation unit is transmitted.

2. OFDM dispensing devices as claimed in claim 1, wherein

The amplitude of the code element changed for transmission path estimation is that wherein x is just whole based on the square root function of variable x Number.

3. OFDM dispensing devices as claimed in claim 1,

The map unit exports multiple signal sequences, wherein the frame structure position of the code element changed for transmission path estimation Put mutually different,

The Fourier inverse transformation unit each carries out inverse fourier transform to the plurality of signal sequence, generates multiple OFDM codes Metasequence,

The transmitting element has multiple antennas, and using the plurality of antenna the plurality of OFDM symbol sequence is sent.

4.OFDM reception devices, including：

Receiving unit, receives OFDM symbol sequence；

Fourier transformation unit, the OFDM symbol sequence to being received carries out Fourier transform, output signal sequence, the signal Sequence is based on frame structure, and the frame structure includes being assigned to multiple element positions of time period and frequency carrier, the frame structure Including the code element and data symbols that change for transmission path estimation, wherein the code element changed for transmission path estimation and The amplitude of the data symbols depends on one or more modulation systems and one or more MIMO methods；

Transmission path changes estimation unit, the code changed for transmission path estimation included using the signal sequence Unit is transmitted the estimation of path distortion, and exports estimated distortion value；

Compensating unit, is compensated using distortion of the estimated distortion value to the signal sequence；

Demodulating unit, the signal sequence to having carried out distortion compensation is demodulated data recovery.

5. OFDM transmission/receptidevice device as claimed in claim 4, wherein,

The amplitude of the code element changed for transmission path estimation is that wherein x is just whole based on the square root function of variable x Number.

6. OFDM transmission/receptidevice device as claimed in claim 4,

The receiving unit has multiple antennas, and the OFDM symbol sequence for being received is multiple OFDM symbol sequences,

The Fourier transformation unit each carries out Fourier transform to the plurality of OFDM symbol sequence, generates multiple signal sequences Row,

The transmission path change estimation unit to being contained in each signal sequence in the estimation changed for transmission path The corresponding transmission path in frame structure position of code element estimated, export it is multiple obtained by estimated values,

Distortion of the compensating unit using each estimated value to each signal sequence is compensated,

The demodulating unit is demodulated and is restored to data to the described each signal sequence for having carried out distortion compensation.