CN102546484A

CN102546484A - Signal channel training method and signal channel training receiver device based on beacon frame

Info

Publication number: CN102546484A
Application number: CN2010105951289A
Authority: CN
Inventors: 陈小元; 刘铁; 焦金良
Original assignee: SHANGHAI BWAVETECH Corp
Current assignee: SHANGHAI BWAVETECH Corp
Priority date: 2010-12-17
Filing date: 2010-12-17
Publication date: 2012-07-04
Anticipated expiration: 2030-12-17
Also published as: CN102546484B

Abstract

The invention discloses a signal channel training method based on a beacon frame, which includes: performing initial synchronous search for the beacon frame and computing initial estimation of frequency offset of a sampling clock after access terminal equipment is powered on; starting tracking estimation and compensation of frequency offset of the sampling clock based on initial estimation of frequency offset of the sampling clock; estimating in-band narrow-band interference and computing the coefficient of a notching filter according to the frequency point position of the in-band narrow-band interference; performing second power control processing for digital baseband signals processed by means of low-pass filtering and notching filtering; starting coarse carrier frequency offset estimation and compensation; starting fine carrier frequency offset estimation and compensation; starting signal channel frequency response estimation; and leading a receiver to enter a signal channel condition tracking state. The invention further discloses a signal channel training receiver device based on the beacon frame. The success probability of synchronous detection for the beacon frame can be increased, and signal channel estimation precision can also be improved.

Description

Channel training method based on beacon frame and receiver device

Technical Field

The invention relates to the field of digital communication, in particular to a channel training method based on a beacon frame in a broadband access communication system based on a coaxial cable. The invention also relates to a channel training receiver arrangement based on beacon frames.

Background

The broadband access technology based on the coaxial cable utilizes the existing coaxial cable network to realize the network access and data transmission with high speed and high reliability without any adjustment to the existing network, and has obvious comprehensive advantages in the aspects of access network modification cost and access bandwidth compared with other access technologies such as xDSL, Ethernet access and even FTTH access. Typical broadband coaxial cable-based access technologies are HiNOC, HomePlug AV, MoCA, and HomePNA, among others.

The broadband access technology based on the coaxial cable generally adopts an OFDM (orthogonal frequency division multiplexing) carrier modulation mode on a physical layer, and adopts a QAM (quadrature amplitude modulation) modulation mode on each subcarrier of the OFDM. In the MAC (Media Access Control) layer, a TDD (Time Division Duplex) mode is usually used as a Duplex mode, and a TDMA (Time Division Multiple Access) technology is used as a Multiple Access technology.

Further, in order to avoid collision between access communications of different users, a fully coordinated scheduling manner is usually adopted, that is, all transmission events of the entire access sub-network are scheduled by one central node. One access subnetwork occupies one channel of the coaxial cable, and one access subnetwork is composed of one central node device and a plurality of access terminal devices. In the physical layer, the access subnetwork can be regarded as a bus type network of a medium sharing type, and the central node device and the access terminal device are both connected to the same sharing physical medium. The central node device makes scheduling decision for the downlink transmission opportunity and the uplink transmission opportunity of each access terminal device according to the arrival status of the downlink data frame, the uplink data transmission reservation application status of each access terminal device and other requirements of management control such as speaking priority, user bandwidth limitation and the like, and notifies each access terminal. Each access terminal thus knows when it can transmit data and when it can receive data. Based on such a fully coordinated scheduling mechanism, collisions between communications between different access terminal devices and the central node device may be completely avoided.

In order to make the above-mentioned fully coordinated TDD/TDMA scheme scheduling mechanism possible, complete timing synchronization must be maintained between the central node device and each access terminal device in a sub-network. In general, a beacon frame is periodically transmitted by the center node device as a timing reference between all devices in one subnet, and each access terminal device performs calibration of a local clock based on synchronization detection and reception of the beacon frame to keep the local clock synchronized with the clock of the center node device.

Various interferences existing in a communication channel, non-ideal characteristics of devices at a transmitting end and a receiving end, and the like may cause difficulties in signal reception processing of a receiver. For example, multipath (or echo) interference existing in a communication channel, signal power attenuation during signal transmission, thermal noise of devices at a transmitting end and a receiving end, a sampling clock frequency deviation and a carrier frequency deviation caused by a frequency difference of a local oscillator used for generating a clock signal at the transmitting end and the receiving end, and the like. The receiving end must estimate the channel time domain impulse response or frequency domain response, the received signal input power, the sampling clock frequency deviation and the carrier frequency deviation caused by the multipath interference, and compensate or eliminate the channel time domain impulse response or frequency domain response, the received signal input power, the sampling clock frequency deviation and the carrier frequency deviation, so that better receiving and demodulating performance can be obtained. The estimation of the receiver on the channel time domain or frequency domain response and the estimation and compensation of the input signal power, the sampling clock frequency deviation, the carrier frequency deviation and the like are generally referred to as channel training, the receiver obtains a signal with proper average power, sampling clock frequency deviation and carrier frequency deviation which are basically eliminated after the channel training, and then equalization and demodulation processing are carried out on the basis of the channel time domain or frequency domain response obtained by the channel training, so that effective information can be correctly recovered.

Generally, after being powered on, an access terminal device searches for a beacon frame on a channel, performs channel training after finding the beacon frame, and starts bidirectional data communication with a central node after completing the channel training. The high error probability of frame synchronization detection of beacon frames and the insufficient accuracy of channel training are the biggest limiting factors of the performance of broadband access communication systems based on coaxial cables. The deviation of the frame synchronization detection result of the beacon frame can cause the packet loss rate of the data transmission of the system to increase. If the frame synchronization detection result has serious deviation, the communication can not be carried out at all, and at the moment, the system can only carry out frame synchronization detection again after reset, so that the time for accessing the terminal equipment to the network is increased, and the network access experience of a user is reduced. On the other hand, the accuracy of channel training is not enough, which results in that the distortion of the signal spectrum caused by power deviation, time-frequency offset and channel response in the communication system is not sufficiently compensated, these residual errors can have a serious influence on the error code performance of the communication system, and the large residual error can result in a great reduction in the packet loss rate, thereby reducing the communication efficiency of the system and finally reducing the network access experience of the user. Therefore, in order to ensure stable and efficient data communication in a broadband access communication system based on a coaxial cable, a beacon frame synchronization detection technique with a high success rate and a channel training technique with high accuracy are required.

Disclosure of Invention

Aiming at the problems, the invention provides a channel training method based on a beacon frame, which can improve the success probability of frame synchronization detection and a channel training result with high precision; to this end, the present invention further provides a channel training receiver apparatus based on beacon frames.

In order to solve the above technical problem, the channel training method based on beacon frame of the present invention comprises the following steps:

step one, the central node equipment periodically sends beacon frames;

step two, after the access terminal equipment is powered on, carrying out initial synchronous search of the beacon frames, and judging whether the synchronous positions of the beacon frames are successfully detected or not based on the synchronous detection results of a plurality of continuous beacon frames; if the frame header synchronization position of the beacon frame is successfully found, calculating initial estimation of sampling clock frequency offset and turning to the third step; if the frame header synchronous position of the beacon frame is not found within a certain time, judging that the detection fails, and exiting the channel training process;

starting sampling clock frequency offset tracking estimation and compensation based on initial estimation of sampling clock frequency offset, and starting up and down beat adjustment of a time base counter; simultaneously starting a first power control process; after the first power control is converged, turning to step four; if the first power control is not converged within a certain time, judging that the convergence fails, and exiting the channel training process;

step four, estimating the in-band narrow-band interference, and if the in-band narrow-band interference is detected, turning to step four; otherwise, directly jumping to the step five;

calculating the coefficient of a notch filter according to the frequency point position of the in-band narrow-band interference, setting the calculated coefficient of the notch filter to the notch filter for notch filtering processing, and filtering out the in-band narrow-band interference of the corresponding frequency point by the notch filter in a time domain;

step six, performing second power control processing on the digital baseband signals after low-pass filtering and notch filtering processing, and adjusting the average power of the digital baseband signals after low-pass filtering and notch filtering processing to the optimal input average power required by a subsequent processing module; after the second power control processing is converged, turning to step seven; if the second power control processing is not converged within a certain time, determining that the convergence fails, and exiting the channel training process;

step seven, starting carrier frequency offset rough estimation and compensation processing to ensure that the carrier frequency offset is converged to a smaller range from an initial large value; stopping the carrier frequency offset rough estimation and the compensation processing after the carrier frequency offset rough estimation and the compensation processing are judged to be converged, and turning to the step eight; if the carrier frequency offset rough estimation and the compensation processing are not converged within a certain time, judging that the convergence fails and quitting the channel training process;

step eight, starting fine frequency offset estimation and compensation processing, and turning to the step nine after judging that the carrier frequency offset fine estimation and the compensation processing are converged; if the carrier frequency offset fine estimation and compensation processing is not converged within a certain time, determining that the convergence fails, and exiting the channel training process;

step nine, starting channel frequency domain response estimation processing, and switching to step ten after judging that the channel frequency domain response estimation processing is converged; if the channel frequency domain response estimation processing is not converged within a certain time, determining that the convergence fails, and exiting the channel training process;

step ten, the receiver enters a channel condition tracking state.

The channel training receiver device based on the beacon frame comprises:

the A/D converter is used for sampling and carrying out analog-to-digital conversion on the analog signal to generate a digital signal;

the first power estimation module is used for estimating the power of the digital signal output by the A/D converter to obtain a first power estimation signal;

the carrier frequency offset correction and down-conversion module is used for carrying out carrier frequency offset correction and digital down-conversion processing on the input digital signal based on the carrier frequency offset signal;

the notch filter is used for carrying out notch filtering processing of specified frequency points on the input digital baseband signals and filtering out the detected in-band narrow-band interference;

an interpolation filter for performing rate conversion on the input digital baseband signal processed by the notch filter in an interpolation mode and converting the digital signal with the sampling rate corresponding to the sampling rate of the A/D converter into data with the symbol rate;

the low-pass filter is used for carrying out low-pass filtering processing on the digital baseband signal which is output by the interpolation filter and subjected to rate conversion so as to filter out an out-of-band interference signal;

the second power estimation module is used for estimating the average power of the output signal of the low-pass filter again to obtain a second power estimation signal;

the digital gain control module is used for carrying out power adjustment on the signal output by the low-pass filter based on the second power control signal so that the average power of the output signal is in an optimal input average power state for a subsequent processing module;

the frame head detection and frame head time sequence control module is used for carrying out frame head detection based on the output signal of the digital gain control module, decomposing effective data frame signals into OFDM symbols from the frame head after detecting the frame head, removing the cyclic protection prefix and outputting the OFDM symbols to the FFT module;

the coarse frequency offset estimation module is used for carrying out coarse estimation on carrier frequency offset based on OFDM data of a time domain to obtain a carrier frequency offset coarse estimation signal;

an FFT module for performing discrete Fourier transform on the received data according to OFDM symbol unit, and transforming the OFDM data of time domain into the OFDM data of frequency domain;

the fine frequency offset tracking estimation module is used for carrying out fine estimation on carrier frequency offset based on the OFDM data of the frequency domain to obtain a carrier frequency offset fine estimation signal;

the channel estimation module is used for estimating channel frequency domain response based on frequency domain data of the OFDM symbol part carried in the beacon frame;

the equalization module divides the data of each subcarrier of the frequency domain OFDM data by the channel frequency domain response estimation result of the corresponding subcarrier based on the channel frequency domain response estimation result, and compensates the phase deflection and amplitude attenuation of the data symbol caused by channel transmission;

the channel decoding module is used for carrying out channel decoding processing on the data after the equalization processing to generate a decoding signal;

the signal-to-noise ratio estimation module is used for estimating the signal-to-noise ratio of each subcarrier by using the data result output by the equalization module;

the channel training controller is responsible for controlling the whole channel training process including the synchronous detection of the frame head of the beacon frame, namely determining when to start the modules, carrying out comprehensive processing according to the processing results processed by the modules to generate corresponding feedback control signals, carrying out convergence judgment on the processing results of the modules through statistical analysis, and carrying out corresponding process control.

The invention can obtain the beacon frame synchronization result with extremely high reliability by adopting a mode of confirming based on the beacon frame synchronization search result for a plurality of times continuously, and avoids the reduction of the system performance caused by the beacon frame synchronization detection error. Meanwhile, the invention adopts a multi-stage segmented beacon frame-based channel training method, which can effectively avoid the reduction of channel estimation precision caused by mutual coupling between different channel deviation estimation processes and provide a high-precision channel training result; and each stage has independent locking detection judgment, so that the current error processing can be terminated at the earliest time for retrying when a detection error occurs, and the system access time performance can be effectively improved. The invention can greatly improve the data communication performance of the broadband access communication system based on the coaxial cable.

Drawings

The invention will be described in further detail with reference to the following detailed description and accompanying drawings:

fig. 1 is a diagram of a typical beacon frame format;

fig. 2 is a schematic diagram of a broadband access subnetwork based on coaxial cable;

FIG. 3 is a block diagram of a receiver apparatus according to an embodiment of the present invention;

FIG. 4 is a flow chart of channel training based on beacon frames;

FIG. 5 is a signal power spectrum with single frequency interference;

FIG. 6 is a schematic diagram of an algorithm for coarse carrier frequency offset estimation;

FIG. 7 is a block diagram of one embodiment of the fine frequency offset tracking estimation block of FIG. 3;

fig. 8 is a block diagram of an embodiment of the channel estimation module of fig. 3.

Detailed Description

A typical beacon frame format is shown in fig. 1, and a beacon frame is composed of a frame header and a frame body, where the frame header mainly serves for frame synchronization capture, and therefore usually includes one or more pseudo-random sequences with good correlation characteristics, such as an M sequence. The frame body part of the beacon frame is mainly used for some interaction of network control information, such as the interaction of control information required when the access terminal device accesses the network, or some broadcast of common information, and generally consists of a few OFDM symbols in consideration of the improvement of bandwidth utilization efficiency. Considering that the beacon frame needs to be received and demodulated under a severe condition before channel training and compensation, a low-order differential modulation mode insensitive to carrier frequency offset is usually adopted, and DBPSK (differential binary phase shift keying) or DQPSK (differential quadrature phase shift keying) and the like are commonly used. For convenience of explanation, the DQPSK modulation scheme is assumed, but those skilled in the art should understand that the method is not limited to be applied to the DQPSK modulation scheme, and can be applied to any low-order differential modulation scheme or low-order non-differential modulation scheme with suitable modifications without departing from the basic idea of the present invention.

Fig. 2 is a schematic diagram of an access subnetwork of a broadband coaxial cable-based access system. As shown, an access subnetwork is composed of a central node device 201 and a plurality of access terminal devices (i.e.,

access node devices

221, 222 … 22k shown in the figure), all connected to the same coaxial cable. The central node equipment is arranged according to the period T_beaconThe beacon frame 211 is periodically transmitted, and the access terminal device maintains clock synchronization with the central node device based on detection and reception of the beacon frame, and acquires downlink control information transmitted by the central node device 201.

Fig. 3 is a system block diagram of an embodiment of a beacon frame based channel training receiver apparatus according to the present invention. As shown, the receiver apparatus comprises: an a/D (analog/digital) converter 302, a first power estimation module 303, a carrier frequency offset correction and down-conversion module 304, a notch filter 305, an interpolation filter 306, a low pass filter 307, a second power estimation module 308, a digital gain control module 309, a frame header detection and frame header timing control module 310, a coarse frequency offset estimation module 316, an FFT (fast fourier transform) module 311, a fine frequency offset tracking estimation module 317, a channel estimation module 318, an equalization module 312, an SNR (signal to noise ratio) estimation module 313, a channel training controller 315, and a channel decoding module 314.

The signal transmitted from the transmitting end reaches the receiving end after being transmitted through a coaxial cable channel, and is subjected to channel selection filtering, amplification and down-conversion to generate a low-intermediate frequency or baseband analog signal 301. The channel selection filtering, amplification, and down-conversion processes will be collectively referred to as analog signal processing in the following description. In some systems, radio frequency transmission is used, and the analog signal processing is also commonly referred to as radio frequency processing. Some systems use low intermediate frequency or baseband signal transmission, in which case no rf processing is required.

The a/D converter 302 samples and analog-to-digital converts the analog signal 301 to generate a digital low-intermediate frequency signal or a baseband signal.

The first power estimation module 303 performs power estimation on the digital signal output by the a/D converter 302, and sends an obtained first power estimation result 322 to the channel training controller 315 for comprehensive processing to obtain a first power control signal 320, and outputs the first power control signal 320 to the analog signal processing part for input signal gain control. The first power estimation module 303 may perform power estimation using all received digital signals corresponding to the beacon frame, or may perform power estimation using only a portion of the received digital signals. As a preferred aspect of the invention, the first power estimation is performed using only the portion of the received digital signal corresponding to the frame header portion of the beacon frame.

And a carrier frequency offset correction and down-conversion module 304, which performs carrier frequency offset correction and digital down-conversion processing on the input digital signal. The digital down-conversion process may be skipped if the input signal to the a/D converter 302 is already a baseband signal. After carrier frequency offset correction and digital down-conversion processing, a digital baseband signal without carrier frequency offset or with little residual carrier frequency offset is obtained.

And a notch filter 305 for performing notch filtering processing of a designated frequency point on the input digital baseband signal, so as to filter out the detected in-band narrowband interference and reduce the adverse effect of the in-band narrowband interference on subsequent signal processing.

The interpolation filter 306 rate-converts the input digital baseband signal processed by the notch filter 305 by interpolation based on the sampling timing offset signal 344, and converts the digital signal having a sampling rate corresponding to the sampling rate of the a/D converter 302 into data at the symbol rate. In performing the rate transformation, a variety of commonly used interpolation algorithms can be used, and first-order linear interpolation, FIR filter interpolation, gaussian interpolation, lagrange interpolation, and the like are common. As a preferable scheme of the invention, a third-order Lagrange interpolation algorithm is adopted for interpolation processing. The sampling timing offset signal 344 is generated by the channel training controller 315 based on the results of the sampling clock frequency offset tracking estimation.

The low-pass filter 307 performs low-pass filtering on the digital baseband signal after rate conversion output by the interpolation filter 306, so as to filter out-of-band interference signals.

The second power estimation module 308 performs average power estimation on the output signal of the low pass filter 307, and the obtained second power estimation result 321 is sent to the channel training controller 315 for comprehensive processing to obtain a second power control signal 323, and is output to the digital gain control module 309 for digital signal gain control. The second power estimation module 308 may perform power estimation using all or only a portion of the received digital signals corresponding to the beacon frame. As a preferred aspect of the present invention, the second power estimation is performed using only the portion of the received digital signal corresponding to the frame header portion of the beacon frame.

The digital gain control module 309 adjusts the power of the signal output from the low-pass filter 307 based on the second power control signal 321, so that the average power of the output signal is in the optimum input average power state for the subsequent processing module. Although the first power estimation block 303 and the associated analog signal power adjustment result in an input signal with the best input average power expected by the receiver. However, the signal may contain in-band narrowband interference and out-of-band interference signals, and after the notch filtering processing and the low-pass filtering processing are performed to filter out these interference signals, the average power of the signal may be already lower than the optimal input average power expected by the subsequent signal processing module, so that the input signal needs to be power-adjusted again by the second power estimation module 309 and the related digital gain control so as to meet the requirement of the subsequent signal processing module for the optimal input average power.

The frame header detection and frame header timing control module 310 performs frame header detection based on the output signal of the digital gain control module 309, decomposes the effective data frame signal into OFDM symbols from the frame header after detecting the frame header, and outputs the OFDM symbols to the FFT module 311 after removing the cyclic protection prefix.

And a coarse frequency offset estimation module 316 for performing coarse estimation of the carrier frequency offset based on the time domain OFDM data. The obtained carrier frequency offset rough estimation result 341 is sent to the channel training controller 315 and the carrier frequency offset fine estimation result 342 for comprehensive processing to obtain a carrier frequency offset signal 340, and is output to the carrier frequency offset correction and down-conversion module 304 for carrier frequency offset compensation. The characteristic of the coarse frequency offset estimation and compensation processing is that the capture and convergence range is very large, so that the coarse frequency offset estimation and compensation processing is usually used for eliminating the large initial carrier frequency offset in the initial stage of channel training, so that the carrier frequency offset is converged to a small range. However, the estimation accuracy of the coarse frequency offset estimation is relatively low, so after the coarse frequency offset estimation and the compensation processing, there may still be a considerable residual carrier frequency offset, and further more accurate carrier frequency offset estimation and compensation processing is required.

The FFT module 311 performs discrete fourier transform on the received data in OFDM symbol units, and transforms OFDM data in the time domain into OFDM data in the frequency domain.

The fine frequency offset tracking estimation module 317 performs fine estimation of carrier frequency offset based on the OFDM data in the frequency domain, and sends a fine estimation result 342 of carrier frequency offset to the channel training controller 315. In the channel training controller 315, the carrier frequency offset fine estimation result 342 and the carrier frequency offset coarse estimation result 341 are comprehensively processed to obtain a carrier frequency offset signal 340, and the carrier frequency offset signal is fed back to the carrier frequency offset correction and down-conversion module 304 for carrier frequency offset correction. The coarse frequency offset estimation and compensation process has reduced the initial carrier frequency offset to a smaller range before the fine carrier frequency offset estimation and compensation process is performed. After the fine estimation and compensation processing of the carrier frequency offset, the residual carrier frequency offset is further reduced to a range which has no adverse effect on the subsequent channel estimation and equalization processing.

The channel estimation module 318 estimates the channel frequency domain response based on the frequency domain OFDM data. In the invention, the channel frequency domain response is estimated based on the data of the OFDM symbol part carried in the beacon frame, and the estimation result of the channel frequency domain response is used for equalization processing to correct the distortion of the signal spectrum caused by channel transmission.

The equalizing module 312 compensates for phase deflection and amplitude attenuation of data symbols caused by channel transmission based on the channel frequency domain response estimation result, specifically by dividing data of each subcarrier of the frequency domain OFDM data by the channel frequency domain response estimation result of the corresponding subcarrier.

The channel decoding module 314 performs channel decoding processing on the equalized data to generate a decoded signal 319. In order to reduce the influence of the transmission error on data communication as much as possible, a sending end usually performs error correction code encoding on a sending data sequence, and the essence of the error correction code encoding is to add redundancy to the sending data sequence, that is, to use channel transmission resources with more than one bit to transmit information of one bit. At the receiving end, even if a part of the data has transmission errors, the transmission errors can be corrected by using the redundant information and a decoding algorithm corresponding to an error correction coding mechanism. Common error correcting codes are convolutional codes, reed solomon codes, BCH codes, Turbo codes, and LDPC (low density parity check) codes, which are used individually or jointly in a serial manner. Further processing of the decoding result signal 319 can extract valid user data information transmitted by the transmitting end.

A signal-to-noise ratio (SNR) estimation module 313, which performs signal-to-noise ratio estimation of each subcarrier by using the data result output by the equalization module 312, and these signal-to-noise ratio estimation results can be used to observe and monitor the communication quality of the current system, and also can be used to determine the appropriate modulation and coding scheme of each subcarrier in adaptive modulation.

The channel training controller 315 is responsible for controlling the whole channel training process including the frame header synchronization detection of the beacon frame, that is, determining when to start what module, performing comprehensive processing according to the processing results of each module to generate a corresponding feedback control signal, performing convergence judgment on the processing results of each module through statistical analysis, and performing corresponding process control, which is described in detail below with respect to the channel training process. The channel training controller may be a pure hardware-implemented circuit module, may be a microcontroller capable of executing software, or may be a combination of a microcontroller capable of executing software and a hardware-implemented circuit module. In a communication system, some signal processing tasks have a heavy processing load and a high real-time requirement, but have a low control flexibility requirement, and are suitable for being implemented by pure hardware circuit modules. While other signal processing tasks may require less real-time performance and may be less burdensome to process, while for higher control flexibility, these tasks may be more suitable for implementation in software executing in a microprocessor. As a preferred aspect of the present invention, the channel training controller 315 is implemented by a combination of a microcontroller that can execute software, plus a hardware-implemented circuit module.

A preferred embodiment of the receiver apparatus of the present invention for performing channel training based on the beacon frame periodically transmitted by the central node is shown in fig. 4, and a flow of the receiver apparatus for performing channel training based on the present invention is described below with reference to fig. 4. The whole process is controlled by the channel training controller 315.

Step one, the central node equipment periodically sends beacon frames;

and step two, after the receiver is powered on, searching a frame synchronization head of the beacon frame based on a correlation method.

After the receiver is powered on, an analog signal 301 output from the analog signal processing system is sampled and converted by an a/D converter 302 to generate a digital signal, and is processed by a carrier frequency offset correction and down-conversion module 304 to generate a digital baseband signal. At this time, since carrier frequency offset estimation is not performed yet, the carrier frequency offset estimation value is an initial value 0, and therefore, it is substantially equivalent to not performing carrier frequency offset compensation. The digital baseband signal is processed by the notch filter 305, the interpolation filter 306, the low pass filter 307 and the digital gain control module 309 and then output to the frame header detection and frame header timing control module 310. At this stage, since the processing such as the in-band narrow-band interference estimation, the sampling time offset estimation, and the second power estimation has not been performed, the notch filtering processing, the interpolation filtering processing, and the digital gain control are all processed based on default parameters, and substantially no effective processing is performed, and only the low-pass filtering processing filters out the out-of-band interference signal.

The frame header detection and frame header timing control module 310 performs beacon frame header synchronization detection based on the low-pass filtered digital signal. The beacon frame header synchronization detection is carried out based on a frame synchronization sequence correlation method, and when a correlation peak value is found to be larger than a preset threshold, the frame header of a beacon frame is considered to be found. The correlation method may adopt an autocorrelation method or a cross-correlation method. In general, regardless of robustness of detection performance or complexity of detection implementation, the cross-correlation method has an advantage over the auto-correlation method, so detection based on the cross-correlation method is more widely adopted. The specific implementation method of frame synchronization can be referred to in the present applicant's chinese patent application "frame synchronization method and apparatus for burst oriented communication system" (application number: 201010539745.7).

Since channel training is not performed under the initial access condition, a large sampling clock frequency deviation (hereinafter, abbreviated as "sampling time deviation") and a large carrier frequency deviation (hereinafter, abbreviated as "carrier frequency deviation") may exist, the input power of the received signal may be far from the optimal input power, and in addition, large in-band narrowband interference may exist, so that a beacon frame header synchronization detection error is easily caused.

In order to improve the detection success probability, the invention adopts a mode of confirming based on continuous multiple synchronous search results. The basic idea is as follows:

since the beacon frames occur according to a nominal fixed period, if two consecutive beacon frames are correctly detected, the time difference between the two beacon frame headers should theoretically be equal to the nominal fixed period. Considering that there is a sampling clock frequency deviation at this time, so that there is a certain deviation between the time difference between two adjacent beacon frame headers and the nominal fixed period, if the deviation between the time difference between the beacon frame headers obtained by two synchronous searches and the nominal fixed period is less than a certain threshold, it can be considered as a related detection result. And an initial estimate of the sampling time offset can be obtained from the deviation between the time difference of the two beacon frame headers and the nominal fixed period. Let the nominal period of the beacon frame be T_beaconIf the time of two consecutive frame headers obtained by frame header detection is T1, T2 and the decision threshold is deltaTH, the following condition | T2-T1-T is satisfied_beaconIf | ≦ deltaTH, judging that the results of the two frame header detections are correlated and correct; otherwise, the two detection results are considered to be not credible. If the results of the two frame header detections are determined to be correlated and correct, the initial estimation of the sampling time offset can be calculated as follows:

<math> <mrow> <mi>clk</mi> <mo>_</mo> <mi>ofst</mi> <mo>_</mo> <mi>ppm</mi> <mo>=</mo> <mfrac> <mrow> <mi>t</mi> <mn>2</mn> <mo>-</mo> <mi>t</mi> <mn>1</mn> <mo>-</mo> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mrow> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mfrac> <mo>·</mo> <msup> <mn>10</mn> <mn>6</mn> </msup> <mrow> <mo>(</mo> <mi>ppm</mi> <mo>)</mo> </mrow> </mrow> </math>

the above calculation results are in ppm (i.e. parts per million, and the following description refers to the offset when sampling is the unit), if the calculation results are greater than 0, it means that the clock frequency of the receiver is faster than that of the transmitter, otherwise it means that the clock frequency of the receiver is slower than that of the transmitter.

However, comparing the results only twice is still prone to errors, since it is not easy to determine a suitable threshold value, which must be determined according to the maximum sampling time offset that can occur in the system. The threshold is prone to false detection if it is determined to be too small and false alarm if it is determined to be too large. As a preferable aspect of the present invention, the number of results of the comparison confirmation is increased to 3 or more times for solving the problem of a judgment error that may be caused due to an improper selection of the threshold. Taking three times as an example, assume that the beacon frame period is T_beaconAssuming a sampling time offset of δ, the clock drift caused by this sampling time offset in one beacon frame period is Δ δ · T_beacon·10^-6. Further assuming that the synchronization times of the detected three consecutive beacon frame headers are t1, t2 and t3, respectively, it can be concluded that the beacon frame header has been correctly detected if the following relationships are satisfied between them:

|t2-t1-T_beacon|≤deltaTH1

|t3-t2-T_beacon|≤deltaTH1

|(t3-t2)-(t2-t1)|≤deltaTH2

wherein the setting of deltaTH1 depends on the maximum initial sampling offset delta that the system may exhibit_maxCan be according to delta_maxAnd T_beaconCalculating to obtain: deltaTH1 ═ delta_max·T_beacon/10⁶。

The setting of deltaTH2 is independent of initial sampling timing, and an empirical value can be obtained through simulation or system test. The physical meanings of the first two conditions are that the difference between the frame header synchronization times of two adjacent beacon frames (i.e. the first difference between the frame header synchronization times of the beacon frames) should be less than a preset first-order difference threshold, and the third condition is that the second-order difference between the frame header synchronization times of a plurality of continuous beacons should be less than a preset second-order difference threshold. Under ideal conditions, the second order difference of frame header synchronization time of consecutive beacon frames should be 0. Because the second-order difference threshold is independent of the initial sampling time offset, the threshold setting and the detection result are not influenced by the initial sampling time offset, and compared with the judgment only using the first-order difference, a more accurate judgment result can be obtained. Based on the t1, t2, and t3, an initial estimate of the sample time offset may also be calculated as follows:

<math> <mrow> <mi>clk</mi> <mo>_</mo> <mi>ofst</mi> <mo>_</mo> <mi>ppm</mi> <mo>=</mo> <mfrac> <mn>1</mn> <mn>2</mn> </mfrac> <mo>·</mo> <mrow> <mo>(</mo> <mfrac> <mrow> <mi>t</mi> <mn>2</mn> <mo>-</mo> <mi>t</mi> <mn>1</mn> <mo>-</mo> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mrow> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mfrac> <mo>+</mo> <mfrac> <mrow> <mi>t</mi> <mn>3</mn> <mo>-</mo> <mi>t</mi> <mn>2</mn> <mo>-</mo> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mrow> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mfrac> <mo>)</mo> </mrow> <mo>·</mo> <msup> <mn>10</mn> <mn>6</mn> </msup> <mrow> <mo>(</mo> <mi>ppm</mi> <mo>)</mo> </mrow> </mrow> </math>

although in this preferred embodiment, deltaTH1 still needs to be decided according to the maximum sampling time offset that may occur in the system, since there is a second-order differential decision based on deltaTH2, deltaTH1 may be set slightly larger, and even if a false alarm occurs in the decision based on deltaTH1, the false alarm is eliminated by the second-order differential decision based on deltaTH 2. Therefore, the beacon frame synchronization detection result obtained based on the preferred scheme has higher reliability.

The above listed conditions are based on the frame header time of three continuous beacons to make a decision, the decision method based on the frame header time of more than three beacon frames is the same as the above decision method, the more the number of beacon frames used for decision is, the more reliable decision result can be obtained, and the cost is that the time of the initial frame synchronous detection is correspondingly lengthened.

Based on N times of beacon frame header detection, the synchronization time of the detected continuous N (more than or equal to 3) beacon frame headers is assumed to be t respectively₁、t₂，…，t_N-1And if the following relation is satisfied between them, the beacon frame header is considered to be detected:

|t_k-t_k-1-T|≤deltaTH1，k＝2，...N

|(t_k-t_k-1)-(t_k+1-t_k)|≤deltaTH2，k＝2，...N-1

the first decision threshold deltaTH1 and the second decision threshold deltaTH2 are set as described above.

Meanwhile, the initial estimation of the sampling clock frequency offset is calculated according to the following method:

<math> <mrow> <mi>clk</mi> <mo>_</mo> <mi>ofst</mi> <mo>_</mo> <mi>ppm</mi> <mo>=</mo> <mfrac> <mn>1</mn> <mrow> <mi>N</mi> <mo>-</mo> <mn>1</mn> </mrow> </mfrac> <mo>·</mo> <munderover> <mi>Σ</mi> <mrow> <mi>k</mi> <mo>=</mo> <mn>2</mn> </mrow> <mi>N</mi> </munderover> <mrow> <mo>(</mo> <mfrac> <mrow> <msub> <mi>t</mi> <mi>k</mi> </msub> <mo>-</mo> <msub> <mi>t</mi> <mrow> <mi>k</mi> <mo>-</mo> <mn>1</mn> </mrow> </msub> <mo>-</mo> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mrow> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mfrac> <mo>)</mo> </mrow> <mo>·</mo> <msup> <mn>10</mn> <mn>6</mn> </msup> <mrow> <mo>(</mo> <mi>ppm</mi> <mo>)</mo> </mrow> </mrow> </math>

in the second step, if the synchronous detection result of the beacon frame header meeting the above conditions is not found within a certain time, the initial synchronous search of the beacon frame is judged to fail, the initial synchronous search of the beacon frame is exited, and the channel training flow is also exited; otherwise, considering that the synchronization time of the beacon frame header is successfully detected, calculating an initial estimation value of the sampling time offset according to the above method, and then turning to the third step.

Since the beacon frame appears periodically, after the synchronization time of the beacon frame is successfully detected in step two, the subsequent search for the head of the beacon frame only needs to be performed in an interval around the predetermined time. For example, if the last detected beacon frame synchronization time is T0 and the beacon frame period is T, it is only necessary to search for the next beacon frame synchronization time within a certain interval before and after the T0+ T time. The range of intervals may be determined based on an estimate of the residual sample time offset.

And step three, starting sampling clock frequency offset tracking estimation and compensation based on the initial estimation of the sampling clock frequency offset, starting up and down beat adjustment of a time base counter, and starting first power control processing.

And a substep 3-1 of starting sampling clock frequency offset tracking estimation and compensation and up-down beat adjustment of a time base counter.

The specific adjusting method is referred to the Chinese patent application of the present applicant as "time offset estimation and correction method of receiver in burst communication system" (application number: 201010588997.9).

Substep 3-2, a first power control process is initiated.

The first power control processing includes first power estimation processing, and first power gain adjustment is performed based on a first power estimation result.

The receiver continues to periodically detect and receive beacon frames. For each detected beacon frame, the average power estimation of the digital signal output by the a/D converter 302 corresponding to the detected beacon frame portion is performed by the first power estimation module 303. As another aspect of the present invention, the first power estimation may also be performed on the digital baseband signal after being processed by the carrier frequency offset correction and down-conversion module 304. Assuming that the digital signal used for the first power estimation is r (k), the digital intermediate frequency signal r (k) output by the a/D converter 302 is a real signal, and the digital baseband signal r (k) after being processed by the carrier frequency offset correction and down-conversion module 304 is a complex signal. The first power estimation module 303 performs the received signal average power estimation by:

where N is the number of data samples used for average power estimation.

The first power estimate may be made for the entire body of the beacon frame or for a portion of the beacon frame. The more samples are used, the higher the estimation accuracy, and the number of samples used in a specific implementation can be determined according to a trade-off between the requirement for estimation accuracy and the computational complexity.

Let P be a first power estimation result estimated based on received data corresponding to the ith beacon frame_aver(l) The channel training controller 315 will P_aver(l) And a preset reference target power P_refA comparison is made to determine the amount of power gain adjustment. In order to give consideration to both the power gain control convergence speed and the gain adjustment precision, the invention adopts a multi-stage control method, wherein each stage corresponds to an adjustment step length and a corresponding adjustment triggering condition. The specific control method comprises the following steps:

average power P of received signal of beacon frame_aver(l) And a reference target power P_refPower error value P_error(l)＝P_aver(l)-P_refCounting, if the power error value P is M1 times continuously_error(l) If the power gain value is larger than the preset threshold PdeltatH1, the power gain value is adjusted downwards by the gain step size gainstep₁(ii) a If the power error value is less than the preset threshold-PdeltatH 1M 1 times in succession, the power gain value is adjusted upward by the gain step gainstep₁. If the power error value P is M2 times in succession_error(l) If the power gain value is larger than the preset threshold PdeltatH2, the power gain value is adjusted downwards by the gain step size gainstep₂(ii) a If the power error value is less than the preset threshold-PdeltatH 2M 2 times in succession, the power gain value is adjusted upward by the gain step gainstep₂. … if M is consecutive_NThe power error value P_error(l) Greater than a preset threshold PdeltatH_NThen the power gain value is adjusted downward by the gain step gainstep_N(ii) a If M is continuous_NWhen the power error value is smaller than the preset threshold PdeltatH_NThen adjust the power gain value up by the gain step gainstep_N. Otherwise, the current power gain value is not adjusted. Wherein M1 is more than M2 and less than … and MN, PdeltatH1 is more than PdeltatH2 is more than … is more than PdeltatH_N，gainstep₁＞

gainstep₂＞…＞gainstep_N. In order to avoid repeated adjustment caused by simultaneous satisfaction of multiple conditions, the priority and adjustment step length gainstep of the adjustment of each stage are set_1～NThe sizes are the same. The adjustment condition judgment is performed in the order of high priority in each level of control, and the judgment of the adjustment condition of low priority is performed only when the adjustment condition of high priority is not satisfied. Once the adjustment condition for a certain level is satisfied and power gain adjustment is performed, all statistical variables are cleared and the next round of power gain adjustment is started from the beginning.

In the above-described manner, the power value and the gain value are both in units of decibels (dB) in the logarithmic domain. Therefore, the addition and subtraction operations are used in connection with the power comparison and gain adjustment. It should be understood by those skilled in the art that the unit of decibels is merely for convenience of description, and the description can be fully equivalent based on the power gain value in the real number domain, in which case the multiplication and division operations are used for power comparison and gain adjustment.

Based on the power gain (k) obtained by the above calculation, the channel training controller 315 generates a first power control signal 320 and outputs the first power control signal to the analog signal processing portion for controlling, for example, the gain of the variable gain amplifier, thereby realizing the power control of the received signal.

The more the number of stages of power control, the more accurate power control can be obtained, and the disadvantage is that the complexity of implementation is increased correspondingly. As a preferred aspect of the present invention, the first power control employs two-stage regulation in consideration of the trade-off between power control accuracy requirements and implementation complexity. In the two-stage adjustment embodiment, the power difference threshold deltaTH1 and the gain step gainstep1 are used for coarse power gain adjustment, mainly for the convenience that the receiver can quickly converge to the vicinity of the reference target power when the power deviation is large, and therefore, the power difference threshold deltaTH1 and the gain step gainstep1 can take larger values. As a preferred embodiment of the present invention, PdeltatH1 and gainstep1 can be 6 dB each.

The power difference threshold PdeltaTH2 and the gain step gainstep2 are used for fine adjustment of power gain, and are mainly used for facilitating that a receiver can track small changes of received signal power when power deviation is small, ensuring that the received signal power can be converged within a small range near a reference target power, and therefore a small value can be obtained. For example, as for a general receiver, the fluctuation of the received signal power around 1 db does not have a significant influence on the receiving performance, the PdeltaTH2 and the gainstep2 may be 1 db respectively as a preferred embodiment of the present invention.

The larger the values of M1 and M2 are obtained, the lower the probability of erroneous adjustment, but the larger the power adjustment delay is, and M1 and M2 may be respectively set to 2 and 3 as a preferable aspect of the present invention.

In sub-step 32, the channel training controller 315 counts the power gain adjustment status, and if no power gain adjustment occurs in consecutive M11 frames, it is determined that the first power control process has entered the convergence state, and the procedure goes to step three; otherwise, if the first power control process still does not enter the convergence state after M12 frames, it is determined that the first power control convergence fails, and the receiver exits the channel training procedure. As a preferred embodiment of the present invention, M11 and M12 may be taken as 3 and 30, respectively.

The above substeps 21 and 22 are independent of each other, and may be performed simultaneously or sequentially in any order.

And step four, after the first power control processing is converged, carrying out in-band narrow-band interference estimation.

Based on that the FFT module 311 performs Fast Fourier Transform (FFT) on the time domain data 330 of the OFDM symbol portion of the beacon frame body to obtain corresponding frequency domain data, the channel training controller 315 determines whether there is in-band narrowband interference in the effective channel bandwidth according to the frequency domain data. In general, since the data itself is random, the frequency domain representation of OFDM can be approximately seen as flat within the effective channel bandwidth, especially when averaging the frequency domain data of multiple OFDM symbols. When single-frequency interference or in-band narrowband interference exists, the corresponding frequency point in the frequency domain will show a large peak, because the fast fourier transform can effectively collect the single-frequency or in-band narrowband interference energy onto a certain frequency band, that is, the FFT processing shows the effect of having processing gain on the single-frequency or narrowband signal. The relationship between the gain and the number of FFT points N _ FFT brought by this FFT transform process is:

fft_gain_dB＝10*log10(N_FFT)

for example, if 256 FFT points are taken, the FFT gain is approximately 24 db.

Fig. 5 shows a signal power spectrum after fast fourier transform for a single frequency Interference with a frequency of 2MHz and a signal Interference ratio (SIR: signal Interference Noise) of 0 db, for a 16MHz channel with 256 total subcarriers, 210 effective subcarriers, and a frequency of 2 MHz. Single frequency interference can be seen as an ideal special case of narrowband interference. As can be seen from the figure, the power density at 2MHz is approximately 24 db higher than the power density at other frequency points.

Based on the above facts, after the OFDM symbol portion of the beacon frame is subjected to fast fourier transform, the signal amplitude of each subcarrier is compared with the average amplitude amp _ aver of all effective subcarriers, and if the difference between the signal amplitude amp (i) of the ith subcarrier and the amp _ aver is greater than a predetermined in-band narrowband interference detection threshold nbi _ detect _ TH, it can be determined that the frequency point corresponding to the subcarrier has in-band narrowband interference. The in-band narrow-band interference detection threshold nbi _ detect _ TH can be obtained by simulation or system testTo an empirical value, it can also be determined based on the performance requirements of the system against in-band narrowband interference and the processing gain provided by the FFT. For example, suppose the requirement of anti-single frequency interference of the system is (C/I)_NBIAnd the processing gain provided by the FFT of the system is FFT _ gain _ dB, then:

nbi_detect_TH＝(C/I)_NBI+fft_gain_dB+ΔNBI

as a preferred embodiment of the present invention,. DELTA._NBICan be taken to be between 3 and 3 decibels.

As a preferred embodiment of the present invention, in order to improve the determination accuracy, the fast fourier transform results of a plurality of OFDM symbols are averaged and then the above decision is performed.

In the step, if the presence of the in-band narrow-band interference is detected, turning to a fifth step; otherwise, directly jumping to the step six.

Step five, according to the frequency point position of the in-band narrow-band interference, the channel training controller 315 calculates the coefficient of the notch filter, and sets the calculated notch filter coefficient to the notch filter 305 for effective notch filtering processing. The notch filter 305 filters out the in-band narrowband interference of the corresponding frequency point in the time domain.

Notch filter 305 may be implemented with either a finite impulse response filter or an infinite impulse response filter. The finite impulse response filter has the advantages of stable work and the defect that the number of filter taps required by the same notch characteristic requirement is large; the infinite impulse response filter has an advantage that a desired notch filtering characteristic can be realized with a relatively small number of taps, but has a disadvantage that a phenomenon of unstable operation easily occurs, and thus attention is required in coefficient design, particularly in fixed-point design. If the method is realized by a finite impulse response filter, determining the frequency domain response requirement of the notch filter according to the frequency point position of the detected single frequency or narrow-band interference, sampling the frequency domain response of the notch filter, and then performing inverse Fourier transform on the sampling of the frequency domain response to obtain the time domain impulse response of the notch filter. If implemented as an infinite impulse response filter, the notch filter coefficients can be calculated based on the position of the pole-zero and the Z-transform. In any of the methods, a notch filter for notching a plurality of frequency points can be realized by one filter. For example, in an infinite impulse response notch filter, one zero corresponds to one notch frequency point, and thus if a plurality of frequency points need to be notched, a plurality of zeros may be set accordingly. For each notch frequency point, the notch width and attenuation can be determined according to the design requirements of the system, and the factors and the frequency of the notch frequency point determine the coefficient of the notch filter. The above two calculation methods of calculating the notch filter coefficients are common knowledge to an engineer skilled in the art, and thus a detailed description of the calculation method of the notch filter coefficients is omitted herein.

After the coefficients of the notch filter are calculated and set in the notch filter 305, the process proceeds to step six.

And step six, performing second power control processing on the received data after filtering processing of the notch filter and the low-pass processor.

And the second power control processing comprises second power estimation processing, comprehensive processing is carried out on the basis of the second power estimation signal to generate a second power control signal, and digital gain adjustment is carried out on the basis of the second power control signal, so that the input signal power of a subsequent module is in an optimal working point state.

The purpose of the aforementioned first power control process is to bring the input signal power of the a/D converter 302 to an optimum operating point that takes into account the trade-off between quantization noise and overflow noise of the a/D conversion. When there is a large out-of-band interference and in-band narrow-band interference, since these are filtered by the low pass filter and the notch filter, although the input signal level of the a/D converter 302 is in the optimum operating point state, the signal point level becomes smaller for the subsequent processing module after the low pass filtering and the notch filtering. This may result in a decrease in reception performance. It is therefore necessary to perform digital gain adjustment on the signal after low-pass filtering and notch filtering.

A second power estimate P of the received signal estimated based on the data received in the first beacon frame_aver，2(l) The channel training controller 315 will P_aver，2(l) And a preset second reference target power P_ref，2A comparison is made to determine the amount of gain adjustment. For each detected beacon frame, the average power of the low-pass filtered and notch filtered signal is estimated. Assuming the signal here is denoted r2(k), N is the number of data samples used to make the second power estimate. The second power estimation module 308 performs the received signal average power estimation by:

the second power estimate may be made for the entire body of the beacon frame or for a portion of the beacon frame. The more sampling points are adopted

The higher the estimation accuracy, the number of sampling points used in the specific implementation can be determined according to the trade-off between the requirement for estimation accuracy and the computational complexity.

In order to give consideration to both the convergence speed of gain control and the precision of gain adjustment, a multi-stage gain adjustment method is adopted, and each stage corresponds to an adjustment step length and a corresponding adjustment triggering condition. The specific adjustment method is as follows:

average power P of received signal of beacon frame_aver，2(l) And a reference power P_ref，2Power error value P_error，2(l)＝P_aver，2(l)P_ref，2Counting, if the power error value P is continuously L1 times_error，2(l) If the power gain value is larger than the preset threshold dagcTH1, adjusting the power gain value downwards by a gain step dagcstep 1; if the power error value is less than the preset threshold, dagcTH1, L1 consecutive times, the power gain value is adjusted up by the gain step, dagcstep 1. If the power error value P is continuously L2 times_error，2(l) If the power gain value is larger than the preset threshold dagcTH2, adjusting the power gain value downwards by a gain step dagcstep 2; if the power error value is less than the preset threshold, dagcTH2, L2 consecutive times, the power gain value is adjusted up by the gain step, dagcstep 2.… if L is continuous_NThe power error value P_error，2(l) Is greater than a preset threshold dagcTH_NThen adjust the power gain value downward by the gain step length dagcstep_N(ii) a If L is continuous_NThe power error value is smaller than the preset threshold-dagcTH_NThen adjust the power gain value upward by the gain step dagcstep_N. Otherwise, the current power gain value is not adjusted. Wherein L1 < L2 < … < L_N，dagcTH1＞dagcTH2＞…＞dagcTH_N，dagcstep1＞dagcstep2＞

…＞dagcstep_N. In order to avoid repeated adjustment caused by simultaneous satisfaction of multiple conditions, the priority of adjustment of each stage is set to be the same as the gain step size. The adjustment condition judgment is performed in the order of high priority in each level of control, and the judgment of the adjustment condition of low priority is performed only when the adjustment condition of high priority is not satisfied. Once the adjustment condition for a certain level is met, and power gain adjustment is performed, all statistical variables are cleared,the power gain adjustment for the next round from scratch.

In the above-described manner, the power value and the gain value are both in units of decibels (dB).

Based on the gain dagc _ gain (k) calculated above, a second power control signal 323 is generated by the channel training controller 315 for digital power gain adjustment by the digital gain control module 309.

The more the number of stages of power control, the more accurate power control can be obtained, and the disadvantage is that the complexity of implementation is increased correspondingly. As a preferred solution of the present invention, the second power control employs two-stage regulation in view of the trade-off between power control accuracy requirements and implementation complexity. In the implementation of two-stage adjustment, the power difference threshold dagcTH1 and the gain step dagcstep1 are used for the second coarse power gain adjustment, mainly to facilitate that the receiver can quickly converge to the vicinity of the reference target power when the power deviation is relatively large (i.e. the narrowband interference power is relatively large or the out-of-band interference is relatively large), so that a relatively large value can be taken. As a preferred embodiment of the present invention, dagcTH1 and dagcstep1 may be 6 decibels, respectively.

The power difference threshold dagcTH2 and the gain step dagcstep2 are used for the second fine power gain adjustment, which is mainly to facilitate that the receiver can track small changes in the received signal power when the power deviation is small, and ensure that the received signal power can converge to a small range near the reference target power, so that a small value can be obtained. For example, since the fluctuation of the received signal power around 1 db does not have a significant influence on the receiving performance for a general receiver, the dagcTH2 and the dagcstep2 may be 1 db respectively as a preferred embodiment of the present invention.

The larger the values of L1 and L2 are obtained, the lower the probability of erroneous adjustment, but the larger the power adjustment delay is, and as a preferable aspect of the present invention, L1 and L2 may be taken as 2 and 3, respectively.

In this step, the channel training controller 315 counts the digital power gain adjustment status, and if no adjustment of any gain step occurs in consecutive L11 frames, it is determined that the second power control process has entered the convergence state, and the process goes to step seven; on the contrary, if the second power control process still does not enter the convergence state after the L12 frame, it is determined that the convergence of the second power control process fails, and the receiver exits the channel training procedure. As a preferred embodiment of the present invention, L11 and L12 can be taken as 3 and 15, respectively.

And step seven, starting coarse frequency offset estimation and compensation processing.

Under ideal conditions, after down-conversion processing is performed at a receiving end, a band-pass signal is moved to a baseband, and the center frequency of the signal should be at a zero frequency, that is, a direct-current point. Because the oscillation frequency of the carrier frequency sources (such as crystal oscillators) at the transmitting end and the receiving end usually has a deviation, and the influence of other non-ideal factors on the channel, after the receiving end is subjected to down-conversion processing, the center frequency of the signal spectrum deviates from a direct current point, and the deviation is called as carrier frequency deviation. For coherent demodulation, if the carrier frequency offset cannot be compensated, the demodulation performance will be seriously affected.

In order to consider both the convergence range and speed of carrier frequency offset estimation and compensation, and the tracking accuracy or residual error performance after convergence, carrier frequency offset estimation and compensation are generally divided into two stages, i.e., coarse estimation and compensation, and fine estimation and compensation. The former generally requires a larger capture range and a faster convergence rate, but the allowable estimation error is also larger, and a larger compensation step size is generally adopted, so that the method is suitable for the initial capture stage of carrier frequency offset estimation and compensation; the latter generally requires higher estimation accuracy, and usually adopts smaller compensation step length, so that the convergence speed is slower and the capture range is smaller, and the method is suitable for fine tracking and adjustment after the carrier frequency offset has converged within a certain range.

In the embodiment of the present invention, the coarse carrier frequency offset estimation is performed in the time domain by the coarse frequency offset estimation module 316, and the received data using the frame header of the beacon frame is performed by using a delay correlation technique.

Coarse frequency offset estimation algorithm processing method referring to fig. 6, in the figure and in the following description ()^*Which means the conjugation of complex numbers.

The received signal of the frame synchronization sequence corresponding to the frame header of the beacon frame is marked as r (k), the original frame synchronization sequence transmitted at the transmitting end is marked as M (k), and the received signal is changed into M (k) e after modulation^jφ(k)The length of the frame sync sequence is denoted as N. The carrier frequency offset rough estimation and compensation are carried out by the following steps:

and 7-1, removing the modulation information in the received signal. The modulated frame synchronization sequence M (k) e^jφ(k)The conjugate multiplication corresponding to the symbol is performed with the corresponding received signal r (k), as shown in the following formula:

r′(k)＝M^*(k)·e^jφ(k)·r(k)k＝0，...，N-1

step 7-2, taking two continuous signals from the received signal r' (k) after the modulation information is removed, and performing delay correlation corresponding to the symbol, as shown in the following formula:

y(k)＝(r′(k))^*·r′(k+K)k＝m，...，n

where 0 ≦ m, N ≦ N-1 defines the range for delay-correlated two consecutive signals taken from the received signal r' (K) that are separated by a distance K, which may even partially overlap. The longer the length of the continuous signal is, the higher the estimation accuracy is, and the smaller the distance between two continuous signals is, the larger the capture range of the carrier frequency offset rough estimation is.

Step 7-3, calculating the phase of the delay correlation result y (k), and then calculating the average to obtain the carrier frequency offset coarse estimation value cfe (i) based on the current beacon frame, as shown in the following formula:

<math> <mrow> <mi>cfe</mi> <mrow> <mo>(</mo> <mi>i</mi> <mo>)</mo> </mrow> <mo>=</mo> <mfrac> <msub> <mi>f</mi> <mi>s</mi> </msub> <mrow> <mn>2</mn> <mi>πK</mi> </mrow> </mfrac> <mi>E</mi> <mrow> <mo>(</mo> <mi>angle</mi> <mrow> <mo>(</mo> <mi>y</mi> <mrow> <mo>(</mo> <mi>k</mi> <mo>)</mo> </mrow> <mo>)</mo> </mrow> <mo>)</mo> </mrow> </mrow> </math>

where E () represents averaging and angle () represents phase of the complex signal.

And 7-4, carrying out statistical judgment on the carrier frequency offset rough estimation result 341 obtained based on each beacon frame, namely cfe (i), so as to obtain a compensation amount cfe _ fofst for carrier frequency offset compensation. The specific method comprises the following steps: initializing cfe _ fofst to 0 at the beginning; if the carrier frequency offset coarse estimation value cfe (i) is greater than the threshold cfeTH1 for K1 times continuously, adjusting the compensation amount cfe _ fofst upwards by a compensation step cfestep 1; if the estimated carrier frequency offset value cfe (i) is less than the threshold-cfeTH 1 for K1 times, the compensation amount cfe _ fofst is adjusted downward by the compensation step cfestep 1. If the carrier frequency offset coarse estimation value cfe (i) is greater than the threshold cfeTH2 for K2 times continuously, adjusting the compensation amount cfe _ fofst upwards by a compensation step cfestep 2; if the estimated carrier frequency offset value cfe (i) is less than the threshold cfeTH2 for K2 times, the compensation amount cfe _ fofst is adjusted downward by the compensation step cfestep 2.… if K is consecutive_NThe carrier frequency offset coarse estimation value cfe (i) is greater than the threshold value cfeTH_NThen, the compensation amount cfe _ fofst is adjusted downward by the compensation step cfestep_N(ii) a If K is continuous_NThe carrier frequency offset coarse estimation value cfe (i) is less than the threshold-cfeTH_NThen, the compensation amount cfe _ fofst is adjusted upward by the compensation step cfestep_N. Otherwise, the compensation amount cfe _ fofst is not adjusted. Wherein K1 < K2 < … < K_N，cfeTH1＞cfeTH2＞…＞cfeTH_N，cfestep1＞cfestep2＞…＞cfestep_N. To avoid multiple conditions being met simultaneously resulting in repeated adjustmentsThe priority of each stage of adjustment is set to be the same as the step size. And in each level of regulation, the regulation condition judgment is carried out according to the high-low order of the priority, and the judgment of the regulation condition with low priority is carried out only when the regulation condition with high priority is not satisfied. Once the adjustment condition of a certain priority is satisfied and the carrier frequency offset compensation amount adjustment is performed, all statistical variables are cleared, and the coarse frequency offset estimation and the corresponding compensation amount adjustment of the next round are started from the beginning.

The carrier frequency offset compensation cfe _ fofst obtained through the above calculation and the carrier frequency offset compensation ffe _ fofst obtained through the fine frequency offset estimation are synthesized to obtain a carrier frequency offset signal 340, and the carrier frequency offset signal 340 is set to the carrier frequency offset correction and down-conversion module 304 for controlling the carrier frequency offset correction.

In this step, the channel training controller 315 counts the adjustment status of the carrier frequency offset compensation amount cfe _ fofst, and if no step adjustment occurs in any continuous K11 frames, it is determined that the adjustment of the carrier frequency offset compensation amount cfe _ fofst has already entered a convergence state, the coarse frequency offset estimation processing is stopped, and then the process goes to step eight; and if the carrier frequency offset compensation amount cfe _ fofst adjustment still does not enter a convergence state after K21 frames, judging that the coarse frequency offset estimation and the compensation convergence fail, and exiting the channel training process by the receiver. As a preferred embodiment of the present invention, K11 and K21 may be taken as 10 and 30, respectively.

And step eight, starting fine frequency offset estimation and compensation processing.

In OFDM systems, fine frequency offset estimation is typically performed in the frequency domain. In the presence of the pilot frequency in the frequency domain, fine estimation of the carrier frequency offset can be performed based on the pilot frequency. Some systems do not have frequency domain pilot, for example, a coaxial cable-based wired communication system is generally superior due to channel conditions, and all subcarriers are used to transmit effective data in order to avoid loss of spectral efficiency caused by the pilot. In this case, a decision-directed method is usually adopted to perform fine estimation of carrier frequency offset. The specific implementation depends on the frame body structure and modulation scheme of the beacon frame. As a preferred embodiment of the present invention, a method for implementing fine carrier frequency offset estimation in a decision-directed manner according to the present invention is described below by taking a case where a frame body portion of a beacon frame includes two OFDM symbols and a DQPSK modulation manner is adopted as an example, and a structure of the fine frequency offset estimation module 317 is shown in fig. 7. The FFT module 311 transforms the time domain signal after the power gain adjustment, the sampling time offset compensation, and the carrier frequency offset compensation into frequency domain data in OFDM units, outputs the frequency domain data to the fine frequency offset tracking estimation module 317, and performs fine frequency offset tracking estimation according to the following steps.

Step 8-1, performing fast fourier transform on the beacon frame body, and transforming the time domain data to the frequency domain to obtain frequency domain data x (l, m), where l is 0, 1 is an OFDM symbol sequence number in the beacon frame body, and m is a subcarrier sequence number in one OFDM symbol;

step 8-2, DQPSK demodulation is carried out on the frequency domain data x (l, m) of the OFDM symbol to obtain a demodulation result bit stream z (n);

step 8-3, carrying out DQPSK coding mapping on z (n) again to obtain w (l, m);

step 8-4, calculating the phase difference phi (l, m) between x (l, m) and w (l, m) as x^*(l，m)·w(l，m)；

Step 8-5, calculating the average value of the phase rotation amount of the corresponding sub-carrier of every two adjacent symbols of the N OFDM, as shown in the following formula:

<math> <mrow> <mi>φ</mi> <mo>_</mo> <mi>diff</mi> <mrow> <mo>(</mo> <mi>i</mi> <mo>)</mo> </mrow> <mo>=</mo> <mfrac> <mn>1</mn> <mrow> <mi>sizeof</mi> <mrow> <mo>(</mo> <mi>Ω</mi> <mo>)</mo> </mrow> </mrow> </mfrac> <munderover> <mi>Σ</mi> <mrow> <mi>n</mi> <mo>=</mo> <mn>2</mn> </mrow> <mi>N</mi> </munderover> <munder> <mi>Σ</mi> <mrow> <mi>m</mi> <mo>&Element;</mo> <mi>Ω</mi> </mrow> </munder> <mrow> <mo>(</mo> <mi>φ</mi> <mrow> <mo>(</mo> <mi>n</mi> <mo>,</mo> <mi>m</mi> <mo>)</mo> </mrow> <mo>-</mo> <mi>φ</mi> <mrow> <mo>(</mo> <mi>n</mi> <mo>-</mo> <mn>1</mn> <mo>,</mo> <mi>m</mi> <mo>)</mo> </mrow> <mo>)</mo> </mrow> </mrow> </math>

where i denotes the number of beacon frames, N denotes the number of OFDM symbols in a beacon frame, Ω denotes the range of subcarriers selected to participate in the calculation, and sizeof (Ω) denotes the number of subcarriers included in Ω. The signal conditions of the subcarriers, which are usually at the edges and center of the spectrum, are more susceptible to the harsh elements; in addition, when the in-band narrowband interference exists, some subcarriers around the in-band narrowband interference frequency point can also be seriously affected. In order to ensure the accuracy of the fine estimation of the carrier frequency offset, the use of these subcarriers should be avoided to participate in the calculation. As a preferred scheme of the present invention, on one hand, subcarriers with better quality are selected according to empirical values obtained by system simulation or test, and subcarriers at the edges and the center points of frequency bands which are easily affected by severe factors are excluded; on the other hand, according to the detection result of the in-band narrow-band interference, a plurality of subcarriers at the left and right of the in-band narrow-band interference frequency point are also excluded.

And 8-6, carrying out statistical judgment on the carrier frequency offset fine estimation value phi _ diff (i) obtained based on each beacon frame to obtain a compensation amount ffe _ offset for carrier frequency offset compensation. The specific method comprises the following steps: if the fine carrier frequency offset estimation value phi _ diff (i) is greater than the threshold ffeTH1 for Q1 times continuously, adjusting the compensation amount ffe _ fofst up by the compensation step length ffesep 1; if the fine carrier frequency offset estimate phi _ diff (i) is smaller than the threshold-ffeTH 1 for Q1 times consecutively, the compensation amount ffe _ fofst is adjusted downward by the compensation step length ffesep 1. If the fine estimated carrier frequency offset value φ _ diff (i) is greater than the threshold ffeTH2 for Q2 times continuously, the compensation amount will beffe _ fofst adjusts the compensation step size ffesep 2 upward; if the fine carrier frequency offset estimate phi _ diff (i) is smaller than the threshold-ffeTH 2 for Q2 times consecutively, the compensation amount ffe _ fofst is adjusted downward by the compensation step length ffesep 2.… if Q continues_NThe fine estimated value of the carrier frequency deviation phi _ diff (i) is greater than the threshold ffeTH_NThen the compensation amount ffe _ fosst is adjusted upward by the compensation step length ffesep_N(ii) a If Q is continuous_NThe fine estimated value of the carrier frequency deviation phi _ diff (i) is smaller than the threshold value-ffeTH_NThen the compensation amount ffe _ fosst is adjusted downward by the compensation step length ffesep_N. Otherwise, the compensation amount cfe _ fofst is not adjusted. Wherein Q1 < Q2 < … < Q_N，ffeTH1＞ffeTH2＞…＞ffeTH_N，ffestep1＞ffestep2＞…＞ffestep_N. In order to avoid repeated adjustment caused by simultaneous satisfaction of multiple conditions, the priority of each stage of adjustment is set to be the same as the size of the compensation step. The adjustment condition judgment is performed in the order of high priority in each level of adjustment, and the judgment of the adjustment condition of low priority is performed only when the adjustment condition of high priority is not satisfied. Once the adjustment condition of a certain priority is satisfied and the carrier frequency offset compensation adjustment is performed, all the statistical variables are cleared, and the carrier frequency offset compensation adjustment of the next round is started from the beginning.

In this step, the channel training controller 315 counts the adjustment status of the carrier frequency offset compensation ffe _ fofst, and if no adjustment of any compensation step length occurs in any consecutive Q11 frames, it is considered that the fine carrier frequency offset estimation and compensation process has entered the convergence state, and the process goes to step nine; if the fine carrier frequency offset estimation and compensation processing still does not enter a convergence state after the Q21 frame, the carrier frequency offset compensation quantity ffe _ offset adjustment is judged to fail, and the receiver exits the channel training process. As a preferred embodiment of the present invention, Q11 and Q21 can be taken to be 3 and 15, respectively.

The carrier frequency offset compensation ffe _ offset obtained through the above calculation and the carrier frequency offset compensation cfe _ offset obtained through the coarse frequency offset estimation are synthesized to obtain a carrier frequency offset signal 340, which is set to the carrier frequency offset correction and down-conversion module 304 for controlling the carrier frequency offset correction.

And step nine, starting channel frequency domain response estimation processing.

In OFDM systems, channel frequency domain response estimation is typically performed in the frequency domain. In a wired communication system using a coaxial cable or the like, since channel conditions are generally good, a pilot dedicated for channel training is not inserted in the frequency domain in order to improve spectrum use efficiency. In this case, channel estimation is usually performed in a decision-directed manner. Typically, the signal-to-noise ratio estimation for each subcarrier is performed simultaneously with the channel estimation.

The FFT module 311 transforms the time domain signal after the power gain adjustment, the sampling time offset compensation, and the carrier frequency offset compensation into frequency domain data in OFDM units, and outputs the frequency domain data to the channel estimation module 318 and the equalization module 312 for channel frequency domain response estimation. In the scheme of the invention, the channel frequency domain response estimation is carried out in a decision-directed manner based on the frame body part of the beacon frame only. And performing signal-to-noise ratio estimation of each subcarrier at the same time of channel frequency domain response estimation. The specific implementation of the channel frequency domain response estimation and the signal-to-noise ratio estimation depends on the frame body structure and modulation of the beacon frame. As a preferred embodiment of the present invention, the following describes an implementation method of channel frequency domain response estimation in a decision-directed manner according to the present invention by taking a case that a frame body portion of a beacon frame includes one or more OFDM symbols and a DQPSK modulation manner as an example, and a specific structure of the channel estimation module 317 is shown in fig. 8.

Step 9-1, carrying out FFT (fast Fourier transform) on the frame body part of the beacon frame to obtain frequency domain data

Wherein i represents the frame number of the beacon frame, n represents the OFDM symbol number in the beacon frame body, and k represents the subcarrier number in the OFDM symbol.

Step 9-2, pair

Using to the last oneChannel estimation values obtained up to beacon frameCarrying out equalization to obtain an equalization result

Wherein,

step 9-3, pairDQPSK demodulation is carried out to obtain demodulation result

Step 9-4, pair

DQPSK modulation is carried out again to obtain DPQSK symbol sequence

Step 9-5 based on

And

processed by the unmodulating information

Obtained after removal of modulation information

I.e. representing the residual channel frequency domain response information for each subcarrier.

Step 9-6, averaging the residual channel frequency domain responses of a plurality of OFDM symbols to obtain an average residual channel frequency domain response, wherein the calculation mode is as follows:

where N is the number of OFDM symbols contained in the beacon frame. However, step 9-6 may be omitted if the beacon frame contains only one OFDM symbol.

Step 9-7, the average residual channel frequency domain response and the channel estimation value until the last beacon frame are obtained

Synthesizing to obtain the channel estimation value of the current frame

And 9-8, performing channel frequency domain response estimation based on each beacon frame, updating the channel frequency domain response, and judging whether the channel frequency domain response estimation processing is converged.

As a preferred aspect of the present invention, the convergence determination of the channel frequency domain response estimation process may adopt the following manner: an empirical value of the number of FRAMEs required for convergence of a channel frequency domain response estimate, such as CHE _ ACQ _ FRAME, is determined by simulation. In actual operation, after the channel frequency domain response estimation is started, the channel frequency domain response estimation and the signal-to-noise ratio estimation are judged to enter the convergence state after the CHE _ ACQ _ FRAME FRAME processing, and the step ten is carried out. In this case there is no result of convergence failure.

As another preferred aspect of the present invention, the convergence determination of the channel frequency domain response estimation process may adopt the following manner: comparing the 'difference' between the estimated frequency domain response result of the previous frame and the processed update result of the current frame, and if the 'difference' is smaller than a preset threshold value and the continuous LOCK _ DECISION frames meet the condition, judging that the channel frequency domain response estimation and the signal-to-noise ratio processing enter a convergence state. Here, LOCK _ DECISION may be an empirical value obtained based on simulation or actual test, and may be adjusted according to actual operating channel conditions. The "gap" may be measured in a variety of ways, one example of which is Mean Square Error (MSE). For example, assuming that the number of subcarriers is N, the channel estimation values of two frames before and after are respectivelyAnd

where i denotes the frame number and k denotes the subcarrier number.

As a preferred embodiment of the present invention, the mean square error can be expressed as follows:

as another preferred embodiment of the present invention, the mean square error may also take another expression:

corresponding to different 'difference' metrics, the preset threshold value mseTH for judging whether locking is different. When the conditions that the mse estimated value is less than or equal to mseTH are met in the detection of the continuous LOCK _ DECISION beacon frames, the channel frequency domain response estimation processing is judged to be converged, and the step ten is carried out; if the condition of mse < ═ mseTH is not satisfied after a certain time, the convergence of the channel estimation process is determined to fail, and the channel training procedure is exited.

The preset threshold value can be obtained through simulation or according to an actual test result, and is adjusted according to the channel condition in actual working.

The signal-to-noise ratio estimation of each subcarrier can be carried out while the channel frequency domain response estimation is carried out. There are many algorithms for estimating the snr of the sub-carrier in the OFDM system, and for engineers skilled in the art to find feasible implementation methods from the existing literature, the details of which are not repeated here.

Step ten, after the channel frequency domain response estimation processing enters a convergence state, the receiver enters a channel condition tracking state. On the other hand, the system may start normal data communication, for example, information exchange with the central node for accessing the network may be started. And in the normal data communication process, the received data frame is equalized and demodulated by using the current channel frequency domain response estimation value.

In a channel tracking state, a receiver periodically and continuously detects and receives each beacon frame, performs first power estimation processing, in-band narrowband interference detection processing, second power estimation processing, sampling clock frequency offset estimation processing, carrier frequency offset fine estimation processing, channel frequency domain response estimation and updating processing on the basis of each detected beacon frame, updates corresponding compensation values according to estimation results, and performs corresponding compensation processing on received data on the basis of the updated compensation values. The estimation processes and the compensation value updating process are the same as those described in the above steps (but it is no longer necessary to determine whether the convergence of the estimation processes is successful or failed), and are not described herein again.

In the case where the channel characteristics have slowly time-varying characteristics, detection estimation is performed once on a per detected beacon frame basis to update various estimation values for compensating for time-varying conditions in which processing is sufficient to track the channel.

The present invention has been described in detail with reference to the embodiments and examples, and the purpose of the present invention is to facilitate understanding of the contents of the present invention and to implement the same, but the present invention is not limited to the contents. Those skilled in the art will understand that: various alternatives, variations and modifications are possible without departing from the spirit and scope of the invention and the appended claims. Various alterations, changes and modifications are intended to be within the scope of the claims without departing from the principles of the invention.

Claims

1. A channel training method based on beacon frames comprises the following steps:

step one, the central node equipment periodically sends beacon frames;

starting sampling clock frequency offset tracking estimation and compensation based on initial estimation of sampling clock frequency offset, starting up and down beat adjustment of a time base counter, and simultaneously starting first power control processing; after the first power control is converged, turning to step four; if the first power control is not converged within a certain time, judging that the convergence fails, and exiting the channel training process;

step four, estimating the in-band narrow-band interference, and if the in-band narrow-band interference is detected, turning to step five; otherwise, directly jumping to the step six;

step six, performing second power control processing on the digital baseband signals after low-pass filtering and notch filtering processing, and adjusting the average power of the digital baseband signals after low-pass filtering and notch filtering processing to the optimal input average power required by a subsequent processing module; after the second power control processing is converged, turning to step seven; if the second power control is not converged within a certain time, judging that the convergence fails, and exiting the channel training process;

step seven, starting carrier frequency offset rough estimation and compensation processing to ensure that the carrier frequency offset is converged to a smaller range from an initial large value; stopping the carrier frequency offset rough estimation and the compensation processing after the carrier frequency offset rough estimation and the compensation processing are judged to be converged, and then turning to the step eight; if the carrier frequency offset rough estimation and compensation processing is not converged within a certain time, determining that the convergence fails and exiting the channel training process;

step eight, starting carrier frequency offset fine estimation and compensation processing, and turning to the step nine after judging that the carrier frequency offset fine estimation and the compensation processing are converged; if the carrier frequency offset fine estimation and compensation processing is not converged within a certain time, determining that the convergence fails, and exiting the channel training process;

step ten, the receiver enters a channel condition tracking state.

2. The channel training method of claim 1, wherein: the beacon frame initial synchronization search described in the step two adopts the following method to make validity judgment,

suppose the beacon frame period is T_beaconAnd the synchronization moments of the detected two consecutive beacon frame headers are t1 and t2 respectively, and the beacon frame header is considered to be detected if the following relationship is satisfied between the detected two beacon frame headers:

|t2-t1-T_beacon|≤deltaTH

wherein, the decision threshold deltaTH is based on the maximum initial sampling time offset delta possibly occurring in the system_maxAnd T_beaconCalculating to obtain:

deltaTH＝δ_max·T_beacon/10⁶；

<math> <mrow> <mi>clk</mi> <mo>_</mo> <mi>ofst</mi> <mo>_</mo> <mi>ppm</mi> <mo>=</mo> <mfrac> <mrow> <mi>t</mi> <mn>2</mn> <mo>-</mo> <mi>t</mi> <mn>1</mn> <mo>-</mo> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mrow> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mfrac> <mo>·</mo> <msup> <mn>10</mn> <mn>6</mn> </msup> <mrow> <mo>(</mo> <mi>ppm</mi> <mo>)</mo> </mrow> <mo>.</mo> </mrow> </math>

3. the channel training method of claim 1, wherein: the beacon frame initial synchronization search described in the step two adopts the following method to make validity judgment,

suppose the beacon frame period is T_beaconThe detected synchronization time of the continuous N beacon frame headers is t respectively₁、t₂，…，t_N-1And if the following relation is satisfied between them, the beacon frame header is considered to be detected:

|t_k-t_k-1-T|≤deltaTH1，k＝2，...N

|(t_k-t_k-1)-(t_k+1-t_k)|≤deltaTH2，k＝2，...N-1

wherein N is more than or equal to 3, the first decision threshold deltaTH1 is set according to the maximum initial sampling time offset delta which can occur in the system_maxAnd T_beaconCalculating to obtain: deltaTH1 ═ delta_max·T_beacon/10⁶；

The second decision threshold deltaTH2 is set independently of the initial sampling time offset, and an empirical value can be obtained through simulation or system test;

<math> <mrow> <mi>clk</mi> <mo>_</mo> <mi>ofst</mi> <mo>_</mo> <mi>ppm</mi> <mo>=</mo> <mfrac> <mn>1</mn> <mrow> <mi>N</mi> <mo>-</mo> <mn>1</mn> </mrow> </mfrac> <mo>·</mo> <munderover> <mi>Σ</mi> <mrow> <mi>k</mi> <mo>=</mo> <mn>2</mn> </mrow> <mi>N</mi> </munderover> <mrow> <mo>(</mo> <mfrac> <mrow> <msub> <mi>t</mi> <mi>k</mi> </msub> <mo>-</mo> <msub> <mi>t</mi> <mrow> <mi>k</mi> <mo>-</mo> <mn>1</mn> </mrow> </msub> <mo>-</mo> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mrow> <msub> <mi>T</mi> <mi>beacon</mi> </msub> </mfrac> <mo>)</mo> </mrow> <mo>·</mo> <msup> <mn>10</mn> <mn>6</mn> </msup> <mrow> <mo>(</mo> <mi>ppm</mi> <mo>)</mo> </mrow> <mo>.</mo> </mrow> </math>

4. the channel training method of claim 1, wherein: step three, the first power control processing comprises first power estimation processing and first power gain adjustment based on a first power estimation result;

the first power estimation processing is to perform average power estimation on digital signals of a beacon frame part detected by digital intermediate frequency signals after analog-to-digital conversion, or perform average power estimation on digital signals of a beacon frame part detected by digital baseband signals after carrier frequency offset correction and down-conversion processing, and a first power estimation result is calculated in the following way:

where N is the number of data samples used for average power estimation, and r (k) is the data sample corresponding to the received signal portion of the beacon frame;

the first power gain adjustment mode is as follows: average power P of received signal estimated based on digital signal corresponding to the first beacon frame part_aver(l) And a reference target power P_refPower error value P_error(l)＝P_aver(l)-P_refCounting, if the power error value P is M1 times continuously_error(l) If the power gain value is larger than the preset threshold PdeltatH1, the power gain value is adjusted downwards by the gain step size gainstep₁(ii) a If the power error value is less than the preset threshold-PdeltatH 1M 1 times in succession, the power gain value is adjusted upward by the gain step gainstep₁(ii) a If the power error value P is M2 times in succession_error(l) If the power gain value is larger than the preset threshold PdeltatH2, the power gain value is adjusted downwards by the gain step size gainstep₂(ii) a If the power error value is less than the preset threshold-PdeltatH 2M 2 times in succession, the power gain value is adjusted upward by the gain step gainstep₂(ii) a … if M is consecutive_NThe power error value P_error(l) Greater than a preset threshold PdeltatH_NThen the power gain value is adjusted downward by the gain step gainstep_N(ii) a If M is continuous_NThe power error value is smaller than the preset threshold-PdeltatH_NThen adjust the power gain value up by the gain step gainstep_N(ii) a Otherwise, the current power gain value is not adjusted;

wherein M1 < M2 < … < M_N，PdeltaTH1＞PdeltaTH2＞…＞PdeltaTH_N，gainstep₁＞gainstep₂＞…＞gainstep_N(ii) a Priority and adjusting step length gainstep of each level control₁～gainstep_NThe sizes are the same; judging the adjusting conditions according to the high-low order of the priority in each level of control, and judging the adjusting conditions with low priority only when the adjusting conditions with high priority are not satisfied; once the adjustment condition for a certain level is satisfied and power gain adjustment is performed, all statistical variables are cleared and the next round of power gain adjustment is started from the beginning.

5. The channel training method of claim 1, wherein: the method for determining whether the first power control converges in step three is to count the power gain adjustment condition, and if no step adjustment occurs in consecutive M11 frames, the first power control is considered to have entered the convergence state, otherwise, if the first power control still does not enter the convergence state after M12 frames, the first power control fails to converge.

6. The channel training method of claim 1, wherein: performing fast Fourier transform on the time domain data of the OFDM symbol part of the beacon frame body to obtain corresponding frequency domain data, comparing the signal amplitude of each subcarrier with the average amplitude amp _ aver of all effective subcarriers, and if the difference between the signal amplitude amp (i) of the ith subcarrier and the amp _ aver is greater than a preset in-band narrowband interference detection threshold nbi _ detect _ TH, considering that the in-band narrowband interference exists at the frequency point corresponding to the subcarrier; wherein nbi _ detect _ TH can be obtained through simulation or system test, or can be determined according to the performance requirement of the system for resisting the in-band narrow-band interference and the processing gain provided by FFT.

7. The channel training method of claim 6, wherein: and averaging the fast Fourier transform results of a plurality of OFDM symbols, and then judging the in-band narrow-band interference estimation.

8. The channel training method of claim 1, wherein: step six, the second power control processing comprises second power estimation processing and second power gain adjustment based on a second power estimation result;

the second power estimation process calculates a second average power as follows:

wherein N is the number of data samples used for average power estimation, and r2(k) is the data samples corresponding to the received signal portion of the beacon frame after notch processing and low pass filtering processing;

the second power gain adjustment mode is as follows: second average power P of received signal estimated based on data received in the first beacon frame_aver，2(l) And a preset second reference target power P_ref，2Power error value P_error，2(l)＝P_aver，2(l)-P_ref，2Counting, if the power error value P is continuously L1 times_error，2(l) If the power gain value is larger than the preset threshold dagcTH1, adjusting the power gain value downwards by a gain step dagcstep 1; if the power error value is less than the preset threshold-dagcTH 1L 1 consecutive times, the power gain value is adjusted up by the gain step dagcstep 1; if the power error value P is continuously L2 times_error，2(l) If the power gain value is larger than the preset threshold dagcTH2, adjusting the power gain value downwards by a gain step dagcstep 2; if the power error value is less than the preset threshold-dagcTH 2L 2 consecutive times, the power gain value is adjusted up by the gain step dagcstep 2; … if L is continuous_NThe power error value P_error，2(l) Is greater than a preset threshold dagcTH_NThen adjust the power gain value downward by the gain step length dagcstep_N(ii) a If L is continuous_NThe power error value is smaller than the preset threshold-dagcTH_NThen adjust the power gain value upward by the gain step dagcstep_N(ii) a It is composed ofThe current power gain value is not adjusted under the rest conditions;

wherein, L1 < L2 < … < L_N，dagcTH1＞dagcTH2＞…＞dagcTH_N，dagcstep1＞dagcstep2＞…＞dagcstep_N；

The priority of each level of control is the same as the size of the gain step, the adjustment condition judgment is carried out in each level of control according to the priority order, and the adjustment condition judgment with low priority is carried out only when the adjustment condition with high priority is not satisfied; once the adjustment condition for a certain level is satisfied and power gain adjustment is performed, all statistical variables are cleared and the next round of power gain adjustment is started from the beginning.

9. The channel training method of claim 1, wherein: the method for judging whether the second power control is converged in the sixth step is to count the adjustment condition of the gain of the second power control, and if no adjustment of any gain step length occurs in the continuous L11 frames, the second power control is considered to have entered the convergence state, otherwise, if the second power control still does not enter the convergence state after the L12 frames, the second power control is judged to have failed to be adjusted.

10. The channel training method of claim 1, wherein: the method for carrier frequency offset rough estimation and compensation processing in step seven comprises the following steps,

the received signal of the frame synchronization sequence corresponding to the frame header of the beacon frame is marked as r (k), the original frame synchronization sequence transmitted at the transmitting end is marked as M (k), and the received signal is changed into M (k) e after modulation^jφ(k)Length of frame sync sequence is marked as N' ()^*It is shown that the conjugate is calculated for a complex number,

step1, removing the modulation information in the received signal, and synchronizing the modulated frame synchronization sequence M (k) e^jφ(k)The conjugate multiplication corresponding to the symbol is performed with the corresponding received signal r (k), as shown in the following formula:

r′(k)＝M^*(k)·e^jφ(k)·r(k)k＝0，...，N-1

step2, two sections of continuous signals are taken from the received signal r' (k) after the modulation information is removed, and delay correlation corresponding to symbols is carried out, as shown in the following formula:

y(k)＝(r′(k))^*·r′(k+K)k＝m，...，n

where 0 ≦ m, N ≦ N-1 defines a range for two consecutive signals taken from the received signal r' (K) for delay correlation, separated by a distance K, which may partially overlap;

step 3, calculating the phase of the delay correlation result y (k), and then calculating the average to obtain a carrier frequency offset coarse estimation value cfe (i) based on the current beacon frame, as shown in the following formula:

wherein E () represents averaging, and angle () represents phase of complex signal;

step 4, performing statistical decision in the following manner on the carrier frequency offset coarse estimation value cfe (i) obtained based on each beacon frame to obtain a compensation amount cfe _ offset for carrier frequency offset compensation:

initializing cfe _ fofst to 0 at the beginning; if K1 times of continuous coarse carrier frequency offset estimation values cfe (i) are greater than a threshold value cfeTH1, adjusting the compensation amount cfe _ fofst upwards by a compensation step size cfestep 1; if K1 times of the carrier frequency offset coarse estimation value cfe (i) is less than the threshold-cfeTH 1, adjusting the compensation amount cfe _ fofst downwards by the compensation step cfestep 1; if K2 consecutive times that the carrier frequency offset coarse estimate cfe (i) is greater than the threshold cfeTH2,the compensation amount cfe _ fofst is adjusted upward by the compensation step cfestep 2; if K2 times of the carrier frequency offset coarse estimation value cfe (i) is less than the threshold-cfeTH 2, adjusting the compensation amount cfe _ fofst downwards by the compensation step cfestep 2; … if K is consecutive_NThe carrier frequency offset coarse estimation value cfe (i) is greater than the threshold value cfeTH_NThen, the compensation amount cfe _ fofst is adjusted downward by the compensation step cfestep_N(ii) a If K is continuous_NThe carrier frequency offset coarse estimation value cfe (i) is less than the threshold-cfeTH_NThen, the compensation amount cfe _ fofst is adjusted upward by the compensation step cfestep_N(ii) a The compensation amount cfe _ fofst is not adjusted under other conditions;

wherein K1 is more than K2 is more than K …_N，cfeTH1＞cfeTH2＞…＞cfeTH_N，cfestep1＞cfestep2＞…＞cfestep_N(ii) a The priority of each level of adjustment is the same as the step size, the adjustment condition judgment is carried out in each level of adjustment according to the priority order, and the judgment of the adjustment condition with low priority is carried out only when the adjustment condition with high priority is not satisfied; once the adjustment condition of a certain priority is satisfied and the carrier frequency offset compensation amount adjustment is performed, all statistical variables are cleared, and the coarse frequency offset estimation and the corresponding compensation amount adjustment of the next round are started from the beginning.

11. The channel training method of claim 1, wherein: seventhly, the method for judging the carrier frequency offset rough estimation and the compensation processing convergence is to count the adjustment condition of the carrier frequency offset compensation amount cfe _ fofst, and if no adjustment of the carrier frequency offset compensation amount cfe _ fofst occurs in the continuous K11 frames, the adjustment of the carrier frequency offset compensation amount cfe _ fofst is considered to enter the convergence state, and the rough frequency offset estimation processing is stopped; and if the carrier frequency offset compensation amount cfe _ fofst adjustment still does not enter a convergence state after K21 frames, determining that the carrier frequency offset rough estimation and compensation processing fails to converge.

12. The channel training method of claim 1, wherein: the method for fine estimation and compensation of carrier frequency offset in step eight comprises the following steps,

step1, performing fast fourier transform on a beacon frame body, and transforming time domain data to a frequency domain to obtain frequency domain data x (l, m), wherein l is 0, 1 is an OFDM symbol sequence number in the beacon frame body, and m is a subcarrier sequence number in one OFDM symbol;

step2, demodulating the equalization processing result x (l, m) to obtain a demodulation result bit stream z (n);

step 3, coding and mapping z (n) again to obtain w (l, m);

step 4, calculating the phase difference phi (l, m) between x (l, m) and w (l, m) as x^*(l，m)·w(l，m)；

Step 5, calculating an average value of phase rotation amounts of corresponding subcarriers between every two adjacent symbols of the N OFDM symbols, as shown in the following formula:

wherein, i represents the number of the beacon frame, N is the number of OFDM symbols in the beacon frame, Ω represents the range of subcarriers selected to participate in the calculation, and sizeof (Ω) represents the number of subcarriers included in Ω;

step 6, carrying out statistical decision on the carrier frequency offset fine estimation value phi _ diff (i) obtained based on each beacon frame according to the following mode to obtain the compensation amount ffe _ offset for carrier frequency offset compensation: ffe _ fosst is initially initialized to 0; if the fine carrier frequency offset estimation value phi _ diff (i) is greater than the threshold ffeTH1 for Q1 times continuously, adjusting the compensation amount ffe _ fofst up by the compensation step length ffesep 1; if the fine carrier frequency offset estimation value phi _ diff (i) is less than the threshold value-ffeTH 1 for Q1 times continuously, adjusting the compensation amount ffe _ fofst downwards by a compensation step length ffesep 1; if the fine carrier frequency offset estimation value phi _ diff (i) is greater than the threshold ffeTH2 for Q2 times continuously, adjusting the compensation amount ffe _ fofst up by the compensation step length ffesep 2; if the fine carrier frequency offset estimation value phi _ diff (i) is less than the threshold-ffeTH 2 for Q2 times continuously, adjusting the compensation amount ffe _ fofst downwards by the compensation step length ffesep 2; … if Q continues_NThe fine estimated value of the carrier frequency deviation phi _ diff (i) is greater than the threshold ffeTH_NThen the compensation amount ffe _ fosst is adjusted upward by the compensation step length ffesep_N(ii) a If Q is continuous_NThe fine estimated value of the carrier frequency deviation phi _ diff (i) is smaller than the threshold value-ffeTH_NThen the compensation amount ffe _ fosst is adjusted downward by the compensation step length ffesep_N(ii) a The compensation amount cfe _ fofst is not adjusted under other conditions;

wherein Q1 < Q2 < … < Q_N，ffeTH1＞ffeTH2＞…＞ffeTH_N，ffestep1＞ffestep2＞…＞ffestep_N(ii) a The priority of each level of adjustment is the same as the step size, the adjustment condition judgment is carried out in each level of adjustment according to the order of the priority, and the judgment of the adjustment condition with low priority is carried out only when the adjustment condition with high priority is not satisfied; once the adjustment condition of a certain priority is satisfied and the carrier frequency offset compensation adjustment is performed, all the statistical variables are cleared, and the carrier frequency offset compensation adjustment of the next round is started from the beginning.

13. The channel training method of claim 12, wherein: and after the subcarriers in the middle of the frequency domain and at two ends of the frequency domain and the subcarriers in a certain range around the frequency point position of the narrow-band interference are removed from all the effective subcarriers, the residual subcarriers are utilized to carry out fine estimation on the frequency offset of the carrier.

14. The channel training method of claim 1, wherein: the method for judging the carrier frequency offset fine estimation and compensation processing convergence in the step eight is to count the adjustment condition of the carrier frequency offset compensation amount ffe _ fofst, and if no adjustment of the carrier frequency offset compensation amount ffe _ fofst occurs in the continuous Q11 frames, the carrier frequency offset fine estimation and compensation processing are considered to enter the convergence state; and if the carrier frequency offset fine estimation and compensation processing still does not enter a convergence state after the Q21 frame, judging that the carrier frequency offset fine estimation and compensation processing fails to converge.

15. The channel training method of claim 1, wherein: step nine the method adopted by the channel frequency domain response estimation process comprises the following steps,

step1, carrying out FFT (fast Fourier transform) on the frame body part of the beacon frame to obtain frequency domain data

Wherein i represents the frame number of the beacon frame, n marks the OFDM symbol number in the beacon frame body, and k represents the subcarrier number in the OFDM symbol;

step2, pair

Using channel estimates obtained up to the last beacon frameCarrying out equalization to obtain an equalization result

Wherein,

step 3, pair

Carrying out differential quadrature phase shift keying DQPSK demodulation to obtain demodulation result

Step 4, pair

Carrying out DQPSK modulation again to obtain DPQSK symbol sequence

Step 5, based on

Andprocessed by the unmodulating information

Obtained after removal of modulation information

Namely representing the residual channel frequency domain response information of each subcarrier;

step 6, averaging the residual channel frequency domain responses of a plurality of OFDM symbols to obtain an average residual channel frequency domain response, wherein the calculation mode is as follows:

wherein, N is the number of OFDM symbols contained in the beacon frame;

step 7, the average residual channel frequency domain response and the channel estimation value until the last beacon frame are obtained

Synthesizing to obtain the channel estimation value of the current frame

And 8, estimating channel frequency domain response based on each beacon frame, and after updating the channel frequency domain response, judging whether the channel frequency domain response estimation processing is converged.

16. The channel training method of claim 15, wherein: the convergence judgment of the channel frequency domain response estimation processing adopts the following method, an empirical value CHE _ ACQ _ FRAME of the FRAME number required by the convergence of the channel frequency domain response estimation is determined through simulation, and after the channel frequency domain response estimation is started, if the CHE _ ACQ _ FRAME FRAME is processed, the channel frequency domain response estimation and the signal-to-noise ratio estimation are judged to enter a convergence state.

17. The channel training method of claim 15, wherein: the convergence judgment of the channel frequency domain response estimation processing adopts the following method, the channel frequency domain response estimation result of the previous frame and the 'difference' of the updating result after the current frame is processed are compared, if the 'difference' is smaller than a certain preset threshold value and the continuous LOCK _ DECISION frames meet the condition, the channel frequency domain response estimation and the signal-to-noise ratio processing are judged to enter a convergence state, wherein the LOCK _ DECISION is an empirical value obtained based on simulation or actual test.

18. The channel training method of claim 17, wherein: the difference is mean square error, and the following calculation mode is adopted:

wherein, the number of the sub-carriers is N, and the channel estimation values of the two frames are respectively

And

i denotes a frame number and k denotes a subcarrier number.

19. The channel training method of claim 17, wherein: the difference is mean square error, and the following calculation mode is adopted:

wherein, the number of the sub-carriers is N, and the channel estimation values of the two frames are respectivelyAnd

i denotes a frame number and k denotes a subcarrier number.

20. A beacon frame based channel training receiver apparatus, comprising:

an interpolation filter for performing rate conversion on the digital baseband signal processed by the notch filter in an interpolation mode and converting the digital signal with a sampling rate corresponding to the sampling rate of the A/D converter into data with a symbol rate;