CN100541998C - switching power supply circuit - Google Patents

Abstract

The invention discloses a switching power supply circuit comprising a switch element, a converter transformer, a secondary side rectification and smoothing circuit, a switch element control unit, a choke coil, a primary side series resonant circuit, a primary side parallel resonant circuit and a series circuit. The switching element enables switching of a direct current (DC) voltage to transform the voltage into an alternating current (AC) voltage. The choke coil is supplied with voltage and connected to one winding end of the primary winding and one terminal of the switching element. The primary side series resonant circuit connects the primary side series resonant capacitor and has a resonant frequency dominated by leakage inductance. The primary side parallel resonance circuit connects the primary side parallel resonance capacitor to the switching element in parallel and has a resonance frequency. A series circuit is formed by a clamping capacitor and an auxiliary switching element conducting during the non-conducting period of the switching element.

Description

switching power supply circuit

technical field

The present invention relates to switching power supply circuits.

Background technique

As types of so-called soft-switching power supplies using a resonant converter, a current resonance type and a voltage resonance type have been widely known. Currently, a half-bridge type current resonant converter formed of two transistor switching elements has been widely used because it can be easily put into practical use.

However, since the characteristics of, for example, high breakdown voltage switching elements are currently being improved, problems related to breakdown voltage associated with putting voltage resonant converters into practical use are also being solved. Furthermore, it is known that a single-ended voltage resonant converter formed by a switching element of one transistor is superior to a current resonant forward converter of one transistor in terms of noise components of the DC output voltage line and input feedback noise.

FIG. 16 illustrates a configuration example of a switching power supply circuit including a single-ended voltage resonance converter. This voltage resonance converter is combined with a secondary side series resonance circuit formed by a leakage inductance L2 of a secondary winding and a secondary side series resonance capacitor C2 to be described later, and is therefore called a multiple resonance converter.

In the switching power supply circuit of Fig. 16, the voltage from the commercial AC power supply AC is rectified and smoothed by the rectification and smoothing circuit formed by the bridge rectification circuit Di and the smoothing capacitor Ci, thereby as the voltage across the smoothing capacitor Ci, a DC input is generated Voltage Ei. The line from the commercial current AC has a noise filter comprising a pair of common mode choke coils CMC and two cross-line capacitors CL, which removes common mode noise.

The DC input voltage Ei is input to the voltage resonant converter as a DC input voltage. The voltage resonant converter has a single-ended configuration comprising one transistor switching element Q1 as described above. The voltage resonant converter in this circuit is separately excited. Specifically, a switching element Q1 formed of a MOSFET is driven in a switching manner by the oscillation and drive circuit 2 .

The body diode DD1 of the MOSFET is connected in parallel to the switching element Q1. In addition, a primary-side parallel resonance capacitor Cr is connected in parallel to the channel between the drain and the source of the switching element Q1. The primary side parallel resonance capacitor Cr and the leakage inductance L1 of the primary winding N1 in the isolation converter transformer form a primary side parallel resonance circuit (voltage resonance circuit). As a switching operation of the switching element Q1, the primary side parallel resonance circuit provides a voltage resonance operation.

To drive the switching element Q1 in a switching manner, the oscillation and driving circuit 2 applies a gate voltage as a driving signal to the gate of the switching element Q1. Thus, the switching element Q1 realizes a switching operation with a switching frequency depending on the period of the driving signal.

The isolation converter transformer PIT sends the switching output from the switching element Q1 to the secondary side. As shown in FIG. 17, the isolated converter transformer PIT is constituted by an EE-shaped core formed by combining E-shaped cores CR1 and CR2 composed of ferrite material, for example. In addition, the primary winding N1 and the secondary winding N2 are wound on the bobbin B covering the central magnetic leg of the EE-shaped core, wherein the winding portion is divided into a primary side and a secondary side. In addition, a gap G of about 0.8 to 1.0 mm in length is provided in the center leg of the EE-shaped core of the isolation converter transformer PIT, thereby obtaining a coupling coefficient k of about 0.80 to 0.85 between the primary side and the secondary side. When the coupling coefficient k has such a value, the degree of coupling between the primary and secondary sides can be regarded as weak coupling, making it difficult to reach a saturated state. The value of the coupling coefficient k is a factor when setting the leakage inductance (the inductance of the leakage inductance L1).

The primary winding N1 in the isolation converter transformer PIT is interposed between the switching element Q1 and the positive electrode of the smoothing capacitor Ci, which allows switching output to be delivered from the switching element Q1. In the secondary winding N2 of the isolation converter transformer PIT, an AC voltage induced by the primary winding N1 is generated.

On the secondary side, the secondary side series resonant capacitor C2 is connected in series with one end of the secondary winding N2, therefore, the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C2 form a secondary side series resonant circuit (current resonant circuit).

Furthermore, rectification diodes Do1 and Do2 and a smoothing capacitor Co are connected to this secondary side series resonance circuit as shown in the figure, thereby forming a voltage doubler half-wave rectification circuit. As a voltage across the smoothing capacitor Co, the voltage doubler half-wave rectification circuit generates a DC output voltage Eo having a level twice the secondary winding voltage V3 induced in the secondary winding N2. The DC output voltage Eo is supplied to a load, and is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 detects the level of the DC output voltage Eo input as a detection voltage, and then inputs the obtained detection output to the oscillation and drive circuit 2 . The oscillation and driving circuit 2 outputs a driving signal whose frequency etc. vary depending on the level of the DC output voltage Eo indicated by the detection output, thereby controlling the switching operation of the switching element Q1 so that the DC output voltage Eo is at a predetermined level keep constant. Thereby, stable control of the DC output voltage Eo is realized.

18A to 18C and 19 show experimental results on the power supply circuit having the configuration shown in FIG. 16 . For these experiments, the main components in the power supply circuit of Fig. 16 were designed with the following parameters.

The core of the isolation converter transformer PIT adopts an EER-35 core, and the gap in its central stem is designed to have a gap length of 1 mm. The numbers of turns of the primary winding N1 and the secondary winding N2 are set to 39T and 23T, respectively. The induced voltage level per turn (T) in the secondary winding N2 is set to 3V/T. The coupling coefficient k of the isolation converter transformer PIT is set to 0.81.

The capacitance of the primary-side parallel resonance capacitor Cr is set to 3900 pF (picofarad). The capacitance of the secondary-side series resonance capacitor C2 is set to 0.1 μF (microfarad). Therefore, the primary side parallel resonance frequency fo1 of the primary side parallel resonance circuit is set to 230 kHz (kilohertz), and the secondary side series resonance frequency fo2 of the secondary side series resonance circuit is set to 82 kHz. Therefore, the relative relationship between the parallel resonance frequency fo1 on the primary side and the series resonance frequency fo2 on the secondary side can be expressed as

The rated level of the DC output voltage Eo is 135V. The allowable load power ranges from the maximum load power Pomax of 200W to the minimum load power Pomin of 0W.

18A to 18C are waveform diagrams showing operations of main components in the power supply circuit shown in FIG. 16, which reflect respective switching periods of the switching element Q1. 18A shows switching voltage V1, switching current IQ1, primary winding current I2, secondary winding current I3, and rectified currents ID1 and ID2 applied to switching element Q1 when the load power is the maximum load power Pomax of 200W. FIG. 18B shows voltage V1 , switch current IQ1 , primary winding current I2 and secondary winding current I3 when the load power is an intermediate load power Po of 120W. FIG. 18C shows the switch voltage V1 and the switch current IQ1 when the load power is the minimum load power Pomin of 0W.

The voltage V1 is a voltage obtained across the switching element Q1, and has a waveform similar to that in FIGS. 18A to 18C. Specifically, the voltage level is at zero level during the period TON when the switching element Q1 is in the conduction state (ON state), and a sinusoidal resonance pulse is obtained during the period TOFF when it is in the off state (OFF state). The resonant pulse waveform of the switching voltage V1 indicates that the operation of the primary side switching converter is a voltage resonant operation.

Switching current IQ1 is the current flowing through switching element Q1 (and body diode DD1 ). The switching current IQ1 flows in the illustrated waveform during the period TON, and is at a zero level during the period TOFF.

The primary winding current I2 flowing through the primary winding N1 is a current derived from synthesis between the current flowing as the switching current IQ1 during the period TON and the current flowing to the primary side parallel resonance capacitor Cr during the period TOFF. The rectification currents ID1 and ID2 (shown only in FIG. 18A ) flowing through the rectification diodes Do1 and Do2 operating as a secondary-side rectification circuit have sinusoidal waveforms similar to those shown in the figure. In the waveform diagram, the waveform of the rectified current ID1 more mainly indicates the resonant operation of the secondary-side series resonant circuit than the waveform of the rectified current ID2.

The secondary winding current I3 flowing through the secondary winding N2 has a waveform derived from a synthesis between the waveforms of the rectified currents ID1 and ID2. FIG. 19 shows the switching frequency fs, the lengths of the periods TON and TOFF of the switching element Q1, and the AC-to-DC power conversion efficiency (ηAC→DC) of the power supply circuit shown in FIG. 16 as a function of load.

Referring first to the AC to DC power conversion efficiency (ηAC→DC), it is clearly seen that a high efficiency of 90% or more is obtained in a wide range of load power Po from 50W to 200W. The inventors of the present application have previously confirmed based on experiments that such characteristics are obtained when a single-ended voltage resonant converter is combined with a secondary-side series resonant circuit.

In addition, the switching frequency fs, period TON, and period TOFF in FIG. 19 indicate the switching operation of the power supply circuit in FIG. 16 as a characteristic of constant voltage control with respect to load variation. In this power supply circuit, the switching frequency fs is almost constant with respect to load variation. In contrast, periods TON and TOFF show linear changes with opposite trends, as shown in FIG. 19 . These characteristics indicate that the switching operation is controlled so that the time ratio between the on and off periods is changed with respect to the variation of the DC output voltage Eo, keeping the switching frequency (switching period) almost constant. This control can be considered as a pulse width modulation (PWM) control, where the length of the on and off periods within one switching cycle is varied. That is, the power supply circuit in FIG. 16 uses PWM control to stabilize the DC output voltage Eo.

FIG. 20 schematically shows the constant voltage control characteristics of the power supply circuit shown in FIG. 16 based on the relationship between the switching frequency fs (kHz) and the DC output voltage Eo.

因此次级侧串联谐振频率fo2低于初级侧并联谐振频率fo1，同样如图20所示。The power supply circuit shown in Fig. 16 includes a primary side parallel resonant circuit and a secondary side series resonant circuit, and thus has two resonant impedance characteristics in a composite manner: a resonant impedance corresponding to the primary side parallel resonant frequency fo1 of the primary side parallel resonant circuit characteristics and resonance impedance characteristics corresponding to the secondary side series resonance frequency fo2 of the secondary side series resonance circuit. Since the power supply circuit in Figure 16 has a frequency relationship

The secondary side series resonant frequency fo2 is therefore lower than the primary side parallel resonant frequency fo1 as also shown in FIG. 20 .

The characteristic curve in Fig. 20 shows the constant voltage control characteristic which is dependent on the control of the switching frequency ^ and is assumed based on these resonant frequencies and under the condition of a certain constant input AC voltage VAC. Specifically, characteristic curves A and B correspond to maximum load power Pomax and minimum load power Pomin, respectively, and indicate constant voltage control characteristics related to resonance impedance corresponding to primary side parallel resonance frequency fo1 of the primary side parallel resonance circuit. Characteristic curves C and D correspond to maximum load power Pomax and minimum load power Pomin, respectively, and indicate constant voltage control characteristics related to resonance impedance corresponding to secondary side series resonance frequency fo2 of the secondary side series resonance circuit. Under the characteristics shown in FIG. 20, when the constant voltage control intends to keep the output voltage at the voltage tg which is the rated level of the DC output voltage Eo, the variation range of the switching frequency ^ required for the constant voltage control (necessary control range ) can be represented by the interval indicated by Δfs.

The control range Δfs shown in Fig. 20 is from the frequency of the supplied voltage level tg on the characteristic curve C (corresponding to the secondary side series resonant frequency fo2 and the maximum load power Pomax of the secondary side series resonant circuit), to the characteristic curve The frequency at which the voltage level tg is supplied at B (corresponding to the primary side parallel resonant frequency fo1 and the minimum load power Pomin of the primary side parallel resonant circuit). The range Δfs intersects the characteristic curve D which corresponds to the secondary side series resonant frequency fo2 and the minimum load power Pomin of the secondary side series resonant circuit and the characteristic curve A which corresponds to the primary side parallel resonant circuit The primary side parallel resonant frequency fo1 and the maximum load power Pomax.

Therefore, as a constant voltage control operation, the power supply circuit in Fig. 16 realizes switch drive control based on PWM control in which the time ratio (ratio between periods TON and TOFF) in one switching cycle is changed while the switching frequency fs is kept almost constant. The realization of PWM control is also indicated by Figs. 18A to 18C, where the widths of the periods TOFF and TON vary depending on the load power, while the length of one switching cycle (TOFF+TON) at Pomax=200W and Po=120W is irrespective of the load power Change is almost always constant.

This operation is due to this resonant impedance characteristic of the power supply circuit with respect to the load variation, in which the transition between the two states is achieved within a narrow switching frequency range (Δfs), where in The resonance impedance (capacitive impedance) corresponding to the primary side parallel resonance frequency fo1 of the primary side parallel resonance circuit is dominant in one state, and the secondary side series resonance corresponding to the secondary side series resonance circuit in the other state The resonant impedance (inductive impedance) of frequency fo2 dominates.

The technology related to the present invention is disclosed in, for example, Japanese Patent Application Laid-Open Publication No. 2000-134925.

Contents of the invention

The power supply circuit in Fig. 16 involves the following issues.

Returning to the above-mentioned waveform diagrams of FIGS. 18A to 18C, the switching current IQ1 shown in FIG. 18A when the load power is the maximum load power Pomax is expressed as follows. Specifically, the switching current IQ1 is at a zero level until the end of the period TOFF, that is, the turning-on moment of the switching element Q1. When the period TON starts, a current of negative polarity initially flows through the body diode DD1, and then the polarity is reversed so that the switching current IQ1 flows between the drain and the source of the switching element Q1. Zero-voltage switching (ZVS) is properly performed in a state indicated by this operation.

In contrast, the switching current IQ1 shown in FIG. 18B when the load power is the intermediate load power Po of 120 W shows a waveform in which the noise current flows at the timing immediately before the end of the period TOFF, which is the switching element The on-time of Q1. This waveform indicates abnormal operation where ZVS is not properly implemented.

That is, it is known that a voltage resonance converter including a secondary-side series resonance circuit as shown in FIG. 16 involves an abnormal operation in which ZVS is not properly realized when the load is an intermediate load. It has been confirmed that, in the actual power supply circuit of FIG. 16 , such an abnormal operation occurs, for example, in the load variation range indicated by part A in FIG. 19 .

A voltage resonant converter including a secondary-side series resonant circuit inherently tends to have the characteristic of advantageously maintaining high efficiency with respect to load variations as described above. However, as shown by the switching current IQ1 of FIG. 18B , correspondingly a peak current flows at the turn-on timing of the switching element Q1 . This noise current causes an increase in switching loss, which is a factor that lowers power conversion efficiency.

In addition, the occurrence of such an abnormal operation causes, for example, a shift in the phase gain characteristic of the constant voltage control circuit, which results in a switching operation in an abnormal oscillation state. Therefore, there is currently a strong consensus that it is difficult to put voltage resonant converters into practical use.

In consideration of the above-mentioned problems, one embodiment of the present invention provides a switching power supply circuit having the following configuration. Specifically, the switching power supply circuit includes switching elements, a converter transformer, and a secondary-side rectification and smoothing circuit. The switching element realizes the switching of the DC voltage, thereby converting the DC voltage into an AC voltage. The converter transformer inputs an AC voltage into the primary winding, causing an AC voltage to be generated in the secondary winding. The secondary side rectification and smoothing circuit includes a secondary side rectification element and a secondary side smoothing capacitor for rectifying and smoothing the AC voltage generated in the secondary winding to generate an output DC voltage, and also includes controlling a switching element based on the output DC voltage switching element control unit. The switching power supply circuit further includes a choke coil supplied with a DC voltage through one end and connected through the other end to one winding end of the primary winding in the converter transformer and one terminal of the switching element. The switching power supply circuit further includes a primary side series resonance circuit formed by connecting a primary side series resonance capacitor between the other winding end of the primary winding in the converter transformer and the other terminal of the switching element, And has a resonant frequency dominated by leakage inductance developed in the primary winding in the converter transformer and said primary side series resonant capacitor. The switching power supply circuit also includes a primary-side parallel resonance circuit formed by connecting a primary-side parallel resonance capacitor in parallel with the switching element and having an Resonant frequency. The switching power supply circuit also includes a series circuit formed of a clamp capacitor and an auxiliary switching element, and connected in parallel to the choke coil. The auxiliary switching element conducts when the switching element is in a non-conducting state.

A switching power supply circuit based on the configuration described above includes a switching element, a converter transformer, a secondary side rectification and smoothing circuit, and a switching element control unit. The switching element realizes the switching of the DC voltage, thereby converting the DC voltage into an AC voltage. The converter transformer inputs an AC voltage into the primary winding so that an AC voltage is generated in the secondary winding. The secondary side rectification and smoothing circuit includes a secondary side rectification element and a secondary side smoothing capacitor for rectifying and smoothing the AC voltage generated in the secondary winding to generate an output DC voltage. The switching element control unit controls the switching element based on the output DC voltage. Thus, in this switching power supply circuit, AC power is converted into DC power, and then DC power is converted into AC power through the switching elements controlled by the switching element control unit, so that a predetermined voltage can be obtained on the secondary side through the converter transformer.

In addition, power is supplied to one winding terminal of the primary winding and one terminal of the switching element in the converter transformer via the choke coil. Therefore, the current supplied from the choke coil is a ripple current close to DC current. In addition, by connecting a series resonant capacitor between the other winding end of the primary winding in the converter transformer and the other terminal of the switching element, a series resonant circuit is formed whose resonant frequency is influenced by the primary winding in the converter transformer The leakage inductance and series resonant capacitor dominate. In addition, a parallel resonance circuit is formed whose resonance frequency is governed by the primary-side parallel resonance capacitor connected in parallel to the switching element and the leakage inductance generated in the primary winding. The formation of these resonance circuits can narrow the variable range of the switching frequency of the switching elements.

In addition, the switching power supply circuit includes a series circuit of an auxiliary switching element and a clamp capacitor connected in parallel to the choke coil. The auxiliary switching element is turned on when the switching element is in a non-conducting state, so the voltage applied to the switching element can be clamped.

According to the embodiments of the present invention, an abnormal operation in which a zero-voltage switching (ZVS) operation is not achieved within a range of intermediate load conditions is eliminated from a switching power supply circuit including a parallel resonance circuit on its primary side.

In addition, a ripple current close to a DC current is obtained as a current flowing into a switching converter from a smoothing capacitor in a rectification and smoothing circuit that generates a rectified and smoothed voltage (DC input voltage) from an AC power source . Therefore, a smaller value can be assigned to the capacitance that is a constituent element of the smoothing capacitor and makes it possible to select a general-purpose product as the smoothing capacitor, which provides advantages such as reduction in cost and size of the smoothing capacitor.

In addition, as described above, a reduction in power loss is achieved by a reduction in the amount of current flowing in the power supply circuit, thereby greatly improving the overall power conversion efficiency characteristics. In addition, switching elements of low breakdown voltage can be used.

Description of drawings

FIG. 1 is a circuit diagram illustrating a basic configuration example of a class E switching converter;

FIG. 2 is a waveform diagram illustrating the operation of a class E switching converter;

3 is a circuit diagram illustrating a configuration example of a switching power supply circuit to which a class E switching converter is applied;

4 is a circuit diagram illustrating a configuration example of a power supply circuit as a first embodiment of the present invention;

FIG. 5 is a diagram illustrating a structural example of the isolation converter transformer of the first embodiment;

6A and 6B are waveform diagrams showing operations of main components in the power supply circuit as the first embodiment while reflecting corresponding switching cycles;

7 is a graph showing change characteristics of AC-to-DC power conversion efficiency and switching frequency of the power supply circuit of the first embodiment as a function of load;

8 is a graph showing change characteristics of AC-to-DC power conversion efficiency and switching frequency of the power supply circuit of the first embodiment as a function of AC input voltage;

FIG. 9 is a diagram illustrating a modification of the secondary side circuit of the first embodiment;

FIG. 10 is a diagram illustrating another modification of the secondary side circuit of the first embodiment;

FIG. 11 is a diagram illustrating a modification of the primary side circuit of the first embodiment;

12 is a circuit diagram illustrating a configuration example of a power supply circuit as a second embodiment of the present invention;

FIG. 13 is a graph showing change characteristics of AC-to-DC power conversion efficiency and switching frequency of the power supply circuit of the second embodiment as a function of load;

FIG. 14 is a diagram illustrating a modification of the secondary-side circuit of the second embodiment;

FIG. 15 is a diagram showing another modification of the secondary side circuit of the second embodiment;

FIG. 16 is a circuit diagram illustrating a configuration example of a power supply circuit as a background art;

17 is a diagram illustrating a structural example of an isolation converter transformer of the background art;

18A to 18C are waveform diagrams showing operations of main components in a power supply circuit shown as background art;

19 is a graph showing variation characteristics of AC-to-DC power conversion efficiency, switching frequency, and lengths of on and off periods of switching elements as a function of load in relation to a power supply circuit shown as a background art;

FIG. 20 is a diagram schematically showing a constant voltage control characteristic of a power supply circuit shown as a background art.

Detailed ways

Before explaining the best mode (hereinafter referred to as embodiment) for carrying out the present invention, a switch realizing class-E (class-E) resonant switching operation will be described below as a background art of the embodiment with reference to FIGS. 1 and 2. Basic configuration of a converter (hereinafter also referred to as a class E switching converter).

Figure 1 illustrates the basic configuration of a Class E switching converter. The class E switching converter in this figure has the same configuration as a DC-AC inverter operating in class E resonant mode.

This class E switching converter includes a transistor switching element Q1. In the converter, the switching element Q1 is a MOSFET. A body diode DD1 is connected in parallel to the channel between the drain and the source of the MOSFET switching element Q1. The forward direction of body diode DD1 is from the source of switching element Q1 to its drain.

In addition, a primary-side parallel resonance capacitor Cr is connected in parallel to the channel between the drain and the source of the switching element Q1. The drain of the switching element Q1 is connected in series to the choke coil L10, and is coupled to the positive electrode of the DC power supply Ein via the choke coil L10. The source of the switching element Q1 is connected to the negative terminal of the DC power supply Ein. The drain of the switching element Q1 is connected to one end of the choke coil L11. The other end of the choke coil L11 is connected in series to the primary side series resonance capacitor C11. An impedance Z as a load is inserted between the primary side series resonance capacitor C11 and the negative electrode of the DC power supply Ein. Specific examples of the impedance Z include piezoelectric transformers and high-frequency compatible fluorescent lamps.

A class E switching converter with this configuration can be considered as a form of a composite resonant converter comprising a parallel resonant circuit formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonant capacitor Cr , and a series resonant circuit formed by the inductance of the choke coil L11 and the capacitance of the primary side series resonant capacitor C11. Furthermore, since a Class E switching converter includes one switching element, it can be regarded as equivalent to a single-ended voltage resonant converter.

FIG. 2 shows the operation of the main components in the class E switching converter shown in FIG. 1 .

The switching voltage V1 is a voltage obtained from both ends of the switching element Q1, and has a waveform like the waveform in FIG. Specifically, the voltage level is at zero level during the period TON in which the switching element Q1 is in the ON state, and a sinusoidal pulse is obtained during the period TOFF in which it is in the OFF state. This switching pulse is caused by the resonant operation (voltage resonant operation) of the above-mentioned parallel resonant circuit.

Switching current IQ1 is the current flowing through switching element Q1 (and body diode DD1 ). During the period TOFF, the switch current IQ1 is at a zero level. During the period TON, the switching current IQ1 has a waveform like the illustrated waveform. Specifically, during a certain period from the beginning of the period TON, the switching current IQ1 initially flows through the body diode DD1 and thus has a negative polarity. Then, the polarity of the current is reversed to the positive polarity, so that the switching current IQ1 flows from the drain of the switching element Q1 to its source. The current I2 flowing through the series resonant circuit, which is the output of the class E switching converter, is derived from the combination of the switching current IQ1 flowing through the switching element Q1 (and body diode DD1 ) and the current flowing to the primary side parallel resonant capacitor Cr, And has a waveform including a sine wave component. The waveforms of the switching current IQ1 and the switching voltage V1 indicate that the ZVS operation is realized at the time when the switching element Q1 is turned off, and the ZVS and ZCS operations are realized at the time when the switching element Q1 is turned on.

The input current I1 flowing from the positive pole of the DC power supply Ein to the Class E switching converter through the choke coil L10 has a ripple waveform with a certain average current level as shown in the figure, because the inductance of the choke coil L10 is set larger than the choke coil L10 Inductance of coil L11. This ripple current can be approximated as a DC current.

The inventors of the present application have constructed a power supply circuit to which a class E switching converter is applied based on the basic configuration described above, and have conducted experiments on the power supply circuit. FIG. 3 is a circuit diagram showing a configuration example of such a power supply circuit.

In the switching power supply circuit in FIG. 3, the line from the commercial AC power supply AC is provided with a pair of common mode choke coils CMC and two cross-line capacitors CL. The common mode choke coil CMC and the cross-line capacitor CL form a noise filter that removes common mode noise added on the line from the commercial power supply AC.

The alternating current from the commercial power supply AC is rectified by the bridge rectifier circuit Di, and the rectified output is charged in the smoothing capacitor Ci. That is, the alternating current is rectified and smoothed by the rectifying and smoothing circuit formed by the bridge rectifying circuit Di and the smoothing capacitor Ci to be converted into direct current. Thus, the DC input voltage Ei is obtained as the voltage across the smoothing capacitor Ci. The DC input voltage Ei serves as the DC input voltage at subsequent stages for the switching converter.

In the power supply circuit of FIG. 3 , a switching converter fed with a DC input voltage Ei as a DC input voltage and realizing a switching operation is formed as a class E switching converter based on the basic configuration of FIG. 1 . In this circuit, a high breakdown voltage MOSFET is selected as the switching element Q1. In addition, the Class E switching converter in this circuit is separately excited. Specifically, the oscillation and drive circuit 2 drives the switching elements in a switching manner.

The drain of the switching element Q1 is connected in series to the choke coil L10, and is coupled to the positive electrode of the smoothing capacitor Ci via the choke coil L10. Therefore, in this circuit, the DC input voltage Ei is supplied to the drain of the switching element Q1 and one winding terminal of the primary winding N1 in the isolation converter transformer PIT via the choke coil L10 connected in series. The source of the switching element Q1 is coupled to ground on the primary side. The inductance L10 formed by the choke coil winding N10 serves as a functional component equivalent to the choke coil L10 in the class E switching converter shown in FIG. 1 .

A switch drive signal (voltage) output from the oscillation and drive circuit 2 is applied to the gate of the switch element Q1. Since a MOSFET is selected as the switching element Q1, the switching element Q1 includes a body diode DD1 such that the body diode DD1 is connected in parallel to the channel between the source and the drain of the switching element Q1 as shown. The anode of the body diode DD1 is connected to the source of the switching element Q1, and the cathode thereof is connected to the drain of the switching element Q1. The body diode DD1 forms a channel that allows the reverse flow of the switching current caused by the on/off operation of the switching element Q1 (switching operations of ON and OFF that respectively indicate the conduction state and the non-conduction state are repeated alternately) produced.

In addition, a primary side parallel resonance capacitor Cr is connected in parallel to the channel between the drain and the source of the switching element Q1. The capacitance of the primary side parallel resonance capacitor Cr and the leakage inductance of the leakage inductance L1 formed by the primary winding N1 in the isolation converter transformer PIT form a parallel resonance circuit (voltage resonance circuit) for switching current flowing through the switching element Q1. In this power supply circuit, based on the assumption that the inductance of the choke coil L10 is higher than that of the leakage inductance L1, the influence of the choke coil L10 is not considered for this primary side parallel resonance circuit. However, if the resonance frequency of the resonance circuit formed by the choke coil L10, smoothing capacitor Ci, and primary side parallel resonance capacitor Cr is close to that formed by the primary side parallel resonance resistor Cr and leakage inductance L1 due to any of the following The resonant frequency of the resonant circuit, the contribution of the choke coil L10 to the primary side parallel resonant circuit also needs to be considered: the inductance of the choke coil L10 is close to the inductance of the leakage inductance L1; the primary side series resonant capacitor to be described later The capacitance of C11 is close to that of the primary-side parallel resonance capacitor Cr; the capacitance of the smoothing capacitor Ci is close to that of the primary-side parallel resonance capacitor Cr; and so on. The resonant operation of the primary side parallel resonant circuit provides voltage resonant operation as a switching operation of the switching element Q1. Due to this operation, the switching voltage V1 which is the voltage between the drain and the source of the switching element Q1 has a sinusoidal resonance pulse waveform during the off period of the switching element Q1.

Further, a series circuit formed by a primary winding N1 and a primary-side series resonance capacitor C11 in an isolated converter transformer PIT to be described later is connected in parallel to the channel between the drain and source of the switching element Q1.

Specifically, one winding terminal (for example, the winding end terminal) of the primary winding N1 is connected to the drain of the switching element Q1, and the other winding terminal thereof (for example, the winding start terminal) is connected to one of the primary side series resonance capacitor C11. electrode. The other electrode of the primary side series resonant capacitor C11 which is not coupled to the primary winding N1 is connected to the source of the switching element Q1 which is at the primary side ground potential.

In order to drive the switching element Q1 by, for example, separate excitation, the oscillation and drive circuit 2 generates a drive signal as a gate voltage for driving the MOSFET in a switching manner based on the oscillation circuit and an oscillation signal obtained from the oscillation circuit, and applies the drive signal to to the gate of switching element Q1. Thus, the switching element Q1 continuously performs on/off operations according to the waveform of the driving signal. That is, the switching element Q1 performs a switching operation.

The isolation converter transformer PIT sends the switching output from the primary side switching converter to the secondary side while isolating the primary side and the secondary side with respect to DC transmission therebetween. For this transmission, the primary winding N1 and the secondary winding N2 are wound around the isolation converter transformer PIT.

The isolated converter transformer PIT in the present circuit comprises an EE-shaped core formed by combining an E-shaped core composed, for example, of ferrite material. In addition, as windings, a primary winding N1 and a secondary winding N2 are wound around the center leg of the EE-shaped core, where the winding portion is divided into a primary side and a secondary side.

In addition, a gap of about 1.6 mm in length was provided in the center leg of the EE-shaped core of the isolation converter transformer PIT, thereby obtaining a coupling coefficient k of about 0.75 between the primary side and the secondary side. The value of this coupling coefficient k is usually a value such that the degree of coupling between the primary and secondary sides is regarded as weak coupling, so that it is difficult for the isolation converter transformer PIT to enter a saturated state.

The primary winding N1 in the isolation converter transformer PIT is an element for forming a primary side series resonance circuit in a class E switching converter formed on the primary side, as described later. An AC output depending on the switching output of the switching element Q1 is obtained in the primary winding N1.

On the secondary side of the isolating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. The secondary side series resonance capacitor C2 is connected in series to the secondary winding N2. Thus, the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonance capacitor C2 form a secondary side series resonance circuit. This secondary side series resonance circuit realizes a resonance operation linked to a rectification operation of a secondary side rectification circuit to be described later. Thus, the secondary winding current flowing through the secondary winding N2 has a sinusoidal waveform. That is, current resonance operation is realized on the secondary side.

The secondary side rectification circuit in this power supply circuit is formed as a voltage doubler half-wave rectification circuit by coupling two rectification diodes Do1 and Do2 with a smoothing capacitor Co to the secondary winding N2, the secondary side series resonance as described above Capacitor C2 is connected in series to secondary winding N2. The connection structure of the voltage doubler half-wave rectifier circuit is as follows. The winding end of the secondary winding N2 is coupled to the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the secondary side series resonant capacitor C2. The cathode of the rectification diode Do1 is connected to the anode of the smoothing capacitor Co. The winding start end of the secondary winding N2 and the anode of the rectification diode Do2 are connected to the negative electrode of the smoothing capacitor Co at the secondary side ground potential.

The rectification operation of the voltage doubler half-wave rectification circuit thus formed is as follows. During a period of half a cycle corresponding to one polarity of the AC voltage (secondary winding voltage) induced by the secondary winding N2 across the secondary winding N2, the forward voltage is applied to the rectifier diode Do2, thereby rectifying Diode Do2 conducts. Therefore, the rectified current charges in the secondary side series resonant capacitor C2. Thus, a voltage having the same level as the AC voltage induced in the secondary winding N2 is generated across the secondary-side series resonance capacitor C2. During a period of half a cycle corresponding to the other polarity of the secondary winding voltage V3, the rectifier diode Do1 is supplied with a forward voltage and thus conducts. At this time, the smoothing capacitor Co is charged by the potential generated by the superposition of the secondary winding voltage V3 and the voltage across the secondary-side series resonance capacitor C2.

Thus, a DC output voltage Eo is generated across the smoothing capacitor Co, Eo having a level equal to twice the level of the alternating voltage excited in the secondary winding N2. In this rectification operation, charging of the smoothing capacitor Co is effected only during the period of one half cycle of one polarity of the alternating voltage excited in the secondary winding N2. That is, a rectification operation as voltage doubling half-wave rectification is realized. In addition, this rectification operation can be regarded as an operation for the resonance output of the secondary side series resonance circuit formed by the series connection of the secondary winding N2 and the secondary side series resonance capacitor C2. The thus generated DC output voltage Eo is supplied to a load. Furthermore, the voltage Eo is shunted and output to the control circuit 1 as a detection voltage.

The control circuit 1 supplies the oscillation and drive circuit 2 with a detection output depending on the level change of the input DC output voltage Eo. Oscillating and driving circuit 2 drives switching element Q1 by varying the switching frequency and accordingly the time ratio (conduction angle) between periods TON and TOFF within one switching cycle based on detection output input from control circuit 1 . This operation acts as a constant voltage control operation for the secondary side DC output voltage.

Varying control of the switching frequency and the conduction angle of the switching element Q1 causes changes in the resonance impedance and the effective period of power transmission on the primary and secondary sides in the power supply circuit. These changes cause changes in the amount of power transferred from the primary winding N1 to the secondary winding N2 in the isolated converter transformer PIT, and cause changes in the amount of power that should be supplied to the load from the secondary side rectification circuit. Thus, the level of the DC output voltage Eo is controlled such that its level variation is canceled out. That is, the DC output voltage Eo can be stabilized.

When comparing the switching converter (Q1, Cr, L10, N1, and C11) of FIG. 3 formed on the primary side of the power circuit thus formed with the above-mentioned class E converter shown in FIG. 1, the switching converter of FIG. The converter can be considered as obtained by removing the impedance Z as load from the circuit of FIG. 1 and replacing the winding of the choke coil L11 in the circuit of FIG. 1 with the primary winding N1 (leakage inductance L1) of the isolation converter transformer PIT. In the primary side switching converter of Fig. 3, the primary side parallel resonant circuit is formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonant capacitor Cr, and the primary side series resonant circuit is formed by the isolation converter transformer PIT The leakage inductance L1 of the primary winding N1 and the capacitance of the primary side series resonant capacitor C11 are formed.

Thus, it can be said that the primary-side switching converter of FIG. 3 is formed as a class-E switching converter realizing a class-E switching operation. The switching output (AC output) generated from the switching operation of the primary side switching converter is transferred from the primary winding N1 equivalent to the choke coil L11 to the secondary winding N2 via magnetic coupling in the isolation converter transformer PIT. The transmitted output is rectified on the secondary side, resulting in a DC output voltage Eo. That is, the power supply circuit shown in FIG. 3 is configured as a DC-DC converter including a class E switching converter on its primary side.

In addition, the thus formed primary-side Class E switching converter can also be considered as a soft-switching power supply configured composite resonant converter, where the series circuit of the primary winding N1 and the primary-side series resonant capacitor C11 forming the primary-side series resonant circuit is connected in parallel Connected to switching element Q1 (and body diode DD1), switching element Q1 is connected to choke coil L10 and/or leakage inductance L1 (the degree of contribution of choke coil L10 and leakage inductance L1 depends on the parameters of the components included in the resonant circuit different) together with the primary side parallel resonant capacitor Cr form a voltage resonant converter.

It is generally believed that a power supply circuit comprising a voltage resonant converter on its primary side may in fact be impractical because it involves a narrow control range of load power and may not be able to maintain ZVS operation at light loads. Therefore, the inventors of the present application have conducted experiments on power supply circuits, such as the related art circuit shown in FIG. 16, including a secondary side series resonance circuit combined with a primary side voltage resonance converter and as a secondary side Circuit voltage doubler half wave rectifier circuit. These experiments reveal that the power supply circuit exhibits characteristics that make it easier to implement than related art power supply circuits with voltage resonant converters.

However, the power supply circuit of FIG. 16 involves abnormal operation when the load is an intermediate load. Specifically, as shown in FIG. 18B , the current flows through the switching element Q1 in the forward direction before the end of the off period (TOFF) of the switching element Q1, so that the ZVS operation cannot be realized. Therefore, it is still difficult to put the circuit into practical use even with the configuration in FIG. 16 .

In the respect that the power supply circuit of FIG. 3 is based on the composite resonant switching converter including the configuration of the voltage resonant converter circuit on the primary side as described above, it can be said that the configuration used in the power supply circuit shown in FIG. The configuration of the power circuit shown is similar.

However, experiments on the power supply circuit of FIG. 3 have revealed that in this power supply circuit, there is no abnormal operation in which ZVS cannot be realized when the load is an intermediate load, and normal switching operation is realized over the entire predetermined allowable load power range .

It has been confirmed that the abnormal operation associated with the intermediate load observed in the power supply circuit of FIG. side series resonant circuit combined. This abnormal operation is mainly a result of the interaction between the primary side parallel resonance circuit and the secondary side series resonance circuit (rectifier circuit) forming the voltage resonance converter caused by their simultaneous operation. That is, it can be inferred that the above-mentioned abnormal operation associated with the intermediate load is the result of the circuit configuration itself having a bond between the primary-side voltage resonant converter and the secondary-side series resonant circuit. Based on this conclusion, as an important improvement, the power supply circuit shown in FIG. 3 is designed to have a configuration in which a class E switching converter is applied instead of a voltage resonant converter as the primary side switching converter.

Due to this configuration, in the power supply circuit of FIG. 3 , regardless of the presence or absence of the series resonance circuit on the secondary side, abnormal operation in which ZVS cannot be achieved when the load is an intermediate load is eliminated.

In this manner, the abnormal operation associated with the intermediate load, which has been a problem in the related art such as the power supply circuit of FIG. 16 , is eliminated from the power supply circuit of FIG. 3 .

However, in a circuit including a class E converter combined with a multi-resonance converter, the peak level of the switching voltage V1, which is a resonance pulse voltage generated in the off period of the switching element Q1, is high. Specifically, when the input AV voltage VAC is 264V, the peak level reaches 1600V, so considering that the breakdown voltage of the margin switching element Q1 needs to be about 1800V.

Therefore, as an embodiment of the present invention, a power supply circuit configuration obtained by further improvement from the power supply circuit shown in FIG. 3 is proposed. Specifically, each configuration is provided with a Class E switching converter for eliminating abnormal operation associated with intermediate loads. Furthermore, each configuration is designed to allow the use of switching element Q1 with a low breakdown voltage.

(first embodiment)

As one of the power supply circuits of the embodiment, a configuration example of the power supply circuit according to the first embodiment of the present invention is shown in FIG. 4 . Components in FIG. 4 that are the same as those in FIG. 3 are given the same reference numerals and descriptions thereof will be omitted.

In the power supply circuit shown in FIG. 4, a choke coil PCC (inductor L10) having a choke coil winding N10 is added to the primary side of the voltage resonant converter, thereby realizing class E switching operation. The coupling coefficient between the primary winding N1 and the secondary winding N2 in the isolation converter transformer PIT is set to 0.8 or less, which corresponds to weak coupling. On the secondary side, a secondary side partial voltage resonance capacitor C4 is connected in parallel to the secondary winding N2, thereby constructing a multiple resonant converter deriving a DC output voltage from a full-wave bridge. In addition, a series circuit of clamp capacitor C3 and auxiliary switching element Q2 is connected in parallel to choke coil PCC (inductance L10 ) in the multi-resonant converter.

To control the gate of the auxiliary switching element Q2, an isolated converter transformer auxiliary winding Ng as part of the primary winding in the isolated converter transformer PIT, and resistors Rg1 and Rg2 are provided.

Each of the switching element Q1 and the auxiliary switching element Q2 in the multi-resonant converter section may be any one of MOSFET, IGBT, and BJT. In the following, a circuit in which MOSFETs are used as these elements will be described.

The main components in the power supply circuit in Fig. 4 are interconnected as follows. One winding end of the choke coil winding N10 is connected to the positive electrode of the smoothing capacitor Ci. The other winding end of the choke coil winding N10 is connected to one winding end of the primary winding N1 in the isolation converter transformer PIT, and is connected to the drain of the MOSFET which is one terminal of the switching element Q1. That is, the inductor L10 is connected between the positive electrode of the smoothing capacitor Ci and one winding terminal of the primary winding N1 and the drain of the MOSFET which is one terminal of the switching element Q1. In addition, the primary side series resonance capacitor C11 is connected between the other winding end of the primary winding N1 in the isolation converter transformer PIT and the source of the MOSFET which is the other terminal of the switching element Q1. Furthermore, one electrode of the primary side parallel resonance capacitor Cr is connected to the drain of the MOSFET which is one terminal of the switching element Q1, and the other electrode of the primary side parallel resonance capacitor Cr is connected to the drain of the MOSFET which is the other terminal of the switching element Q1. Source of MOSFET. That is, the switching element Q1 and the primary side parallel resonance capacitor Cr are connected in parallel with each other.

In addition, the isolation converter transformer auxiliary winding Ng is provided such that the voltage from the isolation converter transformer auxiliary winding Ng is divided by the resistors Rg1 and Rg2 and then applied to the gate of the MOSFET serving as the auxiliary switching element Q2. The drain of the auxiliary switching element Q2 is connected to the clamp capacitor C3. That is, the clamp capacitor C3 and the auxiliary switching element Q2 form a series circuit. A series circuit of clamp capacitor C3 and auxiliary switching element Q2 is connected in parallel to choke coil PCC (inductance L10). The isolated converter transformer auxiliary winding Ng is derived from an additional winding of the primary winding N1, so that windings Ng and N1 are integrally interconnected. This structure is only because the source of the MOSFET serving as the auxiliary switching element Q2 is connected to one end of the primary winding N1. Providing the winding Ng as another winding separate from the winding N1 does not create any problems.

In the circuit configuration described above, the primary-side series resonance capacitor C11 is connected between the other winding terminal of the primary winding N1 and the source of the switching element Q1 in the isolation converter transformer PIT. A primary side series resonant circuit is thereby formed, the resonant frequency of which is dominated by the leakage inductance L1 generated in the primary winding N1 of the isolation converter transformer PIT and the primary side series resonant capacitor C11. In addition, the primary side parallel resonance capacitor Cr is connected in parallel to the switching element Q1, thereby forming a primary side parallel resonance circuit whose resonance frequency is determined by the leakage inductance L1 generated in the primary winding N1 and the primary side parallel resonance capacitor Cr dominates. In addition, the primary side includes a series circuit of clamp capacitor C3 and auxiliary switching element Q2, which is connected in parallel with choke coil PCC (inductance L10), and auxiliary switching element Q2 is designed to be in a non-conductive state when switching element Q1 Turn on when down. The auxiliary switching element Q2 includes a body diode DD2 to allow control of on/off switching of current in one direction, and to be in a conductive state for current in the other direction so as to allow bidirectional current passage.

In response to the switching operation of the switching element Q1, due to the voltage resonance operation of the primary side parallel resonance circuit, a charging/discharging current flows to and from the primary side parallel resonance capacitor Cr during a period in which the switching element Q1 is in the off state. In addition, the primary side series resonance circuit achieves a resonance operation such that a resonance current flows through the primary side series resonance capacitor C11, the primary winding N1, and the passage of the switching element Q1 during a period in which the switching element Q1 is in the on state.

The expression that the resonance frequency is "dominated" in the first embodiment indicates that the resonance frequency value of the primary side series resonance circuit is very dependent on the inductance value of the leakage inductance L1 generated in the primary winding N1 and the capacitance value of the primary side series resonance capacitor C11. The resonance frequency value of the primary side parallel resonance circuit is very dependent on the inductance value of the leakage inductance L1 and the capacitance value of the primary side parallel resonance capacitor Cr. This expression also indicates that the influence of other components on the corresponding resonant frequency is relatively small. Strictly speaking, these resonance frequencies are related to the following ratios: the ratio between the capacitance values of the primary side parallel resonance capacitor Cr and the primary side series resonance capacitor C11, the ratio between the capacitance values of the smoothing capacitor Ci and the primary side series resonance capacitor C11 ratio, and the ratio between the inductance values of inductance L10 and leakage inductance L1. However, these ratios are not dominant, so the resonant frequency is not dominated by these ratios.

The primary-side parallel resonance frequency as an example will be specifically described below. Specifically, as an example, not only the primary side parallel resonance capacitor Cr and leakage inductance L1, but also the primary side series resonance capacitor C11 interconnecting the primary side parallel resonance capacitor Cr and leakage inductance L1 also has an influence on the primary side parallel resonance frequency . However, if the capacitance value of the primary side series resonance capacitor C11 is much larger than the capacitance value of the primary side parallel resonance capacitor Cr, the contribution of the primary side series resonance capacitor C11 to the primary side parallel resonance is small, and it can be determined that the primary side parallel resonance frequency is not Dominated by primary side series resonant capacitor C11. As another example, the parallel connection of the series circuit of the inductor L10 and the smoothing capacitor Ci with the leakage inductance L1 has an effect on the primary side parallel resonance frequency. Generally, the capacitance value of the smoothing capacitor Ci is much larger than that of the primary-side parallel resonance capacitor Cr, so the smoothing capacitor Ci can be considered to be short-circuited in terms of AC transmission. However, if the inductance value of inductor L10 is significantly greater than the inductance value of leakage inductance L1, the inductance value resulting from the parallel connection of inductor L10 to leakage inductance L1 is substantially defined by leakage inductance L1. Therefore, it can be confirmed that the primary side parallel resonance frequency is not dominated by the series circuit of the inductor L10 and the smoothing capacitor Ci. It should be noted that stray capacitance components and inductance components generated in components and interconnections are included in the primary side parallel resonance capacitor Cr, primary side series resonance capacitor C11, leakage inductance L1, and inductance L10.

In the circuit configuration described above, the isolation converter transformer auxiliary winding Ng is connected to the primary winding N1 so that the voltage generated in the winding Ng has such a polarity that the auxiliary switching element Q2 is OFF-(non-conductive ON- (conduction-) state in ON- (conduction-) state. Changing the ratio between the resistance values of the resistors Rg1 and Rg2 makes it possible to adjust the length of the period during which the auxiliary switching element Q2 is in the ON- (conducting-) state.

On the secondary side, the isolated converter transformer PIT comprises a secondary winding N2. The secondary side rectifying element includes a plurality of rectifying diodes Do1 to Do4 rectifying the AC voltage output from the secondary winding N2. The rectified voltage generated by the rectifying diodes Do1 to Do4 is charged in the smoothing capacitor Co.

In addition, a secondary side partial voltage resonance capacitor C4 is provided. Therefore, partial voltage resonance is generated so that switching loss can be prevented from occurring at transition points between the on-state and off-state of the rectifier diodes Do1 to Do4, which can further improve the efficiency of the switching power supply circuit.

More detailed features of the switching power supply circuit shown in FIG. 4 will be described below. FIG. 5 illustrates a structural example of the isolation converter transformer PIT included in the power supply circuit of FIG. 4 having the above configuration. The isolation converter transformer PIT includes an EE-shaped core obtained by combining E-shaped cores CR1 and CR2 composed of ferrite material. In addition, a bobbin B formed of resin or the like is provided, and has such a divided shape that the winding portions on the primary side and the secondary side are independent from each other. The primary winding N1 and the isolated converter transformer auxiliary winding Ng are wound around one winding portion of the bobbin B. The secondary winding N2 is wound around another winding portion.

The bobbin B, already thus wound with the primary and secondary side windings, is fitted to the EE cores (CR1, CR2), which results in the primary winding N1, the isolation converter transformer auxiliary winding Ng and the secondary winding N2 on different winding areas Wrapped around the center stem of an EE shaped core. In this way, the entire structure of the isolated converter transformer PIT is completed.

In the central stem of the EE core, a gap G is formed as shown. The coupling coefficient k that provides a weakly coupled state is thus obtained. That is, the degree of weak coupling in the isolation converter transformer PIT in FIG. 4 is higher than that in the power supply circuit shown in FIG. 16 as the related art. Gap G can be formed by making the central legs of E-cores CR1 and CR2 shorter than their corresponding two outer legs. In this embodiment, EER-35 was used as the core member, and the length of the gap G was set to 1.6 mm. The numbers of turns of the primary winding N1, the secondary winding N2, and the isolation converter transformer auxiliary winding Ng are set to 60T, 30T, and 1T, respectively. The coupling coefficient between the primary and secondary sides of the isolation converter transformer PIT itself was set to 0.75.

It is also possible to construct the choke coil PCC by providing a winding around an EE-shaped core having a predetermined shape and size. In this embodiment, ER-28 was used as the core member, the length of the gap G was set to 0.8 mm, and the number of turns of the choke coil winding N10 was set to 50T. Thus, 1 mH (millihenry) was obtained as the inductance value of the inductor L10.

Parameters of main components in the power supply circuit of FIG. 4 were selected as follows so that experimental results on the power supply circuit to be described later were obtained.

The capacitances of the primary side parallel resonance capacitor Cr, primary side series resonance capacitor C11, clamp capacitor C3 and secondary side partial voltage resonance capacitor C4 are selected as follows.

Cr=1500pF

C11 = 0.01μF

C3 = 0.1μF

C4 = 3300pF

The resistance values of resistors Rg1 and Rg2 are chosen as follows:

Rg1＝150Ω(ohm)

Rg2 = 100Ω

The allowable load power range is from the maximum load power Pomax 300W to the minimum load power Pomin 0W (no load). The rated level of the DC output voltage Eo is 175V.

The experimental results for the power supply circuit of Fig. 4 are shown in the waveform diagrams of Figs. 6A and 6B. FIG. 6A shows waveforms of current and voltage in the case of a maximum load power Pomax of 300W and an input AV voltage VAC of 100V. More specifically, FIG. 6A shows the switching voltage V1 as the voltage across the switching element Q1, the switching current IQ1 as the current flowing through the switching element Q1, and the input current I1 as the current flowing through the choke coil PCC. Figure 6A also shows the primary side series resonant voltage V2 as the voltage across the primary side series resonant capacitor C11, the primary winding current I2 as the current flowing through the primary winding N1, and the primary side winding current I2 as the current flowing to the primary side parallel resonant capacitor Cr Parallel resonant current ICr. In addition, FIG. 6A also shows the secondary winding voltage V3 as the auxiliary switching current IQ2 flowing through the auxiliary switching element Q2, the voltage generated in the secondary winding N2, and the secondary winding N2 as the current flowing through the secondary winding N2. Level winding current I3.

Figure 6B shows the switching voltage V1, switching current IQ1, input current I1, primary side series resonance voltage V2, primary winding current I2, primary side parallel Resonant current ICr, auxiliary switch current IQ2, secondary winding voltage V3, and secondary winding current I3.

The basic operation of the power supply circuit in FIG. 4 will be described below with reference to the waveform diagram of FIG. 6A.

The switching element Q1 is supplied with the voltage across the smoothing capacitor Ci as the DC input voltage Ei, and realizes switching operation.

The switching voltage V1 (the voltage between the drain and the source of the switching element Q1) has a waveform depending on the drain of the switching element Q1 associated with the driving of the switching element Q1 caused by the signal from the oscillation and driving circuit 2 On/off of the channel between source and source. Since the auxiliary switching current IQ2 flows to the clamp capacitor C3, the degree to which the switching voltage V1 rises is suppressed. Specifically, the peak level of the voltage V1 is 460V when the input AC voltage VAC is 100V, and is 660V when the voltage VAC is 230V. If the auxiliary switching element Q2 and the clamp capacitor C3 do not exist, a sinusoidal resonance pulse waveform is obtained as the waveform of the switching voltage V1 during the off period. In contrast, in the power supply circuit of FIG. 4, the peak portion of the sinusoidal resonance pulse waveform is clamped. However, the waveform near the rising edge of the clamped sine wave is substantially similar to that of the unclamped sine wave. Therefore, also when the switching voltage V1 is clamped, the advantage of securing the ZVS operation at the moment of turning off the switching element Q1 is sufficiently obtained.

The switching current IQ1 (the current flowing through the switching element Q1 ) is the current flowing through the switching element Q1 (and the body diode DD1 ) from the drain side of the switching element Q1 . Each switching cycle is divided into a period TON in which the switching element Q1 should be in an on state and a period TOFF in which the switching element should be in an off state. The switching voltage V1 has a waveform in which the voltage is at zero level during the period TON and is a resonance pulse during the period TOFF. Due to the resonant operation of the primary-side parallel resonant circuit, this voltage resonant pulse of the switching voltage V1 is obtained as a pulse having a sinusoidal resonant waveform.

The switch current IQ1 is at zero level during the period TOFF. When the period TOFF ends and the period TON begins, that is, at the moment when the switching element Q1 is turned on, initially the switching current IQ1 flows through the body diode DD1 and thus has a negative polarity waveform. Next, the flow direction is reversed so that the switching current IQ1 flows from the drain to the source of the switching element Q1 and thus has a positive polarity waveform.

The input current I1 (the current flowing from the smoothing capacitor Ci to the primary-side switching converter) flows through a combined inductance between the inductance of the inductance L10 formed by the choke coil winding N10 and the inductance of the leakage inductance L1 of the primary winding N1. Thus, the current flowing from the smoothing capacitor Ci to the switching converter is a ripple current.

The primary-side series resonance voltage V2 (the voltage across the primary-side series resonance capacitor C11 ) has an AC waveform that depends on the switching cycle and is close to a sinusoidal waveform.

The primary winding current I2 (the current flowing through the primary winding N1 ) is the current flowing through the primary winding N1 depending on the switching operation of the switching element Q1 . In the circuit of FIG. 4, the primary winding I2 has substantially the same waveform as that obtained from the synthesis between the switching current IQ1 and the primary side parallel resonance current ICr. Due to the on/off operation of the switching element Q1, the resonance pulse voltage as the switching voltage V1 in the period TOFE is applied to the series circuit of the primary winding N1 and the primary side series resonance capacitor C11, the primary winding N1 and the primary side series resonance capacitor C11 C11 forms the primary side series resonant circuit. Thus, the primary-side series resonance circuit realizes a resonance operation, and the primary winding current I2 has an AC waveform including a sine wave component and depending on a switching period.

When the period TON ends and the period TOFF begins, that is, at the moment when the switching element Q1 is turned off, the primary winding current I2 flows to the primary side parallel resonant capacitor Cr with a positive polarity as the primary side parallel resonant current ICr, thereby starting to provide the primary side parallel resonant current ICr. Operation of resonant capacitor Cr charging. In response to this charging, the switching voltage V1 starts to rise from zero level with a sinusoidal waveform, that is to say a voltage resonance pulse rise. When the polarity of the primary side parallel resonant current ICr becomes negative, the state of the primary side parallel resonant capacitor Cr changes from a charged state to a discharged state, which causes the voltage resonant pulse to drop from its peak level. This operation indicates that the ZVS operation caused by the primary-side parallel resonance circuit and the ZCS operation caused by the primary-side series resonance circuit are realized at the timing of turning on and off the switching element Q1. As described above, the primary side parallel resonance current ICr (the current flowing to the primary side parallel resonance capacitor Cr) flows when the switching voltage V1 rises and falls, thereby reducing the switching loss of the switching element Q1.

The auxiliary switching current IQ2 (the current flowing through the auxiliary switching element Q2) flows every time the switching element Q1 is turned off to clamp the switching voltage V1 to prevent an excessive voltage from being applied between the drain and the source of the switching element Q1. Voltage. Specifically, the phases of the primary winding current I2 and the voltage generated in the primary winding N1 are shifted by 90 degrees from the phase of the voltage generated in the isolation converter transformer auxiliary winding Ng. Thus, at the moment when switching element Q1 is turned off, a voltage that turns on auxiliary switching element Q2 is generated across the isolation converter transformer auxiliary winding Ng, and thus auxiliary switching element Q2 is turned on. Accordingly, current flows to the clamp capacitor C3, which prevents the voltage between the drain and the source of the switching element Q1 from rising.

The secondary winding voltage V3 (the voltage across the secondary winding N2, that is, the voltage across the connecting circuit of the secondary winding N2 and the secondary side partial voltage resonance capacitor C4) is clamped at a level having The absolute value of is equal to the DC output voltage Eo in the conduction period of the rectifier diodes Do1 to Do4.

The secondary winding current I3 (the current flowing through the secondary winding N2 ) is a current partially including a sinusoidal waveform.

The characteristics of the power supply circuit of the first embodiment shown in FIG. 4 will be described with reference to FIGS. 7 and 8 . Fig. 7 shows the AC-to-DC power conversion efficiency of the improved class E switching multi-resonant converter of the first embodiment under the load power range of 0W to 300W when the input AC voltage VAC is 100V and the voltage VAC is 230V ( ηAC→DC) and the change of switching frequency fs. The solid line in FIG. 7 indicates the characteristics when the input AC voltage VAC is 100V, and the broken line indicates the characteristics when the voltage VAC is 230V.

Fig. 8 shows the AC-to-DC power conversion efficiency (ηAC→DC) and the switching efficiency of the improved class E switching multi-resonant converter of the first embodiment under the input AV voltage VAC range of 85V to 230V when the load power is 300W Changes in frequency fs.

Referring to Figure 7, when the input AC voltage VAC is 100V, the results are as follows: AC to DC power conversion efficiency reaches 91.0%; and the range of switching frequency fs is from 89.3kHz to 110.0kHz, so the variable range of switching frequency fs is the width of Δfs It is 20.7kHz. In addition, when the input AC voltage VAC is 230V, the results are as follows: AC to DC power conversion efficiency reaches 94.0%; and the range of switching frequency fs is from 132.2kHz to 147kHz, so the width of the variable range Δfs of switching frequency fs is 14.8kHz . When the input AC voltage VAC is 100V and 230V, the width of the variable range Δfs of the switching frequency fs is smaller than the variable range Δfs of the switching frequency fs in the circuit shown in FIG. 16 as the background art. This is because providing the isolation converter transformer auxiliary winding Ng in the isolation converter transformer PIT allows the time ratio between the conduction periods of the switching element Q1 and the auxiliary switching element Q2 (ratio of the period TON to the period TOFF) to respond to the load power and input AC voltage VAC, which can narrow the variable range Δfs.

Referring to FIG. 8, when a load power of 300W is supplied, the switching frequency fs increases as the input AC voltage VAC increases. In the range of input AC voltage VAC from 170V to 220V, the AC-to-DC power conversion coefficient (ηAC→DC) is a high value of 94.5%. Compared with the circuit shown in FIG. 16 as background art, the value of the AC to DC power conversion coefficient (ηAC→DC) is higher at a wider AC input voltage range.

In the power supply circuit shown in FIG. 16 as a related art example, the current flowing into the switching converter from the smoothing capacitor Ci passes through the primary winding N1 in the isolation converter transformer PIT, and then reaches the switching element Q1 and the primary side parallel resonance capacitor Cr. This current flowing from the smoothing capacitor Ci to the switching converter is the primary winding current I2 and has a comparatively high frequency depending on the switching period. That is, the charging current flowing to the smoothing capacitor Ci and the discharging current flowing from the smoothing capacitor Ci have a frequency higher than that of the commercial AC power supply voltage.

Aluminum electrolytic capacitors are often used for constituent elements like the smoothing capacitor Ci because the capacitor Ci is required to have a high breakdown voltage and the like. Compared to other kinds of capacitors, aluminum electrolytic capacitors tend to suffer a reduction in electrolytic capacitance and have an increase in the tangent of the loss angle when operating at high frequencies. Therefore, it is necessary to select a special product with a low equivalent series resistance (ESR) and a large allowable ripple current as an aluminum electrolytic capacitor for the smoothing capacitor Ci. In addition, it is necessary to correspondingly increase the capacitance of the component as the smoothing capacitor Ci. For example, in the configuration of the power supply circuit in FIG. 16, the capacitance needs to be about 1000 μF to handle the maximum load power Pomax of 300 W which is the same as that in the first embodiment. Aluminum electrolytic capacitors compatible with these components are more expensive than general-purpose aluminum electrolytic capacitors, and the increase in capacitance results in higher component prices. Therefore, using such a special capacitor is disadvantageous in terms of cost.

In contrast, in the power supply circuit of the first embodiment in FIG. 4, the current flowing from the smoothing capacitor Ci into the switching converter passes through the series connection of the choke coil winding N10 and the primary winding N1, and then reaches the switching element Q1. Therefore, the current flowing from the smoothing capacitor Ci to the switching converter becomes a DC current as shown by the input current I1 of FIG. 6A. Therefore, the current flowing from the smoothing capacitor Ci to the switching converter is a DC current, so this embodiment does not involve the aforementioned problems of reduction in electrolytic capacitance and increase in tangent of the loss angle. Furthermore, along with this, the ripple of the period with the commercial AC power supply voltage in the DC input voltage Ei is also reduced. For these reasons, in the present invention, a general-purpose aluminum electrolytic capacitor can be selected as the smoothing capacitor Ci. Furthermore, since the ripple voltage is small, the capacitance as a component of the smoothing capacitor Ci can be reduced compared with that in the circuit of FIG. 16 . This embodiment can achieve cost reduction of the smoothing capacitor Ci. In addition, the waveform of the input current I1 is a sinusoidal waveform. This contributes to the realization of the high-frequency noise reduction effect.

In addition, in the circuit of FIG. 4 in which the class E switching converter is applied to the primary side switching converter, there is no abnormal operation associated with the intermediate load regardless of the presence or absence of the secondary side series resonance circuit, and proper ZVS operation. In this abnormal operation phenomenon, as shown in FIG. 18B , the switching element Q1 is turned on so that the positive switching current IQ1 flows at the source of the switching element Q1 before the original turning-on timing of the switching element Q1 (the start timing of the period TON). and drain flow. This behavior of the switching current IQ1 increases switching losses. The present embodiment prevents such a behavior of the switching current IQ1 corresponding to abnormal operation from occurring, thereby eliminating an increase in switching loss. This feature is also a factor for improving power conversion efficiency.

As is apparent from the comparison between the switching current IQ1 of FIGS. 6A and 18A , the switching current IQ1 of FIG. 6A corresponding to the present embodiment has a waveform in which the current peak occurs before the end of the period TON. The waveform of the switching current IQ1 shown in FIG. 6A indicates that the level of the switching current IQ1 is suppressed at the moment when the switching element Q1 is turned off. If the level of the switching current IQ1 is suppressed at the turn-off time, the switching loss at the turn-off time is correspondingly reduced, which improves the power conversion efficiency.

This waveform of the switching current IQ1 is caused by the class E switching operation of the primary side switching converter. In addition, in this embodiment, the waveform of the input current I1 is a ripple waveform. This contributes to realizing the high-frequency noise reduction effect.

Furthermore, the auxiliary switching element Q2 and the clamp capacitor C3 are provided so that the auxiliary switching current IQ2 flows in synchronization with the off period of the switching element Q1. Thus, even when the input AC voltage VAC is 230V, the maximum value of the voltage applied to the switching element Q1 is as low as about 660V. Therefore, the required breakdown voltage of the switching element Q1 can be significantly reduced, which facilitates the selection of the switching element Q1 and can reduce the cost of the switching power supply circuit. If the auxiliary switching element Q2 and the clamping capacitor C3 are not provided, the breakdown voltage of the switching element Q1 needs to be about 1800V. In this case, if a MOSFET is used as the switching element Q1, its on-resistance value is about 7Ω. On the contrary, if the auxiliary switching element Q2 and the clamping capacitor C3 are provided, it is sufficient that the breakdown voltage of the switching element Q1 is as low as 900V. At this time, the on-resistance value of the switch element Q1 is about 1.2Ω. Therefore, losses caused by on-resistance are reduced and AC-to-DC power conversion efficiency is improved. In addition, selection of switching element Q1 is facilitated and cost reduction is allowed. The power consumption of the auxiliary switching element Q2 is small, and its gate drive circuit can be formed only by adding resistors Rg1 and Rg2 and an isolation converter transformer auxiliary winding Ng. Therefore, when considering the cost reduction caused by the reduction of the breakdown voltage of the switching element Q1, there is no increase in the total cost accompanying the provision of the auxiliary switching element Q2, and actually, the cost of the entire device is reduced.

(Variants)

9 and 10 illustrate variations of the secondary-side circuit of the power supply circuit of the first embodiment. Fig. 11 illustrates a variation of its primary side circuit. The circuit shown in Figure 9 is a voltage doubler half-wave rectifier circuit. This circuit provides advantages similar to those of the above-mentioned embodiments, and in particular can provide the advantage of realizing multiplied rectified voltage. The circuit shown in FIG. 10 is a full-wave rectification circuit including a secondary winding N2 and a secondary winding N2' as a winding provided with a center tap. This circuit also provides similar advantages to the above-described embodiments, and in particular can provide the advantage of utilizing two rectifying diodes to realize full-wave rectification.

In the circuit shown in FIG. 11, instead of the isolation converter transformer auxiliary winding Ng in the isolation converter transformer PIT for generating the drive voltage for the auxiliary switching element Q2, a choke coil auxiliary winding Ng added to the choke coil PCC is provided. winding Ng', and the voltage divided by the resistors Rg3 and Rg4 is applied as the gate voltage of the auxiliary switching element Q2. This circuit offers similar advantages to those of the above-described embodiment, and in particular can provide the advantage that the choke coil PCC and the circuitry associated with the auxiliary switching element Q2 can be arranged close to each other. The resistance values of the resistors Rg3 and Rg4 are, for example, 68Ω and 100Ω, respectively.

(second embodiment)

FIG. 12 illustrates a configuration example of a power supply circuit according to a second embodiment of the present invention. Components in FIG. 12 that are the same as those in FIG. 4 are given the same reference numerals and descriptions thereof will be omitted.

In the power supply circuit shown in FIG. 12, a choke coil PCC (inductor L10) having a choke coil winding N10 is added to the primary side of the voltage resonant converter to realize class E switching operation. The coupling coefficient between the primary winding N1 and the secondary winding N2 in the isolation converter transformer PIT is set to 0.8 or less, which corresponds to weak coupling. On the secondary side, a secondary-side series resonant capacitor C4 is connected in series to the secondary winding N2 to construct a multiple resonant converter deriving a DC output voltage from the full-wave bridge. In addition, a series circuit of clamp capacitor C3 and auxiliary switching element Q2 is connected in parallel to choke coil PCC (inductance L10 ) in the multi-resonant converter.

To control the gate of the auxiliary switching element Q2, an isolated converter transformer auxiliary winding Ng is provided as part of the primary winding in the isolated converter transformer PIT, and resistors Rg1 and Rg2 are provided.

Each of the switching element Q1 and the auxiliary switching element Q2 in the multi-resonant converter section may be any one of MOSFET, IGBT and BJT. A circuit in which MOSFETs are used as these elements will be described below.

The main components in the power supply circuit in Fig. 12 are connected to each other as follows. One winding end of the choke coil winding N10 is connected to the positive electrode of the smoothing capacitor Ci. The other winding end of the choke coil winding N10 is connected to one winding end of the primary winding N1 in the isolation converter transformer PIT, and is connected to the drain of the MOSFET which is one terminal of the switching element Q1. That is, the inductor L10 is connected between the positive electrode of the smoothing capacitor Ci and one winding terminal of the primary winding N1 and the drain of the MOSFET which is one terminal of the switching element Q1. In addition, the primary side series resonance capacitor C11 is connected between the other winding end of the primary winding N1 in the isolation converter transformer PIT and the source of the MOSFET which is the other terminal of the switching element Q1. Furthermore, one electrode of the primary side parallel resonance capacitor Cr is connected to the drain of the MOSFET which is one terminal of the switching element Q1, and the other electrode of the primary side parallel resonance capacitor Cr is connected to the drain of the MOSFET which is the other terminal of the switching element Q1. Source of MOSFET. That is, the switching element Q1 and the primary side parallel resonance capacitor Cr are connected in parallel with each other.

In addition, the isolation converter transformer auxiliary winding Ng is provided such that the voltage from the isolation converter transformer auxiliary winding Ng is divided by the resistors Rg1 and Rg2 and then applied to the gate of the MOSFET serving as the auxiliary switching element Q2. The drain of the auxiliary switching element Q2 is connected to the clamp capacitor C3. That is, the clamp capacitor C3 and the auxiliary switching element Q2 form a series circuit. A series circuit of clamp capacitor C3 and auxiliary switching element Q2 is connected in parallel to choke coil PCC (inductance L10). The isolated converter transformer auxiliary winding Ng is derived from an additional winding of the primary winding N1, so that the windings Ng and N1 are integrally connected to each other. This structure is only because the source of the MOSFET serving as the auxiliary switching element Q2 is connected to one end of the primary winding N1. Providing the winding Ng as another winding separate from the winding N1 does not cause any problems.

In the circuit configuration described above, the primary-side series resonance capacitor C11 is connected between the other winding terminal of the primary winding N1 and the source of the switching element Q1 in the isolation converter transformer PIT. A primary side series resonant circuit is thus formed, the resonant frequency of which is dominated by the leakage inductance L1 generated in the primary winding N1 of the isolation converter transformer PIT and the primary side series resonant capacitor C11. In addition, the primary side parallel resonance capacitor Cr is connected in parallel to the switching element Q1, thereby forming a primary side parallel resonance circuit whose resonance frequency is dominated by the leakage inductance L1 generated in the primary winding N1 and the primary side parallel resonance capacitor Cr. In addition, the primary side includes a series circuit of an auxiliary switching element Q2 and a clamp capacitor C3, which is connected in parallel to the choke coil PCC (inductance L10), and the auxiliary switching element Q2 is designed so that when the switching element Q1 is in non-conduction state is turned on. The auxiliary switching element Q2 includes a body diode DD2 to allow control of on/off switching of current in one direction, and to be in a conductive state for current in the other direction so as to allow bidirectional current passage.

In response to switching operation of switching element Q1, charge/discharge current flows to and from primary side parallel resonance capacitor Cr during a period in which switching element Q1 is in the off state due to voltage resonance operation of the primary side parallel resonance circuit. In addition, the primary side series resonance circuit achieves a resonance operation such that a resonance current flows through the primary side series resonance capacitor C11, the primary winding N1, and the passage of the switching element Q1 during a period in which the switching element Q1 is in the on state.

The expression that the resonance frequency is "dominated" in the second embodiment indicates that the resonance frequency value of the primary side series resonance circuit is very dependent on the inductance value of the leakage inductance L1 generated in the primary winding N1 and the capacitance value of the primary side series resonance capacitor C11, And the resonance frequency value of the primary side parallel resonance circuit is very dependent on the inductance value of the leakage inductance L1 and the capacitance value of the primary side parallel resonance capacitor Cr. This expression also indicates that the influence of other components on the corresponding resonant frequency is relatively small. Strictly speaking, these resonance frequencies are related to the following ratios: the ratio between the capacitance values of the primary side parallel resonance capacitor Cr and the primary side series resonance capacitor C11, the ratio between the capacitance values of the smoothing capacitor Ci and the primary side series resonance capacitor C11 ratio, and the ratio between the inductance values of inductance L10 and leakage inductance L1. However, these ratios are not dominant, so the resonant frequency is not dominated by these ratios.

The primary-side parallel resonance frequency as an example will be specifically described below. Specifically, as an example, not only the primary side parallel resonance capacitor Cr and leakage inductance L1, but also the primary side series resonance capacitor C11 interconnecting the primary side parallel resonance capacitor Cr and leakage inductance L1 also has an influence on the primary side parallel resonance frequency . However, if the capacitance value of the primary side series resonance capacitor C11 is much larger than the capacitance value of the primary side parallel resonance capacitor Cr, the contribution of the primary side series resonance capacitor C11 to the primary side parallel resonance is small, and it can be determined that the primary side parallel resonance frequency is not Dominated by primary side series resonant capacitor C11. As another example, the parallel connection of the series circuit of the inductor L10 and the smoothing capacitor Ci with the leakage inductance L1 has an effect on the primary side parallel resonance frequency. Generally, the capacitance value of the smoothing capacitor Ci is much larger than that of the primary-side parallel resonance capacitor Cr, so the smoothing capacitor Ci can be considered to be short-circuited in terms of AC transmission. However, if the inductance value of inductor L10 is significantly greater than the inductance value of leakage inductance L1, the inductance value resulting from the parallel connection of inductor L10 to leakage inductance L1 is substantially defined by leakage inductance L1. Therefore, it can be confirmed that the primary side parallel resonance frequency is not dominated by the series circuit of the inductor L10 and the smoothing capacitor Ci. It should be noted that stray capacitance components and inductance components generated in components and interconnections are included in the primary side parallel resonance capacitor Cr, primary side series resonance capacitor C11, leakage inductance L1, and inductance L10.

In the circuit configuration described above, the isolation converter transformer auxiliary winding Ng is connected to the primary winding N1 so that the voltage generated in the winding Ng has such a polarity that the auxiliary switching element Q2 is OFF-(non-conductive ON- (conduction-) state in ON- (conduction-) state. Changing the ratio between the resistance values of the resistors Rg1 and Rg2 makes it possible to adjust the length of the period during which the auxiliary switching element Q2 is in the ON- (conducting-) state.

On the secondary side, the isolated converter transformer PIT comprises a secondary winding N2. Because the degree of coupling in the isolation converter transformer is set to weak coupling, the secondary winding N2 has a leakage inductance L2 similar to that of the primary winding N1. In addition, the resonance frequency of the secondary side series resonance circuit is governed by the leakage inductance L2 generated in the secondary winding of the isolation converter transformer PIT and the secondary series resonance capacitor C4.

The formation of the secondary side series resonance circuit can narrow the variation range Δfs of the switching frequency fs of the above-mentioned constant voltage control for the switching power supply circuit.

The secondary side series resonance circuit is connected in series to the secondary side rectification and smoothing circuit. The secondary side rectification and smoothing circuit includes a secondary side rectification element and a secondary side smoothing capacitor. The secondary-side rectifying element is formed by a bridge circuit comprising bridge-connected rectifying diodes Do1 to Do4 and having an input side and an output side. A connection node between the rectification diodes Do1 and Do2 and a connection node between the rectification diodes Do3 and Do4 are defined as an input side. The connection node between the rectification diodes Do1 and Do3 and the connection node between the rectification diodes Do2 and Do4 are defined as the output side. A smoothing capacitor Co is connected to the output side of the bridge circuit. This secondary-side rectification and smoothing circuit is a full-wave rectification circuit that rectifies positive and negative voltages generated in the secondary winding N2 and uses the rectified voltage as a load power supply.

More detailed features of the switching power supply circuit shown in FIG. 12 will be described below. A structural example of the isolation converter transformer PIT included in the power supply circuit of FIG. 12 having the configuration described above is the same as that shown in FIG. 5 , and thus description thereof will be omitted.

It is also possible to construct the choke coil PCC by providing a winding around an EE-shaped core having a predetermined shape and size. In this embodiment, ER-28 was used as the core member, the length of the gap G was set to 0.8 mm, and the number of turns of the choke coil winding N10 was set to 50T. Thus, 1 mH (millihenry) was obtained as the inductance value of the inductor L10.

Parameters of main components in the power supply circuit of FIG. 12 were selected as follows so that experimental results on the power supply circuit to be described later were obtained.

The capacitances of the primary side parallel resonance capacitor Cr, primary side series resonance capacitor C11, clamp capacitor C3 and secondary side partial voltage resonance capacitor C4 are selected as follows.

Cr=1000pF

C11 = 0.018μF

C3 = 0.1μF

C4 = 0.056μF

The resistance values of resistors Rg1 and Rg2 are chosen as follows:

Rg1＝120Ω(ohm)

Rg2 = 100Ω

The allowable load power range is from a maximum load power Pomax of 300W to a minimum load power Pomin of 0W (no load). The rated level of the DC output voltage Eo is 175V.

The experimental results on the power supply circuit of FIG. 12 related to the waveforms of the respective currents and voltages are basically the same as indicated by the waveform diagrams of FIGS. 6A and 6B , and thus descriptions thereof will be omitted.

The characteristics of the power supply circuit of the second embodiment shown in FIG. 12 will be described with reference to FIG. 13 . Fig. 13 shows the AC-to-DC power conversion efficiency of the improved class E switching multi-resonant converter of the first embodiment under the load power range of 0W to 300W when the input AC voltage VAC is 100V and the voltage VAC is 230V ( ηAC→DC) and switching frequency^. The solid line in FIG. 13 indicates the characteristics when the input AC voltage VAC is 100V, and the broken line indicates the characteristics when the voltage VAC is 230V.

The characteristics of AC to DC power conversion efficiency (ηAC→DC) and switching frequency fs under the input AV voltage VAC range of 85V to 230V of the improved Class E switching multi-resonant converter of the second embodiment when the load power is 300W and The one shown in FIG. 8 is similar, so its description will be omitted.

Referring to Fig. 13, when the input AC voltage VAC is 100V, obviously favorable results are obtained as follows: the AC-to-DC power conversion efficiency reaches 91.4%; and the switching frequency fs ranges from 86.2kHz to 86.5kHz, so the switching frequency fs The width of the variable range Δfs is 0.3 kHz. In addition, when the input AC voltage VAC is 230V, the results are as follows: the AC-to-DC power conversion efficiency reaches 93.8%; and the switching frequency fs is constant at 128.2kHz, so the width of the variable range Δfs of the switching frequency fs is 0kHz. When the input AC voltage VAC is 100V and 230V, the width of the variable range Δfs of the switching frequency fs is much smaller than the variable range Δfs of the switching frequency fs in the circuit shown in FIG. 16 as the background art. This is because the primary side series resonance circuit, the primary side parallel resonance circuit and the secondary side series resonance circuit are provided, and the isolation converter transformer auxiliary winding Ng is provided in the isolation converter transformer PIT. More specifically, provision of these circuits and winding Ng allows the time ratio between the conduction periods of the switching element Q1 and the auxiliary switching element Q2 (ratio of the period TON to the period TOFF) to respond to changes in the load power and the input AC voltage VAC. changes, which can narrow the variable range Δfs. (variation of secondary side circuit)

14 and 15 illustrate variations of the secondary-side circuits that can be applied to the first and second embodiments. Although the isolation converter transformer auxiliary winding Ng is provided in the isolation converter transformer PIT in the first and second embodiments, illustration of the isolation converter transformer auxiliary winding Ng is omitted in FIGS. 14 and 15 . The circuit shown in Figure 14 is a voltage doubler half-wave rectification circuit and offers the advantage of achieving doubled rectified voltage. In this circuit, the secondary side series resonant circuit is formed by the leakage inductance L2 of the secondary winding N2 and the secondary side series resonant capacitor C4. The secondary side rectification and smoothing circuit is connected in series to the secondary side series resonant circuit.

The secondary-side rectifying element is formed by a series circuit of two rectifying diodes Do1 and Do2 whose terminals of opposite polarity are connected to each other. The secondary-side smoothing capacitor Co is connected to both ends of the series circuit of rectification diodes Do1 and Do2. In this voltage doubler half-wave rectification circuit, during a period of half a cycle in which a voltage of one polarity is generated in the secondary winding N2, the current flows through the rectifier diode Do2, thereby maintaining the DC voltage. During the period of one half cycle of the other polarity, current flows through the rectifying diode Do1, thereby generating a voltage across the secondary-side smoothing capacitor Co. At this time, the DC voltage held by the secondary-side series resonance capacitor C4 is added to the voltage across the secondary-side smoothing capacitor Co, so that the resulting voltage is output as the DC output voltage Eo.

The circuit shown in Figure 15 is a voltage doubler full-wave rectification circuit. Specifically, the circuit of FIG. 15 includes a DC voltage holding capacitor Co' that is absent in the circuit of FIG. 14 . If this DC voltage holding capacitor Co' is not included, the circuit shown in FIG. 15 is the same as a circuit obtained by combining two voltage multiplier half-wave rectification circuits in FIG. 16 with each other. First, a circuit obtained by removing the DC voltage holding capacitor Co' from the circuit of Fig. 15 will be described. After that, the circuit of Fig. 15 including the DC voltage holding capacitor Co' will be described. In the circuit of Fig. 15, the capacitance value of the DC voltage holding capacitor Co' is significantly larger than the capacitance values of the first secondary side series resonance capacitor C4 and the second secondary side series resonance capacitor C4'.

As secondary windings, the first secondary partial winding N2' and the second secondary partial winding N2" are formed using a center tap, the winding direction of the second secondary partial winding N2" is the same as the winding direction of the first secondary partial winding N2' same direction. Specifically, when the center tap is defined as the reference, the voltage developed at the winding end of the first secondary partial winding N2' opposite to the center tap side and at the side of the second secondary partial winding N2" connected to the center tap The voltages developed at the opposite winding ends are in opposite phases.

In addition, the secondary side series resonance circuit is formed by the first secondary side series resonance circuit and the second secondary side series resonance circuit. The resonance frequency of the first secondary side series resonance circuit is governed by the leakage inductance L2' generated in the first secondary partial winding N2' and the first secondary side series resonance capacitor C4. The resonance frequency of the second secondary side series resonant circuit is governed by the leakage inductance L2" developed in the second secondary partial winding N2" and the second secondary side series resonant capacitor C4'. The respective inductance and resistance values of the leakage inductance L2', the first secondary side series resonance capacitor C4, the leakage inductance L2", and the second secondary side series resonance capacitor C4' are set such that the first and second secondary sides are connected in series The resonant circuits have substantially the same resonant frequency.

A secondary side rectification and smoothing circuit is formed by a first secondary side rectification and smoothing circuit and a second secondary side rectification and smoothing circuit. The first secondary side rectifying and smoothing circuit includes rectifying diodes Do1 and Do2 as first secondary side rectifying elements, which are connected in series to the first secondary side, and a secondary side smoothing capacitor Co. side series resonant circuit. The second secondary side rectifying and smoothing circuit includes rectifying diodes Do3 and Do4 as second secondary side rectifying elements, which are connected in series to the second secondary side, and a secondary side smoothing capacitor Co. side series resonant circuit. The secondary side smoothing capacitor Co is connected to both ends of the series circuit of rectification diodes Do1 and Do2, and is connected to both ends of the series circuit of rectification diodes Do3 and Do4. In this way, a voltage doubler full-wave rectification circuit is constructed.

During the period of one half cycle when a voltage of one polarity is generated in the secondary partial windings N2' and N2", the current flows through the rectifier diode Do2, thereby maintaining the DC voltage through the first secondary side series resonant capacitor C4. At During the half-cycle period of the other polarity, current flows through the rectifier diode Do1, so that a voltage is generated across the secondary side smoothing capacitor Co. At this time, the DC voltage held by the first secondary side series resonance capacitor C4 is added to the voltage across the secondary side smoothing capacitor Co, so that the resulting voltage is output as the DC output voltage Eo. Similarly, during the half cycle period of the other polarity, current flows through the rectifier diode Do4 , so that the DC voltage is maintained by the second secondary side series resonant capacitor C4'. During the period of half cycle of said one polarity, the current flows through the rectifier diode Do3, so that the voltage is across the two sides of the secondary side smoothing capacitor Co At this time, the DC voltage held by the second secondary side series resonance capacitor C4' is added to the voltage across the secondary side smoothing capacitor Co, so that the resulting voltage is output as the DC output voltage Eo. In this In this way, voltage doubling is achieved and the circuit of FIG. 15 operates as a voltage doubler full wave rectification circuit, wherein each voltage doubler rectification circuit operates in a full half cycle of both polarities.

The above operation corresponds to the case where the DC voltage holding capacitor Co' is not included. Conversely, if a DC hold capacitor Co' is included, the voltage held by the first secondary side series resonant capacitor C4 and the voltage held by the second secondary side series resonant capacitor C4' are also held by the DC voltage hold capacitor Co', This eliminates the need for the first and second secondary side series resonant capacitors C4 and C4' to maintain the DC voltage. As a result, capacitors C4 and C4' are not required to have favorable DC characteristics, which facilitates component selection. The reason why the first and second secondary side series resonant capacitors C4 and C4' do not need to hold the DC voltage is as follows: Depending on the respective capacitance values, the DC voltage is captured by the DC voltage holding capacitor Co' and the first secondary side series resonant capacitor C4 The voltage is divided, or the voltage is divided by the DC voltage holding capacitor Co' and the second secondary side series resonant capacitor C4', and the capacitance value of the DC voltage holding capacitor Co' is significantly larger than that of the first and second secondary side series resonant capacitors C4 and the capacitance value of C4'.

It should be noted that the present invention is not limited to the configurations shown in the above embodiments. For example, as switching elements (and auxiliary switching elements), for example, insulated gate bipolar transistors (IGBTs) or bipolar transistors may be used instead of MOSFETs. In addition, although the above-described embodiment uses a separately excited switching converter, the present invention can also be applied to a configuration using a self-excited switching converter.

It should be understood by those skilled in the art that various modifications, combinations, sub-combinations and changes may occur depending on design requirements and other factors so long as they are within the scope of the appended claims or the equivalents thereof.

Cross References to Related Applications

The present invention contains subject matter related to Japanese Patent Application JP2005-287759 filed in Japan Patent Office on September 30, 2005 and Japanese Patent Application JP2005-291082 filed in Japan Patent Office on October 4, 2005, the entire contents of which are incorporated by reference here.

Claims (9)

1. switching power circuit comprises:

Switch element is realized the switch of direct voltage being alternating voltage with dc voltage conversion;

Converter transformer comprises elementary winding and secondary winding, and the described alternating voltage that the conversion by described switch element obtains is transfused to described elementary winding, makes to generate an alternating voltage in described secondary winding;

Primary side rectification and smoothing circuit comprise that the alternating voltage that is used for that described secondary winding is generated carries out rectification and level and smooth to produce the primary side rectifier cell and the primary side smmothing capacitor of output dc voltage;

The switch element control unit is based on the described switch element of described output DC pressure-controlled;

Choking-winding is provided described direct voltage by an end, and is connected to a winding terminal of the elementary winding in the described converter transformer and a terminal of described switch element via the other end;

The primary side series resonant circuit, form between another winding terminal by the primary side series resonance capacitor being connected the elementary winding in the described converter transformer and another terminal of described switch element, and have the leakage inductance that produces in the elementary winding that is subjected in the described converter transformer and the resonance frequency of described primary side series resonance capacitor domination;

The primary side antiresonant circuit forms by primary side parallel resonance capacitor and described switch element are connected in parallel, and has and be subjected to the leakage inductance that produces in the described elementary winding and the resonance frequency of described primary side parallel resonance capacitor domination; And

Series circuit is formed by clamp capacitor and auxiliary switch element, and is connected in parallel to described choking-winding,

Wherein, described auxiliary switch element is controlled to conducting in the non-conduction period of described switch element.

2. switching power circuit according to claim 1, wherein

Turning on and off of described auxiliary switch element is by the auxiliary winding control of the converter transformer that twines around described converter transformer.

3. switching power circuit according to claim 1, wherein

Turning on and off of described auxiliary switch element is by the auxiliary winding control of the choking-winding that twines around described choking-winding.

4. switching power circuit according to claim 1 also comprises

The partial resonance capacitor is connected in parallel to the leakage inductance and the described secondary winding that produce in the secondary winding in the described converter transformer.

5. switching power circuit according to claim 1 also comprises

The primary side series resonance capacitor that primary side series resonant circuit, the resonance frequency that it had are subjected to the leakage inductance that produces in the secondary winding in the described converter transformer and are connected in series to described secondary winding is arranged.

6. switching power circuit according to claim 1, wherein

Described primary side rectifier cell is that the bridgt circuit by the rectifier diode with input side and outlet side forms, and serves as full-wave rectifying circuit, and

Described primary side smmothing capacitor is connected to the outlet side of the bridgt circuit of described rectifier diode.

7. switching power circuit according to claim 1, wherein

Described primary side rectifier cell is that the series circuit by two rectifier diodes forms, and the terminal of the opposite polarity of described two rectifier diodes is connected to each other, and

The two ends that described primary side smmothing capacitor is connected to the series circuit of described two rectifier diodes make the voltage multiplier half-wave rectifying circuit be formed.

8. switching power circuit according to claim 1, wherein

Described primary side rectification and smoothing circuit are full-wave rectifying circuits, and it comprises the centre cap in the secondary winding that is located at described converter transformer.

9. switching power circuit according to claim 1 also comprises

Primary side rectification and smoothing circuit, comprise the primary side rectifier cell and the primary side smmothing capacitor that are used to produce by the rectification smooth voltage that alternating current is carried out rectification and smoothly obtain, described direct voltage provides from described primary side rectification and smoothing circuit.

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