CN107112972A - Tunable matching network with phase switching elements - Google Patents

Abstract

Phase-switched tunable impedance networks (PS-TMNs) are described. The PS-TMN has an input coupleable to a source and an output coupleable to a load. The PS-TMN includes one or more phase-switched reactive elements. The controller provides a respective control signal to each of the one or more phase-switched reactances. Each phase-switched reactive element provides a respective selected reactance value in response to a respective control signal provided thereto.

Description

Tunable matching network with phase switching elements

Cross References to Related Applications

This application claims the benefit of the filing date of US Provisional Application No. 62/094,144, filed December 19, 2014, which is hereby incorporated by reference in its entirety, under 35 USC § 119e.

Background technique

Impedance matching networks are commonly used to maximize power transfer in many radio frequency (FR) and microwave systems. For example, in an RF transmitter, an impedance matching network may be used to provide impedance matching from the output impedance of an RF power amplifier (PA) to the impedance of an RF load (eg, an antenna). Such impedance matching increases the transmitted power, reduces power loss and reduces or eliminates the need for additional circuit components such as isolators.

One class of impedance matching networks refers to tunable impedance matching networks (TMNs), sometimes referred to as automatic antenna tuning elements. A conventional TMN can be implemented as a single-element or lumped-element reactive network, where at least one reactive element is a variable (e.g., tunable) component such that the impedance of the variable component at a particular frequency or over a range of frequencies can be modified. The reactive elements within a TMN may be arranged in a circuit topology, such as a ladder network, an L-shaped network, a T-shaped network, or a Pi-network.

Conventional TMNs can be classified as analog (continuously adjustable) or digital (adjustable among a set of discrete values). An analog TMN utilizes a variable reactive element with a reactive value (at a certain frequency or within a range of frequencies), which can be tuned by adjusting the bias conditions. Digital TMNs implement variable reactive elements as digitally switched arrays of static reactive elements. This method allows the adjustment of the impedance of the reactance value in limited and discrete steps.

Analog TMNs are often implemented using varactor diodes (or varactor diode circuits) or microelectromechanical systems (MEMS) varactors. Although analog TMNs allow fast and accurate impedance matching over a wide range of impedances, relatively high bias voltages are required to operate at high power levels.

Digital TMNs are often implemented using CMOS switches, MEMS switches, PIN diodes, or discrete power transistors. Although MEMS switches have low on-state resistance and can operate with low power consumption up to tens of GHz, MEMS switches require large control voltages. Digital TMNs based on PIN diodes and CMOS switches exhibit low to moderate on-state resistance, and thus can handle high power levels at the expense of some resistive power loss. Digital TMNs based on PIN diodes and CMOS switches are advantageous for on-die integration eg for software defined radio (SDR) integrated circuits (ICs) and other on-chip TMNs. However, digital TMNs exhibit limited tuning resolution, and thus limited accuracy achievable with impedance matching. In some high-power applications (such as RF plasma drivers) that require accurate impedance matching over a very wide impedance range, the use of a digital TMN may not be practical due to the large number of digital switches required to achieve the required fine tuning resolution.

Contents of the invention

This Summary is provided to introduce a selection of concepts in a simplified form that are further described below in the Detailed Description. This Summary is not intended to identify key features or essential features of the claimed subject matter, nor is it intended to be used to limit the scope of the claimed subject matter.

It has been recognized that there is a need for a TMN with increased accuracy and/or faster impedance matching relative to existing TMNs. It is also recognized that there is a need for faster impedance matching with increased accuracy and/or with higher tuning bandwidth over a wider impedance range while allowing for higher power levels with low insertion loss TMN needs under operation.

One aspect of the concepts, systems and techniques described herein is directed to a phase-switched tunable impedance network having an input configured to be coupled to a source and having an output configured to be coupled to a load and one or more phase-switched reactive element. A controller coupled to the phase-switched tunable impedance network provides a respective control signal to each of the one or more phase-switched reactances. Each phase-switched reactive element provides a desired reactance value in response to a corresponding control signal provided thereto.

For this particular arrangement, a phase switched TMN with increased accuracy and/or faster impedance matching compared to prior art TMNs is provided. Phase-switched TMNs also provide faster impedance matching with increased accuracy and/or higher tuning bandwidth over a wider impedance range compared to prior art methods, while at the same time allowing operation at higher power levels. In some embodiments, the reactance value of one or more of the one or more phase-switched reactive elements may be the same, and in other embodiments, each of the one or more phase-switched reactive elements may have a different reactance value.

In an embodiment, the controller is internal to the phase switching TMN. In one embodiment, the controller is external to the phase switching TMN. In an embodiment, part of the controller is internal to the phase switching TMN and part of the controller is external to the phase switching TMN.

In an embodiment, setting each phase switching reactance to a corresponding desired reactance value achieves impedance matching between the source and the load. In an embodiment, setting each phase switching reactance to a corresponding desired reactance value achieves a desired impedance transformation between the source and the load. In another embodiment, setting each phase switching reactance to a corresponding desired reactance value achieves a desired first impedance to the source and a desired second impedance to the load.

In an embodiment, each of the one or more phase switching reactive elements includes one or more reactive elements and at least one switch. At least one of the one or more reactive elements is configured to be switched in and out of the reactive network by at least one switch associated therewith.

In an embodiment, at least one associated switch is operable at a switching frequency and a switching phase related to a characteristic (eg frequency) of the RF signal provided by the source based on a corresponding control signal.

In an embodiment, at least one associated switch is operable at a switching frequency and a switching phase related to the frequency of the RF signal provided by the source based on a corresponding control signal. In an embodiment, at least one associated switch is operable at a switching frequency and a switching phase related to the RF signal being processed from the RF source. In at least some cases, the switch is switched once per RF cycle with a timing controlled to provide a desired reactance value. In an embodiment, switching may occur once per RF cycle or twice per RF cycle depending on whether the system is full wave or half wave.

In an embodiment, at least one switch is operable in a half-wave switching configuration to switch on and off once per cycle of the RF signal at the output port of the RF amplifier. In an embodiment, at least one switch is operable in a full wave switching configuration to switch on and off twice per cycle of the RF signal at the output port of the RF amplifier. In an embodiment, the switching frequency and switching phase are selected to provide a phase switched reactance with a desired reactance value. In other embodiments, other relationships to RF frequency may also be used.

In an embodiment, at least one associated switch is turned on and off at the frequency of the RF signal being processed from the RF source and at a timing to provide a phase switched reactance having a desired reactance value.

In an embodiment, at least one associated switch is turned on and off once per RF cycle of the RF source with a timing to provide a phase-switched reactance having a desired reactance value.

In an embodiment, at least one associated switch is turned on and off once per RF cycle of the RF source with a timing to provide phase switched reactance with a desired reactance value and to provide at least one of zero voltage switching and zero current switching of the switch.

In an embodiment, the switching frequency and switching phase are selected to provide a phase switched reactance with a desired reactance value.

In an embodiment, at least one switch is operable to provide at least one of zero voltage switching (ZVS) and zero current switching (ZCS) of the switch.

In an embodiment, the controller is configured to determine the switching frequency and select the switching phase based on at least one of: a feedback circuit, a feedforward circuit, and an adaptive predistortion system. In an embodiment, the adaptive predistortion system includes a look-up table.

In one embodiment, the control signal operates in a half-wave switching configuration based on a signal provided from a source. In another embodiment, the control signal operates in a full wave switching configuration based on a signal provided from the source.

In one embodiment, the adaptive predistortion system includes a look-up table.

In one embodiment, the phase-switched reactance is a capacitive element, and the capacitance value of the phase-switched capacitive element at a desired frequency is related to the physical DC capacitance value of the phase-switched capacitive element and the switching phase. In another embodiment, the phase switching reactance is an inductive element, and the inductance value of the phase switching inductive element at the desired frequency is related to the physical DC inductance value of the phase switching inductive element and the switching phase.

In an embodiment, the tunable impedance network includes a digital reactance matrix with N selectable reactive elements to adjust the effective reactance value of the digital reactance matrix, where N is a positive integer.

In an embodiment, the tunable impedance network includes one or more analog varactor elements.

In an embodiment, the source is at least one of a radio frequency (RF) source, an RF power amplifier (PA), and a switched mode inverter, and the load is at least one of an antenna, a transmission line, and a plasma load.

In an embodiment, an input of the tunable impedance network is coupled to a radio frequency (RF) amplifier system. The tunable impedance network modulates the load impedance of the RF amplifier system to control the power level of the RF amplifier system.

In an embodiment, the tunable impedance network includes one or more filter components to reduce harmonic content coupled to at least one of the input and output.

In another aspect, a method of operating a tunable impedance network is provided. The tunable impedance network includes an input configured to be coupled to a source, an output configured to be coupled to a load, and one or more phase switched reactances. A controller coupled to the tunable impedance network determines a desired impedance value of the tunable impedance network. A controller provides a respective control signal to each of the one or more phase switching reactances. A corresponding desired reactance value for each phase switching reactance is set in response to a corresponding control signal provided thereto.

In one embodiment, setting each phase switching reactance to a corresponding desired reactance value achieves impedance matching between the source and the load. In another embodiment, setting each phase switching reactance to a corresponding desired reactance value achieves a desired impedance transformation between the source and the load. In another embodiment, setting each phase switching reactance to a corresponding desired reactance value achieves a desired first impedance to the source and a desired second impedance to the load.

In an embodiment, each of the one or more phase switching reactances is provided from a combination of one or more reactive elements and at least one switch. At least one of the one or more reactive elements is configured to be switched in and out of the reactive network by at least one switch associated therewith.

In an embodiment, at least one associated switch is operable at a switching frequency and a switching phase based on a corresponding control signal.

In an embodiment, the switching frequency and switching phase are selected to provide a phase switched reactance with a desired reactance value.

In an embodiment, at least one associated switch is operable at a switching frequency and a switching phase related to the frequency of the RF amplifier system based on a corresponding control signal.

In an embodiment, at least one associated switch is selected to provide phase-switched reactance with a desired reactance value and to provide at least one of zero-voltage switching and zero-current switching of said switch every RF cycle of the RF amplifier system. Switch on and off once. In an embodiment, the controller determines the switching frequency and selects the switching phase based on at least one of: a feedback circuit, a feedforward circuit, and an adaptive predistortion system.

In an embodiment, at least one switch is operable at a switching frequency and a switching phase related to the frequency of the RF signal provided by the source based on a corresponding control signal. In an embodiment, at least one switch is operable in a half-wave switching configuration to switch on and off once per cycle of the RF signal at the output port of the RF amplifier. In an embodiment, at least one switch is operable in a full wave switching configuration to switch on and off twice per cycle of the RF signal at the output port of the RF amplifier.

In an embodiment, at least one switch is operable to provide at least one of zero voltage switching (ZVS) and zero current switching (ZCS) of said switch.

In an embodiment, the phase-switched reactance includes a capacitive element, and the capacitance value of the phase-switched capacitive element at a desired frequency is related to the physical DC capacitance value of the phase-switched capacitive element and the switching phase.

In one embodiment, the phase-switched reactance is a capacitive element, and the capacitance value of the phase-switched capacitive element at a desired frequency is related to the physical DC capacitance value of the phase-switched capacitive element and the switching phase. In another embodiment, the phase switching reactance is an inductive element, and the inductance value of the phase switching inductive element at the desired frequency is related to the physical DC inductance value of the phase switching inductive element and the switching phase.

In an embodiment, the tunable impedance network includes a digital reactance matrix with N selectable reactive elements to adjust the effective reactance value of the digital reactance matrix, where N is a positive integer.

In an embodiment, the tunable impedance network includes one or more analog varactor elements.

In an embodiment, the source is at least one of a radio frequency (RF) source, an RF power amplifier (PA), and a switched mode inverter, and the load is at least one of an antenna, a transmission line, and a plasma load.

In an embodiment, an input of the tunable impedance network is coupled to a radio frequency (RF) amplifier system. The tunable impedance network modulates the load impedance of the RF amplifier system to control the power level of the RF amplifier system.

In an embodiment, the tunable impedance network includes one or more filter components to reduce harmonic content coupled to at least one of the input and output.

In another aspect of the concepts, systems and techniques described herein, a radio frequency (RF) amplifier system having an input port and an output port includes an RF amplifier having an input port coupled to the input port of the RF amplifier system and having an output port. A phase-switched tunable impedance network is coupled between the output port of the RF amplifier and the output port of the RF amplifier system. The phase-switched tunable impedance network changes its impedance to modulate the impedance presented to the output port of the RF amplifier.

In an embodiment, the phase-switched tunable impedance network includes one or more phase-switched reactive elements. Each phase switched reactive element receives a corresponding control signal. Each phase switched reactive element is provided with a corresponding desired reactance value in response to a corresponding control signal provided thereto.

In an embodiment, the controller provides a respective control signal to each of the one or more phase switched reactive elements.

In an embodiment, each of the one or more phase switching reactive elements includes one or more reactive elements and at least one switch. At least one of the one or more reactive elements is configured to be switched in and out of the phase-switched tunable impedance network by at least one switch associated therewith.

In an embodiment, at least one associated switch is operable at a switching frequency and a switching phase related to the frequency of the RF signal at the output port of the RF amplifier based on a corresponding control signal.

In an embodiment, at least one switch is operable in a half-wave switching configuration to switch on and off once per cycle of the RF signal at the output port of the RF amplifier.

In an embodiment, at least one switch is operable in a full wave switching configuration to switch on and off twice per cycle of the RF signal at the output port of the RF amplifier.

In an embodiment, each of the one or more phase switched reactive elements includes a capacitor in parallel with the switch. In an embodiment, each of the one or more phase-switched reactive elements includes an inductor in series with a combination of capacitors in parallel with the switch.

In an embodiment at least one switch is operable to provide at least one of zero voltage switching and zero current switching of said at least one switch.

In an embodiment, the switching frequency and switching phase are selected to provide a phase-switched reactive element having a desired reactance value.

In an embodiment, at least one of the one or more phase switched reactive elements is a capacitive element having a capacitance value. The capacitance value of the phase-switched capacitive element at the desired frequency is related to the physical DC capacitance value of the phase-switched capacitive element and the switching phase. In an embodiment, at least one of the one or more phase switched reactive elements is an inductive element having an inductive value. The inductance value of the phase switching inductive element at the desired frequency is related to the physical DC inductance value of the phase switching inductive element and the switching phase.

In an embodiment, the impedance presented by the phase-switched tunable matching network to the output port of the RF amplifier is dynamically adapted to match the impedance of a variable load coupled to the output port of the RF amplifier system to the impedance of the RF amplifier.

In an embodiment, an RF load is coupled to the output port of the RF amplifier system. The RF load is at least one of an antenna, a transmission line, and a plasma load.

In an embodiment, the RF amplifier includes a switching inverter including at least one switching element configured to generate RF power.

In an embodiment, the controller modulates the impedance of the phase-switched tunable impedance network presented to the output port of the RF amplifier such that the RF amplifier maintains zero voltage switching (ZVS) switching at least one switching element of the inverter.

In an embodiment, at least one switch switching each of the one or more phase-switched reactive elements modulates the impedance transformation provided by the phase-switched tunable impedance network.

In an embodiment, the RF amplifier includes a filter coupled to a phase-switched tunable impedance network. The filter has filter characteristics that reduce harmonic content presented to at least one of the output port of the RF amplifier and the output port of the RF amplifier system produced by the phase-switched tunable impedance network. In an embodiment, the filter includes one or more filter components to electrically isolate the DC signal between the phase-switched tunable impedance network and at least one of the RF amplifier and the output port of the RF amplifier system.

In an embodiment, the phase-switched tunable impedance network includes a first parallel-path phase-switching varactor element coupled to a first node of the series-path reactive element. In an embodiment, the phase-switched tunable impedance network further includes a second parallel-path phase-switching variable reactive element coupled to the second node of the series-path reactive element.

In another aspect, a method of operating a radio frequency (RF) amplifier system includes providing an RF signal to an input port of an RF amplifier and amplifying the RF signal by the RF amplifier to provide an amplified RF signal at an output port of the amplifier. Impedance modulation of a phase-switched tunable impedance network coupled between an output port of the RF amplifier and an output port of the RF amplifier system varies an impedance presented to the RF amplifier.

In an embodiment, changing the impedance of the phase-switched tunable impedance network includes receiving a control signal by the phase-switched tunable impedance network. Responsive to the control signal, at least one reactive element is electrically connected to or disconnected from the phase-switching tunable impedance network at the output port of the RF amplifier at a frequency and phase related to the frequency of the RF signal to Provide a phase-switched tunable impedance network with a desired reactance value.

In an embodiment, the phase-switched tunable impedance network includes one or more reactive elements and at least one switch. Electrically connecting or disconnecting at least one reactive element into or from the phase-switched tunable impedance network includes switching at least one of the one or more reactive elements into and out of phase through at least one switch associated therewith The tunable impedance network is switched, and at least one associated switch is operated at a switching frequency and a switching phase based on a corresponding control signal.

In an embodiment, the switching frequency and switching phase are selected based on the frequency of the RF signal at the output port of the RF amplifier to provide a phase switched reactive element with a desired reactance value.

In an embodiment, the impedance presented to the RF amplifier by the phase-switched tunable matching network is dynamically adapted to match the impedance of a variable load coupled to the output port of the RF amplifier system to the impedance of the RF amplifier.

In an embodiment, modulating the load impedance presented to the RF amplifier by the phase-switched tunable matching network controls the power level of the amplified signal provided to the output port of the RF amplifier system.

In an embodiment, the amplifier includes a switching inverter including at least one switching element configured to generate RF power. Zero voltage switching (ZVS) of at least one switching element of the switching inverter is maintained by modulating the impedance of the phase switching tunable impedance network presented to the RF amplifier.

In an embodiment, at least one switch of the phase switched tunable impedance network is switched to provide at least one of zero voltage switching and zero current switching of the at least one switch.

Description of drawings

Other aspects, features, and advantages of the claimed invention will become more fully apparent from the following detailed description, appended claims, and drawings, wherein like reference numerals identify similar or identical elements. Reference numerals introduced in the specification in relation to a drawing may be repeated in one or more subsequent figures without additional description in the specification, in order to provide context for other features.

FIG. 1 is a block diagram of an illustrative tunable impedance matching network (TMN) according to the described embodiments;

2 is a schematic diagram of an illustrative phase-switching variable capacitive element of the TMN of FIG. 1;

3 is a graph of current and voltage versus phase for a control signal of the phase-switched variable capacitive element of FIG. 2 .

4 is a schematic diagram of an illustrative phase-switching variable inductance element of the TMN of FIG. 1;

5 is a graph of current and voltage versus phase for a control signal of the phase switching variable inductance element of FIG. 4 .

Figure 6 is a graph of the normalized effective capacitance (or inductance) of the phase switching element of Figures 2 and 4 compared to the control angle of the phase switching element;

7 is a graph of total harmonic distortion of the phase switching element of FIGS. 2 and 4 compared to the control angle of the phase switching element;

Figure 8 is a graph of current and voltage versus phase for a control signal of a full-wave switched varactor element;

9 is a graph of current and voltage versus phase for a control signal of a full-wave switched variable inductance element;

10A-D are schematic diagrams of illustrative switched reactive elements in accordance with described embodiments;

11 is a schematic diagram of an illustrative phase-switched TMN using a digitally switched capacitor matrix;

12 is a schematic diagram of an illustrative phase-switched TMN using a digitally switched inductance matrix;

Figure 13 is a schematic diagram of an illustrative phase-switched TMN in accordance with the described embodiments;

14 is a Smith chart of the range of load impedances that can be matched by the tuning network of FIG. 13 for an illustrative range of operation;

Figure 15 is a schematic diagram of additional details of the tuning network of Figure 13;

16 is a block diagram of an illustrative topology of a phase switched impedance modulated amplifier according to the described embodiments;

17 is a block diagram of another illustrative topology of a phase switched impedance modulated amplifier according to the described embodiments;

18A-E are schematic diagrams of an illustrative three-switch phase-switched impedance modulated amplifier in accordance with described embodiments;

19 and 20 are schematic diagrams of illustrative two-switch phase-switched impedance modulated amplifiers in accordance with described embodiments;

21 is a schematic diagram of an illustrative phase-switched impedance modulated amplifier within an illustrative operating range;

22 and 23 are Smith charts showing the range of load impedances that may be matched by the phase-switched impedance-modulated amplifier of FIG. 21 for an illustrative range of operation; and

24 is a flowchart of an illustrative process for operating the TMN of FIG. 1 .

detailed description

Table 1 summarizes a list of abbreviations used throughout the specification to aid in understanding the examples:

The embodiments are directed towards a tunable matching network based on phase-switched variable network reactive elements, referred to herein as a phase-switched tunable matching network (PS-TMN). Such PS-TMNs provide fast, high bandwidth, continuous impedance matching over a wide impedance range while operating efficiently at high power levels without requiring high bias voltages or currents. Such PS-TMNs can be used alone or in combination with other matching techniques such as discrete switched reactance banks.

Such a PS-TN can be used in various reconfigurable and adaptive RF systems such as RF front ends for software defined radio (SDR) and cognitive radio (CR) applications over a wide range of frequency bands, in Operates at different bandwidths and according to various communication standards. PS-TMN can also be used in other RF applications such as drivers of RF plasma loads to compensate for fast load changes or in wireless power transfer (WPT) to compensate for impedance mismatches between transmitter and receiver to Maximize the transferred power and/or efficiency.

The embodiments also provide zero voltage switching (ZVS) radio frequency (RF) amplifiers referred to herein as phase switched impedance modulated (PSIM) amplifiers. Such a PSIM amplifier can use PS-TMN to operate over a large frequency range by efficiently modulating the output power over a wide frequency range and/or matching into highly variable loads, such as loads that are variable over a wide impedance range. internal operation.

Referring to FIG. 1 , a radio frequency (RF) system 100 includes a phase-switching tunable matching network (PS-TMN) 112 coupled between a source 102 having an impedance Z _S and a load 114 having an impedance Z _L . In some applications, source 112, control circuit 106, and PS-TMN 112 (and other components of RF system 100) are coupled to a supply voltage (eg, V _DC ) and ground. Control circuit 106 is coupled to PS-TMN 112 and provides control signals to PS-TMN 112 in order to control the operation of PS-TMN 112 . In response to such control signals, PS-TMN 112 provides desired impedance transformation characteristics. It should be appreciated that control circuitry 106 may be an internal component of PS-TMN 112, or may be an external component coupled to PS-TMN 112 or some portion of control circuitry 106 (or the functionality provided by control circuitry 106 may be 112, while other parts of the control circuit 106 may be external to the PS-TMN 112).

In some embodiments, control circuit 106 controls PS-TMN 112 based at least in part on information received from optional feedforward circuit 104 coupled to source 102 and/or optional feedback circuit 110 coupled to load 114. operate. In some embodiments, optional feedforward circuitry 104 includes adaptive predistortion circuitry 107 and control circuitry 106 includes a look-up table (LUT) 108 . For example, as will be described in more detail below, some embodiments may use one or more non-linear control techniques (e.g., by control circuitry 106) to determine appropriate control signals for PS-TMN 112, such as using fixed or adaptable look-up tables (such as LUT 108) to store predetermined control signal information, feedback (such as through feedback circuit 110) and/or feedforward compensation (such as through feedforward circuit 104) to adaptively adjust control signal information, or implement control signals (such as Distortion circuit 107) digital predistortion or other similar techniques.

PS-TMN 112 includes one or more phase switched reactive elements 116(1)-116(N). As will be described in more detail below, the phase-switched reactive elements 116(1)-116 may be implemented using one or more capacitive elements (eg, capacitors), one or more inductive elements (eg, inductors), or a combination of both. (N). Phase switched reactive elements 116(1)-116(N) may be controlled to adjust the effective impedance (Z _S,IN and Z _L,IN ) presented to the terminals of PS-TMN 112 at a desired frequency. The phase-switched reactive elements 116(1)-116(N) are switched, eg, by parallel switches or series switches, and the effective impedance of the phase-switched reactive elements is controlled by adjusting the phase and/or duty cycle of the parallel or series switches. In some embodiments, the desired frequency may be the RF operating frequency of RF source 102 (eg, the frequency of the signal provided from RF source 102 to PS-TMN 112).

By modulating the effective impedance at the desired operating frequency of the RF system 100 (for example, by adjusting the impedance of the phase-switching reactive elements 116(1)-116(N), it is possible to adjust, tune, change or otherwise operate the The impedance presented to source 102 and/or load 114 . For example, phase-switched reactive elements 116(1)-116(N) allow presentation of a desired impedance (Z _S,IN ) from source 102 to PS-TMN 112 and from load 114 to PS-TMN 114 (Z _L,IN ). TMN 112.

The control signal provided to PS-TMN 112 operates to control the timing of switching on and/or off the switches of phase-switched reactive elements 116 ( 1 )- 116 (N) relative to the RF signal provided from source 102 . Switching provides the effective reactance values of the phase-switched reactive elements 116 ( 1 )- 116 (N) that achieve the desired impedance transformation of the PS-TMN 112 . Feedforward information may include information about the effective input impedance of PS-TMN 112, the timing of the RF waveform, prescribed signal levels and/or impedance levels, and the like. Feedback information may include measured information about effective load impedance and/or power reflected from the load, timing of RF waveforms, and the like.

Thus in some embodiments, PS-TMN 112 may be used to provide the desired impedance transformation between source 102 and load 114 . For example, PS-TMN 112 may provide impedance matching between source 102 and load 114 . Alternatively, the impedance of PS-TMN 112 may be adjusted to compensate for changes in the impedance (Z _L ) of load 114 such that source 102 is coupled to a more stable impedance (eg, Z _S,IN ) provided by PS-TMN 112 .

Referring to FIG. 2 , a sinusoidal current source 202 having a current I drives an illustrative phase switching varactor 200 . The phase-switched varactor is shown here as comprising a parallel combination of a capacitor 204 and a switch 206 to provide a phase-switched varactor as a phase-switched varactor 200 . Capacitor 204 has a physical capacitance C ₀ and a voltage V _C . The state of switch 206 is controlled by the characteristics of signal Q. For example, when signal Q has a logic high value, switch 206 provides a low impedance signal path between its terminals (e.g., switch 206 is "on" or "closed"), and when signal Q has a logic low value, switch 206 A high impedance signal path is provided between its terminals (eg, switch 206 is "open" or "off"). Thus, switch 206 can be considered to switch capacitor 204 into the circuit when the switch is open (current I flows into capacitor 204) and out of circuit when the switch is closed (current I flows through the closed switch and bypasses capacitor 204) out.

If switch 206 is always open (off), the effective capacitance C _EFF of phase-switching variable capacitance 200 presented to source 202 is equal to the physical capacitance C ₀ of capacitor 204 . Alternatively, if switch 206 is always on (closed), the low impedance path between the terminals of switch 206 effectively "short-circuits" capacitor 204, and the voltage across capacitor 204 of phase-switching variable capacitor 200 remains at Zero in the sense of being independent of the current I behaves as an infinite capacitor. The effective capacitance C _EFF of capacitor 204 can be theoretically controlled between C ₀ and infinity by controlling the conduction angle of switch 206 during the AC cycle of sinusoidal current source 202 from 0 to 2π. As used herein, the conduction angle is the angle of the sinusoidal signal when switch 206 is turned on. The conduction angle at which the switch is turned on may be fully determined by the switching signal Q (eg, switching angle) or partially determined by the switching signal Q and partially by the circuit waveform (eg, voltage V _C and current I).

Referring to FIG. 3 , illustrative waveforms of current I and capacitor voltage V _C (eg, the voltage of capacitor 204 ) are shown for switch control signal Q as a function of period angle θ. Specifically, for a half-wave switched capacitor, curve 302 shows I(θ), curve 306 shows V _C (θ), and curve 304 shows Q(θ). As shown in FIG. 3, at each cycle of I(θ), α radians after I(θ) transitions from negative to positive switch 206 is opened (turned off) (e.g., switch 206 is turned on/closed until α until the arc enters the positive half cycle of I(θ)). Switch 206 remains open (turned off) until after the capacitor voltage drops to zero. Biasing the switch to its conductive state (eg, turning the switch on or closing the switch) after the capacitor voltage drops to zero ensures zero voltage switching (ZVS) conduction of the switch 206 .

If the switch includes a diode that naturally prevents the voltage from going negative, the timing of the active turn-on switch Q can be relaxed because it will naturally transition "on" when the switch voltage reaches zero, and the active turn-on signal can be in the diode conduction is issued when. Capacitor C ₀ across the switch provides inhibition of the off transition, providing zero voltage switching (ZVS) opening of switch 206 .

As shown in FIG. 3, when I(θ) is a purely sinusoidal current source, switch 206 remains open (turned off) until the conduction angle of the switch is reached (eg, at 2α). Thus, for a half-wave switched capacitor, switch 206 is turned on and off once per cycle of the RF signal from source 102 (eg, I(θ) as shown by curve 302 ).

Adjusting a sets when the switch 206 turns on and off during the cycle (eg, the conduction angle of the switch 206 ) and thus controls the voltage at which the capacitor peaks. Thus, there is a relationship between the switching angle (α) and the magnitude of the fundamental component of V _C (θ) at the switching frequency. Therefore, the effective capacitance C _EFF of capacitor 204 can be expressed as a function of α:

Referring to Figure 4, it is also possible to implement a phase-switched varactor as a switched inductor network, which allows continuous control of its effective inductance at the switching frequency. Such a switched inductor network is shown in FIG. 4 as a phase switched variable inductor 400 and corresponds to the topology dual of the switched capacitor network 200 shown in FIG. 2 . As shown in FIG. 4 , an illustrative switched variable inductor 400 includes a series combination of an inductor 404 and a switch 406 driven by a sinusoidal voltage source 402 having a voltage V . Inductor 404 has a physical inductance L ₀ and an inductor current I _L . The state of the switch 406 is controlled by the signal Q, for example, the switch 406 can be turned on (eg, closed) when the signal Q has a logic high value and can be turned off (eg, off) when the signal Q has a logic low value. Thus, the switch 406 can be considered to switch the inductor 404 into the circuit when the switch is closed (applying the voltage V to the inductor 404) and switching the inductor 404 out of the circuit when the switch is open (no voltage V is applied to the inductor 404). 404).

Similar to the switched capacitor implementation of the phase-switched varactor 400 at the switching frequency, the effective inductance L _EFF of the phase-switched varactor 400 at the switching frequency can be modulated from a base value L ₀ to infinity similarly to the switched capacitor embodiment of the phase-switched varactor 400 described for FIG. 2 . For example, if switch 406 is always on (closed), the effective inductance L _EFF of phase switching varactor 400 seen by source 402 is equal to the physical inductance L ₀ of inductor 404 . Alternatively, if switch 406 is always open (turned off), inductor 404 behaves as an infinite inductor in the sense that the current through inductor 404 remains zero independent of voltage V. The effective inductance _LEFF of the inductor 404 can be ideally controlled between L ₀ and infinity by controlling the conduction angle of the switch 406 during the AC cycle of the sinusoidal voltage source 402 from 0 to 2π.

Referring to FIG. 5 , illustrative waveforms of current I and voltage V _C of capacitor 204 are shown for switch control signal Q as a function of period angle θ. As a result of the properties of topological duality, the voltage waveform of the switched capacitor network shown in Figure 3 is similar to the current waveform of the switched inductor shown in Figure 5, and vice versa.

Specifically, for a half-wave switched inductor, curve 502 shows _IL (θ), curve 506 shows V(θ), and curve 504 shows Q(θ). As shown in FIG. 5, at each cycle of V(θ), α radian switch 406 is turned on (closed) after V(θ) transitions from negative to positive (e.g., switch 406 is open/closed until α until the arc enters the positive half cycle of V(θ)). Switch 406 remains on (closed) until after the inductor voltage drops to zero. Because the switch has an inductor in series with it, zero current switching (ZCS) of the switch can be achieved. Opening the switch at the time when the inductor voltage drops to zero ensures zero current switching (ZCS) of the switch 406 . In duality with a capacitive circuit, utilizing a diode as part of switch Q ensures natural switching (opening) of the switch and relaxes detailed active timing of the switching control waveform's off moment. As shown in FIG. 5, when V(θ) is a purely sinusoidal voltage source, switch 406 remains on (closed) until the conduction angle of the switch is reached (eg, at 2α).

Adjusting α sets when the switch 406 turns on and off during the cycle (eg, controls the conduction angle of the switch 406 ) and thus controls the current at which the inductor peaks. Thus, similar to the phase-switched varactor switched capacitor embodiment described for FIG. 2 , there is a relationship between the switching angle (α) and the magnitude of the fundamental component of _IL (θ) at the switching frequency. Therefore, the effective capacitance _LEFF of inductor 404 can be expressed as a function of α:

As a result of topological duality, the expression (1b) for the effective inductance is the same as the expression (1a) for the effective capacitance. Expression (1a) agrees with the intuitive expectation of infinite effective capacitance when the switch is always on (α=π), and predicts that between C _EFF and C ₀ when the switch is permanently off (α=0). equivalence between them. Expression (1b) agrees with the intuitive expectation of infinite effective inductance when the switch is always in the off state (α=0), and predicts that between _LEFF and _L0 when the switch is permanently on (α=π) equivalence between them. Thus, according to expressions (1a) and (1b), the effective capacitance C _EFF and effective inductance _LEFF at the switching frequency can be modulated by controlling the conduction angle of the switch associated with the capacitor or inductor.

Referring to FIG. 6 , the normalized effective capacitance C _EFF /C ₀ or the normalized effective inductance L _EFF /L ₀ is shown by curve 602 at the switching frequency. For a capacitive circuit, this is the same thing as the normalized admittance Y _EFF /Y ₀ , and for an inductive circuit it is the same thing as the normalized reactance X _EFF /X ₀ . As a result of topological duality, the normalized admittance Y _EFF /Y ₀ of the phase-switched capacitor circuit of FIG. 2 is the same as the normalized reactance X _EFF /X ₀ of the phase-switched inductor circuit of FIG. 4 .

As shown in FIG. 6 , the normalized effective capacitance C _EFF (or inductance L _EFF ) increases rapidly with α and approaches infinity when α approaches π (eg, 180 degrees).

Referring to FIG. 7 , curve 702 shows the total harmonic distortion versus α for capacitor voltage (inductor current) for a purely sinusoidal current (voltage) excitation source. The actual range over which C _EFF or _LEFF can be modulated depends on the amount of harmonic distortion that may be present in the network. As α increases toward π (eg, the conduction angle of the switch increases), the spiral rise of capacitor voltage V _C (eg, curve 306 ) or inductor current _IL (eg, curve 502 ) is limited to a shorter period of time. As shown in Figure 7, this results in a considerable harmonic content of the capacitor voltage for a large Y _EFF /Y ₀ or X _EEF /X ₀ (eg C _EEF /C ₀ or L _EEF /L ₀ ) ratio (eg as α increases total harmonic distortion increases). The amount of harmonic distortion allowed in a given system depends on the regulatory limits on the harmonic currents allowed in the source and/or load and the amount of filtering necessary or required.

Note that FIG. 7 shows the harmonic distortion of the phase-switched varactor (eg, the harmonic distortion of the capacitor voltage of the phase-switched varactor 200 or the harmonic distortion of the inductor voltage of the phase-switched varactor 400), and Not the harmonic content that is actually injected into the source and/or load (eg, source 102 and load 114 ) of the RF system. In some embodiments, a phase-switched varactor (eg, phase-switched varactor 200 or phase-switched varactor 400 ) includes additional filtering components (not shown in FIGS. 2 and 4 ) to reduce the injected Harmonic content into sources and/or loads (eg, source 102 and load 114).

As described with respect to FIGS. 3 and 5 , a phase-switched variable reactance (eg, phase-switched variable capacitor 200 or phase-switched variable inductor 400 ) is half-wave switched, wherein the switches are operated such that the capacitor voltage ( FIG. 3 curve 306) and the inductor current (curve 502 of Figure 5) are unipolar. However, other switching schemes are also possible. For example, FIGS. 8 and 9 show illustrative waveforms of the current I and the voltage V of the switch control signal Q as a function of the period angle θ for the switched capacitor network shown in FIG. 3 and the switched inductor network shown in FIG. 5, respectively. .

Specifically, as shown in FIG. 8 , for a full wave switched capacitor, curve 802 shows I(θ), curve 806 shows V _C (θ), and curve 804 shows Q(θ). As shown in FIG. 9 , for a full wave switched inductor, curve 902 shows _IL (θ), curve 906 shows V(θ), and curve 904 shows Q(θ). When the phase-switching variable capacitor 200 is full-wave switched, the switch (e.g., switch 206) is turned off twice per cycle of I(θ) (e.g., Q(θ) is zero), and the off-period is equal to when I( θ) is zero time as the center. For a purely sinusoidal excitation current I(θ), this produces a bipolar capacitor voltage waveform V _C (θ). Capacitor voltage V _C (θ) has a zero DC average value. Similarly, when phase-switching varactor 400 is full-wave switched, a switch (e.g., switch 406) is turned off twice per cycle of V(θ) (e.g., Q(θ) has a logic high value), then The on-period is centered on the instant when V(θ) is zero. For a purely sinusoidal excitation voltage V(θ), this produces a bipolar inductor current waveform _IL (θ), which also has a zero DC mean value. Thus, for a full-wave switched capacitor (or inductor), switch 206 is turned on and off twice per cycle of the RF signal from source 102 (eg, I(θ) as shown by curve 802 ).

As with half-wave switching (eg, as shown in FIGS. 3 and 5 ), the effective capacitance C _EEF and effective inductance _LEFF at the switching frequency can be modulated by controlling the switching angle α of the switch. For a full-wave switched capacitor, the effective capacitance C _EFF of capacitor 204 can be expressed as a function of α:

Similarly, the effective inductance _LEFF of inductor 404 can be expressed as a function of α:

Therefore, the effective capacitance/inductance that can be achieved for a given switching angle α using a full-wave switched network (such as relations (2a) and (2b)) is that which can be achieved using a half-wave switched network (such as relations (1a) and (1b) ) half of the effective capacitance/inductance achieved. However, for the same switching angle α (ie, the switching angle controlling the overall switch conduction angle), full-wave switching networks inherently produce reduced harmonic content of capacitor voltage and inductor current compared to half-wave switching networks. On the other hand, implementing full-wave switching requires that the switch must operate at twice the operating frequency (eg, to switch twice per cycle). Furthermore, for capacitive modulation, bidirectional blocking switches are required, which can complicate switch implementations with general semiconductor switches.

Relationships (1) and (2) above show that for a purely sinusoidal excitation signal, the effective capacitance and inductance of the switching network shown in Figures 2 and 4 can be based on the switching angle α. For excitation signals that are not purely sinusoidal, the effective reactance can be controlled by appropriate choice of timing or the switching angle α at which the switch turns off (or on), although relations (1) and (2) cannot calculate α the exact value of . Along with the circuit waveform determining the point of zero voltage (or zero current) as the switch is turned on (or off), the switching angle α determines the total conduction angle of the switch during the cycle. For excitation signals that are not purely sinusoidal, an adaptable look-up table (such as LUT 108), feedback circuit 110, or feedforward circuit 104 (including optional digital predistortion circuit 107) can be used to determine the value of α for a given desired effective reactance. desired value.

Phase-switched variable capacitor 200 and phase-switched variable inductor 400 can be used as building blocks for implementing phase-switched variable reactance and other tunable circuits (eg, TMN). In particular, some applications can substantially benefit from variable reactance, the value of which can be controlled over a range across capacitive and inductive reactance and/or by modulating the effective reactance over a more limited range. Adding phase-switched variable capacitor 200 and/or phase-switched variable inductor 400 with additional reactive components can provide a wider range of variable reactance.

10A-10D show illustrative embodiments of phase-switched reactive circuits including capacitive and inductive elements, thereby expanding the range over which the impedance of the phase-switched reactive circuit can be tuned compared to the single-element circuits shown in FIGS. 2 and 4 .

For example, FIG. 10A shows a phase switched reactive circuit 1002 including an inductor 1012 in series with a phase switched capacitor 1013 . Phase switching capacitor 1013 includes a switch 1016 in parallel with capacitor 1014, similarly as described for FIG. 2 . FIG. 10B shows a phase-switched reactive circuit 1004 comprising an inductor 1024 in series with a capacitor 1022 , the series combination of inductor 1024 and capacitor 1022 being arranged in parallel with a phase-switched capacitor 1025 . Capacitor 1022 is not phase switched, and is therefore shown as C _DC . Phase switching capacitor 1025 includes a switch 1028 in parallel with capacitor 1026 , similarly as described for FIG. 2 . FIG. 10C shows a phase switched reactive circuit 1006 comprising a capacitor 1032 in parallel with a phase switched inductor 1033 . Phase switching inductor 1033 includes switch 1036 in parallel with inductor 1034 , similarly as described for FIG. 4 . FIG. 10D shows a phase-switched reactive circuit 1008 comprising an inductor 1042 in parallel with a capacitor 1044 , the parallel combination of inductor 1042 and capacitor 1044 being arranged in series with a phase-switched capacitor 1045 . Inductor 1042 is not phase switched, and is therefore shown as L _DC . Phase switching inductor 1045 includes switch 1048 in parallel with inductor 1046 , similarly as described for FIG. 4 .

Circuit variations other than those shown in FIGS. 10A-10D are also possible, as will be appreciated by those skilled in the art. For example, placing the capacitor in series with the phase switching capacitor provides a net effective impedance having a maximum capacitance equal to the series combination of the capacitor and the physical capacitance of the phase switching capacitor and a minimum capacitance equal to the series combination of the capacitor and the phase switching capacitance value.

As described with respect to FIGS. 6 and 7 , for phase-switching varactor 200 and phase-switching varactor 400 , between their variable reactance ranges and the amount of harmonic content injected into the rest of the system There is a tradeoff between. In other words, the range over which effective reactance may be controlled is limited by the amount of harmonic content that may be tolerated within the system (eg, by source 102 and/or load 114 ). Some embodiments may use additional or external filtering components to reduce the harmonic content injected into the source 102 and/or load 114 . In some embodiments, however, it may not be possible to use additional filtering components.

Referring to Figures 11 and 12, the phase-switched variable capacitor 200 and the phase-switched variable inductor 400 can be combined without the use of additional filtering components by using one or more digitally controlled capacitors or inductor matrices that are not phase switched to reduce the harmonic content. Such a hybrid switching network includes RF switches operating at the RF operating frequency and having a controlled phase and duty cycle with respect to the RF waveform. The hybrid switching network also includes digital switches associated with one or more capacitors or inductors in the switching matrix. Digital switches typically operate at frequencies much lower than the RF frequency, but can operate up to an RF frequency (eg, on a cycle-by-cycle basis) determined by the control bandwidth of the effective reactance C _EFF or _LEFF .

Referring to FIG. 11 , hybrid switched network 1100 includes phase switched reactance (eg, capacitor C ₀ 1116 and parallel switch 1118 ) and digitally controlled capacitor network 1102 . Although shown as a phase-switched variable capacitance (e.g., capacitor _C0 1116 and parallel switch 1118) coupled in parallel with digitally controlled capacitor network 1102 and load 114, in other embodiments, a phase-switched reactance may be implemented as a The network 1102 and load 114 are coupled in parallel with a phase-switched variable inductor (such as that shown in FIG. 4 ) or one of the phase-switched reactive circuits shown in FIGS. 10A-D or other equivalent circuits.

Digitally controlled capacitor network 1102 includes a plurality of capacitors and associated switches shown as capacitors 1104 , 1108 and 1112 and switches 1106 , 1110 and 1114 . In some embodiments, each of capacitors 1104, 1108, and 1112 has a unique capacitance value, allowing the capacitance value of digitally controlled capacitor network 1102 to be varied across a large capacitance range. For example, as shown in FIG. 11, capacitors 1104, 1108, and 1112 can be increased in increments of C ₀ from a phase-switched capacitor base value (eg, C ₀ ) until a maximum capacitance value (eg, (2·2 ^N −1)· C ₀ ), where N is the number of capacitors in the digitally controlled capacitor network 1102 .

Switches 1106, 1110, and 1114 are coupled in series with respective ones of capacitors 1104, 1108, and 1112 and are operable to adjust the capacitance of digitally controlled capacitor network 1102 by connecting (or disconnecting) the respective capacitors. Switches 1106 , 1110 , and 1114 are operable based on one or more control signals from control circuit 106 . As noted, switches 1106 , 1110 , and 1114 typically operate at frequencies less than the RF frequency to adjust the capacitance value of digitally controlled capacitor network 1102 .

Referring to FIG. 12 , hybrid switched network 1200 includes phase switched reactance (eg, capacitor L ₀ 1216 and series switch 1218 ) and digitally controlled inductor network 1202 . Although shown as a phase-switched variable inductance (e.g., capacitor _C0 1216 and series switch 1218) coupled in series with digitally controlled inductor network 1202 and in parallel with load 114, in other embodiments the phase-switched reactance may be implemented as A phase-switched variable capacitor (such as that shown in FIG. 2 ) is either one of the phase-switched reactance circuits shown in FIGS. 10A-D or other equivalent circuits.

Digitally controlled inductor network 1202 includes a plurality of inductors and associated switches shown as inductors 1206 , 1210 and 1214 and switches 1204 , 1208 and 1212 . In some embodiments, each of inductors 1206, 1210, and 1214 has a unique inductance value, allowing the inductance value of digitally controlled inductor network 1202 to be varied across a large inductance range. For example, as shown in FIG. 12, inductors 1206, 1210, and 1214 and 1218 may be increased in increments of _L0 from a phase-switched inductor base value (eg, _L0 ) until a maximum inductance value is reached.

Switches 1204, 1208, and 1212 are coupled in parallel with respective ones of inductors 1206, 1210, and 1214 and are operable to adjust digital The inductance of the inductor network 1202 is controlled. Switches 1204 , 1208 , and 1212 are operable based on one or more control signals from control circuit 106 . As noted, switches 1204 , 1208 , and 1212 typically operate at frequencies less than the RF frequency to adjust the inductance value of digitally controlled inductor network 1202 .

Digitally controlled capacitor network 1102 and digitally controlled inductor network 1202 augment the reactance of the phase switching (e.g. capacitor _C0 1116 and parallel switch 1118 or inductor _L0 1216 and series switch 1218) The reactance can be continuously varied without impact on source 102 and/or load 114 range when too much harmonic content is introduced. For example, the embodiment shown in FIGS. 11 and 12 uses a digitally controlled capacitor network 1102 (or a digitally controlled inductor network 1202 ) to control the base value C ₀ (or L ₀ ) of the switching network 1100 (or 1200 ). A switch, such as switch 1118 or switch 1218, which phase switches the reactance, may be operated to increase the base capacitance C ₀ (or inductance L ₀ ) by a factor determined by relationships (1) and (2) described above.

For example, the effective capacitance _CEFF at the switching frequency of the hybrid switched capacitor network 1100 can be controlled at a lower capacitance value C and an upper capacitance value by half-wave switching an RF switch with a switching angle α varying from ₀ to about π/2 between, as shown in Figure 3. As shown in FIG. 7, RF switching operation with a switching angle α smaller than π/2 (90 degrees) corresponds to a peak harmonic distortion of less than greater than 35%. Thus, hybrid switching networks such as 1100 and 1200 allow continuous control of effective reactance at switching frequencies over a wide range of capacitance (or inductance) with minimal harmonic distortion and the need for adjustable bias voltage or current.

In various embodiments, an RF switch (eg, switch 206 or switch 406 ) of TMN 112 may be implemented as one of a combination of various types of switching elements, eg, based on RF frequency or other operating parameters of RF system 100 . For example, lateral or vertical FETs, HEMTs, thyristors, diodes, or other similar circuit elements may be used.

Phase-switched variable capacitor 200 and phase-switched variable inductor 400 can be used as circuit elements within a more complex phase-switched tunable matching network (PS-TMN) such as a Pi network topology PS-TMN (Pi-TMN), although Other network topologies such as L-shaped networks, T-shaped networks or other similar networks are possible. FIG. 13 shows a schematic diagram of an illustrative RF system 1300 including an RF source 1301 coupled to a Pi-TMN 1302 coupled to an RF load 1303 . Pi-TMN 1302 includes two variable parallel capacitive susceptances B ₁ 1310 and B ₂ 1314 . In the illustrative embodiment, RF source 1301 is typically the output of a power amplifier or another RF system. As shown in FIG. 13 , RF source 1301 can be represented by its Norton equivalent circuit as including current source 1304 in parallel with source resistance R _S 1306 and source susceptance B _S 1308 . Similarly, RF load 1303 may be represented as including load resistance _RL 1318 in parallel with load susceptance BL 1316 _. The source and load impedances Z _S and Z _L can be expressed respectively as:

Z _S ＝(R _S ^-1 +jB _S ) ^-1 (3)

Z _L ＝(R _L ^-1 +jB _L ) ^-1 (4)

Therefore, it can be shown that the _susceptances B1 and B2 required to match the load impedance _ZL to the source impedance _ZS are given by _:

Therefore, the Pi-TMN 1302 can be used to match the load impedance Z _L to the source impedance Z _S by adjusting the values of the variable shunt capacitive susceptance B ₁ 1310 and B ₂ 1314

As shown in Figure 3, an embodiment of the Pi-TMN 1302 includes two variable parallel capacitive susceptances B ₁ 1310 and B ₂ 1314 and a fixed inductive reactance X 1312, although many other implementations of the Pi-TMN are possible, for example using Variable parallel inductive susceptance and fixed capacitive reactance, implementing all three reactance branches as variable components, etc. It should of course be realized that it is also possible to implement an L-segment TMN with one variable parallel path element and one variable series path element. Other types of networks may also be used. As described in more detail below, ground-referenced variable capacitors are well suited for implementation at RF frequencies using phase-switched varactor networks.

Referring to FIG. 14 , an illustrative range of load impedances that may be matched by Pi-TMN 1302 is shown as shaded area 1402 in Smith circle plot 1400 (normalized to R _S ). For example, the impedance value represented by the shaded area 1402 can be formed by having X= _RS and a susceptance B ₁ variable over the range of 1/ _RS to 4/ _RS and in the range of 1/ _RS to 2/ _RS . An illustrative Pi _- TMN implementation of variable susceptance B2. As shown in Figure 14, the Pi-TMN 1302 is capable of matching the impedance of the RF source 1301 to a load impedance that varies over approximately a 10:1 resistance range and a 5:1 resistance range (both capacitively and inductively). To accomplish this, the Pi-TMN 1302 modulates B1 on a ₁ :4 range and B2 on a 1: ₂ range, which can be achieved using a phase-switched varactor network such as shown in Figures 2 and 4 .

FIG. 15 shows an illustrative embodiment of a phase-switched Pi-TMN circuit 1502 that achieves the matching range shown in FIG. 14 for a source impedance of 50Ω (eg, R _S 1506 ). The inductive reactance X is chosen to be equal in value to the Norton equivalent source resistance _RS (eg 50Ω). As shown in FIG. 15 , variable capacitive susceptances B ₁ and B ₂ are implemented as half-wave phase-switched capacitors (phase-switched capacitor 200 of FIG. 2 ). Variable capacitive susceptance B ₁ includes phase switching capacitor C _P2 1514 and FET switch 1512 , which is implemented by switching control signal q2 , with switching angle α ₂ . The variable capacitive susceptance B ₂ includes a phase switching capacitor C _P1 1520 and a FET switch 1522 implemented by a switching control signal q1 with a switching angle α ₁ .

In the illustrative embodiment, the phase-switched Pi-TMN circuit 1502 operates at 27.12 MHz and can be adjusted by properly adjusting the switching angles of the switches (α ₁ and α ₂₁ ) and the phase offset between them (for example, by adjusting the switching Signals q1 and q2) are controlled to match the 50Ω source impedance to a load impedance that varies (both capacitively and inductively) over approximately a 10:1 resistance range and a 5:1 resistance range.

Implementing varactor susceptances B1 and B2 as _a half _- wave FET switched capacitor network provides zero-voltage switching (ZVS) operation of the switches and allows the use of a single ground-referenced switch per varactor (e.g. varactor capacitor FET 1512 of susceptance B ₁ and FET 1522 of variable capacitance susceptance B ₂ ). ZVS operation is desirable in switching systems because it reduces switching power losses and improves overall system efficiency. In addition, the outputs of FETs 1512 and 1522 (drain to source) are in parallel with phase switching capacitors C _P1 and C _P2 and thus can be added to the parallel capacitance and used as part of TMN.

In the illustrative Pi-TMN circuit 1502, the inductive reactance X 1312 shown in FIG. 13 is implemented to include an inductive reactance X 1312 arranged in series between variable susceptances B ₁ and B ₂ arranged as a parallel element (e.g., coupled to ground). A series resonant circuit of inductor L _S2 1516 and capacitor C _S2 1518 . Inductor L _S2 1516 and capacitor C _S2 1518 are selected to have an inductive impedance approximately equal to the source impedance (eg, 50Ω) at the desired frequency.

In the embodiment shown in Figure 15, two additional series resonant circuits are included, one as an input filter and one as an output filter of the Pi-TMN circuit 1502 to limit the amount of harmonic content injected into the source and load. Quantity, as a result of the toggle. For example, capacitor C _S1 1508 and inductor L _S1 1510 act as a series resonant input filter between source 1504 and Pi-TMN circuit 1502 . Similarly, capacitor C _S3 1524 and inductor L _S3 1526 act as a series resonant output filter between load 1528 and Pi-TMN circuit 1502 .

The quality factor Q of the series resonant circuit of LS2 1516 and CS2 1518 controls the interaction between phase switching capacitor C _P1 1520 and phase switching capacitor C _P2 1514 . For example, increasing the quality factor Q (e.g., by increasing the values of _LS2 1516 and _CS2 1518) decreases the interaction between phase-switching capacitor C _P1 1520 and phase-switching capacitor C _P2 1514, while increasing the quality factor Q also decreases The effective bandwidth of the network.

For example, in order for the phase-switched Pi- _TMN circuit 1502 to achieve the matching range shown in FIG. _P1 1520 may have a physical value C ₀ of 130 pF, and phase switching capacitor C _P2 1514 may have a physical value C ₀ of 100 pF. To achieve a desired quality factor by the series resonant circuit between phase switching capacitor C _P1 1520 and phase switching capacitor C _P2 1514, capacitor C _S2 1518 may have a value of 0.01 μF and inductor L _S2 1516 may have a value of 297 nH. To achieve the desired input and output filtering by the series resonant circuit, capacitors C _S1 1508 and C _S3 1526 may have a value of 23.4 pF, and inductors L _S1 1510 and L _S3 1524 may have a value of 1.47 μH. Additionally, FETs 1512 and 1522 may have an on-resistance of 10 mΩ, and the body diode of each FET may have a forward voltage of 0.4V and an on-resistance of 10 mΩ.

The switching of FETs 1512 and 1522 are synchronized to their drain currents based on a switching angle α, which is based on the desired effective capacitance of capacitors CP1 and CP2. As described above for the half-wave phase switched capacitors, FETs 1512 and 1522 are turned off when their drain currents cross from negative to positive, and then turned on again once their respective drain voltages drop to zero. Appropriate values of _a for each FET 1512 and 1522 can be calculated by determining the required B1 and B2 susceptances for the desired load impedance Z _L as given by relationships ( ₅ ) and (6). Once each capacitance value B1 and B2 is known, that value can be plugged in as in relation (1a) (for half _- wave phase _- switched capacitors) or relation (2a) (for full-wave phase-switched capacitors) In C _EFF (C ₀ is a known value as the physical capacitance of the capacitor) to determine the value of α corresponding to the expected susceptance value.

As stated, for phase-switched networks with non-pure sinusoidal current excitation, relations (1) and (2) may not lead to the exact value of α to achieve the desired susceptance. In addition, the non-linearity of the drain-to-source switched capacitance and the mutual interaction of the two switched networks (eg, capacitive susceptances B ₁ and B ₂ ) can also lead to inaccurate calculations of α. Accordingly, some embodiments use non-linear control techniques (e.g., via control circuit 106) to determine an appropriate value for α, such as fixed or adaptable look-up tables (e.g., LUT 108), feedback (e.g., via feedback circuit 110), feed-forward compensation (eg via feedforward circuit 104), digital predistortion of switching angles (eg via predistortion circuit 107) or other similar techniques.

In order to set the correct value of switching control parameter α for each FET 1502 and 1522 of Pi-TMN circuit 1502 to achieve a given impedance, LUT 108 may store predetermined switching angles (e.g., α ₁ and α ₂ ) corresponding to various complex impedances . For example, Table 2 shows an illustrative list of possible load impedances that Pi-TMN circuit 1502 can match to a 50Ω source and corresponding values of switching angles α1 and _α2 for switching control signals _q1 and q2:

Table 2 shows that it is possible for Pi-TMN circuit 1502 to match a 50Ω source impedance to a load impedance that varies resistively by a factor of at least 10:1. Based on the switching angles (α ₁ and α ₂ ) listed in Table 2 and the effective reactance (such as C _EFF /C ₀ or L _EFF /L ₀ ) versus α shown in Figure 6, the 2 of the effective capacitance can be shown. The :1 modulation enables impedance matching to load impedances that resistively vary over a 10:1 range.

Other types of systems may also use the phase switching network described herein. For example, a wide variety of systems can benefit from RF power amplifiers (PAs) that deliver power at specific frequencies or over specific frequency bands. Such a PA can beneficially control output power over a wide range and maintain high efficiency over its entire operating range. Conventional linear amplifiers (eg, Class A, B, AB, etc.) offer the benefits of wide-range dynamic output power control and high-fidelity amplification, but have limited peak efficiency that drops off rapidly as power recedes. On the other hand, switching PAs (e.g. inverters like D, E, F, Φ, etc.) offer peak peak efficiency, but only generate a constant envelope signal (at constant supply voltage) while remaining in switching mode .

One technique for output power control in switching PAs is through load modulation, where the load of the PA is modulated by an external network. In the described embodiment, the load of the PA is controlled by a phase-switched tunable matching network (TMN) (e.g., a network comprising one or more phase-switched variable capacitors 200 or phase-switched variable inductors 400, such as Pi-TMN circuit 1502 )modulation. For example, the impedance transformation of the phase-switched TMN can control the output power of the PA.

Referring to FIG. 16 , such a phase switched impedance modulation (PSIM) amplifier is shown as PSIM amplifier 1600 . PSIM amplifier 1600 includes an RF power amplifier (or inverter) 1602 that generates RF power at a particular frequency or over a particular range of frequencies. The RF PA 1602 is coupled to a power source (eg, voltage VDC and ground) and a phase switching TMN 1604 . Phase-switched TMN 1604 is coupled to RF load 1606, which has load impedance Z _L . The phase switching TMN 1604 is coupled to a controller 1608 that controls the operation of the TMN, eg, by providing control signals to the switches of the TMN based on the switching angle (eg, α) to achieve a desired impedance. Although not shown in Figure 16, in some embodiments, a controller 1608 is coupled to the RF PA 1602 and also controls the operation of the PA. Phase switch TMN 1604 is adaptively controlled to transform the load impedance Z _L to the impedance presented to PA 1602 . For example, phase switch TMN 1604 may control the output power of PA 1602 and/or compensate for frequency and/or load impedance by modulating the load presented to PA 1602 (eg, Z _TMN ) to provide high frequency and desired power to the load.

In various embodiments, the PA 1602 is (1) a switching inverter, (2) an amplitude modulated linear PA, or (3) a combination of these (eg, depending on the desired output). For example, FIG. 17 shows a block diagram of an illustrative PSIM amplifier 1700, but including switch PA 1702 (eg, E, F, or ΦPA class, etc.), which includes a single switch (eg, FET 1706). In other embodiments, other types of PAs may be used, such as linear PAs (eg, Class A, B, AB, or C) or other switched PAs that use more than one switch to convert DC power to RF power (eg, Class D, Inverter D, etc.).

As noted, modulating the effective load impedance Z _TMN seen by the PA looking into the phase switch TMN (eg, TMN 1604 or 1710 ) controls the output power over the operating power range of the PSIM amplifier (eg, amplifiers 1602 and 1702 ). In addition, the operating power range of the PSIM amplifier can also be further extended by using amplitude modulation of the PA drive signal for large output power backoffs.

Some embodiments may also use other power modulation techniques, such as discrete or continuous drain modulation of the power amplifier. The PA's drain modulation modulates (eg switches) the bias voltage applied to the PA's bias terminal. For example, a drain modulation technique can switch the bias voltage among multiple discrete voltage levels or continuously adjust the bias voltage over the entire voltage range.

In addition to performing impedance modulation and output power control of the RF PA, a phase switch TMN (eg, TMN 1604 or 1710) can also compensate for changes in load impedance _ZL . For example, the phase switch TMN can be continuously tuned to compensate for the amplifier's loading as the operating frequency varies by using the phase switch TMN to match the variable load impedance to the desired RF inverter load impedance Z _TMN for a given output power level Changes in network impedance and thus maintain ZVS operation. Thus, a PSIM amplifier, such as PSIM amplifiers 1600 and 1700, dynamically controls the output power it delivers to a widely varying load impedance, such as an RF plasma load, over a large frequency range.

Thus, PSIM amplifiers such as PSIM amplifiers 1600 and 1700 allow (1) efficient dynamic control of output power over a wide power range; (2) impedance matching and the ability to deliver power into a wide range of loads, and (3) Full zero voltage switching (ZVS) operation over the entire frequency range of frequency agile operation.

While the block diagrams of PSIM amplifiers 1600 and 1700 shown in FIGS. The example integrates PS-TMN into the design of RF PA. As a result, such an integrated PSIM amplifier can be viewed as an RF amplifier comprising two or more switches, where a first switch (or set of switches) is primarily responsible for generating RF power from DC input power, while a second switch (or a set of switches) group switch) is primarily responsible for modulating the effective impedance presented to the RF amplifier by the load network. In most embodiments, the second switch (or set of switches) does not convert DC power to RF power (e.g., the second switch provides zero power conversion from DC to RF), although in some embodiments, the second switch Some functional power can be converted from DC to RF or RF to DC.

In most embodiments, the PSIM amplifier may be a zero voltage switching (ZVS) amplifier with switching transistors operating in essentially switch mode and switching on and off at zero voltage switching, enabling high efficiencies to be achieved. In other implementations, a PSIM amplifier may provide switched mode operation (eg, protection operation) over some of its operating range (eg, while delivering high output power) and utilize linear mode operation over other parts of its range.

For example, FIG. 18A shows a schematic diagram of an illustrative topology of a PSIM amplifier 1800A. As shown, _PAIM amplifier _1800A is coupled to DC source 1802 coupled in series with inductor LF, which in turn is coupled to the parallel combination of transistor 1804 and capacitor _CF. Inductor L _F , inductor _CF and FET 1804 typically operate to generate RF output power from a DC source to the rest of the network. Branch reactance X1 is coupled between capacitor _CF and node _N2 coupled to Pi _{- TMN} comprising reactance X2 coupled to the _first phase switching reactance ₍ e.g. FET 1806, branch reactance _XS2 and phase switching variable reactance X _P2 ) and a second phase switching reactance (eg FET 1808 , branch reactance X _S3 and phase switching variable reactance X _P3 ). Branch reactance X1 is coupled between Pi _{- TMN} at node N1 and load impedance _ZL . The branch reactances X ₁ , X ₂ , X ₃ , X _S2 , X _S3 and the phase switching varactors X _P2 and X _P3 can be realized as various reactive networks, depending on the desired functionality of the design.

FIG. 18B shows an illustrative design 1800B of the PSIM amplifier topology shown in FIG. 18A. As shown in FIG. 18B , a phase-switched varactor (comprising FET switches 1806 and 1808 and phase-switched capacitors C _P2 and C _P3 ) is implemented using a half-wave phase-switched capacitor network such as that described for FIGS. 2 and 3 . As shown in FIG. 18B, three switches 1814, 1816, and 1818 are isolated from each other at DC (eg, by capacitors C _S1 , C _S2 , and C _S3 , respectively). FET switch 1814 is responsible for generating all RF power, while FET switches 1816 and 1818 are responsible for transforming and modulating the impedance presented by load ZL to the DC to RF portion of the circuit (eg, at the output port of switch 1814 at node _N2 ).

Figure 18C shows an illustrative design 1800C of the PSIM amplifier topology shown in Figure 18A. Network 1800C is similar to network 1800B, although in network 1800C, phase-switched capacitor networks (eg, FET 1826 and capacitor C _P2 and FET 1828 and capacitor C _P3 ) are connected in series with capacitors C _P4 and C _P5 , respectively. This increases the sensitivity of the PSIM amplifier to changes in the effective reactance of the switched capacitor network.

18D shows an illustrative design 1800D of the PSIM amplifier topology shown in FIG. 18A in which FET switches 1834 and 1836 are DC coupled (eg, via inductor L _S1 ), and thus potentially, one of FET switches 1834 and 1836 or two can be used to convert DC power to RF power and vice versa. FET switch 1838, on the other hand, is DC isolated (eg, by capacitors C _S2 and _{C S3} ), and thus is only used for impedance matching to load impedance ZL.

18E shows an illustrative design 1800E of the PSIM amplifier topology shown in FIG. 18A , where FET switches 1844, 1846, and 1848 are DC coupled (e.g., via inductor L _S2 ), and only the load is DC isolated (e.g., via capacitor C _S3 ). Thus in such an embodiment, all three FET switches 1844, 1846, and 1848 could potentially be used to switch between DC power and RF power and/or be responsible for impedance matching the network to the load, although not necessarily all three. Each function is provided.

As shown in FIG. 18E , the switched capacitor network of capacitor CF and FET switch 1844 is in parallel with the phase switching network of capacitor _{C P2} , inductor L ₂ and FET switch 1846 . As a result, some embodiments may combine these two networks into a single switched reactive network with an input current that matches the sum of the input currents of the two switched reactive networks associated with FETs 1844 and 1846 . Thus, in some embodiments, the three-switch PSIM shown in FIG. 18E may be implemented as a two-switch PSIM such as that shown in FIGS. 19 and 20 .

Referring to FIG. 19 , a schematic diagram of an illustrative topology of a two-switch PSIM 1900 is shown. Two-switch _PSIM 1900 is coupled to RF source 1902 coupled in series with inductor _LF , which in turn is coupled to the parallel combination of FET 1904 and capacitor _CF. Branch reactance X ₁ is coupled between capacitor _CF and a phase-switching reactance network _{including reactance X S2 coupled} between phase-switching reactance X _P2 and FET 1906 . _The branch reactance X2 is coupled between the phase switching reactance network and the load reactance _ZL . The branch reactances X ₁ , X ₂ and X _S2 and the phase-switched variable reactance X _P2 can be realized as various reactance networks, depending on the desired functionality of the design. Either or both of switch FETs 1904 and 1906 may be used to convert between DC power and RF power.

Referring to FIG. 20 , an illustrative implementation of a two-switch PSIM 19 00 is shown having a branch reactance X ₁ implemented as an inductor L _S1 and a capacitor C _S1 . Capacitor CS1 provides DC isolation between FET switches 2004 and 2006 . Thus, FET switch 2004 generates RF power, and FET switch 2006 modulates the impedance presented to the source.

FIG. 21 shows an illustrative implementation of a three-switch PSIM amplifier 2100 . PSIM amplifier 2100 operates over the 20.86 MHz to 27.12 MHz frequency range (a multiple of 1.3 in frequency). Furthermore, the PSIM amplifier 2100 provides the capability of 10:1 dynamic control of the output power delivered to a load having an impedance _ZL of 50Ω, with ±10% impedance variation (resistance and reactance).

PSIM amplifier 2100 includes RF PA (inverter) 2102, Pi-TMN 2104, branch filter 2106 and load impedance Z _L . RF PA 2102 includes FET switch 2108, inductor LF and an output network formed by capacitors _CF and _{C S1} and inductor L _S1 . In the embodiment shown in Figure 21, the RF PA 2102 is a modified Class E inverter with FET switches 2108 that convert between DC power and RF power. Pi-TMN 2104 includes a first phase switching capacitor (eg, C _P2 and FET 2110 ) and a second phase switching capacitor (eg, C _P1 and FET 2112 ). Branch filter 2106 includes inductor _{L S3} and capacitor C _S3 coupled between Pi-TMN 2104 and load ZL.

While the Pi-TMN 2104 maintains the inverter load impedance ZTMN as an approximately resistive load at the operating frequency of the RF PA 2101, the RF PA 2102 maintains zero voltage switching (ZVS) and high efficiency at different output power levels . The RF PA 2102 produces peak RF power when Z _TMN is 50Ω (eg, matched load impedance Z _L ). Dynamic control of the power back-off of the RF PA 2102 can be achieved by the Pi-TMN 2104 modulating Z _TMN .

For operation over the 20.86MHz to 27.12MHz frequency range, the illustrative embodiment of the PSIM amplifier 2100 shown in FIG. 21 uses an inductor LF having a value of _113nH , a capacitor CF having a value of _180pF , A capacitor C _S1 with a value of 3.81 μH, an inductor L _S1 with a value of 3.81 μH, a phase switching capacitor C _P2 with a physical value C ₀ of 152 pF, an inductor L _S2 with a value of 381 nH, a capacitor C _S2 with a value of 0.01 μF , a phase switching capacitor C _P1 with a physical value C ₀ of 152 pF, an inductor L _S3 with a value of 3.81 μH, and a capacitor C _S3 with a value of 15.2 pF. In some embodiments, Pi-TMN 2104 uses half-wave switched capacitor networks (eg, capacitor C _P2 and FET 2110 and capacitor C _P1 and FET 2112 ).

The series reactive network branch formed by capacitor C _S2 and inductor L _S2 has an inductive impedance of 50Ω at a frequency of 20.86 MHz and also DC isolates the two switching networks (e.g. capacitor C _p2 and FET 2110 and capacitor C _p1 and FET 2112) . The impedance of capacitor C _S2 and inductor L _S2 sets the resistance range over which ZTMN of Pi-TMN 2104 can be modulated. The series resonant network formed by capacitor C _S3 and inductor L _S3 provides additional filtering of the load current I _L and prevents DC current and high frequency harmonic content from coupling to the load Z _L . Pi-TMN 2104 can modulate the impedance Z _TMN presented to RF PA 2102 by appropriately driving FET switches 2110 and 2112 , eg, by adjusting the conduction angle of the FETs. By modulating the impedance _ZTMN presented to the RF PA 2102, the Pi- _TMN 2014 can control the output power delivered from the RF PA 2102 to the load ZL.

FIG. 22 shows an illustrative impedance range (eg, shaded area 2202 ) for which Z _TMN of Pi-TMN 2104 can be adjusted at 20.86 MHz. Figure 23 shows an illustrative impedance range (eg, shaded area 2203), Z _TMN of Pi-TMN 2104 can be adjusted at 27.12 MHz. Smith charts 2200 and 2300 are normalized to 50Ω. Shaded areas 2202 and 2302 show that Pi-TMN 2104 can be phase-switched by changing capacitor CP1 over a 1:6 impedance range (e.g., changing switching angle α1 of FET 2102 over approximately 0 degrees to ₁₂₅ degrees) and at 1:10 The phase switching capacitor CP1 is varied over the impedance range (eg, the switching angle α ₂ of the FET 2110 is varied over a range of approximately 0 degrees to 135 degrees) to match the load impedance ZL over a range of 10:1. Additionally, ZTMN can be modulated to account for ±10% variation (resistive and reactive) in load impedance ZL at the operating frequency of RFPA 202 .

To set the correct value of switching angle α1 _of FET 2112 and switching angle _α2 of FET 2110 of Pi-TMN 2104 to achieve a given impedance, LUT 108 may store predetermined switching angles (e.g., α ₁ and α ₂ ). For example, Table 3 shows an illustrative list of possible impedances Z _TMN and corresponding switching angles (eg α ₁ and α ₂ ) that can be matched to a 50Ω load impedance. The values of Table 3 can be determined based on a simulation of PSIM amplifier 2100, where FETs 2110 and 2112 are modeled with an on-state resistance of 10 mΩ and a body diode with a forward voltage of 0.4V. The output power listed in Table 3 includes the power delivered at base and higher frequencies when the PSIM amplifier is supplied with a 48VDC power supply.

As noted, PSIM amplifier 2100 maintains zero voltage switching of all FETs throughout a wide range of output power, load impedance, and operating frequency. For example, in order for the illustrative PSIM amplifier 2100 to deliver 58.6W of output power to a 50Ω load ZL at _20.86MHz with a supply voltage of 48VDC, the TMN 2102 needs to provide a near 1:1 impedance match (e.g., ZL= _{ZTMN =} 50Ω ). Under this operating condition, the required effective parallel capacitances at nodes _N1 and _N2 are equal to the C _P1 and C _P2 capacitances, respectively, and therefore FET switches 2110 and 2112 are off during the entire cycle, and FET switches 2110 and The drain voltage waveform of the 2112 will be sinusoidal.

As another example, in order for the illustrative PSIM amplifier 2100 to deliver 3.50W of output power to a 50Ω load Z _L at 27.12MHz with a supply voltage of 48VDC, the TMN 2102 needs to present an impedance Z _TMN of approximately 500Ω (as shown in Table 3 shown). Under this operating condition, the required effective parallel capacitance at nodes N1 and _N2 is higher _than the C _P1 and C _P2 capacitances, respectively, and therefore FET switches 2110 and 2112 are on during the entire cycle while maintaining ZVS. Despite the high frequency harmonic content of the drain voltage waveforms of FET switches 2110 and 2112, load current IL flowing through load AL should remain nearly sinusoidal. Thus, PSIM amplifier 2100 is capable of providing dynamic output power control while matching to variable loads over the entire range of switching frequencies.

Accordingly, as described herein, various embodiments provide a tunable matching network based on a phase-switched variable network reactive element referred to as a phase-switched tunable matching network (PS-TMN). Such PS-TMNs provide fast, high bandwidth, continuous impedance matching over a wide impedance range while operating efficiently at high power levels without requiring high bias voltages or currents. Such a PS-TMN may be used alone, or may also be used in combination with other matching techniques such as discrete switched reactance banks. The embodiments also provide zero voltage switching (ZVS) radio frequency (RF) amplifiers referred to herein as phase switched impedance modulated (PSIM) amplifiers. Such a PSIM amplifier can operate over a large frequency range using PS-TMN by efficiently modulating the output power over a wide frequency range and matching into highly variable loads (eg matching to a wide impedance range).

Reference herein to "one embodiment" or "an embodiment" means that a particular feature, structure, or characteristic described in connection with the embodiment can be included in at least one embodiment of the claimed subject matter. The appearances of the phrase "in one embodiment" in various places in the specification are not necessarily all referring to the same embodiment, nor are separate or alternative embodiments necessarily mutually exclusive of other embodiments. The same applies to the term "implementation".

As used in this specification, the words "exemplary" and "illustrative" are used herein to mean an illustration, instance, or illustration. Any aspect or design described herein as "exemplary" or "illustrative" is not necessarily to be construed as preferred or advantageous over other aspects or designs. Rather, use of the words "exemplary" and "illustrative" is intended to present concepts in the specific manner in which the words "exemplary" and "illustrative".

Furthermore, the term "or" is intended to mean an inclusive "or" rather than an exclusive "or". That is, unless otherwise specified or clear from context, "X employs A or B" is intended to mean any naturally contained permutation. That is, if X employs A, X employs B, or X employs both A and B, then "X employs A or B" is satisfied under any of the foregoing instances. Furthermore, the articles "a" and "an" as used in this application and the appended claims should generally be construed to mean "one or more" unless specified otherwise or clear from context to refer to a singular form.

To the extent directional terms (eg, upper, lower, parallel, perpendicular, etc.) are used in the specification and claims, these terms are intended merely to help describe the embodiments and are not intended to limit the claims in any way. Such terms need not be exact (eg, exactly perpendicular or exactly parallel, etc.), but instead, it is intended that normal tolerances and ranges apply. Similarly, unless expressly stated otherwise, each value and range is to be construed as approximate as if the word "about," "substantially," or "approximately" precedes the value or range of values.

Some embodiments may be implemented in the form of methods and apparatus for performing those methods. Furthermore, various functions of circuit elements may also be implemented as processing blocks in software programs, as will be apparent to those skilled in the art. The described embodiments can also be implemented in program code embodied in tangible media, such as magnetic recording media, hard drives, floppy disks, magnetic tape media, optical recording media, compact discs (CDs), digital versatile discs (DVDs), solid-state memory, hybrid magnetic and solid-state memory, or any other machine-readable storage medium in which when the program code is loaded into and executed by a machine such as a computer, the machine becomes a means for implementing the claimed invention. The described embodiments can also be implemented in the form of program code, for example whether stored in a storage medium, loaded into a machine and/or executed by a machine or on some transmission medium or carrier, such as on electrical wiring or wiring. , transmission through fiber optics or via electromagnetic radiation, wherein a machine is said to be a means for carrying out the claimed invention when the program code is loaded into a machine mechanism such as a computer and executed by the machine. When implemented on a processing device, the program code segments combine with the processor to provide a unique device that operates similarly to specific logic circuits. Such processing devices may include, for example, general purpose microprocessors, digital signal processors (DSPs), reduced instruction set computers (RISCs), complex instruction set computers (CISCs), application specific integrated circuits (ASICs), field programmable gate arrays (FPGAs), ), programmable logic arrays (PLAs), microcontrollers, embedded controllers, multicode processors, and/or other devices, including combinations of the foregoing. It can also be realized in the form of other sequences of bit streams or signals transmitted electrically or optically through a medium, stored in a magnetic field deformation in a magnetic recording medium, etc., produced using a method and/or apparatus as recited in the claims the examples.

Also for the purposes of this description, the terms "coupled", "coupled", "coupled", "connected", "connected" or "connected" refer to any means known or later developed in the art, Where energy is permitted to transfer between two or more elements, and the insertion of one or more additional elements is contemplated though not required. In contrast, the terms "directly coupled", "directly connected", etc. imply the absence of such additional elements. Signals and corresponding nodes or ports may be referred to by the same names and are interchangeable for purposes herein.

It should be understood that the steps of the methods set forth herein do not necessarily need to be performed in the order recited, and that the order of the steps of such methods should be understood as illustrative only. Likewise, additional steps may be included in such methods, and certain steps may be omitted or combined in methods consistent with various embodiments.

It will be further understood that various changes in the details, materials and arrangements of parts described and shown herein may be made by those skilled in the art without departing from the scope of the following claims.

Claims (68)

1. a kind of tunable impedance network of Phase-switching, the input with the source that is configured to coupled to and with being configured as coupling The output end of load is closed, the tunable impedance network includes：

One or more Phase-switching reactance components；

Controller, its each offer being configured as into one or more of Phase-switching reactance components controls letter accordingly Number, to cause the corresponding control signal in response to providing it, each Phase-switching reactance component provides corresponding choosing Determine reactance value.

2. tunable impedance network according to claim 1, wherein, each Phase-switching reactance is set to the corresponding phase Reactance value is hoped to realize the impedance matching between the source and the load.

3. tunable impedance network according to claim 1, wherein, each Phase-switching reactance is set to the corresponding phase Reactance value is hoped to realize the desired impedance ratio between the source and the load.

4. tunable impedance network according to claim 1, wherein, each Phase-switching reactance is set to the corresponding phase Reactance value is hoped to realize the impedance of expectation first to the source and the impedance of expectation second to the load.

5. tunable impedance network according to claim 1, wherein, it is every in one or more of Phase-switching reactance It is individual including：

One or more reactance components and at least one switch,

Wherein, at least one in one or more of reactance components is configured as by least one switch associated there The reactance network neutralization is switched to switch out from the reactance network.

6. tunable impedance network according to claim 5, wherein, at least one described associated switch is based on described Corresponding control signal, in the relevant switching frequency of the frequency of the RF signals with being provided by the source and can switch to enter under phase Row operation.

7. tunable impedance network according to claim 6, wherein, at least one described switch can match somebody with somebody in half-wave switching Middle operation is put to switch on and off once with each cycle of the RF signals at the output port of the RF amplifiers.

8. tunable impedance network according to claim 6, wherein, at least one described switch can match somebody with somebody in all-wave switching Middle operation is put to switch on and off twice with each cycle of the RF signals at the output port of the RF amplifiers.

9. tunable impedance network according to claim 6, wherein, select the switching frequency and the switching phase with There is provided with the Phase-switching reactance for expecting reactance value.

10. tunable impedance network according to claim 6, wherein, at least one described switch can be operated for carrying For at least one in the zero voltage switching (ZVS) and zero current switching (ZCS) of the switch.

11. tunable impedance network according to claim 6, wherein, the controller is configured as in being based on down lising At least one determine the switching frequency and select the switching phase：Feedback circuit, feed forward circuit and adaptive pre- mistake True system.

12. tunable impedance network according to claim 11, wherein, the self-adapted pre-distortion system includes searching Table.

13. tunable impedance network according to claim 6, wherein, the Phase-switching reactance is capacity cell, and The capacitance of the Phase-switching capacity cell under expected frequency and the physics DC electricity of the Phase-switching capacity cell Capacitance is relevant with the switching phase.

14. tunable impedance network according to claim 6, wherein, the Phase-switching reactance is inductance element, and The inductance value of the Phase-switching inductance element under expected frequency and the physics DC electricity of the Phase-switching inductance element Inductance value is relevant with the switching phase.

15. tunable impedance network according to claim 1, further comprises：

Digital reactance matrix, it includes N number of reactance component that can be selected to adjust the effective reactance of the digital reactance matrix Value, wherein N is positive integer.

16. tunable impedance network according to claim 1, further comprises：

One or more simulation variable reactive elements.

17. tunable impedance network according to claim 1, wherein, the source includes radio frequency (RF) source, RF power amplifications At least one of device (PA) and switch mode inverter.

18. tunable impedance network according to claim 1, wherein, the load includes antenna, transmission line and plasma At least one of body load.

19. tunable impedance network according to claim 1, wherein, the input of the tunable impedance network Radio frequency (RF) amplifier system is coupled to, the tunable impedance network is configured as modulating the load of the RF amplifier systems Impedance is to control the power level of the RF amplifier systems.

20. tunable impedance network according to claim 1, further comprises：

One or more filter parts, it is configured to reduce at least one be coupled in the input and the output end Individual harmonic content.

21. a kind of method for operating tunable impedance network, the tunable impedance network includes being configured to coupled to source Input, the output end for being configured to coupled to load and one or more Phase-switching reactance, methods described include：

The desired impedance value of the tunable impedance network is determined by the controller for being coupled to the tunable impedance network；

From each offer corresponding control signal of the controller into one or more of Phase-switching reactance；

In response to the corresponding control signal provided it, the corresponding expectation reactance of each Phase-switching reactance is set Value.

22. method according to claim 21, wherein, each Phase-switching reactance is set to corresponding expectation reactance value Realize the impedance matching between the source and the load.

23. method according to claim 21, wherein, each Phase-switching reactance is set to corresponding expectation reactance value Realize the desired impedance ratio between the source and the load.

24. method according to claim 21, wherein, each Phase-switching reactance is set to corresponding expectation reactance value Realize the impedance of expectation first to the source and the impedance of expectation second to the load.

25. method according to claim 21, wherein, it is each including one in one or more of Phase-switching reactance Individual or multiple reactance components and at least one switch, methods described further comprise：

At least one in one or more of reactance components is switched to by least one switch associated there described Reactance network is neutralized and switched out from the reactance network.

26. method according to claim 25, further comprises：

Based on corresponding control signal, in the relevant switching frequency of the frequency of the RF signals with being provided by the source and switching At least one described associated switch is operated under phase.

27. method according to claim 26, further comprises：

The selection switching phase has the Phase-switching reactance for expecting reactance value to provide.

28. method according to claim 25, further comprises：

Based on corresponding control signal, in the relevant switching frequency of the frequency of the RF signals with being provided by the source and switching At least one described switch is operated under phase.

29. method according to claim 28, be included in half-wave handover configurations operate at least one described switch with The each cycle of the RF signals at the output port of the RF amplifiers is switched on and off once.

30. method according to claim 28, be included in all-wave handover configurations operate at least one described switch with The each cycle of the RF signals at the output port of the RF amplifiers is switched on and off twice.

31. method according to claim 28, including at least one described switch of operation are electric with provide the switch zero Crush-cutting changes at least one in (ZVS) and zero current switching (ZCS).

32. method according to claim 25, wherein, the Phase-switching reactance includes capacity cell, and the phase The physics DC capacitances of the capacitance of the position switch-capacitor element under expected frequency and the Phase-switching capacity cell and The switching phase is relevant.

33. method according to claim 25, wherein, the Phase-switching reactance includes inductance element, and the phase The physics DC inductance values of the position inductance value of the switching inductance element under expected frequency and the Phase-switching inductance element and The switching phase is relevant.

34. method according to claim 21, wherein, the tunable impedance network includes to select with N number of The digital reactance matrix of reactance component is to adjust the effective reactance value of the digital reactance matrix, and wherein N is positive integer.

35. method according to claim 21, wherein, it is variable that the tunable impedance network includes one or more simulations Reactance component.

36. method according to claim 21, wherein, the source include radio frequency (RF) source, RF power amplifiers (PA) and At least one of switch mode inverter, and wherein, the load is included in antenna, transmission line and plasma load It is at least one.

37. method according to claim 21, wherein, the input of the tunable impedance network is coupled to radio frequency (RF) amplifier system, methods described includes：

Modulate the load impedance of the RF amplifier systems to control the RF amplifier systems by the tunable impedance network Power level.

38. method according to claim 21, further comprises：

Reduced by the one or more filter parts for being coupled to the tunable impedance network and be coupled to the input and institute State the harmonic content of at least one in output end.

39. a kind of radio frequency (RF) amplifier system with input port and output port, the RF amplifier systems include：

RF amplifiers, it has the input port for the input port for being coupled to the RF amplifier systems and with output end Mouthful；And

Phase-switching is tunable impedance network, its output port for being coupling in the RF amplifiers and the RF amplifiers system Between the output port of system, the tunable impedance network of Phase-switching be configured as change its impedance the RF is put with modulating The impedance that the output port of big device is presented.

40. the RF amplifier systems according to claim 39, wherein, the tunable impedance network of Phase-switching includes one Being each configured as in individual or multiple Phase-switching reactance components, one or more of Phase-switching reactance components receives phase The control signal answered, and the corresponding control signal in response to providing it, each Phase-switching reactance component are carried It is provided with corresponding expectation reactance value.

41. RF amplifier systems according to claim 40, further comprise：

Controller, its each offer being configured as into one or more of Phase-switching reactance components controls letter accordingly Number.

42. RF amplifier systems according to claim 40, wherein, in one or more of Phase-switching reactance components Each include：

One or more reactance components；

At least one switch；

Wherein, at least one in one or more of reactance components is configured as by least one switch associated there The tunable impedance network neutralization of the Phase-switching is switched to switch out from the tunable impedance network of the Phase-switching.

43. RF amplifier systems according to claim 42, wherein, at least one described associated switch is based on described Corresponding control signal, can be in the switching frequency relevant with the frequency of the RF signals at the output port of the RF amplifiers Operated under rate and switching phase.

44. RF amplifier systems according to claim 42, wherein, at least one described switch can match somebody with somebody in half-wave switching Middle operation is put to switch on and off once with each cycle of the RF signals at the output port of the RF amplifiers.

45. RF amplifier systems according to claim 42, wherein, at least one described switch can match somebody with somebody in all-wave switching Middle operation is put to switch on and off twice with each cycle of the RF signals at the output port of the RF amplifiers.

46. RF amplifier systems according to claim 42, wherein, in one or more of Phase-switching reactance components Each include the capacitor with switch in parallel.

47. RF amplifier systems according to claim 46, wherein, in one or more of Phase-switching reactance components Each further comprise inductor, the inductor and the capacitor and the combined serial of the switch in parallel.

48. RF amplifier systems according to claim 42, wherein, at least one described switch can be operated for providing It is described at least one switch zero voltage switching and zero current switching at least one.

49. RF amplifier systems according to claim 43, wherein, select the switching frequency and the switching phase with There is provided with the Phase-switching reactance component for expecting reactance value.

50. RF amplifier systems according to claim 49, wherein, in one or more of Phase-switching reactance components At least one be under one of list：

Capacity cell with capacitance, and wherein, the electric capacity of the Phase-switching capacity cell under expected frequency Value is relevant with the physics DC capacitances and the switching phase of the Phase-switching capacity cell；And

Inductance element with inductance value, and wherein, the inductance of the Phase-switching inductance element under expected frequency Value is relevant with the physics DC inductance values and the switching phase of the Phase-switching inductance element.

51. the RF amplifier systems according to claim 39, wherein, by the Phase-switching tunable match network to institute The impedance for stating the output port presentation of RF amplifiers is dynamically adapted to make to be coupled to the described of the RF amplifier systems The impedance matching of the variable load impedance of output port and the RF amplifiers.

52. RF amplifier systems according to claim 51, further comprise the institute for being coupled to the RF amplifier systems The RF loads of output port are stated, wherein, the RF loads are at least one of antenna, transmission line and plasma load.

53. RF amplifier systems according to claim 42, wherein, the RF amplifiers include switching inverter, described Switching inverter includes at least one switching device for being configured as producing RF power.

54. RF amplifier systems according to claim 50, wherein, it is tunable that the controller modulates the Phase-switching The impedance that impedance network is presented to the output port of the RF amplifiers, to cause the RF amplifiers to maintain the switching The zero voltage switching (ZVS) of at least one switching device of inverter.

55. RF amplifier systems according to claim 42, wherein, switch one or more of Phase-switching reactance elements The impedance that at least one each described switch modulation in part is provided by the tunable impedance network of the Phase-switching is converted.

56. the RF amplifier systems according to claim 39, further comprise：

Wave filter, it is coupled to the tunable impedance network of the Phase-switching, and the wave filter has filter characteristic, the filter Ripple device characteristic reduces in the output port of the RF amplifiers and the output port of the RF amplifier systems At least one harmonic content produced by the tunable impedance network of the Phase-switching presented.

57. RF amplifier systems according to claim 56, wherein, the wave filter includes one or more wave filter portions Part, one or more of filter parts are configured as the tunable impedance network of the Phase-switching and the RF amplifiers DC signals between at least one in the output port of the RF amplifier systems are electrically isolated.

58. the RF amplifier systems according to claim 39, wherein, the tunable impedance network of Phase-switching includes：

It is coupled to the first parallel pathways Phase-switching variable reactive element of the first node of tandem paths reactance component.

59. RF amplifier systems according to claim 58, wherein, the tunable impedance network of Phase-switching is further Including：

It is coupled to the second parallel pathways Phase-switching variable reactive element of the Section Point of the tandem paths reactance component.

60. the method for one kind operation radio frequency (RF) amplifier system, methods described includes：

RF signals are provided to the input port of RF amplifiers；

Amplify the RF signals to provide the RF signals of amplification in the output port of the amplifier by the RF amplifiers；With And

Change the phase coupled between the output port and the output port of the RF amplifier systems of the RF amplifiers Position switches the impedance of tunable impedance network to modulate the impedance that the RF amplifiers are presented.

61. method according to claim 60, wherein, change the impedance of the tunable impedance network of Phase-switching Including：

Control signal is received by the tunable impedance network of the Phase-switching；And

In response to the control signal, make at least one reactance component with the output port of the RF amplifiers at RF It is electrically connected under the relevant frequency of the frequency of signal and phase in the tunable impedance network of the Phase-switching or from the phase Switch tunable impedance network to disconnect providing the tunable impedance network of the Phase-switching with expectation reactance value.

62. method according to claim 61, wherein, the tunable impedance network of Phase-switching includes one or more Reactance component and at least one switch, and wherein, at least one reactance component is electrically connected to the Phase-switching tunable Disconnecting in impedance network or from the tunable impedance network of the Phase-switching includes：

At least one for making in one or more of reactance components by least one switch associated there is switched to institute The tunable impedance network neutralization of Phase-switching is stated to switch out from the tunable impedance network of the Phase-switching；And

Based on corresponding control signal in switching frequency and at least one described associated switch of operation under switching phase.

63. method according to claim 62, further comprises：

Selected based on the frequency of the RF signals at the output port of the RF amplifiers switching frequency and The switching phase has the Phase-switching reactance component for expecting reactance value to provide.

64. method according to claim 60, further comprises：

The impedance for making the Phase-switching tunable match network that the RF amplifiers are presented is dynamically adapted to make to be coupled to The impedance matching of the variable load impedance of the output port of the RF amplifier systems and the RF amplifiers.

65. method according to claim 60, wherein, modulation is by the tunable impedance network of the Phase-switching to the RF The load impedance control that amplifier is presented is provided to the letter of the amplification of the output port of the RF amplifier systems Number power level.

66. method according to claim 62, wherein, the amplifier includes switching inverter, the switching inverter At least one switching device including being configured as producing RF power, methods described further comprises：

Described cut is maintained by modulating the impedance of the tunable impedance network of the Phase-switching presented to the RF amplifiers Change the zero voltage switching (ZVS) of at least one switching device of inverter.

67. method according to claim 62, further comprises：

Switch at least one described switch of the tunable impedance network of the Phase-switching to provide at least one switch At least one in zero voltage switching and zero current switching.

68. method according to claim 25, further comprises：

At least one in being listd based under, determines the switching frequency by the controller and selects the switching phase：Instead Current feed circuit, feed forward circuit and self-adapted pre-distortion system.