CN106301052A - A kind of sliding moding structure Delta modulator approach of improvement - Google Patents

Abstract

The present invention relates to an improved sliding mode variable structure Delta modulation method, which is characterized in that it comprises the following steps: step 1, establishing a discrete-time dynamic model of a series resonant inverter controlled by a traditional current-adjustable Delta controller, and step 2 1. Introduce an integrator to improve the output performance of the current adjustable Delta controller; step 3, analyze the switching state of the series resonant inverter based on the sliding mode variable structure control theory, and select an appropriate integral gain value Ki of the integrator; step 4, introduce the integral In the reset link, an appropriate error bandwidth is selected to further expand the upper limit of the integral gain value Ki of the integrator. The present invention further reduces the steady-state bias value of the output current by adding an integrator with a reset function to the traditional current-adjustable Delta controller, while keeping the ripple current content at a low level.

Description

Improved sliding mode variable structure Delta modulation method

Technical Field

The invention belongs to the technical field of power electronic control, relates to high-power switching power supplies such as an induction heating power supply and an Uninterruptible Power Supply (UPS), and particularly relates to an improved sliding mode variable structure Delta modulation method.

Background

With the improvement of the switching speed and the rated capacity of the power semiconductor device, the voltage source type series resonance inverter has attracted more and more attention in the fields of induction heating melting, uninterruptible power supplies and the like. The voltage source series resonance inverter is generally composed of a direct current voltage source, a high frequency inverter, a load matching transformer and an induction coil, and the corresponding method for controlling the output current or the output power of the load is realized by using a thyristor controlled rectifier and an inverter with constant switching frequency. Although this approach is easy to implement in theory, the switching devices in the rectifier typically operate in a hard switching state, further increasing power losses and the size and weight of the device.

In order to solve the above problems, a high-performance current control technique is currently used. The inverter is supplied with direct current by a diode uncontrollable rectifying circuit, and a power switching device in the inverter circuit works in a resonance state to obtain higher output power density. Alternatively, a current-regulated Delta Controller (CRDM) is applied to the generation of the series resonant inverter control pulse train, and this method uses a high gain regulator and thus has a strong robustness and response rate. But CRDM control output current is often accompanied by non-zero steady state errors and offsets, resulting in degraded output power performance.

Disclosure of Invention

The invention aims to overcome the defects of the prior art and provide an improved sliding mode variable structure Delta modulation method which is reasonable in design, small in steady-state offset value of output current and low in ripple current content.

The technical problem to be solved by the invention is realized by adopting the following technical scheme:

an improved sliding mode variable structure Delta modulation method comprises the following steps:

step 1, establishing a discrete time dynamic model of a series resonance inverter controlled based on a traditional current adjustable Delta controller, and obtaining a relational expression between output load current and current ripple amplitude of the series resonance inverter in a sampling period through equivalent analysis of the series resonance inverter controlled based on the traditional current adjustable Delta controller;

step 2, drawing a corresponding relation graph of load output current and current ripple amplitude under different equivalent control inputs according to the relational expression of the output load current variation and the current ripple amplitude of the series resonance inverter in the step 1; in order to reduce the current bias value of the output side of the current adjustable Delta controller, an integrator is introduced to improve the output performance of the current adjustable Delta controller;

step 3, analyzing the switching state of the series resonance inverter based on a sliding mode variable structure control theory, and selecting a proper integrator integral gain value Ki;

and 4, in order to further expand the upper limit value of the integral gain value Ki of the integrator, introducing an integral resetting link on a feedforward channel of the current adjustable Delta controller introduced into the integrator in the step 2, selecting a proper error bandwidth eta, forcibly resetting the integrator when the current error value of the output side is greater than the preset error bandwidth eta, and further expanding the upper limit value of the integral gain value Ki of the integrator by the control method.

Furthermore, the relational expression between the output load current and the current ripple amplitude of the series resonant inverter in step 1 is as follows:

$\begin{array}{c}\mathrm{\Δ}I(k+1)=I_o(k+1)-I_o\left(k\right)\\ =\frac{\mathrm{\π}}{2Q}\{I_{\max }{u}^{*}(k+1)-I_o\left(k\right)\}\end{array}$

wherein u x (k +1) is an equivalent control input, and the corresponding expression is:

$u^{*}(k+1)=\frac{M\left(k\right)+M(k+1)}{2}$

where m (k) is used to describe the operation mode of the inverter, it can be expressed as follows:

in the above formula, Q is the load quality factor; delta I is the current ripple amplitude; i is_maxIs the maximum value of the output current; i is_oTo output a current.

Moreover, the specific method for improving the output performance of the current-adjustable Delta controller by introducing the integrator in the step 2 comprises the following steps: an integrator is added to a feedforward channel of a traditional current adjustable Delta controller, in each half resonance period, the output current value of the series resonance inverter is collected and compared with a reference current value to generate an error signal, and the switching control signal of the inverter at the next moment is generated through adjustment of the integrator to further determine the operation mode of the inverter at the next moment.

The specific method of step 3 is: in order to select a proper integrator gain Ki, a sliding mode variable structure control theory is introduced, a switching state expression based on sliding mode variable structure control is established, a corresponding expression of the integral gain Ki and an equivalent control input u x (k +1) is obtained by establishing a relation between the switching state and the equivalent control input under the control of the sliding mode variable structure, and then different equivalent control inputs are selected to determine that the upper limit value of the integral gain Ki is as follows:

$0<K_i<\frac{\mathrm{\&pi;}}{2QT}$

in the above formula, T is a half resonance period;

moreover, the specific method for selecting the appropriate error bandwidth η in the step 4 is as follows: and (2) combining the relational expression between the output load current and the current ripple amplitude of the series resonance inverter in the step (1), drawing to obtain corresponding curves of equivalent control input and current error values under different reference current values, and further analyzing that the expression of the error bandwidth eta is as follows:

${\mathrm{\&eta;}}_{min}=max\left\{\right|{\mathrm{\&Delta;I}}_{(u^{*}=0.5)}|+|{\mathrm{\&Delta;I}}_{(u^{*}=1)}|,|{\mathrm{\&Delta;I}}_{(u^{*}=0.5)}|+|{\mathrm{\&Delta;I}}_{(u^{*}=0)}\left|\right\}=\frac{1.5\mathrm{\&pi;}}{2Q}{I}_{max}$

the invention has the advantages and positive effects that:

1. the invention provides an improved sliding mode variable structure Delta modulation method, which is characterized in that an integrator with a reset function is added into a traditional current adjustable Delta Controller (CRDM), so that the steady-state offset value of output current is further reduced, the ripple current content is kept at a lower level, and the transient response overshoot is almost negligible. The adjusting performance of the current controller is verified through experiments, and when the current controller is applied to systems such as an induction heating power supply and an uninterruptible power supply, the output power density is high, and the control adjusting range is wide.

2. The invention provides an improved sliding mode variable structure Delta modulation method based on analysis of a dynamic current model of a series resonance inverter in a zero current switching state. The gain and the error bandwidth of the integrator are set by adopting a sliding mode variable structure theory based on a discrete time domain, so that the stability of the output current of the resonant inverter is further improved, and the output power of the series resonant inverter is improved.

Drawings

FIG. 1 is a flow chart of the method of the present invention;

FIG. 2 is a schematic diagram of a series resonant circuit of the present invention;

FIG. 3 is a schematic diagram of the corresponding relationship between the load output current and the current ripple amplitude under different equivalent control inputs;

FIG. 4 is a schematic diagram of a conventional current adjustable Delta controller of the present invention;

FIG. 5 is a schematic diagram of the current adjustable Delta controller incorporating the integral reset of the present invention;

FIG. 6 is a graph of the present invention as I_ref<0.5I_maxSchematic diagram of corresponding current error and control input value;

FIG. 7 shows a schematic diagram of the present invention_ref>0.5I_maxAnd (4) a schematic diagram of the corresponding current error and the control input value.

Detailed Description

The embodiments of the invention will be described in further detail below with reference to the accompanying drawings:

an improved sliding mode variable structure Delta modulation method is shown in figure 1, and comprises the following steps:

step 1, establishing a discrete time dynamic model of a series resonance inverter controlled by a traditional current adjustable Delta controller, and obtaining a relational expression between output load current and current ripple amplitude of the series resonance inverter in a sampling period through equivalent analysis of the series resonance inverter controlled by the traditional current adjustable Delta controller, wherein when the traditional current adjustable Delta controller is adopted for control, the relational expression comprises the following steps:

$\begin{array}{c}\mathrm{\&Delta;}I(k+1)=I_o(k+1)-I_o\left(k\right)\\ =\frac{\mathrm{\&pi;}}{2Q}\{I_{\max }{u}^{*}(k+1)-I_o\left(k\right)\}\end{array}$

wherein u x (k +1) is an equivalent control input, and the corresponding expression is:

$u^{*}(k+1)=\frac{M\left(k\right)+M(k+1)}{2}$

where m (k) is used to describe the operation mode of the inverter, it can be expressed as follows:

in the above formula, Q is the load quality factor; delta I is the current ripple amplitude; i is_maxIs the maximum value of the output current; i is_oIs an output current;

step 2, drawing a corresponding relation schematic diagram of load output current and current ripple amplitude under different equivalent control inputs as shown in fig. 3 according to the relational expression of the output load current variation and the current ripple amplitude of the series resonance inverter in the step 1; as can be seen from fig. 3, when the equivalent control input is a negative value, the load current value gradually decreases and the current ripple amplitude increases corresponding to the inverter operating in the feedback state. In this case, the control input value is only two states of 0 and 1, and the switching speed of the output state or the current change rate cannot be properly adjusted, so that the current ripple content on the output side is high and the current bias is large. Therefore, in order to reduceThe current bias value of the output side of the current adjustable Delta controller is obtained by adding an integrator to a feedforward channel of the traditional current adjustable Delta controller shown in figure 4, and acquiring the output current value I of the series resonance inverter in each half resonance period_oAnd a reference current value I_refComparing to generate an error signal I_eAnd then, a switch control signal of the inverter at the next moment is generated through the adjustment of the integrator, so that the operation mode of the inverter at the next moment is determined, and the output performance of the current adjustable Delta controller is improved.

Step 3, analyzing the switching state of the series resonance inverter based on a sliding mode variable structure control theory, and selecting a proper integrator integral gain value Ki;

the specific method of the step 3 comprises the following steps: in order to select a proper integrator gain Ki, a sliding mode variable structure control theory is introduced, a switching state expression based on sliding mode variable structure control is established, a corresponding expression of the integral gain Ki and an equivalent control input u x (k +1) is obtained by establishing a relation between the switching state and the equivalent control input under the control of the sliding mode variable structure, and then different equivalent control inputs are selected to determine that the upper limit value of the integral gain Ki is as follows:

$0<K_i<\frac{\mathrm{\&pi;}}{2QT}$

in the above formula, T is a half resonance period;

step 4, in order to further expand the upper limit value of the integral gain value Ki of the integrator, an integral resetting link is introduced into a feedforward channel of the current adjustable Delta controller introduced into the integrator in the step 2, then the minimum value of an error bandwidth eta is determined, a smaller eta value needs to be selected in order to reduce overshoot, and the integrator is not reset when the maximum current error value is met under the steady state condition; and selecting a proper error bandwidth eta, and forcibly resetting the integrator when the current error value of the output side is greater than the preset error bandwidth eta, thereby further expanding the upper limit value of the integral gain value Ki of the integrator by the control method.

The invention relates to a current-adjustable Delta controller schematic diagram added with integral reset, which is shown in figure 5;

the specific method for selecting the appropriate error bandwidth η in the step 4 comprises the following steps: and (2) combining the relational expression between the output load current and the current ripple amplitude of the series resonance inverter in the step (1), drawing to obtain corresponding curves of equivalent control input and current error values under different reference current values, and further analyzing that the expression of the error bandwidth eta is as follows:

${\mathrm{\&eta;}}_{min}=max\left\{\right|{\mathrm{\&Delta;I}}_{(u^{*}=0.5)}|+|{\mathrm{\&Delta;I}}_{(u^{*}=1)}|,|{\mathrm{\&Delta;I}}_{(u^{*}=0.5)}|+|{\mathrm{\&Delta;I}}_{(u^{*}=0)}\left|\right\}=\frac{1.5\mathrm{\&pi;}}{2Q}{I}_{max}$

the following describes in detail the determination process of the upper limit value of the integral gain Ki of the integrator in this embodiment:

the initial value of the secondary side output voltage of the known matching transformer is V_sIf the corresponding time t is equal to 0, the current i is output_oAnd tank capacitor voltage v_cThe simplified expression of (c) can be expressed as:

$i_o\left(t\right)=-e^{\mathrm{\&alpha;}t}\frac{v_c\left(0\right)-V_s}{{\mathrm{\&omega;}}_d{L}_{eq}}sin\left({\mathrm{\&omega;}}_{d}t\right)---\left(1\right)$

$v_c\left(t\right)=V_s+e^{-\mathrm{\&alpha;}t}\left(v_c\right(0)-V_s)\frac{{\mathrm{\&omega;}}_r}{{\mathrm{\&omega;}}_d}cos({\mathrm{\&omega;}}_{d}t-\mathrm{\&phi;})---\left(2\right)$

wherein,

$\mathrm{\&alpha;}=\frac{R_{eq}}{2L_{eq}},{\mathrm{\&omega;}}_d={\mathrm{\&omega;}}_r\sqrt{1-{\left(\frac{1}{2Q}\right)}^2},\mathrm{\&phi;}={\sin }^{-1}\left(\frac{1}{2Q}\right),Q=\frac{{\mathrm{\&omega;}}_r{L}_{eq}}{R_{eq}}$

in the above formula, L_eqIs the load equivalent inductance value; r_eqIs the load equivalent resistance value;

since each power switching device always changes its switching state at the moment of the current zero crossing, the switching frequency of the switching device is always equal to the load resonant frequency. The value V of the voltage at the input terminal of the series resonant circuit shown in FIG. 2_dcThe inverter switching states are determined as follows:

$v_s=\left\{\begin{array}{ccc}{V}_{dc}& if& (Q_1,Q_2)areON\\ 0& if& (Q_1,Q_2)or(Q_3,Q_4)areON\\ -V_{dc}& if& (Q_3,Q_4)areON\end{array}\right\}---\left(3\right)$

in the above formula, Q₁、Q₂、Q₃、Q₄Respectively corresponding to the states of power switching devices in the series resonance inverter;

defining a discrete variable m (k) describing the operating mode of the inverter can be expressed as follows:

in conjunction with equation (4), equation (3) can be written as:

v_s(t)＝V_dcM(k)sign(i_o(t))for kT＜t＜(k+1)T (5)

in the above formula, T ═ pi/ω_dHalf a resonance period;

suppose the absolute value of the peak value of the load output current is I_oIn a half resonance period, the instantaneous capacitor voltage of the switch is V_cWith the above two discrete variables as state quantities, it can be obtained from equations (1), (2) and (5):

$I_o\left(k\right)=\left|i_o(kT+\frac{T}{2})\right|=\frac{V_{dc}M\left(k\right)+V_c\left(k\right)}{{\mathrm{\&omega;}}_d{L}_{eq}}\exp \left(\frac{-\mathrm{\&pi;}}{4Q}\right)---\left(6\right)$

$\begin{array}{c}{V}_c(k+1)=\left|v_c(kT+T)\right|\\ =V_c\left(k\right)\exp \left(\frac{-\mathrm{\&pi;}}{2Q}\right)+(1+\exp (\frac{-\mathrm{\&pi;}}{2Q}\left)\right)V_{dc}M\left(k\right)\end{array}---\left(7\right)$

in the above formula, the load quality factor Q>>1；

Substituting equation (7) into equation (6) can obtain the current I in discrete time_o(k+1)：

I_o(k+1)＝ΦI_o(k)+u^*(k+1) (8)

The value of the equivalent control input u x (k +1) ranges from {1, 0.5, 0, -0.5, -1}, and the maximum value of the output current flowing through the load coil can be obtained by equation (8):

$I_{max}=\frac{\mathrm{\&Gamma;}}{(1-\mathrm{\&Phi;})}=\frac{4V_{dc}}{{\mathrm{\&pi;R}}_{eq}}---\left(9\right)$

in a sampling period, the change of the load current determines the amplitude of the current ripple, and the relational expression is as follows:

$\begin{array}{c}\mathrm{\&Delta;}I(k+1)=I_o(k+1)-I_o\left(k\right)\\ =(\mathrm{\&Phi;}-1)I_o\left(k\right)+{\mathrm{\&Gamma;u}}^{*}(k+1)\\ =\frac{\mathrm{\&pi;}}{2Q}\{I_{\max }{u}^{*}(k+1)-I_o\left(k\right)\}\end{array}---\left(10\right)$

wherein,

$\mathrm{\&Phi;}=\exp \left(\frac{-\mathrm{\&pi;}}{2Q}\right)=1-\frac{\mathrm{\&pi;}}{2Q},\mathrm{\&Gamma;}=\exp \left(\frac{-\mathrm{\&pi;}}{4Q}\right)\frac{2V_{dc}}{R_{eq}Q}=(1-\frac{\mathrm{\&pi;}}{4Q})\frac{2V_{dc}}{R_{eq}Q},u^{*}(k+1)=\frac{M\left(k\right)+M(k+1)}{2}$

the switching state s (k) based on the sliding mode variable structure control can be expressed as:

S(k)＝I_e(k)+K_iz(k) (11)

z(k+1)＝z(k)+TI_e(k) (12)

in the above formula, I_eIs the current error value; z (k) is an integral gain coefficient;

can be combined to form (8), (11) and (12) to obtain S (k);

$\begin{array}{c}S(k+1)-S\left(k\right)=(1-\mathrm{\&Phi;})I_o\left(k\right)-{\mathrm{\&Gamma;u}}^{*}(k+1)+K_i{\mathrm{TI}}_e\left(k\right)\\ =-\mathrm{\&Delta;}I(k+1)+K_i{\mathrm{TI}}_e\left(k\right)\end{array}---\left(13\right)$

when S (K +1) -S (K) is 0, K is obtained by solution_i；

$K_i=\frac{\mathrm{\&Delta;}I(k+1)}{{\mathrm{TI}}_e\left(k\right)}=\frac{\mathrm{\&pi;}}{2QT}\{1+\frac{I_{max}{u}^{*}(k+1)-I_{ref}}{i_e\left(k\right)}\}---\left(14\right)$

When the control input signal value M (k +1) ═ 1, u × k +1 may take on {1, 0.5 };

when u (K +1) ═ 1, K_iSatisfies the following conditions:

0＜K_i＜K₁(15)

wherein, K₁Is when u x (k +1) ═ 1 and I_e(k)>At 0 time corresponds to K_iA value;

when the control input signal value M (k +1) ═ 0, u × k +1 may take on {0, 0.5 };

when u (K +1) ═ 0, K_iSatisfies the following conditions:

0＜K_i＜K_o(16)

wherein, K_oIs when u x (k +1) ═ 0 and I_e(k)<K corresponding to formula (12) at 0_iThe value is obtained.

When u (K +1) ═ 0 or 1, K can be obtained from formulas (14) and (15)_iSatisfies the following conditions:

$0<K_i<\frac{\mathrm{\&pi;}}{2QT}\&lsqb;1+min\{\frac{I_{max}-I_{ref}}{\left|I_e\left(k\right)\right|},\frac{I_{ref}}{\left|I_e\left(k\right)\right|}\}\&rsqb;---\left(17\right)$

however, when u (K +1) ═ 0.5, K_iCannot be solved, this time corresponds to S (k)<0，I_o(k)<0.5 or S (k)>0，I_o(k)>0.5, in order to reduce the current ripple in this case, the value of the next time S (k +2) should satisfy:

S(k){S(k+2)-S(k+1)}＜0 (18)

at this time, u^*A possible value of (k +2) is {1, 0}, and M (k +2) ═ M (k +1) is satisfied;

when I is satisfied by the above formula (17)_ref＝I_maxOr I_refWhen 0, it can be found that:

$0<K_i<\frac{\mathrm{\&pi;}}{2QT}---\left(19\right)$

in this embodiment, FIGS. 6 and 7 are views of the present invention, respectively_ref<0.5I_maxWhen and when I_ref>0.5I_maxAnd (4) a schematic diagram of the corresponding current error and the control input value. Wherein, I_refFor reference current value, I_maxThe maximum value of the output current is analyzed, and the error bandwidth η is expressed as:

${\mathrm{\&eta;}}_{min}=max\left\{\right|{\mathrm{\&Delta;I}}_{(u^{*}=0.5)}|+|{\mathrm{\&Delta;I}}_{(u^{*}=1)}|,|{\mathrm{\&Delta;I}}_{(u^{*}=0.5)}|+|{\mathrm{\&Delta;I}}_{(u^{*}=0)}\left|\right\}=\frac{1.5\mathrm{\&pi;}}{2Q}{I}_{max}---\left(20\right)$

the working principle of the invention is as follows:

the invention is suitable for high-power switching power supplies such as an induction heating power supply, an Uninterruptible Power Supply (UPS) and the like, and firstly provides a discrete time dynamic model based on a series resonance inverter for simplifying and effectively analyzing a current adjustable Delta Controller (CRDM). And analyzing to obtain the corresponding change condition of the output current value of the resonant inverter and the reason for generating output current ripples under different switching states.

Then when a current adjustable Delta controller with integral reset is added to control the resonant inverter, the response time of the output state switching speed or the current change rate is further shortened, the current ripple content on the output side is correspondingly reduced, and the current bias is reduced.

And then, the switching state of the inverter is controlled and analyzed by combining a sliding mode variable structure, a proper integral gain value and an error bandwidth value are selected, the current value of the output side of the inverter circuit acquired in each half resonant period is compared with a reference current value, a generated error signal is output through a PI controller, and the operation mode of the inverter at the next moment is determined through the regulation of a regulator.

It should be emphasized that the embodiments described herein are illustrative rather than restrictive, and thus the present invention is not limited to the embodiments described in the detailed description, but also includes other embodiments that can be derived from the technical solutions of the present invention by those skilled in the art.

Claims (5)

1. an improved sliding mode variable structure Delta modulation method, is characterized in that: comprise the following steps: Step 1. Establish a discrete-time dynamic model of the series resonant inverter controlled by the traditional current-adjustable Delta controller, and through the equivalent analysis of the series-resonant inverter based on the traditional current-adjustable Delta controller In the sampling period, when the traditional current adjustable Delta controller is used to control, the relational expression between the output load current and the current ripple amplitude of the series resonant inverter; Step 2. According to the relationship expression between the output load current variation and the current ripple amplitude of the series resonant inverter in step 1, draw the correspondence between the load output current and the current ripple amplitude under different equivalent control inputs Relationship diagram; and in order to reduce the output side current bias value of the current adjustable Delta controller, an integrator is introduced to improve the output performance of the current adjustable Delta controller; Step 3. Analyze the switching state of the series resonant inverter based on the sliding mode variable structure control theory, and select an appropriate integral gain value Ki of the integrator; Step 4, in order to further expand the upper limit of the integral gain value Ki of the integrator, introduce an integral reset link on the feedforward channel of the current adjustable Delta controller introduced into the integrator in the step 2, and select a suitable error bandwidth η, when the current error value on the output side is greater than the preset error bandwidth η, the integrator is forced to reset. With this control method, the upper limit of the integral gain value Ki of the integrator is further expanded. 2. A kind of improved sliding mode variable structure Delta modulation method according to claim 1, is characterized in that: the relational expression between the output load current of the series resonant inverter of described step 1 and current ripple amplitude The formula is: $\begin{array}{c}\mathrm{<span\; class="notranslate">\; \&Delta;</span>}\mathrm{<span\; class="notranslate">\; I</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span>\\ <span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; \&pi;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; Q</span>}}<span\; class="notranslate">\; \{</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; max</span>}}{\mathrm{<span\; class="notranslate">\; u</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \}</span>\end{array}$ Among them, u*(k+1) is the equivalent control input, and the corresponding expression is: ${\mathrm{<span\; class="notranslate">\; u</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; m</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; m</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; 2</span>}}$ Among them, M(k) is used to describe the operation mode of the inverter, which can be expressed as follows: In the above formula, Q is the load quality factor; ΔI is the current ripple amplitude; I _max is the maximum value of the output current; I _o is the output current. 3. A kind of improved sliding mode variable structure Delta modulation method according to claim 1 or 2, it is characterized in that: the introduction integrator of described step 2 improves the output performance of current adjustable Delta controller The specific method is: in An integrator is added to the feed-forward channel of the traditional adjustable current Delta controller. In each half of the resonance cycle, the output current value of the series resonant inverter is collected and compared with the reference current value to generate an error signal, and then passed through the integrator The adjustment to generate the switch control signal of the inverter at the next moment, and then determine the operation mode of the inverter at the next moment. 4. a kind of improved sliding mode variable structure Delta modulation method according to claim 1 or 2, is characterized in that: the concrete method of described step 3 is: in order to select suitable integrator gain Ki, introduce sliding mode variable structure Control theory, establish the switch state expression based on the sliding mode variable structure control, and establish the relationship between the switch state and the equivalent control input under the sliding mode variable structure control, and obtain the integral gain Ki and the equivalent control input u*(k+ 1), and then by selecting different equivalent control inputs, the upper limit value of the integral gain Ki is determined as: $\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; <</span>{\mathrm{<span\; class="notranslate">\; K</span>}}_{\mathrm{<span\; class="notranslate">\; i</span>}}<span\; class="notranslate">\; <</span>\frac{\mathrm{<span\; class="notranslate">\; \&pi;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; Q</span>}\mathrm{<span\; class="notranslate">\; T</span>}}$ In the above formula, T is half the resonant cycle. 5. a kind of improved sliding mode variable structure Delta modulation method according to claim 1 or 2, is characterized in that: the concrete method of choosing the suitable error bandwidth n of described step 4 is: combine the described series connection of step 1 The relational expression between the output load current of the resonant inverter and the current ripple amplitude is plotted to obtain the corresponding curves of the equivalent control input and the current error value under different reference current values, and then the error bandwidth η is analyzed The expression is: ${\mathrm{<span\; class="notranslate">\; \&eta;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}\mathrm{<span\; class="notranslate">\; i</span>}\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; m</span>}\mathrm{<span\; class="notranslate">\; a</span>}\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; \{</span><span\; class="notranslate">\; |</span>{\mathrm{<span\; class="notranslate">\; \&Delta;I</span>}}_{<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; u</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0.5</span>}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; |</span><span\; class="notranslate">\; +</span><span\; class="notranslate">\; |</span>{\mathrm{<span\; class="notranslate">\; \&Delta;I</span>}}_{<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; u</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; |</span><span\; class="notranslate">\; ,</span><span\; class="notranslate">\; |</span>{\mathrm{<span\; class="notranslate">\; \&Delta;I</span>}}_{<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; u</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0.5</span>}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; |</span><span\; class="notranslate">\; +</span><span\; class="notranslate">\; |</span>{\mathrm{<span\; class="notranslate">\; \&Delta;I</span>}}_{<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; u</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; |</span><span\; class="notranslate">\; \}</span><span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1.5</span>}\mathrm{<span\; class="notranslate">\; \&pi;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; Q</span>}}{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}\mathrm{<span\; class="notranslate">\; a</span>}\mathrm{<span\; class="notranslate">\; x</span>}}<span\; class="notranslate">\; .</span>$