CA2238680C - Multicode spread spectrum communications system - Google Patents

Abstract

MultiCode Spread Spectrum (MCSS) is a modulation scheme that assigns a number N of Spread Spectrum (SS) codes to an individual user where the number of chips per SS code is M. When viewed as Direct Sequence Spread Spectrum, MCSS requires up to N correlators (or equivalently up to N Matched Filters) at the receiver with a complexity of the order of NM operations. In addition, a non ideal communication channel can cause InterCode Interference (ICI), i.e. interference between the N SS codes. In this patent, we introduce three new types of MCSS. MCSS Type I allows the information in a MCSS signal to be detected using a sequence of partial corrrelations with a combined complexity of the order of M operations. MCSS Type II allows the information in a MCSS signal to be detected in a sequence of low complexity parallel operations which reduce the ICI. MCSS Type III allows the information in a MCSS signal to be detected using a filter suitable for ASIC implementation or on Digital Signal Processor, which reduces the effect of multipath. In addition to low complexity detection and reduced ICI, MCSS has the added advantage that it is spectrally efficient.

Description

TITLE OF THE INVENTION: MultiCode Spread Spectrum Communications System NAMES OF THE INVENTORS: Michel T. Fattouche, Hatim Zaghloul, Paul R. Milligan and David L. Snell Field of the Invention: The invention deals with the field of multiple access communications using Spread Spectrum modulation. Multiple access can be classified as either random access, polling, TDMA, FDMA, CDMA or any combination thereof. Spread Spectrum can be classified as Direct Sequence, Frequency-Hopping or a combination of the two.

Background of the Invention: Commonly used spread spectrum techniques are Direct Sequence Spread Spectrum (DSSS) and Code Division Multiple Access (CDMA) as explained respectively in Chapters 13 and 15 of "Digital Communication" by J.G. Proakis, Third Edition, 1995, McGraw Hill. DSSS
(See Simon M.K. et al., "Spread Spectrum Communications Handbook,"
Revised Edition, McGraw-Hill, 1994 and see Dixon, R.C., "Spread Spectrum systems with commercial applications," Wiley InterScience, 1994) is a communication scheme in which information symbols are spread over code bits (generally called chips). It is customary to use noise-like codes called pseudo-random noise (PN) sequences. These PN sequences have the property that their auto-correlation is almost a delta function. In other words, proper codes perform an invertible randomized spreading of the information sequence. The advantages of this information spreading are:
1. The transmitted signal can be buried in noise and thus has a low probability of intercept.

2. The receiver can recover the signal from interferers (such as other transmitted codes) with a jamming margin that is proportional to the spreading code length.

3. DSSS codes of duration longer than the delay spread of the propagation channel can lead to multipath diversity implementable using a Rake receiver.

4. The FCC and Industry Canada have allowed the use of unlicensed low power DSSS systems of code lengths greater than or equal to 10 (part 15 rules) in some frequency bands (the ISM bands).

It is the last advantage (i.e. advantage 4. above) that has given much interest recently to DSSS.
An obvious limitation of DSSS systems is the limited throughput they can offer. In any given bandwidth, W, a code of length M will reduce the cffective bandwidth to W/M. To increase the overall bandwidth efficiency, system designers introduced Code Division Multiple Access (CDMA) where multiple DSSS communication links can be established simultaneously over the same frequency band provided each link uses a unique code that is noise-like, i.e. provided the cross-correlation between codes is almost null.
Examples of CDMA is the next generation of digital Cellular communications in North America: "the TIA Interim 3"tandard IS-95," (see QUALCOMM Inc., "An overview of the application of Code Division Multiple Access (CDMA) to digital cellular systems and personal cellular networks," May 21, 1992 and see Viterbi, A.J., "CDMA, Principles of Spread Spectrum Communications,"
Addison-Wesley, 1995) where a Base Station (BS) communicates to a number of Mobile Stations (MS) simultaneously over the same channel. The MSs share one carrier frequency during the mobile-to-base link (also known as the reverse link) which is 45MHz away from the one used by the BS during the base-to-mobile link (also known as the forward link). During the forward link, the BS transceiver is assigned N codes where N is less than or equal to M
and M is the number of chips per DSSS code. During the reverse link each MS is assigned a unique code.

CDMA problems are:
1. The near-far problem on the reverse link: an MS transmitter "near" the BS
receiver can overwhelm the reception of codes transmitted from other MSs that are "far" from the BS.
2. Synchronization on the reverse link: synchronization is complex (especially) if the BS receiver does not know in advance either the identity of the code being transmitted, or its time of arrival.

Summary of the Invention: We have recognized that low power DSSS
systems would be ideal communicators provided the problems of CDMA
could be resolved. In order to avoid both the near-far problem and the synchronization problem that exist on the reverse link of a CDMA system, we have opted in this patent to use only the forward link at all times for MCSS

Types I and II. This is achieved within a specified channel by allowing only one transceiver to transmit at a time within a certain coverage area. Such a transceiver is forced during transmission to act as the BS in transmit mode while the remaining transceivers are forced to act as MSs in receive mode. In this patent, we refer to such a modulation scheme as MultiCode Spread Spectrum (MCSS).
On the other hand, both the near-far problem and the synchronization problem that exist on the reverse link of a CDMA system are reduced drastically by using MCSS Type III. In this case, each user is assigned one code and each code is assigned a guard time such that it starts to transmit only after a given amount of time relative to any adjacent codes. By forcing the users to have separate start times, MCSS Type III forces the codes to be (quasi) orthogonal as long as the guard time between adjacent codes is long enough.
When viewed as DSSS, a MCSS receiver requires up to N correlators (or equivalently up to N Matched Filters) (such as in QUALCOMM Inc,."An overview of the application of Code Division Multiple Access (CDMA) to digital cellular systems and personal cellular networks, May 21, 1994 and as in Viterbi, A.J., "CDMA, Principles of Spread Spectrum Communications,"
Addison-Wesley, 1995) with a complexity of the order of NM operations.
When both N and M are large, this complexity is prohibitive. In addition, a nonideal communication channel can cause InterCode Interference (ICI), i.e.
interference between the N SS codes at the receiver. In this patent, we introduce three new types of MCSS. MCSS Type I allows the information in a MCSS signal to be detected using a sequence of partial correlations with a combined complexity of the order of M operations. MCSS Type II allows the information in a MCSS signal to be detected in a sequence of low complexity parallel operations while reducing the ICI. MCSS Type III allows the information in a MCSS signal to be detected in a sequence of low complexity Multiply and Accumulate (MAC) operations implementable as a filter, which reduce the effect of multipath. In addition to low complexity detection and ICI
reduction, our implementation of MCSS has the advantage that it is spectrally efficient since N can be made approximately equal to M. In DSSS, N=1 while in CDMA typically N < 0.4M.

Description of the Invention:

The description of the invention consists of six parts. The first three parts correspond to trie transmitter for each one of the three types of MCSS
introduced in this patent, while the last three parts correspond to the receiver for each one of the three types of MCSS.

Description of the Transmitter for MCSS Type I:
Figure 1 illustrates a block diagram of the transmitter for MCSS Type I
with an input of V frames of B data symbols each, every VBT seconds and an output of P frames of M multicoded SS symbols each, every PMTc seconds where T is the duration of one data symbol and Tc is the duration of one chip in a spread spectrum code. The data symbols can be either analog or digital.
If digital, they belong to an alphabet of finite size. If analog, they correspond to the samples of an analog signal.
Figure 1 is described as follows:
= The first block in Figure 1 is a serial-to-parallel converter (101) with an input of B data symbols and an output of one frame of B data symbols, every BT seconds.
= The second block is a 2 Dimensional (2D) shift register (102) with an input of V frames of B data symbols each (input by shifting the frames from left to right V times) and an output of Q frames of B data symbols each, every VBT seconds.
= When the data symbols are analog, the third block (103) in Figure 1 corresponds to an analog pulse modulator with several possible modulation schemes such as Pulse Amplitude Modulation (PAM), Pulse Position Modulation (PPM), Pulse Frequency Modulation (PFM), etc. When the data symbols are digital, the third block is a channel encoder/modulator (103) with an input of Q frames of B data symbols each and an output of P
frames of J modulated symbols each, every QBT seconds. The channel encoder/modulator performs two functions: (1) to encode and (2) to modulate the data symbols. The first function offers protection to the symbols against a non ideal communication channel by adding redundancy to the input sequence of data symbols while the second function maps the protected symbols into constellation points that are appropriate to the communication channel. Sometimes it is possible to perform the two functions simultaneously such as in the case of Trellis Coded Modulation (TCM). For simplicity, we assume throughout the patent that the two functions are performed simultarieously and refer to the block performing the two functions as the channel encoder/modulator.
Different types of channel encoders are available:
= If the 2D shift register (102) is operated wi*'i V=Q, then the encoder performs block encoding, otherwise if V<Q, the encoder performs convolutional encoding. Furthermore, if B>J then the encoder is a trellis encoded modulator either with block encoding if V=Q or with convolutional encoding with V<Q.
= If B=J, the code rate is Q/P, i.e. the encoder takes Q data symbols in and generates P encoded data symbols out where P>Q. Furthermore, if V<Q then (V-1) is the constraint length of the convolutional encoder.
= If the 2D shift register (102) is operated with B> 1, then it can act as an interleaver which interleaves the data symbols prior to the channel encoder (103), otherwise if B=1 the channel encoder does not rely on interleaving. Another possible form of interleaving is to interleave the coded data symbols after the channel encoder (not shown in Figure 1).
Different types of modulators are available such as: Binary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying (QPSK), Multilevel Phase Shift Keying (MPSK), Quadrature Amplitude Modulation (QAM), Frequency Shift Keying (FSK), Continuous Phase Modulation (CPM), Amplitude Shift Keying (ASK), etc. All amplitude and frequency modulation schemes can be demodulated either coherently or noncoherently. All phase modulation schemes can de demodulated either coherently or differentially. In the latter case, differential encoding is required in the modulator such as in Differential BPSK (DBPSK), Differential QPSK (DQPSK), Differential MPSK (DMPSK), etc.
Even though the output of the channel encoder/modulator (103) corresponds to an encoded and modulated data symbol, we will refer to it of as a `modulated symbol'.

= The fourth block is a spreader type I(104) with an input of P frames of J
modulated symbols each and an output of P frames of N spread spectrum symbols each, of length M chips per spread spectrum symbol, every PMTc seconds. The spreader type I(104) is explained further below in Figures 2-

5.

6 = The fifth block is a 3 Dimensional (3D) shift register (105) with an input of P frames of N spread spectrum symbols each (input by shifting the PN
symbols from inside to outside M chip times), and an output of M frames of N chips each (output by shifting MN chips from left to right P times) every PMTc seconds.
= The sixth block is a set of M adders (106). Each adder has an input of N
chips and an output of one multicoded SS symbol, every MTc seconds.
= The seventh block is a parallel-to-serial converter (107) with an input of one frame of M multicoded SS symbol and an output of M multicoded SS
symbol every MTc seconds.

The spreader type I(104) in Figure 1 is described further in Figure 2 with an input of P frames of J modulated symbols each, generated by the channel encoder/modulator (103) in Figure 1, and an output of P frames of N spread spectrum symbols each, of length M chips per spread spectrum symbol. Figure 2 is described as follows:
= The first block in Figure 2 is a set of P converters (201) with an input of one frame of J modulated symbols per converter, and an output of one frame of N subsets of modulated symbols per converter. The ith subset contains a number J; of modulated symbols where Jl +J2 +...+JN =J and i=1,...,N.
= The second block is a set of N computing means (202) with an input of one subset of modulated symbols per computing means, and an output of one spread spectrum symbol, of length M chips per computing means.

The set of N computing means (202) in Figure 2 is described further in Figure 3 which displays only the ith computing mean where i=1,...,N. The ith computing mean has as an input the ith subset of modulated symbols, and as an output the ith spread spectrum symbol of length M chips. Figure 3 is described as follows.
= The first block in Figure 3 is the ith mapper (301) with two inputs and one output. The two inputs are: (1) the ith subset of modulated symbols which contains a number J; of modulated symbols, and (2) Li spread spectrum codes of length M chips each. The output is the ith spread spectrum symbol. The ith mapper chooses from the set of L; spread spectrum codes the code corresponding to the ith subset of modulated symbols to become

7 the ith spread spectrum code representing an invertible randomized spreading of the ith subset of modulated symbols.
= The second block in Figure 3 is the ith source (302) of L; spread spectrum codes with an output of Li spread spectrum codes of 1 ngth M chips each.
The ith source (302) can be thought of as eithei a lookup table or a code generator. Two different implementations of the ith source are shown in Figures 4 and 5.

Remarks on the "invertible randomized spreading":
1. In this patent, the invertible randomized spreading of a signal using a spreader is only invertible to the extent of the available arithmetic precision of the machine used to implement the spreader. In other words, with finite precision arithmetic, the spreading is allowed to add a limited amount of quantization noise.
2. Moreover, the randomized spreading of a signal is not a perfect randomization of the signal (which is impossible) but only a pseudo-randomization. This is typical of spread spectrum techniques in general.
3. Finally, in some cases such as over the multipath communication channel, it is advantageous to spread the signal over a bandwidth wider than 25% of the coherence bandwidth of the channel. In this patent, we refer to such a spreading as wideband spreading. In the indoor wireless channel, 25% of the coherence bandwidth ranges from 2MHz to 4MHz. In the outdoor wireless channel, 25% of the coherence bandwidth ranges from 30KHz to 60KHz. In other words, in this patent wideband spreading corresponds to a spreading of the information signal over a bandwidth wider than 30KHz over the outdoor wireless channel and wider than 2MHz over the indoor wireless channel, regardless of the bandwidth of the information signal and regardless of the carrier frequency of modulation.

The ith source (302) of Figure 3 can also be generated as in Figure 4 as a set of Li transforms with an input of one preset sequence of length M chips per transform and an output of one spread spectrum code of length M chips per transform. In other words, the ith source of spread spectrum codes could be either a look-up table containing the codes such as in Figure 3 or a number of transforms generating the codes such as in Figure 4.

8 The ith source (302) of Figure 3 can also be generated as in Figure 5 as two separate blocks.
= The first block (501) consists of a set of L; transforms with an input of one preset sequence of length M chips per transform and an output of one spread spectrum code of length M chips per transform.
= The second block is a randomizing transform (502) with an input of L;
transformed codes of length M chips each generated by the first block (501) and an output of Li spread spectrum codes of length M chips each.
= The randomizing transform consists of two parts. The first part is a randomizing look-up table (503) which contains a set of M preset values:
al,;, a2,;,...,aM,;. The second part multiplies each transformed symbol from the set of transformed symbols generated by the first transform (501) by the set of M preset values generated by the randomizing look-up table (503). The multiplication is performed chip-by-chip, i.e. the kth chip in the ith transformed symbol is multiplied by the kth value ak ; in the set of M
preset values for all values of k=1, ..., M.

Description of the Transmitter for MCSS Type II:
Figure 6 illustrates a block diagram of the transmitter for MCSS Type II
with an input of VB data symbols every VBT seconds and an output of PM
multicoded SS symbols every PMTc seconds. Figure 6 is described as follows:
= The first block in Figure 6 is a serial-to-parallel converter (601) with an input of B data symbols and an output of one frame of B data symbols, every BT seconds.
= The second block is a 2 Dimensional (2D) shift register (602) with an input of V frames of B data symbols each (input by shifting the frames from left to right V times) and an output of Q frames of B data symbols each, every VBT seconds.
= The third block is a channel encoder/modulator (603) with an input of Q
frames of B data symbols each and an output of P frames of J modulated symbols each, every QBT seconds. The function of the channel encoder/modulator is exactly the same as the --`:annel encoder/modulator (103) described above for MCSS type I in Figure 1.
= The fourth block is a spreader type II (604) with an input of P frames of J
modulated symbols each and an output of P frames of M multicoded SS

9 symbols each, every PMTc seconds. The spreader type II is explained further below in Figures 7-9.
= The fifth block is a 2 Dimensional (2D) shift register (605) with an input of P frames of M multicoded SS symbols each, and an output of P frames of M multicoded SS symbols each (output by shifting the M frames from left to right P times) every PMTc seconds.
= The sixth block is a parallel-to-serial converter (606) with an input of one frame of M multicoded SS symbols and an output of M multicoded SS
symbols every MTc seconds.

The spreader type II (604) in Figure 6 is described further in Figure 7 with an input of P frames of J modulated symbols each, generated by the channel encoder/modulator (603) in Figure 6, and an output of P frames of M
multicoded SS symbols each. Figure 7 is described as follows:
= The first block in Figure 7 is a set of P converters (701) with an input of one frame of J modulated symbols per converter, and an output of one frame of M subsets of modulated symbols per converter. The ith subset contains a number of J; of modulated symbols where J, +J2 +...+JM =J and i=1,...,M.
= The second block is a set of P M-point transforms (702) with an input of M
subsets of modulated symbols per transform, and an output of a frame of M
multicoded SS symbols per transform. The P M-point transforms perform the invertible randomized spreading of the M subsets of modulated symbols.

The set of P M-point transforms (702) in Figure 7 is described further in Figure 8 which displays only the ith M-point transform where i=1,...,N. The input of the ith transform is the ith subset of J; modulated symbols, and the output is the ith frame of M multicoded SS symbols. In Figure 8, the ith M-point transform is the randomizing transform (801) similar to the randomizing transform (502) in Figure 5 with the set of preset values given as: a1,;, aZ.;, ..., aM,;. In this case, the kth preset value ak,; multiplies the kth subset of Jk modulated symbols to generate the kth multicoded SS symbol.

The ith M-point transform (801) in Figure 8 can further include a second M-point transform (902) as described in Figure 9.

= The first M-point transform (901) is the ith randomizing transform with an input of the ith subset of J; modulated symbols, and an output of the ith frame of M transformed symbols.
= The second M-point transform (902) is the ith second M-point transform with an input of the ith frame of transformed symbols, and an output of the ith frame of M multicoded SS symbols.

Description of the Transmitter for MCSS Type III:
Figure 10 illustrates a block diagram of the transmitter for MCSS Type III with an input of a stream of data symbols and an output of a stream of multicoded SS symbols. Figure 10 is described as follows:
= The first block is a channel encoder/modulator (1001) with an input of a stream of data symbols and an output of a stream of modulated symbols.
The function of the channel encoder/modulator is similar to the channel encoder/modulator for MCSS types I and II (103) and (603) respectively except its operation is serial. Such a representation is commonly used in textbooks to implicitly imply that the data rate of the output stream of modulated symbols could be different from the input stream of data symbols. In other words, the channel encoder/modulator can add redundancy to the input stream of data symbols to protect it against channel distortion and noise. The type of redundancy varies depending on the type of encoding used. In block encoding, the redundancy depends only on the current block of data. In convolutional encoding, it depends on the current block and parts of the previous block of data. In both types of encoding trellis coding can be used which modulates the modulated symbols output from the encoder.
Even though Figure 10 does not contain an interleaver, it is possible to include one either before the channel encoder/modulator or after.
= The second block is a spreader type III (1002) with an input of a stream of modulated symbols and an output of a stream of multicoded SS symbols.
The spreader type III is further explained in Figures 11-13.
= The third block is a ramper (1003) with an input of multicoded SS symbols and an output of a ramped multicoded SS symbols. The ramper is further explained in Figure 14.

The spreader type III (1002) in Figure 10 is described further in Figure 11 as two blocks with an input of a stream of modulated symbols, generated by the channel encoder/modulator (1001) in Figure 10, and an output of a stream of multicoded SS symbols.
= The first block is a randomizer (1101) with an input of a stream of modulated symbols and an output of a randomized modulated symbols. The randomizer is described further in Figure 12.
= The second block is a computing means (1102) with an input of the stream of randomized modulated symbols and an output of a stream of multicoded SS
symbols. The computing means is described further in Figure 13.

In Figure 12 the randomizer (1101) from Figure I 1 is described further as two parts.
= The first part is a chip-by-chip multiplier (1201) with two inputs and one output. The first input is the stream of modulated symbols and the second input is a stream of preset values output from a randomizing lookup table (1202). The output is the product between the two inputs obtained chip-by-chip, i.e. the kth randomized modulated symbols is obtained by multiplying the kth modulated symbol with the kth preset value ak.
= The second part is the randomizing lookup table (1202) which is the source of a stream of preset values: ...,ak,ak+l,... As mentioned before, the randomizing sequence is only pseudo-randomizing the modulated symbols.
In Figure 13 the computing means (1102) from Figure 11 is described further as a filter which performs the invertible randomized spreading of the stream of modulated symbols.

Figure 14 illustrates the ramper (1003) in Figure 10 as a mixer with two inputs and one output. The first input is the stream of multicoded SS symbols, the second input is a linearly ramping carrier frequency ej2if `Z which ramps the multicoded SS stream over the time `t' thereby generating a stream of ramped multicoded SS symbols where j= -1--l and f o is a constant.

Description of the Receiver for MCSS Type I:
Figure 15 illustrates a block diagram of the receiver for MCSS type I & II
with an input of PM multicoded SS symbols, every PMTc seconds and an output of VB estimated data symbols, every VBT seconds. Figure 15 is described as follows:
= The first block in Figure 15 is a serial-to-parallel converter (1501) with an input of M multicoded SS symbols and an output of one frame of M
multicoded SS symbols every MTc seconds.
= The second block is a 2 Dimensional (2D) shift register (1502) with an input of one frame of M multicoded SS symbols each (input by shifting the frame from left to right P times) and an output of P frames of M
multicoded SS symbols each, every PMTc seconds.
= The third block is a despreader type I(1503) with an input of P frames of M multicoded SS symbols each and an output of P frames of J despread symbols each every PMTc seconds. The despreader type I is further explained below.
= The fourth block is a channel decoder/demodulator (1504) with an input of P frames of J despread symbols each and an output of V frames of B
estimated data symbols each, every VBT seconds. The channel decoder/demodulator performs two functions: (1) to map the despread symbols into protected data symbols and (2) either to detect errors, or to correct errors, or both. Sometimes, the two functions can be performed simultaneously. In this case, the channel decoder/demodulator performs soft-decision decoding, otherwise, it performs hard-decision decoding. By perfonning the two function, the channel encoder/demodulator accepts the despread symbols and generates estimated data symbols = The fifth block is a 2 Dimensional (2D) shift register (1505) with an input of V frames of B estimated data symbols each, and an output of V frames of B estimated data symbols (output by shifting the V frames from left to right) every VBT seconds. If the 2D shift register (102) is operated with B>1, then it might act as an interleaver. In this case, the receiver requires a de-interleaver which is accomplished using the 2D shift register (1505).
= The sixth block is a parallel-to-serial converter (1506) with an input of one frame of B estimated data symbols and an output of B estimated data symbols, every VBT seconds.

The despreader type I(1504) in Figure 15 is described further in Figure 16 with an input of P frames of M multicoded SS symbols each from the received sequence of multicoded SS symbols, and an output of P frames of J despread symbols each. Figure 16 is described as follows:
= The first block in Figure 16 is a set of P parallel-to-serial converters (1601) with an input of one frame of M multicoded SS s,,mbols per converter, and an output of M multicoded SS symbols per converter.
= The second block is a set of N computing means (1602) each having the same input of M multicoded SS symbols and an output of one computed value per computing means.
= The third block is a detector (1603) with an input of N computed values and an output of J despread symbols per detector. When the data symbols are digital, the detector can make either hard decisions or soft decisions.
When the data symbols are analog, L; is necessarily equal to 1 for i=l,...,N
and the detector is not required.

The set of N computing means (1602) in Figure 16 is described further in Figure 17 which displays only the ith computing mean where i=1,...,N. The ith computing mean has as an input the M multicoded SS symbols, and as an output the ith computed value. Figure 17 is described as follows.
= The first block in Figure 17 is a set of L; partial correlators (1701). The nth partial correlator has two inputs where n=1,2,...,L;. The first input consists of the M multicoded SS symbols and the second input consists of the nth spread spectrum code of length M chips out of the ith source of Li spread spectrum codes. The output of the nth partial correlator is the nth partially correlated value obtained by correlating parts of the first input with the corresponding parts of the second input.
= The second block is the ith source (1702) of L; spread spectrum codes with an output of L; spread spectrum codes of length M chips each.
= The third block is the ith sub-detector (1703) with an input of Li partially correlated values and an output of the ith computed value. The ith sub-detector has two tasks. First using the L; partially correlated values it has to obtain the full correlation between the M multicoded SS symbols and each one of the Li spread spectrum codes of length M chips obtained from the ith source (1702). Then, it has to select the spread spectrum code corresponding to the largest correlation. Such a detected spread spectrum code together with the corresponding full correlation value form the ith computed value.

The detector (1703) in Figure 16 takes all the computed values from each one of the N computing means and outputs J despread symbols. Based on the function of each sub-detector, one can say that the detector (1603) has two tasks at hand. First, it has to map each detected spread spectrum code into a first set of despread symbols, then it has to map each full correlation value into a second set of despread symbols. In other words, the first set of despread symbols correspond to spread spectrum codes that form a subset of the spread spectrum codes corresponding to the second set of despread symbols.

It is also possible to have several layers of sub-detectors completing different levels of partial correlations and ending with N spread spectrum codes corresponding to the largest full correlation values per computing means. In this case, the tasks of the detector are first to map each detected spread spectrum code (obtained through the several layers of sub-detection) into sets of despread symbols, then to map each full correlation value into a final set of despread symbols.

Description of the Receiver for MCSS Type II:
Figure 15 illustrates a block diagram of the receiver for MCSS Type II
with an input of PM multicoded SS symbols every PMTc seconds and an output of VB estimated data symbols every VBT seconds. Figure 15 illustrates also the block diagram of the receiver for MCSS Type I and has been described above.

The despreader type II (1504) in Figure.15 is described further in Figure 18 with an input of P frames of M multicoded SS symbols each, and an output of P frames of J despread symbols each. Figure 18 is described as follows:
= The first block in Figure 18 is a set of P M-point transforms (1801) with an input of one frame of M multicoded SS symbols per transformer, and an output of M transformed symbols per transformer.
= The second block is a set of P detectors (1802) with an input of M
transformed symbols per detector, and an output of J despread symbols per detector. Once again the detector can either make soft decisions or hard decisions.

Description of the Receiver for MCSS Type III:
Figure 19 illustrates a block diagram of the receiver for MCSS Type III
with an input of a stream of ramped multicoded SS symbols and an output of a stream of estimated data symbols. Figure 19 is descri;.ed as follows:
= The first block in Figure 19 is a de-ramper (1901) with an input of the stream of ramped multicoded SS symbols and an output of an estimated stream of multicoded SS symbols. The de-ramper is further described in Figure 20.
= The second block is a de-spreader Type III (1902) with an input of the estimated stream of multicoded SS symbols and an output of a stream of detected symbols. The de-spreader type II is further explained in Figure 21-23.
= The third block is a channel decoder/demodulator (1903) with the input consisting of the stream of detected symbols, and an output of a stream of estimated data symbols. It is clear from Figure 19 that no de-interleaver is included in the receiver. As mentioned above, if an interleaver is added to the transmitter in Figure 10, then Figure 19 requires a de-interleaver.

Figure 20 illustrates the deramper (1901) in Figure 19 as a mixer with two inputs and one output. The first input is the ramped multicoded SS symbols and the second input is a linearly ramping carrier frequency which deramps the ramped multicoded SS stream thereby generating an estimated stream of multicoded SS symbols.

The despreader type III (1902) in Figure 19 is described further in Figure 21 as three blocks.
= The first block is a computing means (2101) with an input of an estimated stream of multicoded SS symbols and an output of a stream of randomized despread symbols. Figure 22 describes the computing means (2101) in Figure 21 as a filter (2201) which performs the despreading process.
= The second block is a de-randomizer (2102) with an input of a stream of randomized despread symbols and an output of a stream of despread symbols.
The de-randomizer (2102) is described further in Figure 23.
= The third block is a detector (2103) with an input of a stream of despread symbols and an output of a stream of detected symbols. When the detector is a hard-decision detector it makes a decision on the despread symbols such that the detected values takes a finite number of values out of a predetermined alphabet of finite size. When the detector is a soft-decision detector the detected symbols are the same as the despread symbols.

The de-randomizer (2102) is described further in Figure 23 as two parts.
= The first part is a chip-by-chip multiplier (2301) with two inputs and an output. The first input is a stream of randomized despread data symbols and the second input is a stream of preset values output from a de-randomizing lookup table (2302). The output is the chip-by-chip product between the two inputs, i.e. the kth despread symbol is obtained as the product between the kth randomized despread symbol and the kth preset value bk.
= The second part is a de-randomizing lookup table (2302) which outputs a stream of preset values: ...,bk,bk+l ,...

Preferred Embodiments of the Invention:
From the above description of the invention, it is clear that the contribution of the invention is primarily in the spreader in the transmitter and in the despreader in the receiver for each one of the three type of MCSS
introduced in the patent. The secondary contribution of the patent resides in the channel encoder/modulator and in the extra components that can be used in both the transmitter and in the receiver for each three types such as: the ramping and de-ramping of the signal and diversity techniques. For these reasons, we have separated the preferred embodiments of the invention into three parts. Each part corresponds to the spreader and the despreader for each one of the three types of MCSS and its extras.

Preferred Embodiments of the Spreader/Despreader for MCSS Type I:
= In Figure 1, the spreader Type I(104) performs an invertible randomized spreading of the modulated symbols which carry either digital information or analog information, and in Figure 15 the despreader Type I(1503) performs a reverse operation to the spreader Type I(104) within the limits of available precision (i.e. with some level of quantization noise).
= In Figure 1, the spreader Type I(104) performs an invertible randomized spreading of the modulated, and in Figure 15 the despreader Type I(1503) performs a reverse operation to the spreader Type I(104) while taking into account the effects of the communications channel such as noise, distortion and interference. The effects of the channel are sometimes unknown to the receiver (e.g. over selective fading channels which cause intersymbol interference). In such cases, the channel has to be estimated using for example a pilot signal known to the receiver as in "MultiCode Direct Sequence Spread Spectrum," by M. Fattouche and H. Zaghloul, US patent #5,555,268, Sept. 1996.
= In Figure 2, if Jk=O for any k=1,...,N then the output of the kth computing means is the all zeros spread spectrum codes of length M chips.
= In Figure 2, if the modulated symbols are M-ary symbols, then a preferred value for L; is M to the power of J;. In other words, by choosing one spread spectrum code out of L; codes, J; symbols of information are conveyed.
= In Figure 3, a preferred function for the ith mapper is to choose one spread spectrum code (out of the Li available codes) based on one part of the ith subset of Ji modulated symbols while the second part of the subset is used to choose the symbol that multiplies the chosen spread spectrum code. In other words, assuming that the kth spread spectrum code Sk is chosen by the ith mapper (301) (out of the Li available codes) based on the first part of the ith subset of Ji modulated symbols and that the symbol ~ is chosen to multiply Sk based on the second part of the ith subset of J; modulated symbols, then the ith spread spectrum symbol out of the ith mapper (301) is Sk~. This is equivalent to spreading ~ over Sk.
= In Figure 3, ~ can be chosen as a DBPSK symbol, a DQPSK symbol, a DMPSK symbol, a QAM symbol, a FSK symbol, a CPM symbol, an ASK
symbol, etc.
= In Figure 3, the Li spread spectrum codes, out of the ith source (302) of L;
available spread spectrum codes, correspond to Walsh codes. Each Walsh code in Figure 3 is generated in Figure 4 as the output of an M-point Walsh transform where the input is a preset sequence of length M chips with (M-1) chips taking a zero value while one chip taking a unity value.
= In Figure 3, the Li spread spectrum codes, out of the ith source (302) of Li available spread spectrum codes, correspond to randomized Walsh codes.
Each Walsh code generated in Figure 4 as the output of an M-point Walsh transform is randomized in Figure 5 using a chip-by-chip multiplier where the kth chip of each Walsh code is multiplied by the preset value ak ; output from the ith randomizing lookup table.

= In Figure 5, the M preset values { al ;,a2i ,...,aM;} are chosen such that their amplitudes: Iat,;I,Ia2;I,...,laM,;I are all equal to unity.
= In Figure 3, a preferred value for L; is 2 and a preferred value for M is 10 with the two preferred spread spectrum codes out of the ith source (302) taking the values:

{ CI,C2 IC3 ,C4.+C5.,C6,,C7,,C8,,C9 X10} and { C 1 ,C2 ,C3 ,C4 C5 ,-C6 1-C7 ,-C8 ,-C9 ,-C 10 } (1) = In equation (1), preferred values for the chips `c1,c2,c3,C4,C5,C6,c7,cS,c9,clo' are `1,-1,1,1,1,j, j,j,j,j' which we refer to as the 'Wi-LAN codes Type I'.
Preferred Embodiments of the Spreader/Despreader for MCSS Type II:
= In Figure 6, the spreader Type II (604) performs an invertible randomized spreading of the modulated symbols which carry either digital information or analog information, and in Figure 15 the despreader Type II (1503) performs a reverse operation to the spreader Type II (604) within the limits of available precision (i.e. with some level of quantization noise).
= In Figure 6, the spreader Type II (604) performs an invertible randomized spreading of the modulated, and in Figure 15 the despreader Type II (1503) performs a reverse operation to the spreader Type 11 (604) while taking into account the effects of the communications channel such as noise, distortion and interference. As mentioned above, the effects of the channel are sometimes unknown to the receiver (e.g. over selective fading channels which cause intersymbol interference). In such cases, the channel has to be estimated using for example a pilot signal known to the receiver as in "MultiCode Direct Sequence Spread Spectrum," by M. Fattouche and H.
Zaghloul, US Patent #5,555,268, Sept. 1996.
= Two preferred types of pilot signals can be used to estimate the effects of the channel on the information-bearing data symbols:
1. Pilot Frames inserted either before, during or after the Data frames of M multicoded SS symbols; and 2. Pilot Symbols inserted within each data frame of M multicoded SS
symbols.
Pilot frames estimate the long term effects of the channel, while pilot symbols estimate the short term effects of the channel.
= When channel estimation is used in the receiver as mentioned above, it is possible to use coherent detection with phase modulation, such as BPSK, QPSK and MPSK, after removing the effects of the channel from the phase of the received signal. On the other hand, if the effects of the channel are not removed, differential detection is selected instead with differentially-encoded phase modulation such as DPSK, DQPSK and DMPSK.
= Furthermore, when channel estimation is used in the receiver as mentioned above, it is possible to use amplitude modulation together with coherent detection of phase modulation, such as ASK and QAM, after removing the effects of the channel from the phase and the amplitude of the received signal. On the other hand, if the effects of the channel are not removed, differential detection is selected instead with differentially-encoded phase and amplitude modulation such as Differential QAM using the star constellation.
= A preferred modulation technique is QAM when the channel is estimated and its effects removed.
= Another preferred modulation technique is DMPSK when the effects of the channel are not removed. In this case, a reference symbol is chosen at the beginning of each frame output from the channel modulator/modulator (603).
= In Figure 6, a preferred channel encoder/modulator (603) is a Reed-Solomon channel encoder used for encoding M-ary symbols and for correcting errors caused by the channel at the receiver. If the data symbols are binary, it is preferred to choose to combine several input bits into one symbol prior to encoding. A preferred technique to combine several bits into one symbol is to combine bits that share the same position within a number of consecutive frames. For example, the kth bit in the nth frame can be combined with the kth bit in the (n+l)th frame to form a dibit, where k=1,...,Q.
= In Figure 6, if the data symbols are M-ary, a preferred value for B is unity when using a Reed-Solomon encoder, i.e. no interleaver is required in this case.
= In Figure 7, preferred values for J1,J2,...,JM are unity.
= In Figure 8, preferred values for { al,;, a2,;,...,aM,; } are such that their amplitudes: iat,; I,la2,; I,...,IaM,;I are all equal to unity.
= In Figure 9, preferred ith second M-point transform (902) is a Discrete Fourier Transform (DFT).

= When J1=J2=...=JM=1, Iat,;I=Ia2.;I=...=IaM,;I=l and the ith second M-point transform is a DFT, the MCSS transmitter is similar to the one in the issued patent: "Method and Apparatus for Multiple Access between Transceivers in Wireless Communications using OFDM Spread Spectrum," by M.
Fattouche and H. Zaghloul, US Patent # 5,282,222, Jan. 25 1994.
= The generated spread spectrum codes using = J1=J2=...=JM=1, = Ia1,;I=1a2,;I=...=IaM,;1=l, = the ith second M-point transform as a DFT, and = the channel encoder as a Reed-Solomon encoder without an interleaver are referred to as the 'Wi-LAN codes Type II'.
= Another preferred embodiment of the ith second M-point transform (902) is a Circular FIR (CFIR) filter of length M coefficients which performs an M-point circular convolution between each block of M modulated symbols and its own coefficients. In this case, a preferred embodiment of the M-point transform (1801) is also a CFIR filter of length M coefficients which performs the inverse operation of the spreading CFIR filter by performing an M-point circular convolution between each block of M multicoded SS
symbols and its own coefficients. When the channel is estimated, the despreading CFIR filter can also invert the effects of the channel using either 1. a linear algorithm such as Zero Forcing Equalization (ZFE) and Minimum Mean Square Equalization (MMSE); or 2. a nonlinear algorithm such as Decision Feedback Equalization (DFE) and Maximum Likelihood (ML).
The effect of a nonideal frequency-selective communication channel is to cause the multicodes to loose their orthogonality at the receiver. In the case when ZFE is employed, the CFIR filter acts as a decorrelating filter which decorrelates the M multicoded symbols from one another at the receiver thereby forcing the symbols to be orthogonal.
An advantage of using CFIR filter for spreading and despreading the data symbols is that IF-sampling can be inherently employed in the MCSS
receiver without increasing the complexity of the digital portion of the receiver since interpolation and decimation filters can be included in the CFIR filters.

Preferred Embodiments of the Spreader/Despreader for MCSS Type III:
= In Figure 10, the spreader Type III (1002) performs an invertible randomized spreading of the stream of modula+ed symbols which carry either digital information or analog information, and in Figure 19 the despreader Type I(1902) performs a reverse operation to the spreader Type III (1002) within the limits of available precision (i.e. with some level of quantization noise).
= In Figure 10, the spreader Type III (1002) performs an invertible randomized spreading of the stream of modulated symbols, and in Figure 19 the despreader Type III (1902) performs a reverse operation to the spreader Type III (1002) while taking into account the effects of the communications channel such as noise, distortion and interference. As mentioned above, the effects of the channel are sometimes unknown to the receiver (e.g. over selective fading channels which cause intersymbol interference). In such cases, the channel has to be estimated using for example a pilot signal known to the receiver as in "MultiCode Direct Sequence Spread Spectrum," by M. Fattouche and H. Zaghloul, US Patent #5,555,268 Sept. 1996.
= A preferred randomizer (1101) in Figure 11 is a trivial one with no effect on the modulated symbols.
= Another preferred randomizer (1101) is one where the preset values out of the randomizing lookup table (1202) :{.... ak_l,ak,ak+l, } have amplitudes which are equal to unity.
= In Figure 13, a preferred filter is a Finite Impulse Response (FIR) filter with the coefficients obtained as the values of a polyphase code.
= In Figure 13, a preferred filter is an FIR filter with the coefficients obtained as approximations to the values of a polyphase code.
= In Figure 13, a preferred filter is an FIR filter with the following 16 coefficients:
{1,1,1,1, forming its impulse response where j=V--I. The 16 coefficients correspond to the following polyphase code:

e'00(0),ej10(0),e j20(0), e j30(0) ejoe(1) _ile(1) ej20(1),ej36(1), ,C , j0A(2) e.10(2) _j20(2) ej30(2)e C , ,C e e 6(3)'C_je(3)'ej2e(3)'ej3A(3)}
where 0(0)=0, 0(1)=27c/4, 9(2)=4n/4, 8(3 )=6n/4, and j= -\[-1 .
= In Figure 13, another preferred filter is an FIR filter with 64 coefficients corresponding to the following polyphase code:
e'OA(0),ea 1A(0),ej20(0), e j30(0), e' 40(0), e'50(0),e'60(0),ej7A(0) ~09(1) 0(1), ej20(1),ej30(1)'C_j49(1)' ej50(1)'ej69(1)' ej70(1), e,00(2)ej 10(2)ej20(2)'ej30(2)'d 40(2)' d 50(2)'ej69(2)'ej79(2), e'00(3)'ej 10(3),ej20(3), e j30(3), e 40(3) e'50(3),ej68(3),ej79(3) , d 00(4) 10(4)20(4) ej30(4)'d40(4), d 50(4), ej60(4)'ej7e(4)~
, >
ejo9(5) ~ 10(5) ej20(5)ej30(5)id 40(5)' ej50(5)'ej69(5)'ej70(5)' ej09(6)~ej10(6)~ej20(6) ej30(6)~ej40(6), ej50(6)'ej60(6)'ej78(6), doe(7) ~10(7) ej2e(7) C_j30(7)~e40(7) ej50(7)~ej69(7)~e.i70(7)}

where 0(0)=0, 0(1)=27t/8, 0(2)=47r/8, 0(3)=67r/8, 0(4)=8n/8, 0(5)=107/8, 0(6)=12TE/8, 0(7)=147c/8, and j= [--I .
= In general, a preferred filter in Figure 13 with M coefficients corresponding to a polyphase code can be obtained as the concatenation of the rows of an \[M- x-\[M- matrix (assuming -vrM- is an integer) with the coefficient in the ith row and kth column equal to e'(i-1)e(k-1) where 0(k)=2nk/-,[M-, and j=-\[--I.
= Another preferred filter in Figure 13 with M coefficients corresponding to a binary approximation of a polyphase code can be obtained as the concatenation of the rows of an ~M x-N matrix with the coefficient in the ith row and kth column determined as follows:
= when (i-1)0 (k-1) is an integer number of Tz/2, the coefficient is equal to ej(`-1)e(k-1) where 0(k)=2-xk/-,,rM-, otherwise = when (i-1) 6(k-1) is not an integer number of Tc/2, the coefficient is equal to e' "/2 where n is an integer number which minimizes the value:
(nn/2 - (i-1)8(k-l))2.
We refer to the spread spectrum code corresponding to the coefficients of a filter representing a binary approximation of a polyphase code as discussed above as the 'Wi-LAN code Type III' .

For example when M=64, the above procedure produces the following filter coefficients:
{1,1,1,1,1,1,1,1, 1, l,j,j,- l,-1,-j,-j, 1,j,-1,-j,1,j,-1,-j, 1,- l,j,-j,-1, l,-j,j, = A preferred filter in Figure 21 performs a reverse operation to the filter (1301) in Figure 13.
= Another preferred filter in Figure 21 performs a matching filtering operation to the filter (1301) in Figure 13.
= A preferred de-randomizer (2102) in Figure 21 is one where the preset values out of the de-randomizing lookup table (2302) :{...,bk_,,bk,bk+l.===}
performs a reverse operation to the randomizer (1101) in Figure 11.
= Another preferred de-randomizer (2102) in Figure 21 is one where the preset values out of the de-randomizing lookup table (2302):
{...,bk_t,bk,bk+i,... } are equal to the reciprocal of the preset values out of the randomizing lookup table (1202) in Figure 12, i.e. bk= 1/ak for all values of k.
= A preferred diversity technique for MCSS Type III is shown in Figure 24 where we have L branches with one de-ramper (2401) per branch. Each de-ramper linearly de-ramps the received signal using a linearly deramping carrier frequency of fixed slope and unique intercept. Each intercept corresponds to a unique time of arrival of the different multipath components. The outputs of the L de-rampers are then combined in the combiner (2402) using any appropriate combining technique such as: co-phasing combining, maximum ratio combining, selection combining, equal gain combining, etc. The output of the combiner is then despread using the de-spreader (2403) and input into the channel decoder/demodulator (2404) to generate the estimated data symbols.
= A preferred value for fo in Figure 14 is 1/(2tiMTc) where ti is the relative delay between the first arriving radio signal and the second arriving radio signal at the receiver, M is the number of coefficients in the spreading filter (1301) in Figure 13 and T, is the duration of one chip (or equivalently it is the unit delay in the spreading filter (1301)). In other words, the symbol rate at both the input and the output of the spreading filter (1301) is 1/ Tc.

A person skilled in the art could make immaterial modifications to the invention described in this patent document without departing from the essence of the invention that is intended to be covered by the scope of the claims that follow.

Claims (21)

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

1. A transceiver for transmitting a first stream of data symbols, the transceiver comprising:
a first converter for converting the first stream of data symbols into plural sets of B data symbols each;
a channel encoder/modulator for encoding plural sets of B data symbols into plural sets of J modulated symbols;
a spreader for spreading plural sets of J modulated symbols into plural sets of N
spread spectrum symbols of length M chips each, wherein the spreader includes a second converter for converting each one of the plural sets of J modulated symbols into N
subsets of modulated symbols; and a set of N computing means for operating on each one of the N subsets of modulated symbols to produce the N spread spectrum symbols whereby the ith set of computing means operates on the ith subset of modulated symbols to produce the ith spread spectrum symbol;
and M combiners for combining each set of the plural sets of N spread spectrum symbols into M multicoded spread spectrum symbols;
and a third converter for converting the plural sets of M multicoded spread spectrum symbols into a first stream of multicoded spread spectrum symbols for transmission.

2. The transceiver of claim 1 in which the ith computing means includes:
a source of available spread spectrum codes; and a modulator to choose for each ith subset of modulated symbols one spread spectrum code from the source of available spread spectrum codes to become the spread spectrum code representing the ith subset of modulated symbols, thereby spreading each subset of modulated symbols over a separate spread spectrum code.

3. The transceiver of claim 2 in which the spread spectrum codes are generated by operation of a non-trivial transform on a sequence of input signals.

4. The transceiver of claim 3 in which the non-trivial transform comprises a Walsh transform and a randomizing transform.

5. A transceiver for transmitting a first stream of data symbols, the transceiver comprising:
a first converter for converting the first stream of data symbols into plural sets of B data symbols each;
a channel encoder/modulator for encoding plural sets of B data symbols into plural sets of J modulated symbols;
a spreader for spreading plural sets of J modulated symbols into plural sets of M
multicoded spread spectrum symbols, wherein the spreader includes;
a second converter for converting each one of the plural sets of J modulated symbols into M subsets of modulated symbols; and a transformer for operating on the M subsets of modulated symbols to generate M
multicoded spread spectrum symbols as output, the M multicoded spread spectrum symbols corresponding to spreading each subset of modulated symbols over a separate spread spectrum code;
and a third converter for converting the plural sets of M multicoded spread spectrum symbols into a first stream of multicoded spread spectrum symbols for transmission.

6. The transceiver of claim 5 in which the transformer applies a first transform corresponding to a randomizing transform of the M data symbols.

7. The transceiver of claim 6 in which the first transform is followed by a second transform corresponding to a Fourier transform.

8. The transceiver of claim 5 in which the channel encoder/modulator comprises a Reed-Solomon encoder.

9. The transceiver of claim 5, in which the J modulated symbols comprises pilot symbols.

10. The transceiver of claim 5 in which plural sets of M multicoded spread spectrum symbols correspond to plural sets of M pilot symbols.

11. The transceiver of claim 6 in which the first transform is followed by a second transform corresponding to a circular finite impulse response filter.

12. A transceiver for receiving a first stream of data symbols, the transceiver comprising:
means for receiving a sequence of multicoded spread spectrum symbols, the multicoded spread spectrum symbols having been generated by spreading a first stream of data symbols;
a first converter for converting the received stream of multicoded spread spectrum symbols into plural sets of M multicoded spread spectrum symbols each;
a despreader for despreading plural sets of M multicoded spread spectrum symbols to produce plural sets of J despread symbols, wherein the despreader includes a set of N computing means to operate on the set of M multicoded spread spectrum symbols, from the received sequence of multicoded spread spectrum symbols, to generate a set of N computed values; and a detector for operating on the set of N
computed values to. generate a set of J despread symbols;
a channel decoder/demodulator for decoding plural sets of J despread symbols into plural sets of B estimated data symbols of the second stream of data symbols; and a second converter for converting the plural sets of the B estimated data symbols into a stream of estimated data symbols of the first stream of data symbols.

13. The transceiver of claim 12 in which the ith computing means comprises: a set of partial correlators for partially correlating the set of M multicoded spread spectrum symbol with corresponding parts of each spread spectrum code from the ith source of available spread spectrum codes to generate partially correlated values; and a sub-detector for operating on the partially correlated values to produce the ith computed value.

14. A transceiver for receiving a first stream of data symbols, the transceiver comprising:
means for receiving a sequence of multicoded spread spectrum symbols, the multicoded spread spectrum symbols having been generated by spreading the first stream of data symbols;
a first converter for converting the received stream of multicoded spread spectrum symbols into plural sets of M multicoded spread spectrum symbols each;
a despreader for despreading plural sets of M multicoded spread spectrum symbols to produce plural sets of J despread symbols, wherein the despreader includes a non trivial inverse transformer for inverse transforming M multicoded spread spectrum symbol from the received sequence of multicoded spread spectrum symbols into M

transformed symbols; and a detector for operating on the M transformed symbols to produce J despread symbols;
a channel decoder/demodulator for decoding plural sets of J despread symbols into plural sets of B estimated data symbols of the second stream of data symbols; and a second converter for converting the plural sets of the B estimated data symbols into a stream of estimated data symbols of the second stream of data symbols.

15. The transceiver of claim 14 in which the non-trivial inverse transformer inverse transforms M multicoded spread spectrum symbol from the received sequence of multicoded spread spectrum symbols into M transformed symbols; and inverts the effects of the channel using pilots relying on a linear algorithm.

16. The transceiver of claim 14 in which the non trivial inverse transformer corresponds to a circular finite impulse response filter.

17. The transceiver of claim 1 in which the spreader is based on Wi-LAN codes Type I.

18. The transceiver of claim 5 in which the spreader is based on Wi-LAN codes Type II.

19. The transceiver of claim 15 in which the pilots are selected from the group of pilot symbols and pilot frames.

20. The transceiver of claim 3 in which the non-trivial transform comprises a Fourier transform and a randomizing transform.

21. The transceiver of claim 14 in which the non-trivial inverse transformer inverse transforms M multicoded spread spectrum symbol from the received sequence of multicoded spread spectrum symbols into M transformed symbols; and inverts the effects of the channel using pilots relying on a non-linear algorithm.