BRPI0015969B1 - Method for deriving at least three audio signals from two input audio signals - Google Patents

Abstract

Various equivalent adaptive audio matrix arrangements are disclosed, each of which includes a feedback-derived control system that automatically causes the cancellation of undesired matrix crosstalk components in the matrix output. Each adaptive audio matrix arrangement includes a passive matrix that produces a pair of passive matrix signals in response to two input signals. A feedback-derived control system operates on each pair of passive matrix signals, urging the magnitudes of pairs of intermediate signals toward equality. Each control system includes variable gain elements and a feedback and comparison arrangement generating a pair of control signals for controlling the variable gain elements. Additional control signals may be derived from the two pairs of control signals for use in obtaining more than four output signals from the adaptive matrix.

Description

Patent Descriptive Report for "METHOD FOR DERIVING AT LEAST THREE AUDIO SIGNALS FROM TWO INPUT AUDIO SIGNALS".

TECHNICAL FIELD The invention relates to the processing of audio signals. In particular, the invention relates to "multidirectional" (or "multichannel") audio decoding using an "a-daptative" (or "active") audio matrix method, which derives three streams (or " signals "or" channels ") of audio signals from a pair of streams (or" signals "or" channels ") of input audio signals. The invention is useful for audio signal retrieval, wherein each signal is associated with a direction and has been combined into not many signals by a coding matrix. While the invention is described in terms of such deliberate matrix coding, it should be understood that the invention need not be used with any particular matrix coding and is also useful for generating pleasant directional effects of originally recorded material for two channel playback.

BACKGROUND OF THE INVENTION Coding and decoding of audio arrays are well known in the prior art. For example, in so-called "4-2-4" audio matrix coding and decoding, four source signals, typically associated with four cardinal directions (such as, for example, left, center, right and surrounding, or front left, front right, rear left and rear right) are amplitude - phase encoded in two signals. The two signals are transmitted or stored and then decoded by an amplitude - phase matrix decoder to retrieve the approximations of the four original source signals. Decoded signals are approximations, because matrix decoders suffer from the well-known disadvantage of crosstalk between decoded audio signals. Ideally, decoded signals should be identical to source signals, with infinite separation between signals. However, the inherent crosstalk in matrix decooders results in a separation of only 3 dB between signals associated with adjacent directions. An audio matrix, in which the matrix characteristics do not vary, is known in the art as a "passive" matrix. To overcome the crosstalk problem in matrix decoders, it is known in the prior art to adaptively vary the characteristics of the decoding matrix to improve the separation between the decoded signals and to approximate the source signals more closely. A well known example of such an active matrix decoder is the Dolby Pro Logic decoder, described in U.S. Patent 4,799,260, the patent of which is incorporated herein by reference. Patent 260 cites various prior art patents, many of which describe various other types of adaptive matrix decoders. Other prior art patents include the same patents, including U.S. patents 5,625,696, 5,644,640, 5,504,819, 5,428,687 and 5,172,415. Each of these patents is also incorporated herein entirely by reference. Although adaptive matrix decoders are intended to reduce crosstalk in reproduced signals and more closely replicate source signals, the prior art has done so in many ways, many of which are complex and cumbersome, which fail to recognize the desirable relationships between intermediate signals, which can be used to simplify the decoder and improve decoder accuracy. Accordingly, the present invention is directed to methods and apparatus which have hitherto recognized and employed unappreciated relationships between intermediate signals in adaptive array decoders. Exploitation of these relationships enables unwanted crosstalk components to be easily canceled, particularly by using self-cancellation arrangements using negative feedback.

According to a first aspect of the invention, the invention is a method for deriving at least three output audio signals from input audio signals, in which four audio signals are derived from the two audio signals. input by a passive matrix, which produces two pairs of audio signals in response to the two audio signals: a first pair of derived audio signals representing directions extending on a first axis (such as "left" and "right" signals). ") and a second pair of derived audio signals representing directions extending on a second axis (such as" central "and" surrounding "signals), said first and second axes being substantially mutually orthogonal to each other. Each pair of the derived audio signals is processed to produce the respective first and second pairs (the left / right and center / surrounding pairs, respectively) of intermediate audio signals, so that the relative amplitudes of the audio signals on each pair of intermediate audio signals are driven towards equality. A first output signal (such as the left output signal Lsaida)> representing a first direction extending on the axis of the pair of derived audio signals (the left / right pair) from which the first pair (the left / right pair) of intermediate signals is produced, is produced by at least the same polarity combination of at least one component of each of the second pair (the center / surrounding pair) of intermediate audio signals. A second output signal (such as the right output signal Output) representing a second direction extending on the axis of the pair of derived audio signals (the left / right pair), of which the first pair (the left / right pair) of intermediate signals is produced, is produced by at least the opposite polarity combination of at least one component of each of the second pair (the center / surrounding pair) of the intermediate audio signals. A third output signal (such as the center output signal Output or the surrounding output signal Output), representing a first direction extending on the pair axis (the center / surrounding pair) of derived audio signals, of which the second pair (the center / surrounding pair) of the intermediate signals is produced, it is produced by at least the same or opposite polarity combination of at least one component of each of the first pair (the left / right pair) of signals audio speakers. Optionally, a fourth output signal (such as the surrounding output signal Exit if the third output signal is the input output signal Csaic, or Output if the third output signal is Exit) representing a second direction if extending on the axis of the (center / surrounding) pair of derived audio signals, from which the second (center / surrounding) pair of intermediate signals is produced, is produced at least by combining, with the opposite polarity, if the third output signal is produced by combining the same polarity, or by combining the same polarity, if the third output signal is produced by combining the opposite polarity, at least one component of each of said pair (the left / right pair) of intermediate audio signals.

The relations so far not valued between the decoded signals is that by pushing towards the equalities of the intermediate audio signals in each pair of intermediate audio signals, the unwanted crosstalk components in the decoded output signals are eliminated substantially. The principle does not require complete equality to achieve substantial cancellation of crosstalk. Such processing is readily and preferably implemented by use of negative feedback arrangements which act to cause automatic cancellation of unwanted crosstalk components. The invention includes embodiments having equivalent topologies. In each embodiment, as described above, the intermediate signals are derived from a passive matrix operating in pair of input signals, and these intermediate signals are driven equally. In embodiments incorporating a first topology, an intermediate signal cancellation component is combined with passive matrix signals (from the passive matrix operating on input or other signals) to produce output signals. In one embodiment employing a second topology, pairs of intermediate signals are combined into output signals. Other aspects of the present invention include derivation of additional control signals to produce additional output signals. It is a primary object of the invention to achieve a measurable and noticeably high degree of crosstalk cancellation under a wide variety of input signal conditions, using a circuit set without any special requirements for accuracy, and not requiring any unusual complexity in the control path, both of which are found in the prior art. It is another object of the invention to achieve such high performance with a simpler or lower cost circuit set than prior art circuit sets.

BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 is a functional and schematic diagram of a prior art passive decoding matrix useful in understanding the present invention. Figure 2 is a functional and schematic diagram of a prior art active matrix decoder useful in understanding the present invention, wherein the variably scaled versions of the passive matrix outputs are summed with the unchanged passive matrix outputs. in linear combiners. Figure 3 is a schematic and functional diagram of a feedback derived control system according to the present invention for the right and left VCAs and the sum and difference VCAs of Figure 2 and for the VCAs in other embodiments. of the present invention. Figure 4 is a schematic and functional diagram showing an arrangement according to the present invention equivalent to the combination of Figures 2 and 3, in which the output combiners generate the passive matrix output signal components in response to the input signals Lt and Rt instead of receiving them from the passive matrix from which the cancellation components are derived. Figure 5 is a schematic and functional diagram according to the present invention showing an arrangement equivalent to the combination of Figures 2 and 3 and Figure 4. In the configuration of Figure 5, the signals to be kept equal are the signals applied to the output bypass combiners and the feedback circuits for VCA control, the outputs of the feedback circuits including the passive matrix components. Figure 6 is a schematic and functional diagram according to the present invention showing an arrangement equivalent to the combination of Figures 2 and 3, Figure 4 and Figure 5, wherein the gain of the variable gain circuit (1 -g), provided by a VCA and a subtractor, is replaced by a VCA whose gain varies in the opposite direction of the VCAs in the VCA and subtractor configurations. In this embodiment, the passive matrix components are implicit. In other embodiments, the passive matrix components are explicit. [0018] Figure 7 is an idealized graph plotting the left and right VCA gains gi and gr of the Lt / Rt re-limitation derived control system (vertical axis) against the gradual compounding angle α (horizontal axis). Figure 8 is an idealized graph plotting the gains of the sum and difference VCAs, gc and gs, of the sum / difference feedback derived control system (vertical axis) against the gradual compounding angle α (horizontal axis). ). Figure 9 is an idealized graph plotting the left / right control and inverted sum / difference control voltages for a gradation in which the maximum and minimum values of the control signals are ± 15 volts (vertical axis) against the angle. of gradual composition α (horizontal axis). Figure 10 is an idealized graph plotting the lowest of the curves in Figure 9 (vertical axis) against the gradual composition angle α (horizontal axis). Figure 11 is an idealized graph plotting the lowest of the curves in Figure 9 (vertical axis) against the gradual compounding angle α (horizontal axis) for the case where the sum / difference voltage has been scaled by 0.8 before taking the lowest of the curves. Figure 12 is an idealized graph plotting the gains of the rear left and rear right VCAs, gib and grt>, from the rear left / rear right feedback derived control system (vertical axis) against the gradual compounding angle α (horizontal axis). Figure 13 is a functional and schematic diagram of a portion of an active matrix decoder according to the present invention in which six outputs are obtained. Figure 14 is a functional and schematic diagram showing the derivation of six cancellation signals for use in a six output active matrix decoder, such as that of Figure 13. Figure 15 is a circuit diagram schematic showing a practical circuit incorporating aspects of the present invention.

Best Mode for Conducting the Invention A passive decoding matrix is shown, functionally and schematically, in Figure 1.  The following equations refer to the outputs for inputs, Lt and Rt ("total left" and "total right"): (The symbol "*" in these and other equations throughout this document indicates multiplication. ) [0028] The center output is the sum of the inputs, and the surrounding output is the difference between the inputs.  Both have, in addition, a gradation, this gradation being arbitrary, and is selected to be 1/2, for ease of explanation.  Other gradation values are possible.  The Csaida output is obtained by applying Lt and Rt with a scale factor of +1/2 on a linear combiner 2.  The SSaida output is obtained by applying L, and Rt with scale factors of +1/2 and -1/2, respectively, in a linear combiner 4.  The passive matrix of Figure 1 thus produces two pairs of audio signals, the first pair being Output and Output.  The second pair is CSA. da and Ssaida · In this example, the cardinal directions of the passive matrix are designated "left", "central", "right" and "surrounding".  Adjacent cardinal directions extend on mutually orthogonal axes, so that for these direction marks, left is adjacent to center and surrounding, surrounding is adjacent to left and right, etc.  It is to be understood that the invention is applicable to any orthogonal 2: 4 decoding matrix.  A passive matrix decoder derives n audio signals from m audio signals, where n is greater than m, according to an invariable ratio (for example, in Figure 1, Output is always 1/2 * ( Bypass + Output)) · In contrast, an active matrix decoder derives n audio signals according to a variable ratio.  One way to configure an active matrix decoder is to combine the signal-dependent signal components with the output signals of a passive matrix.  For example, as shown functionally and schematically in Figure 2, four voltage controlled amplifiers (VCAs) 6, 8, 10 and 12, transmitting variably scaled versions of the passive matrix outputs, are summed with the unchanged passive matrix outputs (i.e. that is, the two inputs themselves together with the two outputs of combiner 2 and 4) into linear combiner 14, 16, 18 and 20.  Because VCAs have their inputs derived from the left, right, center and surrounding outputs of the passive matrix, respectively, their gains can be projected gi, gr, gc and gs (all positive).  The output signals from VCAs constitute cancellation signals and are combined with passively derived outputs having crosstalk in the directions from which the cancellation signals are derived, to improve the directional performance of the crosstalk-eliminating matrix decoder.  Note that, in the arrangement of Figure 2, the passive matrix paths are still present.  Each output is the combination of the respective passive matrix output plus the output of the two VCAs.  VCA outputs are selected and scaled to provide the desired crosstalk cancellation for the respective passive matrix output, whereas crosstalk components occur at outputs representing adjacent cardinal directions.  For example, a central signal has crosstalk in passively decoded left and right signals, and a surrounding signal has crosstalk in passively decoded left and right signals.  Consequently, the left signal output must be combined with the cancellation signal components derived from the passively decoded center and surrounding signals, and similarly for the other four outputs.  The manner in which the signals are scaled, polarized and combined in Figure 2 provides the desired crosstalk elimination.  By varying the respective VCA gain in the range from zero to one (for the scaling example of Figure 2), unwanted crosstalk components in passively decoded outputs can be eliminated.  The arrangement of Figure 2 has the following equations: If all VCAs had zero gains, the arrangement would be the same as in the passive matrix.  For any equal values of all VCA gains, the arrangement in Figure 2 is the same as in the passive matrix apart from a constant scaling.  For example, if all VCAs had gains of 0.1: The result is the passive matrix scaled by a factor of 0.9.  Thus it will be apparent that the precise value of quiescent VCA gain described below is not critical.  Consider an example.  For cardinal directions (left, right, center and surrounding) only, the respective inputs are Lt only, Rt only, Lt = Rt (same polarity), and Lt = -Rt (opposite polarity), and the corresponding desired outputs are Exit only, Exit only, Exit only and Exit only.  In each case, ideally, one output should transmit a signal, and the remaining ones should not transmit any.  Upon inspection, it is evident that if VCAs can be controlled, so that the one corresponding to the desired cardinal direction has a gain of 1 and those remaining less than 1, then at all outputs except that desired, the signals of the VCAs will cancel unwanted exits.  As explained above, in the configuration of Figure 2, VCA outputs act to cancel the crosstalk components in the adjacent cardinal directions (for which the passive matrix has crosstalk).  Thus, for example, if both inputs are fed with equal phase signals, so that Rt = Lt = (as it were) 1, and if as a consequence gc = 1 and gi, gr and gs are all zero or close to zero, you can: [0038] The only output is the desired output.  A similar calculation will show that the same applies to the case of a single sign from one of the other three cardinal directions.  Equations 5, 6, 7, and 8 may be equivalent written as follows: In this arrangement, each output is the combination of two signals.  Both Output and Output involve the sum and difference of the input signals and the gains of the sum and difference VCAs (the VCAs whose inputs are derived from the center and surrounding directions, the pair of orthogonal directions to the left and right directions).  Both outputs and outputs involve the actual input signals and the left and right VCA gains (the VCAs whose respective inputs are derived from the left and right directions, the pair of orthogonal directions to the center and surrounding directions).  Consider a non-cardinal direction, whose Rt is fed the same signal as Lt, with the same polarity but attenuated.  This condition represents a signal placed somewhere between the left and center cardinal directions, and should therefore transmit the Outputs and Outputs with little or no Output and Output.  For Output and Output, this zero output can be obtained if both terms are equal in magnitude but of opposite polarity.  For Output, the ratio for this cancellation is [0044] quantity of [1/2 * (Lt + Rt) * (1 - gc)] = quantity of [1/2 * (Lt - Rt) * (1 - gs)] (Eq.  13) For Output, the corresponding ratio is magnitude of [1/2 * Lt * (1 - g,)] = quantity of [1/2 * Rt * (1 - gr)] (Eq.  14) A consideration of a gradually (or simply positioned) composite signal between any two adjacent cardinal directions will reveal the same two relationships.  In other words, when the input signals represent a sound composed gradually between any two adjacent outputs, these magnitude ratios will ensure that the sound emerges from the outputs corresponding to those two adjacent cardinal directions, and that the other two outputs transmit nothing.  To obtain this result substantially, the magnitudes of the two terms in each of equations 9-12 must be driven into equality.  This can be achieved by trying to keep the relative quantities of the two signal pairs within the active matrix equal: [(Lt + Rt) * (1 - gc)] = [(Lt - Rt) * (1 - gs)] (Eq.  15) and magnitude of [Lt * (1 - gi)] = quantity of [Rt * (1 - gr) j (Eq.  16) The desired relationships shown in equations 15 and 16 are the same as those in equations 13 and 14, but with the scheduling omitted.  The polarity with which the signals are combined and their scaling can be taken care of when the respective outputs are obtained with the combiner 14, 16, 18 and 20 of Figure 2.  [0048] The invention is based on the discovery of such relationships of equal magnitude quantities not yet appreciated and preferably, as described below, the use of self-action feedback control to maintain these relationships.  From the above discussion concerning the cancellation of unwanted crosstalk signal components and the requirements for cardinal directions, it can be deduced that for the scaling used in this explanation, the maximum gain for a VCA must be unitary.  Under quiescent undefined or "runaway" conditions, VCAs must adopt a small gain, effectively providing the passive matrix.  When a pair's VCA gain needs to increase from its quiescent value towards unity, the other pair's gain may remain at quiescent gain, or it may move in the opposite direction.  A convenient and practical relationship is to keep the product of the pair's gains constant.  Using analog VCAs, whose gain in dB is a linear function of their control voltages, this happens automatically if a control voltage is applied equally (but with an effective opposite polarity) to both of a pair.  Another alternative is to keep the sum of the pair's gains constant.  Of course, the invention may be implemented digitally or in software instead by using analog components.  Thus, for example if the quiescent gain is 1 / a, a practical relationship between the two pair gains should be their product, so that a typical value for "a" may be in the range from 10 to 20.  Figure 3 shows, functionally and schematically, a feedback derived control system for the left and right VCAs (6 and 12, respectively) of Figure 2.  It receives input signals Lt and Rt, processes them to derive intermediate signals Lt * (1 -gi) and Rt * (1 - gr), compares the magnitude of intermediate signals, and generates an error signal in response to any in magnitude, the error signal causing the VCAs to reduce the difference in magnitude.  One way of obtaining this result is to rectify the intermediate signals to derive their quantities and apply the two signals of magnitude to a comparator whose output controls the gains of VCAs with a polarity such that, for example, an increase in the Lt signal increases. gi and diminishes gr.  Circuit values (or their equivalents in digital or software implementations) are selected such that when the comparator output is zero, the quiescent amplifier gain is less than unity (for example, 1 / a).  In the analog domain, a practical way to implement the comparison function is to convert the two quantities into the logarithmic domain, so that the comparator subtracts them rather than determines their relationship.  Many analog VCAs have gains proportional to a control signal exponent so that they inherently and conveniently take the antilogarithm of the logarithmic comparator control outputs.  In contrast, however, if implemented digitally, it may be more convenient to divide the two quantities and use the resulting ones as the direct multipliers or splitters for VCA functions.  More specifically, as shown in Figure 3, the Lt input is applied to the "left" VCA 6 and to a linear combiner input 22, where it is applied with a +1 scaling.  The left VCA output 6 is applied to combiner 22 with a -1 scaling (thereby forming a subtractor), and the combiner 22 output is applied to a full wave rectifier 24.  The input Rt is applied to the right VCA 12 and to an input of a linear combiner 26 where it is applied with a +1 scaling.  The left VCA output 12 is applied to combiner 26 with a -1 scaling (thereby forming a subtractor), and the combiner output 26 is applied to a full wave rectifier 28.  Rectifier outputs 24 and 28 are applied, respectively, to the non-inverted and inverted inputs of an operational amplifier 30 operating as a differential amplifier.  The output of amplifier 30 provides a control signal in the nature of an error signal, which is applied without inversion to the gain control input of VCA 6, and with polarity inversion to the gain control input of VCA 12.  The error signal indicates that the two signals, whose quantities are to be equalized, differ in magnitude.  This error signal is used to "govern" the VCAs in the correct direction to reduce the difference in magnitude of intermediate signals.  The outputs for combos 16 and 18 are taken from the outputs of VCA 6 and VCA 12.  Thus, only one component of each intermediate signal is applied to the output combiner, ie Ltgr and -Rtgi.  For steady-state signal conditions, the difference in magnitude may be reduced to a negligible amount by providing sufficient loop gain.  However, it is not necessary to reduce the differences in magnitude to zero or a negligible amount to achieve substantial crosstalk cancellation.  For example, a loop gain sufficient to reduce the difference in dB by a factor of 10 theoretically results in a worst case crosstalk better than 30 dB below.  For dynamic conditions, the time constants in the feedback control arrangement should be selected to drive the equalities in a way that is essentially inaudible for most signal conditions.  The details of selecting time constants in the various embodiments described are beyond the scope of the invention.  Preferably, the circuit parameters are selected to provide about 20 dB of negative feedback and so that the gains of the VCAs cannot increase above the unit.  VCA gains may range from some small value (eg, 1 / a2, much smaller than unit) upwards but not exceeding unit for the scaling examples described here in conjunction with the arrangements of the Figures. 2, 4 and 5.  Due to the negative feedback, the arrangement of Figure 3 will act to keep the signals input to the rectifiers approximately equal.  Since exact gains are not critical when they are small, any other relationship that forces one pair's gain to a small value when the other increases towards unity will yield similar acceptable results.  The feedback derived control system for the central and surrounding VCAs (8 and 10, respectively) of Figure 2 is substantially identical to the arrangement of Figure 3 as described, but not receiving Lt and Rt, but the sum and the their differences and applying their outputs from VCA 6 and VCA 12 (constituting a component of their intermediate signal) in combiner 14 and 20.  Thus, a high degree of crosstalk cancellation can be achieved under a wide variety of input signal conditions by using a circuit set without any special requirements for accuracy while employing a simple control route, which is integrated into the signal path.  The feedback derived control system operates to process passive matrix audio signal pairs, so that the relative amplitude quantities of the intermediate audio signals in each pair of intermediate audio signals are driven equally.  [0060] The feedback derived control system shown in Figure 3 controls the gains of the two VCAs 6 and 12 inversely to drive the inputs to rectifiers 24 and 28 in the same direction.  The degree to which these two terms are driven into equality depends on the characteristics of the rectifiers, the comparator 30 following them, and the gain / control ratios of the VCAs.  The higher the loop gain, the closer the equality, but an equality push will occur regardless of the characteristics of these elements (provided, of course, that the polarities of the signals are such as to reduce level differences).  In practice, the comparator may not have infinite gain, but can be understood as a finite gain subtractor.  If the rectifiers are linear, that is, if their outputs are directly proportional to the input quantities, the comparator or subtractor output is a function of the signal voltage or current difference.  If, instead, the rectifiers respond to the logic of their input quantities, that is, at the level expressed in dB, a subtraction made at the comparator input is equivalent to taking the ratio of the input levels.  This is beneficial in that the result is then independent of the absolute signal level, but depends only on the difference in signal expressed in dB.  Considering the levels of the source signals expressed in dB to more closely reflect human perception, this means that other things being equal to the loop gain are independent of the level of sound intensity, and therefore that the degree The impulse thrust towards equality is also independent of the intensity level of the absolute sound.  At the same very low level, of course, the lo-garitic rectifiers will stop operating precisely and thus will be an input limit below which the drive for equality will stop.  However, the result is that control can be maintained over a range of 70 dB or greater, without the need for extraordinarily high loop gains for high input signal levels, with potential problems with loop stability resulting.  Similarly, VCAs 6 and 12 may have gains, which are, directly or inversely, proportional to their control voltages (ie, multipliers or dividers).  This would have the effect that when the gains are very small, small absolute variations in control voltage would cause large variations in the gain expressed in dB.  For example, consider a VCA with a maximum unit gain as required in the feedback derived control system configuration, and a control voltage Vc ranging from, for example, 0 to 10 volts, so that the gain can be expressed as A = 0.1 * Vc.  When you are near the maximum, a variation of 100 mV (millivolts) of, for example, 9. 900 to 10. 000 mV transmits a gain variation of 20 * log (10. 000/9. 900) or about 0.09 dB.  When Vc is much smaller, a 100 mV range of, for example, 100 to 200 mV, transmits a gain gain of 20 * log (200/100) or 6 dB.  Therefore, the effective loop gain and therefore the response speed would vary greatly depending on whether the control signal is large or small.  Again, there may be problems with loop stability.  [0063] This problem can be eliminated by using VCAs whose gain in dB is proportional to the control voltage, or expressed differently, whose voltage or current gain is dependent on the control voltage exponent or antilogarithm.  A small variation in control voltage, such as 100 mV, will then generate the same variation in dB when the control voltage is within its range.  These devices are readily available as analog ICs, and the feature, or an approximation of it, is easily obtainable in digital implementations.  The preferred embodiment therefore employs logarithmic rectifiers and exponentially controlled variable gain amplification, imparting an almost uniform equality thrust (considered in dB) over a wide range of input levels and ratios of the two. input signals.  Once in human hearing, the perception of direction is not often constant, it is desirable to apply some frequency weighting to the signals input to the rectifiers so as to emphasize those frequencies that contribute most to the human sense of direction and not emphasize those that can lead in the wrong direction.  As a result, in practical embodiments, rectifiers 24 and 28 in Figure 3 are preceded by empirically derived filters, providing a response that attenuates low frequencies and very high frequencies, and provides a soft boost response across the middle of the audible range.  Note that these filters do not alter the frequency response of the output signals, they merely change the control signals and VCA gains in feedback derived control systems.  An arrangement equivalent to the combination of Figures 2 and 3 is shown functionally and schematically in Figure 4.  It differs from the combinations of Figures 2 and 3 in that the output combiners generate passive matrix output signal components in response to the Lt and Rt input signals rather than receiving them from the passive matrix from which they are derived. the cancellation components.  The arrangement provides the same results as the combination of Figures 2 and 3, provided that the sum coefficients are essentially the same in the passive matrices.  Figure 4 incorporates the feedback arrangements described in conjunction with Figure 3.  More specifically, in Figure 4, the Lt and Rt inputs are first applied to a passive matrix, which includes combiners 2 and 4, as in the passive matrix configuration of Figure 1.  The Lt input, which is also the "left" output of the passive matrix, is applied to the "left" VCA 32 and an input of a linear combiner 34 with +1 scaling.  The left VCA 32 output is applied to a combiner 34 with a stagger of -1 (thereby forming a subtractor).  Input Rt, which is also the "right" output of the passive matrix, is applied to the "right" VCA 44 and an input of a linear combiner 46 with a +1 scaling.  The right VCA output 44 is applied to a combiner 46 with a stagger of -1 (thereby forming a subtractor).  The outputs of the combiners 34 and 46 are the signals Lt * (1 - gi) and Rt * (1 - gr), respectively, and it is desired to keep the magnitude of these signals equal or to push them towards equality.  To obtain this result, these signals are preferably applied to a feedback circuit as shown in Figure 3 and described in conjunction therewith.  The feedback circuit then controls the gain of VCAs 32 and 44.  In addition, further referring to Figure 4, the "central" passive matrix output of combiner 2 is applied to the "central" VCA 36 and an input of a linear combiner 38 with a +1 scaling.  The output of the central VCA 36 is applied to combiner 38 with a scaling of -1 (thereby forming a subtractor).  The "surrounding" output of the passive matrix of combiner 4 is applied to the "surrounding" VCA 40 and an input of a linear combiner 42 with a +1 scaling.  The output of the surrounding VCA 40 is applied to combiner 42 with a stagger of -1 (thereby forming a subtractor).  The outputs of combiner 38 and 42 are the signals [1/2 * (Lt + Rt) * (1 - gc)] and [1/2 * (Lt - Rt) * (1 - gs)], respectively, and you want keep the greatness of these signs equal, or push them towards equality.  To obtain this result, these signals are preferably applied to a feedback circuit as shown in Figure 3 and described in conjunction therewith.  The feedback circuit then controls the gain of VCAs 38 and 42.  The Output, Output, Output and Output output signals are produced by the combiner 48, 50, 52 and 54.  Each combiner receives the output of two VCAs (the outputs of the VCAs constituting a component of the intermediate signals, the quantities of which are sought to be kept equal), to provide cancellation signal components and each or both input signals to provide components. of the passive matrix signals.  More specifically, the input signal Lt is applied with a +1 scaling on the Output 48 combiner, a +1/2 scaling on the Output 50 combiner, and a +1/2 scaling on the Output 52 combiner. .  The input signal Rt is applied with a +1 scaling on the Output 54 combinator, a +1/2 scaling on the Output 50 combinator, and a -1/2 scaling on the 52 Output combiner.  The left VCA 32 output is applied with a -1/2 scaling on Output 50 combiner, and also with a -1/2 scaling on Output 52 combiner.  The right VCA output 44 is applied with a -1/2 scaling on the Output 50 combiner, and with a +1/2 scaling on the Output 52 combiner.  The output of the central VCA 36 is applied with a -1 scaling on the Output 48 combinator, and with a -1 scaling on the Output 54 combinator.  The output of the surrounding VCA 40 is applied with a -1 scaling on the Output 48 AC, and with a +1 scaling on the Output 54 AC.  It will be noted that in several of the figures, for example Figures 2 and 4, it may initially appear that cancellation signals do not oppose passive matrix signals (for example, some of the cancellation signals are applied to same polarity as the passive matrix signal is applied).  However, in operation, when a cancellation signal becomes significant, it will have a polarity that is not opposed to the passive matrix signal.  Another arrangement equivalent to the combination of Figures 2 and 3 and Figure 4 is shown functionally and schematically in Figure 5.  In the configuration of Figure 5, the signals that will be kept the same are the signals applied to the output bypass combiners and the feedback circuits for VCA control.  These signals include the components of the passive matrix output signals.  In contrast, in the arrangement of Figure 4, the signals applied to the feedback loop output combiners are the output signals from the VCAs and exclude the passive matrix components.  Thus, in Figure 4 (and the combinations of Figures 2 and 3), the passive matrix components must be explicitly combined with the feedback circuit outputs, while in Figure 5, the feedback circuit outputs include the components of the passive matrix. passive matrix and are sufficient by themselves.  It will also be noted that in the arrangement of Figure 5, the intermediate signal outputs instead of the VCA outputs (each of which is only a component of the intermediate signal) are applied to the output combiner.  Nevertheless, the configurations of Figures 4 and 5 (together with the combination of Figures 2 and 3) are equivalent, and if the sum coefficients are accurate, the outputs of Figure 5 are the same as those of Figure 4 (and combination of Figures 2 and 3).  In Figure 5, the four intermediate signals, [1/2 * (Lt + Rt) * (1 - gc)], [1/2 * (Lt - Rt) * (1 - gs)], [1 / 2 * Lt * (1 - g,)] and [1/2 * Rt * (1 - gr)], in equations 9, 10, 11 and 12, are obtained by processing the passive matrix outputs and are then added. or subtracted to derive the desired outputs.  The signals are also fed to the rectifiers and comparators of the two feedback circuits, as described above in conjunction with Figure 3, the feedback circuits desirably acting to keep the signal pair quantities equal.  The feedback circuits of FIG. 3, as applied to the configuration of FIG. 5, have their outputs for output combiners taken from outputs of combiners 22 and 26 instead of VCAs 6 and 12.  Referring still to Figure 5, the connections between combiner 2 and 4, VCAs 32, 36, 40 and 44, and combiner 34, 38, 42 and 46 are the same as in the arrangement of Figure 4.  Also, in both arrangements of Figures 4 and 5, the combiner outputs 34, 38, 42 and 46 are preferably applied to the two feedback control circuits (the combiner outputs 34 and 36 to a first of these circuits for generate the control signals for VCAs 32 and 44, and the outputs of combiner 38 and 42 for one second of these circuits to generate the control signals for VCAs 36 and 40).  In Figure 5, the output of combiner 34, the signal Lt * (1 - gi), is applied with a +1 scaling to the Output 58 combiner, and with a +1 scaling to the Output 60 combiner.  The output of combiner 46, the signal Rt * (1 - gr), is applied with a +1 scaling to the Output 58 combiner, and with a -1 scaling to the Output 60 combiner.  The output from combiner 38, the signal [1/2 * (Lt + Rt) * (1 - gc), is applied to Output combiner 56 with a +1 scaling, and Output 62 combiner with a + scaling. 1.  The output of combiner 42, the signal [1/2 * (Lt - Rt) * (1 - gs), is applied to the Output 56 combiner with a +1 scaling, and the Output 62 combiner with a - scaling. 1.  Unlike the prior art adaptive matrix decoders whose control signals are generated from the inputs, the invention preferably employs a closed loop control in which the magnitudes of the signals providing the outputs are measured and feedback to provide the adaptation.  In particular, unlike prior art open loop systems, the desired cancellation of unwanted signals in non-cardinal directions does not depend on an accurate similarity of signal and control path characteristics, and closed loop configurations greatly reduce the need for precision in the circuitry.  Ideally, apart from the shortcomings of the circuits, the "maintenance of equal quantities" configurations of the invention are "perfect" in the sense that any source fed into the Lt and Rt inputs with known relative amplitudes and polarity will produce signals. desired outputs and negligible signals from others.  "Known relative ranges and polarity" means that the inputs Lt and Rt represent a cardinal direction or a position between adjacent cardinal directions.  Considering equations 9, 10, 11 and 12 again, it will be noted that the overall gain of each variable gain circuit incorporating a VCA is a subtractive arrangement in the form (1 - g).  Each VCA gain can range from a small amount up to but not exceeding the unit.  Correspondingly, the gain of the variable gain circuit (1 - g) may range from very close to unity to zero.  Thus, Figure 5 can be redrawn like Figure 6, in which each VCA and associated subtractor have been replaced by only one VCA, whose gain varies in the opposite direction to that of the VCAs in Figure 5.  Thus, each gain of the variable gain circuit (1 - g) (implemented, for example, by a VCA having a gain "g", whose output is subtracted from an output of the passive matrix, as in Figures 2/3, 4 and 5), is replaced by a gain of the corresponding variable gain circuit "h" (implemented, for example, by an autonomous VCA having a gain "h" acting on a passive matrix output).  If the gain characteristics "(1 - g)" are the same as the gain "h", and if the feedback circuits act to maintain equality between the magnitude of the required signal pairs, the configuration of Figure 6 is equivalent to the configuration. of Figure 5, and will transmit the same outputs.  In fact, all the described configurations, the configurations of Figures 2/3, 4, 5 and 6 are equivalent to each other.  Although the configuration in Figure 6 is equivalent and works exactly the same as all previous configurations, note that the passive matrix does not appear explicitly, but is implied.  In the quiescent or ungoverned condition of the previous configurations, the gains of VCAs g fall to small values.  In the configuration of Figure 6, the corresponding ungoverned condition occurs when all VCA h gains increase to or near their maximums.  Referring to Figure 6 more specifically, the "left" output of the passive matrix, which is also the same as the input signal Lt, is applied to a "left" VCA 64 having a gain h |, for produce the intermediate signal Lt * h ,.  The "right" output of the passive matrix, which is also the same as the input signal Rt> is applied to a "right" VCA 70 having an hr gain to produce the intermediate signal Rt * hr.  The "central" passive matrix output of combiner 2 is applied to a "central" VCA 66 having a gain hc, to produce an intermediate signal 1/2 * (Lt + Rt) * hc.  The "surround" output of the passive array of combiner 4 is applied to a "surround" VCA 68 having an hSt gain to produce an intermediate signal 1/2 * (Lt - Rt) * hs.  As explained above, the gains of the VCAs h operate inversely with the gains of the VCAs g, so that the gain characteristics h are the same as the gain characteristics (1 - g).

Control Voltage Generation An analysis of the control signals developed in conjunction with the embodiments described herein is useful in better understanding the present invention and in explaining how the teachings of the present invention can be applied to derive five or more flows. of audio signals, each associated with a direction, of a pair of audio input signal streams. In the following analysis, the results will be illustrated by considering an audio source, which is gradually clockwise composed around the listener in a circle, starting from the back and going via the left, front center , right and back again. The variable α is a measure of the angle (in degrees) of the image relative to a listener, 0 degree being at the rear and 180 degrees at the center front. The quantities of the Lt and R inputs are related to α by the following expressions: There is a one-to-one mapping between the parameter α and the ratio of the quantities and the polarities of the input signals; The use of α leads to a more convenient analysis. When α is 90 degrees, Lt is finite and Rt is zero, that is, only left. When α is 180 degrees, Lt and Rt are equal with the same polarity (center front). When α is 0, Lt and Rt are equal, but with opposite polarities (central rear). As explained below, particular values of interest occur when Lt and Rt differ by 5 dB and have opposite polarity; This produces α values of 31 degrees on each side of zero. In practice, the front left and right speakers are generally placed a little further forward by ± 90 degrees from the center (eg ± 30 to 45 degrees) so that α does not effectively represent the angle to the listener, but is an arbitrary parameter to illustrate gradual composition. The figures to be described are arranged so that the intermediate part of the horizontal axis (a = 180 degrees) represents the central front and the left and right ends (a = 0 and 360) represent the rear. As discussed above in conjunction with the description of Figure 3, a convenient and practical relationship between the gains of a pair of VCAs in a feedback derived control system keeps their product constant. With exponentially powered controlled VCAs so that as the gain of one increases the gain of the other drops, this happens automatically when the same control signal feeds both of the pair, as in the mode of Figure 3. [0083] Denoting the input signals by Lt and Rt, adjusting the product of the gains of VCAs g, and gr equal to 1 / a2, and assuming a sufficiently large loop gain that the resulting equilibrium thrust is complete, the system of Feedback-derived control of Figure 3 adjusts the gains of VCAs, so that the following equation is satisfied: Furthermore, Clearly, in the first of these equations, the absolute quantities of Lt and Rt are irrelevant. The result depends on their relation Lt / Rt, calling it X. Substituting gr from the second equation into the first one gives a quadratic equation in gi, which has the solution (the other root of the quadratic does not represent a real system): Plotting g, and gr against the gradual compounding angle a, gives Figure 7. As might be expected, g increases from a very low value to a maximum of the unit when the input represents only left (a = 90), and then falls back to a low center forward value (a = 180). In the right half, g, remains very small. Similarly and symmetrically, gr is small except in the middle part of the right half of the gradual composition, increasing to unity when α is 270 degrees (right only). [0086] The above results are for the Lt / Rt feedback derived control system. The sum / difference feedback derived control system acts in exactly the same way, producing gc sum gain and difference gain plots. gs, as shown in Figure 8. Again, as expected, the sum gain increases for the center front unit, dropping to a low value elsewhere, while the difference gain increases for the rear unit. In the feedback derived control system, VCA gains depend on the control voltage exponent, as in the preferred embodiment, so the control voltage depends on the logarithm of the gain. Thus, from the above equations, equations for the Lt / Rt and sum / difference control voltages can be derived, that is, the comparator output of the feedback derived control system, the comparator 30 of Figure 3. Figure 9 shows the left / right and sum / difference control voltages, which are inverted (ie effectively difference / sum), in a mode in which the maximum and minimum values of the control signals are ± 15 volts. Of course, other escalations are possible. The curves in Figure 9 intersect at two points, one where the signals represent an image somewhere to the listener's left rear and the other somewhere in the front half. Due to the inherent symmetries in the curves, these crossing points are exactly midway between the α values corresponding to the adjacent cardinal directions. In Figure 9, they occur at 45 and 225 degrees. The prior art (e.g., US Patent 5,644,640 to the present inventor James W. Fosgate) shows that it is possible to derive from the two main control signals, another control signal that is larger (more positive) or smaller. (less positive) of the two, although this prior art derives the main control signals in a different way and makes different use of the resulting control signals. Figure 10 illustrates a signal equal to the smallest of the curves in Figure 9. This derived control reaches a maximum when α is 45 degrees, that is, the value when the two original curves intersect. It may not be desirable for the maximum of the derived control signal to increase to its maximum precisely at α = 45. In practical embodiments, it is preferable that the derived cardinal direction representing the left rear be closer to the rear, ie have a value that is less than 45 degrees. The precise position of the maximum can be moved by decking (adding or subtracting a constant) or scaling one or both of the left / right and sum / difference control signals, so that their curves intersect at preferred values. a, before taking the most positive or most negative function. For example, Figure 11 shows the same operation as Figure 10, except that the sum / difference voltage was scaled by 0.8, with the result that the maximum then occurs at α = 31 degrees. In exactly the same way, by comparing the inverted left / right control with the inverted sum / difference, and employing similar scaling or scaling, a second new control signal can be derived, the maximum of which occurs at a predetermined position corresponding to the right rear. from the listener to a desired and predetermined α (for example, 360 - 31 or 329 degrees, 31 degrees across from zero, symmetrical with the left rear). It is a reverse left / right of Figure 11. Figure 12 shows the effect of applying these derived control signals to the VCAs in such a way that the most positive value provides a unit gain. Like the left and right VCAs that provide unit-increasing gains in the left and right cardinal directions, so these left and right rear-rear VCA gains increase for the unit when a signal is placed at predetermined locations (in this and -example, α = 31 degrees on each side of zero), but remain too small for all other positions. Similar results can be obtained with linearly controlled VCAs. The curves for the main control voltages versus the gradual composition parameter α will be different, but will intersect at points that can be selected by proper scaling or scaling, so that other control voltages for specific image positions other than those four Initial cardinal directions can be derived by a less-than operation. Clearly, it is also possible to invert the control signals and derive new ones by taking the largest (most positive) instead of the least (most negative). Modifying the main control signals to move their crossing point before taking the largest or smallest may alternatively consist of a nonlinear operation instead of or beyond a decking or a staggering. It will be apparent that the modification provides for the generation of other control voltages, the maximums of which lie at any desired ratio, and the relative quantities and polarities of Lt and Rt (the input signals).

An adaptive matrix with more than four outputs Figures 2 and 4 show that a passive matrix may have adaptive cancellation terms added to cancel unwanted crosstalk. In these cases, there are four possible cancellation terms derived via four VCAs, and each VCA achieved a maximum, generally unitary gain for a source in one of four cardinal directions and corresponding to a dominant output from one of the four outputs (left, center, right and rear). The system was perfect in the sense that a signal of gradual composition between two adjacent cardinal directions produced little or nothing of the outputs other than those corresponding to the two adjacent cardinal outputs. [0096] This principle can be extended to enable systems with more than four outputs. In such cases, the system is not "perfect", but unwanted signals can still be sufficiently canceled that the result is audibly undisturbed by crosstalk. See, for example, the six-output matrix of Figure 13. Figure 13, a functional and schematic diagram of a portion of a passive matrix according to the present invention, is a useful aid in explaining the manner in which More than four outputs are obtained. Figure 14 shows the derivation of six useful cancellation signals in Figure 13. Referring first to Figure 13, there are six outputs: Front Left (Out), Front Center (Out), Front Right (Out), Center Back (or surrounding) (Exit), rear right (R Out) and rear left (L Out) · For the three front and surrounding outputs, the initial passive matrix is the same as that of the four output system described above (one direct Lt input , the combination of Lt plus Rt halved and applied to a linear combiner 80 to produce the center front, the combination of Lt minus Rt halved and applied to a linear combiner 82 to produce the center rear, and an input Rt direct). There are two additional rear outputs, the left rear and the right rear, resulting from applying Lt with a stagger of 1 and Rt with a stagger of -ba and a linear combiner 84, and applying Lt with a stagger of -b and Rt with a scaling from 1 to a linear combiner 86, corresponding to the different combinations of the inputs according to the equations LB output = Lt -b * Rt and RB output = Rt - b * Lt. Here, b is a positive coefficient typically less than 1, for example 0.25. Note the symmetry that is not essential to the invention, but would be expected in any practical system. In Figure 13, in addition to the passive matrix terms, the output linear combiner (88, 90, 92, 94, 96 and 98) receive multiple active cancellation terms (on lines 100, 102, 104, 106 , 108, 110, 112, 114, 116, 118, 120 and 122) as required to cancel the passive matrix outputs. These terms consist of entries and / or combinations of entries multiplied by VCA gains (not shown), or combinations of entries and entries multiplied by VCA gains. As described above, VCAs are controlled so that their gains increase to the unit for one cardinal entry condition and are substantially lower for the other conditions. [0099] The configuration of Figure 13 has six cardinal directions, provided by the Lt and Rt inputs at relative quantities and defined polarities, each of which should result in signals only from the appropriate output, with substantial signal cancellation at the other five outputs. For an input condition representing a gradually compound signal between two adjacent cardinal directions, the outputs corresponding to those cardinal directions must transmit signals, but the remaining outputs should transmit little or nothing. Thus, it is expected that for each output, in addition to the passive matrix there will be many cancellation terms (in practice, more than the two shown in Figure 13), each corresponding to the unwanted output for an input corresponding to each one. from the other cardinal directions. In practice, the arrangement of Figure 13 may be modified to eliminate the central rear outlet Ssai. (thereby eliminating the combiner 82 and 94), so that the center rear is merely an intermediate composed gradually between the left rear and the right rear instead of a sixth cardinal direction. For either the six output system of Figure 13 or its five output alternative, there are six possible cancellation signals: the four derivatives via the two pairs of VCAs, which are part of the left feedback derived control systems. / right and sum / difference and two more derivatives via the rear left and rear right VCAs, controlled as described above (see also the embodiment of Figure 14, described below). The gains of the six VCAs are according to Figure 7 (left gi and right gr), Figure 8 (sum gc and difference gs) and Figure 12 (rear left g! B and right rear grb) · Cancel signals are summed with terms of the passive matrix, using calculated or otherwise selected coefficients to minimize unwanted crosstalk, as described below. We arrive at the cancellation mix coefficients required for each cardinal output, considering the input signals and the VCA gains for each of the other cardinal directions, remembering that these VCA gains increase only to unity for the signals in the card. corresponding cardinal direction, and fall off the unit very quickly as the image moves away. Thus, for example, in the case of the left exit, one needs to consider the signal conditions for the center front, only the right, the right rear, the center rear (not a real cardinal direction in the case of five exits) and rear left. Consider in detail the left output, Output, for the modification of five outputs in Figure 13. Contains the passive matrix term, Lt. To cancel the output, when the input is in the center, when Lt = Rt and gc = 1, the term -1/2 * gc * (Lt + Rt) is required, just as in the four-output system of Figures 2 or 4. To cancel when the input is in the center rear, or anywhere in between center rear and front right (therefore including right rear), -1/2 * gs * (Lt + Rt) is required, exactly again as in the four-output system of Figures 2 or 4. To cancel when the input represents the left rear, a left rear VCA signal is required whose gain, g! b, varies as in Figure 12. This can clearly convey a significant cancellation signal when the input is in the rear region left. Since the left rear can be considered somewhere between the left front, represented only by Lt, and the center rear, represented by -1/2 * (Lt - Rt), the left rear VCA is expected to operate in a combination of these signals. Various combinations may be used, but by using a sum of the signals that have already passed the left and difference VCAs, that is, gi * Lt and 1/2 * gs * (Lt - Rt), the combination varies. a-accord with the position of the composite signals gradually in the but not exactly left rear region, providing better cancellation for those gradual compositions as well as the cardinal left rear itself. Note that in this left rear position, which can be considered as intermediate between left and right, both gi and gs have finite values less than unity. Therefore, the expected equation for Output will be: The coefficient x may be derived empirically or from a consideration of accurate VCA gains, when a source is in the region of the left rear cardinal direction. The term [Lt] is the term of the passive matrix. The terms 1/2 * gs * (Lt + Rt), -1/2 * gs * (Lt + Rt) and 1/2 * x * g! B * ((gi * Lt + gs * 1/2 * ( Lt - Rt)) represent cancellation terms (see Figure 14), which can be combined with Lt in the linear combiner 88 (Figure 13), to derive the output audio signal Output. As explained above, there may be more than two crosstalk cancellation terms entries than the two (100 and 102) shown in Figure 13. [00106] The Equation for Output is derived similarly, or by symmetry: [00107] The term [Rt] is the matrix term The terms -1/2 * gc * (U + Rt), 1/2 * gs * (Lt + Rt) and -1/2 * x * grb * ((gr * Rt + gs * 1/2 * (Lt -Rt)) represent cancellation terms (see Figure 14), which can be combined with Rt in linear combiner 98 (Figure 13), to derive the output audio signal R · As explained above, there may be more than two crosstalk cancellation terms entries than the two (120 and 122) shown in Figure 13. [00108] A sa central front exit, Exit, contains the passive matrix term, 1/2 * (Lt + Rt), plus the left and right cancellation terms, as for the four-output system, -1/2 * gi * Lt e - 1/2 * gr * Rt: [00109] There is no need for explicit cancellation terms for the left rear, center rear or right rear since they are effectively compounded gradually via the rear (surrounding, four-way exit) and those already canceled. The term [1/2 (Lt + Rt)] is the term of the passive matrix. The terms -1/2 * gi * Lt and -1/2 * gr * Rt represent the cancellation terms (see Figure 14), which can be applied to entries 100 and 102, and combined with a staggered version of Lt and Rt on linear combiner 90 (Figure 13), to derive the output audio signal Output- [00110] For the left rear output, the passive matrix, as expressed above, is L, - b * Rt. For a single left input , when gi = 1, clearly the required cancellation term is therefore -gi * Lt. For a single right entry, when gr = 1, the cancellation term is + b * gr * Rt. For a central front entry, when Lt = Rt and gc = 1, the unwanted output of the passive terms, Lt - b * Rt, can be canceled by (1 - b) * gc * (Lt + Rt). The right rear cancellation term is -grt> * (gr * Rt - 1/2 gs * (Lt - Rt)), the same as the term used for Exit, with an optimized coefficient y, which can be successfully achieved, empirically or calculated from the gains of VCAs in the left or right rear conditions. Thus, Similarly, [00111] With respect to equation 24, the term [Lt - b * Rt] is the passive matrix term and the terms -gi * Lt, + b * gr * Rt, -1/2 * (1 - b) * gc * (Lt + Rt) and y * grb * (gr * Rt - gs * 1/2 * (Lt - Rt)) represent the cancellation terms (see Figure 14), which can be combined with Lt -bRt on linear combiner 92 (Figure 13), to derive the LB output audio signal. As explained above, there may be more than two crosstalk cancellation terms entries than the two (108 and 110) shown in Figure 13. [00112] With respect to equation 25, [Rt - b * Lt] is the term of the passive matrix and the components -gr * Rt, b * Lt * gi, -1/2 * (1 - b) * gc * (Lt + Rt) and -y * gib * (gi * Lt + gs * 1 / 2 * (Lt - Rt)) represent the cancellation terms (see Figure 14), which can be combined with Rt - b * Lt in linear combiner 96 (Figure 13), to derive the RB output audio signal. As explained above, there may be more than two crosstalk cancellation terms entries than the two (116 and 118) shown in Figure 13. In practice, all coefficients may require adjustments to compensate for finite loop gains. and other imperfections of feedback-derived control systems that do not accurately transmit equal signal levels, and other combinations of the six cancellation signals may be employed. These principles can of course be extended to embodiments having more than five or six outputs. However, other control signals may be derived by additional application of scaling, decalking or nonlinear processing of the two main control signals of the left / right feedback and sum / difference portions of the feedback derived control systems, allowing the generation of additional cancellation signals via the VCAs whose gains increase to a maximum at other desired predetermined values of α. The synthesis process of considering each output in the presence of signals in each of the other cardinal directions successively will produce the appropriate terms and coefficients to generate additional outputs. Referring now to Figure 14, the input signals Lt and Rt are applied to a passive matrix 130, which produces a left matrix signal output from input Lt, a right matrix signal output from input Rt, a center output of a linear combiner 132, whose input is Lte Rt, each with a scale factor of +1/2, and a surrounding output of a linear combiner 134, whose input is Lt and Rt with scale factors of + 1 / 2 and -1/2, respectively. The cardinal directions of the passive matrix are designated "left", "central", "right" and "surrounding". Adjacent cardinal directions extend on mutually orthogonal axes, so that for these direction marks, left is adjacent to central and surrounding; surrounding is adjacent left and right, etc. The left and right passive matrix signals are applied to a first pair of variable gain circuits 136 and 138 and associated feedback derived control system 140. The central and surrounding passive array signals are applied to a second pair of variable gain circuits 142 and 144 and associated feedback derived control system 146. The "left" variable gain circuitry 136 includes a voltage controlled (VCA) amplifier 148 having a gi gain and a linear combiner 150. The output of the VCA is subtracted from the left passive matrix signal at combiner 150, so that the overall gain of the variable gain circuit is (1 - gt) and the variable gain circuit output at the combiner output constituting an intermediate signal is (1 - gi) * Lt. The output signal from VCA 148, constituting a cancellation signal, is gi * Lt. [00118] The "right" variable gain circuit 138 includes a voltage controlled amplifier (VCA) 152 having a gr gain and a linear combiner 154. The output of the VCA is subtracted from the right passive matrix signal at combiner 154, so that the overall gain of the variable gain circuit is (1 - gr) and the output of the variable gain circuit at the combiner output, constituting an intermediate signal, is (1 - gr). gr) * Rt. The output signal from the VCA 152, gr * Rt, is a cancellation signal. Intermediate signals (1 - gr) * Rt and (1 - gi) * Lt constitute a first pair of intermediate signals. The relative magnitudes of this first pair of intermediate signs are to be driven towards equality. This is done by the associated feedback derived control system 140 described below. The "central" variable control gain circuit 142 includes a voltage controlled amplifier (VCA) 156 having a gc gain and a linear combiner 158. The output of the VCA is subtracted from the central passive matrix signal at combiner 158, so the overall gain of the variable gain circuit is (1 - gc) and the variable gain circuit output at the combiner output, constituting an intermediate signal, is 1/2 * (1 - gc) * (Lt + Rt ). The VCA 156 output signal, 1/2 * gc * (Lt + Rt), constitutes a cancellation signal. Surrounding variable gain circuit 144 includes a voltage-controlled amplifier (VCA) 160 having a gr gain and a linear combiner 162. The output of the VCA is subtracted from the central passive matrix signal at combiner 162, so that the overall gain of the variable gain circuit is (1 - gs) and the variable gain circuit output at the combiner output constituting an intermediate signal is 1/2 * (1 - gs) * (Lt - Rt). The VCA 160 output signal, 1/2 * gs * (Lt - Rt), constitutes a cancellation signal. The intermediate signals 1/2 * (1 - gc) * (Lt + Rt) and 1/2 * (1 - gs) * (Lt - Rt) constitute a second pair of intermediate signals. It is also desired that the relative magnitudes of this first pair of intermediate signals be driven towards equality. This is done by the associated feedback derived control system 146 described below. Feedback derived control system 140 associated with the first intermediate signal pair includes filters 164 and 166, receiving outputs from combiner 150 and 154, respectively. The respective filter outputs are applied to the logarithmic rectifiers 168 and 170, which rectify and produce the logarithm of their inputs. Rectified and log outputs are applied with opposite polarities to a linear combiner 172, whose output, constituting a subtraction of its inputs, is applied to a non-inversion amplifier 174 (devices 172 and 174 correspond to the magnitude comparator 30 of the Figure 3). Subtraction of signals transformed into logarithms provides a comparison function. As mentioned above, this is a practical way to implement a comparison function in the analog domain. In this case, VCAs 148 and 152 are of the type inherently taking the antilogarithm of their control inputs, thereby taking the antilogarithm of the logarithmic comparator control output. The output of amplifier 174 constitutes a control signal for VCAs 148 and 152. As mentioned above, if implemented digitally, it may be more convenient to divide the two quantities and use the resulting ones as direct multipliers for VCA functions. As noted above, filters 164 and 166 can be empirically derived, providing a response that attenuates low frequencies and very high frequencies, and provides a slightly increasing response from the midpoint of the audible range. These filters do not alter the frequency response of the output signals, but merely change the control signals and VCA gains in feedback derived control systems. Feedback-derived control system 146 associated with the second intermediate signal pair includes filters 176 and 178, receiving outputs from VCAs 158 and 162, respectively. The respective filter outputs are applied to the logarithmic rectifiers 180 and 182, which rectify and produce the logarithm of their inputs. The rectified outputs and transformed into logarithms are applied, with opposite polarities, to a linear combiner 184, whose output, constituting a subtraction of their inputs, is applied to a noninverting amplifier 186 (devices 184 and 186 correspond to the magnitude comparator 30 of Figure 3). Feedback-derived control system 146 operates in the same manner as control system 140. The output of amplifier 186 constitutes a control signal for VCAs 158 and 162. [00123] Additional control signals are derived from control signals from the feedback derived control systems 140 and 146. The control signal of the control system 140 is applied to the first and second scaling, decking, reversing, etc. functions. 188 and 190. The control signal of the control system 146 is applied to the first and second scaling, deviation, inversion, etc. functions. 192 and 194. Functions 188, 190, 192 and 194 may include one or more of polarity inversion, amplitude deviation, amplitude scaling, and / or nonlinear processing described above. Also according to the above descriptions, the lower or upper values of the outputs of functions 188 and 192 and of functions 190 and 194 are taken by the lower or upper functions 196, and 198, respectively, to produce other control signals, which are applied. to a left rear VCA 200 and a right rear VCA 202, respectively. In this case, additional control signals are derived in the manner described above to provide adequate control signals for generating a rear left cancellation signal and a rear right cancellation signal. The input to the left rear VCA 200 is obtained by additively combining the left and surrounding cancellation signals into a linear combiner 204. The input to the right rear VCA 202 is obtained by subtractively combining the right and surrounding cancellation signals into one. linear combiner 204. Alternatively and not particularly preferred, the inputs for VCAs 200 and 202 may be derived from the left and surrounding passive matrix outputs and the right and surrounding passive matrix outputs, respectively. The left rear VCA output 200 is the left rear cancellation signal gib * 1/2 * ((gi * Lt + gs (Lt - Rt)). The right rear VCA output 202 is the rear right cancellation signal grb * 1/2 * ((gr * Lt + gs (Lt - Rt)). [00124] Figure 15 is a schematic circuit diagram showing a practical circuit incorporating aspects of the present invention. The resistor values shown are in ohms. When not indicated, capacitor values are in microfarads. [00125] In Figure 15, "TL074" is a Texas Instruments (high input impedance) low-noise JFET multi-purpose input operational amplifier intended for high fidelity and audio pre-amplification device details are generally available in the published literature A data sheet can be found on the Internet at 'htpp: //www.ti.com/sc/docs/products /analog/tl074.html ». [00126]" SSM-2120 "in Figure 1 5 is a monolithic integrated circuit intended for audio applications. It includes two VCAs and two level detectors, allowing logarithmic control of gain or attenuation of signals presented to level detectors, depending on their magnitude. Device details are generally available in the published literature. A datasheet can be found on the Internet at 'htpp: //www.analog.com/pdf/1788_c.pdf'. [00127] The table below lists the terms used in this document for the labels on the outputs of the VCAs and for the vertical transmitter labels of Figure 15. 00128] In Figure 15, the labels on the wires going to the output matrix resistors are intended. to conduct signal functions, not their sources. Thus, for example, the few top wires leading to the front left output are as follows: Note that in Figure 15, whatever the polarity of the terms of the VCAs, the matrix itself is able to invert any terms ( U2C, etc). In addition, "servo" in Figure 15 refers to the feedback derived control system described herein. The present invention may be implemented using analog, hybrid / digital and / or digital signal processing, in which functions are performed in software and / or unalterable software. Analog terms such as VCA, rectifier, etc. are intended to include their digital equivalents. For example, in a digital embodiment, a VCA is performed by multiplication or division.

Claims (35)

A method for deriving at least three audio output signals from two input audio signals comprising the steps of: deriving four audio signals from said two input audio signals, wherein the four audio signals are derived with one passive matrix, which produces two pairs of audio signals in response to two audio signals, a first pair of derived audio signals representing directions extending on a first axis and a second pair of derived audio signals representing directions extending into a second axis, said first and second axes being mutually orthogonal between them; producing a first output signal representing a first direction extending on the axis of the pair of derived audio signals from which the first intermediate signal pair is produced, said first output signal being produced at least by combination with the same polarity. at least one component of each of said second pair of intermediate audio signals; producing a second output signal representing a second direction extending on the axis of the pair of derived audio signals from which the first intermediate signal pair is produced, said second output signal being produced at least by combination with the opposite polarity at least one component of each of said second pair of intermediate audio signals; and producing a third output signal representing a first direction extending on the axis of the pair of derived audio signals from which the second intermediate signal pair is produced, said third output signal being produced at least by combination with it. the opposite polarity of at least one component of each of said first pair of intermediate audio signals; characterized in that it further comprises the step of processing each of said derived audio signal pairs to produce the respective first and second intermediate audio signal pairs, wherein the relative amplitudes of the audio signals in each pair are of intermediate audio signals are driven towards equality. A method according to claim 1, further comprising the steps of: producing a first output signal, including combining a component of each of said second pair of intermediate audio signals with an audio signal. passive matrix, said first direction, said component constituting a cancellation signal opposite said passive matrix audio signal; producing a second output signal, including combining a component of each of said second pair of intermediate audio signals with a passive matrix audio signal, representing said second direction, said component constituting a cancellation signal opposite said signal passive matrix audio; producing a third output signal including combining a component of each of said first pair of intermediate audio signals with a passive matrix audio signal representing said third direction, said component constituting a cancellation signal opposite said signal passive matrix audio, and optionally; producing a fourth output signal, including combining a component of each of said first pair of intermediate audio signals with a passive matrix audio signal, representing said fourth direction, said component constituting a cancellation signal opposite said signal audio from the passive array. A method according to claim 2, characterized in that the matrix audio signals representing said first, second, third and optionally fourth directions, respectively, are produced by said passive matrix. Method according to claim 2, characterized in that the matrix audio signals representing said first, second, third and optionally fourth directions, respectively, are produced in various linear combiners, which also combine the passive matrix audio signals with those of said signal components. Method according to claim 1, characterized in that the respective output signals are produced by combining said intermediate signal pairs. Method according to claim 1, 2 or 5, characterized in that the processing includes feedback of each pair of intermediate audio signals for use in controlling the relative amplitudes of the respective pair of intermediate audio signals. A method according to claim 6, characterized in that the processing includes applying each derived audio signal to a respective variable gain circuit, wherein the gain of each variable gain circuit associated with each pair of audio signals. derived audio is controlled in response to the amplitudes of the variable gain circuit outputs in the respective pair. Method according to claim 7, characterized in that each variable gain circuit includes a voltage controlled amplifier (VCA) having a gain g, in combination with a subtractive combiner, the gain of the variable gain circuit. The resulting signal is (1 - g), and said cancellation signals are taken from the outputs of said voltage controlled amplifiers. Method according to claim 7, characterized in that each variable gain circuit comprises a voltage controlled amplifier (VCA) having a gain g, the resulting variable gain circuit gain is g, and said cancellation signals are taken from the outputs of said voltage controlled amplifiers. Method according to claim 7, characterized in that the gain of each variable gain circuit is low for quiescent input signal conditions, so that said signal outputs are the signals produced by said matrix. passive. Method according to claim 7, characterized in that the gain of each variable gain circuit is high for quiescent input signal conditions, so that said signal outputs are the signals produced by said matrix. passive. Method according to claim 7, characterized in that the gains of the variable gain circuitry associated with each pair of derived audio signals are controlled by applying the outputs of the respective variable gain circuitry in the pair to a frequency comparator. quantities, which generates a control signal that controls the gains of the variable gain circuits. Method according to claim 12, characterized in that the respective magnitude comparators control the gains of the variable gain circuitry associated with the pairs of derived audio signals, so that for some input signal conditions , an increase in output magnitude of one variable gain circuit relative to the other causes a decrease in gain of the variable gain circuit having the output increased. Method according to claim 13, characterized in that the respective magnitude comparators control the gain of the variable gain circuitry associated with the pair of derived audio signals, so that for some signal conditions input, an increase in the output magnitude of one variable gain circuit over the other causes no variation in the gain of the variable gain circuit having no increased output. Method according to claim 13, characterized in that the respective quantity controllers control the gains of the variable gain circuits with the pairs of derived audio signals, so that, for some conditions of the input signals, An increase in the output magnitude of one variable gain circuit relative to another also makes the product of the gains of the variable gain circuitry constant. Method according to claim 12, characterized in that the respective magnitude comparators control the gains of the variable gain circuitry with the pairs of derived audio signals, so that, for some conditions of the input signals, an increase in the output magnitude of one variable gain circuit over the other causes an increase in gain of the variable gain circuit having the output increased. Method according to claim 16, characterized in that the respective magnitude comparators control the gains of the variable gain circuits with the pairs of derived audio signals, so that for some conditions of the input signals, An increase in the output magnitude of one variable gain circuit relative to the other also causes no variation in the gain of the variable gain circuit without the output being increased. Method according to claim 16, characterized in that the respective quantity controllers control the gains of the variable gain circuits with the pairs of derived audio signals, so that, for some conditions of the frequency signals. input, an increase in the output magnitude of one variable gain circuit relative to the other also makes the gain product of the variable gain circuitry constant. Method according to claim 12, characterized in that the gain of said variable gain circuitry in dB are linear functions of its control voltages, each magnitude comparator has a finite gain and the output of each circuit of Variable gain is applied to a quantity comparator via a rectifier that transmits an output signal proportional to the logarithm of its input. Method according to claim 19, characterized in that each rectifier is preceded by a filter having a response that attenuates the low frequencies and the very high frequencies, and provides a slightly upward response to the intermediate part of the range. audible. The method of claim 12 further comprising deriving one or more additional control signals from the two control signals controlling the variable gain circuitry associated with each pair of passive matrix audio signals, wherein said one or more additional control signals are derived by modifying one or both control signals and generating the smallest or largest of the two modified control signals. Method according to claim 21, characterized in that one or both of said control signals are (are) modified by polarity inversion, amplitude deceleration, amplitude scaling and / or nonlinear processing. the respective signal. A method according to claim 21, further comprising one or more variable gain circuits, receiving as an input the combination of two of said various cancellation signals or the combination of the two passive matrix signals, wherein said one or more additional control signals control those respective ones of said one or more variable gain circuits, so that the circuit gain increases to a maximum when said input signals represent a different direction from directions extending in said first and second axes, and generating one or more cancellation signals by controlling said one or more additional variable gain circuits with respective respective one or more additional control signals. Method according to claim 23, characterized in that at least five output signals are produced by combining each of the at least five passive matrix audio signals with two or more of said various cancellation signals and of said one or more additional cancellation signals, the cancellation signals opposing each passive matrix audio signal, so that the passive matrix audio signal is canceled by the cancellation signals when said input audio signals represent signals associated with directions other than the direction represented by the passive matrix audio signal. Method according to claim 12, characterized in that the magnitude of the audio signals in a first pair of intermediate audio signals can be represented by: the magnitude of [(Lt + Rt) * (1 - gc )], or, the magnitude of [(Lt + Rt) * (hc)], and the magnitude of [(Lt - Rt) * (1 - gs)], or, the magnitude of [(Lt + Rt) * (hs)], and the magnitude of the audio signals in the other pair of intermediate audio signals can be represented by: the magnitude of [(Lt) * (1 - gi)], or the equivalent of the magnitude of [(Lt) * (h |)], and the magnitude of [(Rt) * (1 - gr)], or, the equivalent of the magnitude of [(Rt) * (hr)], where Lt and Rt are a pair of audio signals produced by said passive matrix, Lt + Rt and Lt - Rt are the other pair of audio signals produced by said passive matrix, (1 - gc) and hc are the gain of a variable gain circuit. associated with the passive matrix's Lt + Rt output, (1 -gs) and hs are the gain of an ass variable gain circuit. with the output Lt - Rt of the passive matrix, (1 - gi) and h | are the gain of a variable gain circuit associated with the passive matrix output Lt, and (1 -gr) and hr are the gain of a variable gain circuit associated with the passive matrix output Rt. A method according to claim 1, further comprising the step of: producing a fourth output signal representing a second direction extending on the axis of said pair of derived audio signals, of which the second pair intermediate signal output is produced, said third output signal being output at least by combination with the opposite polarity if the third output signal is output by combination with the same polarity, or at least by combination with the same polarity if the third output signal is produced by combining at least one opposite polarity of each of said pair of intermediate audio signals. A method for deriving at least three audio signals, each associated with a direction, from two input audio signals comprising the steps of: generating with a passive matrix, in response to said two input audio signals, various passive matrix signals, including two pairs of passive matrix audio signals, a first pair of passive matrix audio signals, representing directions extending on a first axis, and a second pair of passive matrix audio signals representing directions extending on a second axis, said first and second axes being mutually orthogonal between them; deriving various cancellation signals from said intermediate audio signal pairs; and producing at least three output signals by combining each of the at least three passive matrix audio signals with two or more of the various cancellation signals, the cancellation signals opposing each passive matrix audio signal of so that the passive matrix audio signal is canceled by the cancellation signals, when said input audio signals represent signals associated with directions other than the direction represented by the passive matrix audio signal; characterized in that it further comprises the step of processing each of said passive matrix audio signal pairs to produce the respective first and second intermediate audio signal pairs, so that the relative amplitudes of the audio signals in each pair of intermediate audio signals are driven towards equality; Method according to claim 27, characterized in that the processing includes the feedback of each pair of intermediate audio signals for use in controlling the relative amplitudes of the respective pairs of intermediate audio signals. A method according to claim 28, wherein said processing includes applying each passive matrix signal to said two pairs of passive matrix audio signals to a variable gain circuit, each circuit including a controlled amplifier. (VAC) having a gain g in combination with a subtractive combiner, wherein the gain of the resulting variable gain circuit is (1 - g), and said cancellation signals are taken from the outputs of said amplifier controlled by voltage. Method according to claim 29, characterized in that the gains of the variable gain circuits associated with each pair of passive matrix audio signals are controlled by applying the outputs of the respective variable gain circuitry of each pair to a quantity comparator, which generates a control signal that controls the gains of the variable gain circuits. Method according to claim 30, characterized in that the outputs of the respective variable gain circuit of each pair are applied to a quantity comparator via a rectifier, the rectifiers transmit signals proportional to the logarithm of their inputs, The comparator has a finite gain, and the VCA gains in dB are linear functions of its control voltages. The method of claim 30, further comprising deriving one or more additional control signals from the two control signals, which control the variable gain circuits associated with each pair of matrix audio signals. wherein said one or more additional control signals are derived by modifying one or both control signals, and generate the smallest or largest of an unmodified control signal and a control signal. of two modified control signals. A method according to claim 32, characterized in that one or both of said control signals are modified by polarity inversion, amplitude deceleration, amplitude scaling and / or processing. of the respective signal. A method according to claim 32, further comprising one or more variable gain circuits, receiving as an input the combination of two of said various cancellation signals or the combination of the two passive matrix signals in said one or more additional control signals control those respective ones of said one or more variable gain circuits, so that the circuit gain increases to a maximum when said input signals represent a direction different from the directions extending on said first and second axes, and generating one or more control cancellation signals from said one or more additional variable gain circuits with respective respective one or more additional control signals. A method according to claim 34, characterized in that at least five output signals are produced by combining each of the at least five passive matrix audio signals with two or more of said various cancellation signals and of said one or more additional cancellation signals, the cancellation signals opposing each passive matrix audio signal, so that the passive matrix audio signal is canceled by the cancellation signals when said input audio signals represent signals associated with directions other than the direction represented by the passive matrix audio signal.