MX2007003409A

MX2007003409A - Contactless multiposition switches using capacitive touch sensors.

Info

Publication number: MX2007003409A
Application number: MX2007003409A
Authority: MX
Inventors: Bahar N Wadia; David W Caldwell; Rick L Oppor; Donald Charles Mueller; Tony V Gioiosa; Bob B Campbell; Thomas M Schreiber; Mark R Matthews; Dan Eavenson; Anthony Russo; Andy Livingston; Scott Roberts
Original assignee: Touchsensor Tech Llc
Priority date: 2004-09-24
Filing date: 2005-09-26
Publication date: 2008-03-04
Also published as: CN101065904A; AU2005289529A2; AU2005289529A1; EP1803221A1; CA2581515A1; WO2006036950A1; CA2581515C; NZ554621A; KR101186393B1; KR20070064647A; JP2008515153A; WO2006036950A8; BRPI0515584A; AU2005289529B2

Abstract

A touch switch apparatus for detecting the presence of an object such as a human appendage, the apparatus having a touch pad, an electric field generated about the touch pad and also having a preferably integrated and local control circuit connected to the touch pad and to a controlled device. Practical applications for touch switch apparatus, including use of touch switch apparatus in connection with other structure, in particular multiposition contactless switches, to emulate mechanical switches.

Description

MULTIPLE NON-CONTACT POSITION SWITCHES USING CABLE TACTICAL SENSORS FIELD OF THE INVENTION The present invention relates to touch switches (i.e., switches that are operated, for example, by the touch of a finger or a touch pad, also referred to herein as touch sensors or effect sensors). field) and related control circuits and practical applications for them.

BACKGROUND OF THE INVENTION Mechanical switches have been used for a long time to control apparatus of all types, including home electrical appliances, machine tools, automobiles and related systems and all kinds of other domestic and industrial equipment. Mechanical switches are typically mounted on a substrate and require some type of penetration through the substrate. These penetrations as well as the penetrations in the switch itself, can allow dirt, water or other contaminants to pass through the substrate or get trapped inside the switch, thus generating short electrical circuits or other malfunctions. Touch switches are often used to replace conventional mechanical switches. Unlike mechanical switches, touch switches contain no moving parts that break or wear. In addition, the touch switches can be mounted or formed on a continuous substrate sheet, for example a switch panel, without the need to open the substrate. The use of touch switches instead of mechanical switches can therefore be advantageous, particularly in environments where contaminants are likely to be present. Tactile switch panels are also easier to clean compared to typical mechanical switch panels because they can be made without openings in the substrate that can allow the penetration of contaminants. Known touch switches typically comprise a touch pad having one or more electrodes. The touch pads communicate with control or interconnection circuits which are often complicated and far from the touch pads. A signal is usually provided to one or more of the electrodes comprising the touch pad, creating an electric field around the affected electrodes. The control / interconnection circuits detect alterations in the electric fields and generate a response which is produced for use by the controlled device. Although touch switches solve many problems related to mechanical switches, the designs of known touch switches are not perfect. For example, many known touch switches may exhibit malfunction when contaminants such as water or other liquids are present in the substrate. The contaminant can act as a conductor for the electric fields created around the touch pads, which causes undesired switch operations. This represents a problem in areas where such contaminants are commonly found, such as the kitchen and some factory environments. Existing designs of touch switches also suffer from problems related to crosstalk, that is, interference between electric fields around adjacent touch pads. Crosstalk may cause an erroneous touch switch to be triggered or may cause two switches to be operated simultaneously by a touch next to a single touch pad. Many known tactile switch designs are also susceptible to non-proposed actuation due to electrical noise or other interferences that affect the same touch pad or electrodes extending from the touchpad to its associated control circuit. This problem can be aggravated in applications where the touchpad is a relatively large distance away from the control circuits, as is often the case with conventional tactile switch designs. Existing touch switch designs commonly require complicated control circuits in order to interconnect with the devices they control. It is likely that these control circuits are made up of a large number of separate components which occupy considerable space on a circuit board. Due to their physical size, the control circuits are typically located at a substantial distance from the touch pads themselves. The physical size of the control / interconnection circuits and their remoteness from the touch pads can aggravate many of the problems discussed above such as crosstalk and susceptibility to electrical noise and interference. The size and the distance also complicate the tactile switch panel design in general resulting in increased production costs and complexity. Some known tactile switch designs require a grounded electrode separate from the touchpad to the interconnect / control circuit or to the controlled device. Some devices using conventional mechanical switches and do not require, or may not easily accommodate, these electrodes connected to ground. The adaptation of such devices for use with such touch switches may require the addition of special grounding supplies, and therefore the design and time of production, complexity and costs are increased. These requirements for grounded electrodes can prevent simple direct replacement of conventional mechanical switch panels with touch switch panels. Recent improvements in touch switch design include techniques that decrease the input and output impedance of the touch switch itself, making it highly immune to false fluctuations due to contaminants and external noise sources. The patent of E.U.A. No. 5,594,222 discloses a low impedance touch switch design which is less susceptible to malfunction in the presence of contaminants and electrical noise compared to many previous designs. Although this approach has several advantages over the prior art, there are certain attributes that may limit its application. For example, the resulting touch switch may be sensitive to temperature variations. To the extent that the temperature variations in the outputs are small in relation to the legitimate signal changes and are small in relation to signal variations induced by variations of transistors, then a single transistor or other amplifying device will be very satisfactory. However, this technique may require the use of additional circuits for interconnection with the controlled device which increases the costs and complexity of the entire touch switch design. In applications where there is little dynamic range to allow compensation and where temperature changes are significant in relation to legitimate signal changes, a different approach may be more appropriate to eliminate or reduce the effects of temperature. In addition, although the low impedance solution of this technique can differentiate between contaminants with a certain finite amount of impedance and a human touch with a certain finite amount of impedance, this technique may not be sufficient to differentiate between extremely low levels of impedance. Such a situation may exist when the entire tactile switch (for example in both indoor and outdoor electrodes) is covered with a large amount of contaminant. A similar situation can exist, essentially of zero impedance when a conductive material, such as a metal tray, completely covers the touch switch. The patent of E.U.A. No. 6,310,611, assigned to the same beneficiary of the present application and incorporated herein by reference in its entirety, discloses a touch switch apparatus having a differential measurement circuit which corrects many of the problems related to common mode disturbances affecting the touch switches. For example, a touch switch having a two-electrode touch pad can be configured to generate an electric field around each electrode. A common mode alteration, such as a contaminant that substantially covers both electrodes, is likely to affect the electric field around each of the electrodes in a substantially equal manner. Each electrode provides a signal proportional to the alteration of the differential measurement circuit. Since the signals from the electrodes are therefore considered to be substantially equal, the differential measurement circuit does not detect a differential and does not respond to a common mode alteration. On the other hand, if the field around only one of the electrodes is altered, the signal provided by that electrode to the differential measurement circuit will likely be substantially different than that provided by another, unaffected electrode. The differential circuit can respond by providing an output based on different degrees of stimulation in the first and second electrodes, which can cause a switch drive based on the particular stimulation state of the electrodes or can provide information based on Many states of stimulation on the electrodes. Although the differential measurement circuit solution solves many problems known in the prior art, it is relatively complex and can be expensive to design and manufacture. A differential measurement circuit typically comprises many more parts than a more conventional control circuit. It is likely that additional parts require more space in a touch switch panel. As such, the control circuit is likely to be even further away from the touch pad than it could be with a non-differential circuit design, which requires long electrodes between the touch pad and its control circuit. This can actually aggravate concerns regarding electrical interference. In addition, when a differential measurement circuit is constructed, the coincidence of the components becomes important. Proper matching of the components presents an additional manufacturing burden and is likely to add costs. In addition, when differential sensing techniques are used, the resulting signals are relatively small compared to the dynamic range of absolute signal changes of the electrodes, especially in low impedance applications. Therefore, the resulting signal can be altered by noise and other environmental effects. Proper damping of the differential signal may typically require the use of additional components to build a switch or a shock absorber. In addition, when a stimulus such as a pulse signal is applied from a remote control circuit, the pulse signal may be altered. The stimuli that generate circuits such as pulse generating circuits typically require many components and occupy physical space that can interfere with the sensing electrodes. Therefore, the signal generating circuits need to be physically located away from the sensing electrodes if they occupy physical space that can inadvertently alter or polarize the sensing electrodes, which can effectively reduce the operation of the sensor signal to noise ratio. Although the above improvements can reduce undesired actuation of the switches as a result of crosstalk between switches and the effects of electrical interference on other control circuits, they do not completely eliminate these problems. In addition, they do not solve the need for separate circuits connected to ground in certain touch switch applications or solve the concerns related to them. furtherIt would be advantageous if the aforementioned characteristics can be implemented using a physical structural form as small as possible. Typically, the operation of a field effect sensor does not require the application of force or physical displacement of a structural member by a user, as would be the case, for example, with a mechanical push button, a tilting circuit or a rotary switch . Although this is a desirable attribute in many applications, in other applications it may be desirable for a user to apply force or physically move a switch member in order to provide the user with the physical perception that the switch has changed state. In some applications, it may be desirable to provide a switch mechanism that has the advantages offered by the field effect sensors while retaining the mechanical feel of a conventional mechanical switch.

BRIEF DESCRIPTION OF THE INVENTION The present invention provides a touch switch apparatus comprising a touchpad and a control circuit that is located near the touchpad. The touch pad and the control circuit can be mounted on a dielectric substrate. The control circuit is small compared to the overall size of the device. In a preferred embodiment, the control circuit is substantially reduced to one or more integrated circuits. The compact physical condition of the control circuit in the integrated circuit mode reduces the sensitivity of the touch switch to common mode interference and crosstalk and interference between adjacent touch switches. The integrated circuit solution also provides a better match and balance of the control circuit components. The touch switch of the present invention can be configured in a variety of preferred embodiments. In some embodiments, the touch switch may emulate a conventional contact-type or mechanical contact-type switch. In other embodiments, the touch switch can emulate a momentary contact type of the mechanical switch. In addition, in other embodiments, the touch switch may provide multiple outputs in relation to the detection on the sensing electrodes. In a preferred embodiment, the touchpad has a first electrode and a second electrode proximate the first electrode. At least one of the electrodes is electrically coupled to the local control circuit. The first and second electrodes and the local control circuit are typically placed on the same surface of a substrate, opposite the side of the substrate to be used as the touch surface. However, they do not need to be coplanar and can be placed on opposite sides of a substrate. In an alternative embodiment, the touch pad has a single electrode which is electrically coupled to the local control circuit. In other alternative embodiments, the touchpad may have more than two electrodes. In a preferred embodiment, the control circuit includes a means for generating a signal and providing it to the touch pad to create an electric field around one or more of the electrodes comprising the touch pad. Alternatively, said signal may be generated elsewhere and may be provided to one or more of the electrodes to create one or more electric fields around it. The control circuit detects alterations to the electric fields in response to stimuli to them, such as the tip of the finger of a user making contact with or approaching the substrate adjacent to the touch switch. The control circuit responds selectively to said field alterations by generating a control signal for use by a controlled device, such as a home electrical appliance or an industrial machine. In a preferred embodiment, the control circuit senses and responds to the electrical potential between the first and second electrodes in response to the introduction of a stimulus in proximity to either the first electrode, the second electrode or both. Such a differential measurement circuit provides rejection of common mode signals (i.e., signals that may tend to alter both electrodes approximately equally) such as temperature, electrical noise, variations in the power supply and other inputs . The differential measurement circuit also provides for the rejection of common mode signals resulting from the application of contaminants to the substrate adjacent to the touch switch. In a preferred embodiment, a signal is applied to the first electrode and a second electrode. The signal can be generated from inside the control circuit or from another part. An electric potential develops at each electrode and, consequently,, an electric field is generated around each of the electrodes. Two matching transistors are distributed in a differential measurement circuit, where the first transistor is connected to the first electrode and the second transistor is connected to the second electrode. Each transistor output is connected to a peak detector circuit, and the output of each peak detector circuit in turn is provided to a decision circuit. Each transistor output is altered when the electric field around its corresponding electrode is altered for example when the electrode is touched or when approaching a user. The peak detector circuits respond to changes in the transistor outputs and provide signals that correspond to the peak potentials from the transistors to the decision circuit. The decision circuit uses peak potentials in a predetermined manner to provide an output for use by other portions of the control circuit. In a preferred embodiment, the internal and external electrodes are operably associated with the inputs to the decision circuit for example when an alteration of an electric field around a first electrode is greater than the degree of alteration of an electric field around a second electrode , the decision circuit will provide a high level output. Conversely, the decision circuit will provide a low level output when an alteration to the electric field around the second electrode is greater than the degree of alteration of an electric field around the first electrode. When the fields around both electrodes are distributed in a more or less equal way, the decision circuit will provide a low level output. The first condition can be created, for example, when the tip of one finger substantially covers the first electrode but not the second electrode. The second condition can be generated, for example, when the tip of a finger or a contaminant substantially covers the second electrode but not the first electrode. The third condition can be generated, for example, when a contaminant or an object such as a metal tray covers both the first and the second electrodes. The output of the decision circuit is provided to other circuit components such as an electrical latch, which selectively causes a control signal to be transmitted from the control circuit, depending on the output state of the decision circuit. In a preferred embodiment, the high level output of the decision circuit eventually causes a control signal to be transmitted from the control circuit, while no control signal will be transmitted in response to a low level output. In an alternative embodiment, a low level output of the decision circuit causes a control signal to be transmitted from the control circuit, while a control signal will not be transmitted in response to a high level output. The touch switch apparatus of the present invention can be used to perform almost any function which can be performed by a mechanical switch, such as turning a device on or off, adjusting the temperature or setting a clock or timer It can be used instead of, and can solve the problems associated with existing touch switches. It can also be used as a direct replacement for membrane type mechanical switches. The touch switch apparatus of the present invention is suitable for use in environments where temperature variations are extreme, where substantial amounts of contaminants may be present where metal objects may be placed on or above the touch pad. The present invention provides input circuit portions for more effectively communicating signals between the touch pad electrodes and the logic and decision circuits. In a preferred embodiment, these input portions of the control circuit include active devices and peak detection circuits in various configurations for converting high frequency transient pulses into DC signals. These modes can eliminate the need for more complicated AC processing circuits and can allow the use of a DC processing circuit which will reduce the size and cost of the integrated circuits of the touch switch assemblies. In addition, these preferred embodiments may be capable of discharging the electric fields associated with the peak detection circuits, which correspond to the electric fields at the input electrodes. In other preferred embodiments, the negative effects of the parasitic capacitance caused by the junction pad and the wire junction configuration are compensated for by incorporating dissipative capacitance in the input portions of the control circuits mentioned above. The dissipation according to these embodiments of the present invention can eliminate imbalances in the differential measurement circuit by parasitic capacitance and can in this way provide more consistent electrical information moving within the decision circuit. In other preferred embodiments, protection of control circuits from damage caused by eddy current and sometimes high electrostatic potential of the touch pad input electrodes is provided by active blocking device configurations at the input portions of the circuit of control. Other preferred embodiments can provide statistical filtering and sampling in high noise environments and other environments. In addition, other preferred embodiments provide the linearization of input signal sent to decision circuits using differential measurement techniques. The present invention also provides inverter circuits which facilitate direct replacement in the membrane and other mechanical switches with touch detection switches. In the preferred embodiments, this inverter circuit configuration can provide isolation of the inherent leakage current paths that develop from substrates with impurities used to manufacture the control and integrated circuits of the touch switch assemblies. It is also an object of the present invention to provide an analog output that takes advantage of the input configurations of the circuits used by the invention. A further objective of the invention is to provide ways to detect capacitive inputs. The present invention also relates to practical applications for touch switches. Although the touch switches described herein are particularly suitable for use in connection with many of the applications disclosed herein, other touch switches and sensors, for example capacitive sensors and field effect sensors as described in the patents of USA No. 5,594,222 and 6,310,611, the descriptions of which are incorporated herein by reference, may also be used in said applications.

BRIEF DESCRIPTION OF THE DRAWINGS Various features, advantages and other uses of the present invention will become more apparent with reference to the following detailed description and drawings, in which: Figure 1 is a perspective drawing of the components of a preferred embodiment of a touch switch of the present invention; Figure 2 is a cross-sectional view of a two electrode touch pad and an integrated circuit chip of the present invention; Figure 3 is a plan view of one embodiment of a touch switch apparatus of the present invention; Figure 4 is an electrical schematic representation of a touch switch control circuit configured for a preferred mode of operation; Figure 5 is a schematic electrical representation of a tactile switch control circuit configured for an alternative preferred mode of operation; Figure 6 is a schematic electrical representation of a tactile switch control circuit configured for another alternative preferred mode of operation; Fig. 7 is an electrical schematic representation of a tactile switch control circuit configured for another additional alternative preferred mode of operation; Figure 8 is a cross-sectional view of an alternative embodiment of a touchpad of the present invention; Figure 9 is a cross-sectional view of another alternative embodiment of a touchpad of the present invention; Figure 10 is a diagrammatic representation of a mode of a touch switch panel using a plurality of tactile switches in the form of a matrix; Figures HA to 11D are electrical schematic representations of input circuits for tactile switch control circuits that are compatible with the circuits shown in Figures 4 to 7; Figures 12A to 12H are schematic electrical representations of input circuits for tactile switch control circuits of Figures HA to 11D wherein the active devices serve as current sources; Figures 13A to 13H are schematic electrical representations of input circuits for tactile switch control circuits of Figures 12A to 12H with different combinations of active devices; Figures 14A to 14D are electrical schematic representations of input circuits for tactile switch control circuits of Figures HA to 11D having active square root extraction devices; Figures 15A to 15D are electrical schematic representations of input circuits for tactile switch control circuits of Figures 14A to 14D having different active square root extraction devices; Figure 16 is a schematic electrical representation of input circuits for the touch switch control circuit of Figure 15A having dissipative capacitance provided by the capacitors; Figure 17A is a schematic electrical representation of input circuits for the touch switch control circuit of Figure 16 wherein the dissipative capacitance is provided by the suppression capacitance of the diodes at the inputs; Figure 17B is a diagram of a touch switch assembly showing a possible configuration where the electrodes are close to the integrated circuit; Figure 18A shows a configuration that provides negative feedback directly in an input circuit; Fig. 18B shows a common gate configuration with front end dissipative capacitance and illustrating how the input configuration can be different from a common source configuration, as shown for all of the previous drawings; Figure 18C shows the configuration of Figure 18B but with the suppression diodes; Figure 18D shows the configuration of Figure 18B but in the single-electrode format and using two dissipative capacitors and illustrates the matching of the cost-effective integrated circuit; Figure 18E shows the configuration of Figure 18D but with the suppression diodes; Figure 19 is a schematic electrical representation of output circuits for the integrated circuit of a touch switch control circuit; Figures 20A to 20D are schematic representations of tactile cell arrays for use with various modes of operation; Figures 21A to 21F are schematic representations of MOSFET blocking devices; Fig. 22 is a schematic of a way of configuring a membrane array or other mechanical switches and the addressing and synchronization therefor; Fig. 23 is a schematic of Fig. 22 wherein the switches are touch switch assemblies having two connections to the address lines of the array configuration; Figures 24A to 24B are electrical schematic representations of some characteristics of the output circuit shown in Figure 9 communicating with a touch switch control circuit; Figure 25A shows a possible configuration of the active devices that constitute an inverter circuit, according to the present invention; Figures 25B to 25C are schematic representations of the inverter circuit according to the present invention; Figures 26A to 26C show a capacitive switch apparatus for use with the integrated circuit of the present invention, wherein the circuit shown in Figure 26D can respond to the capacitance between two electrodes that change due to a change in the distance between them; Fig. 26D shows a circuit according to the present invention for use with the application described with reference to Figs. 26A to 26C; Figures 27A to 27D show a capacitive liquid detecting switch apparatus for use with the integrated circuit of the present invention, wherein the circuit shown in Figure 27E can respond to a change in the relative dielectric constant of an electrode; Fig. 27E shows a circuit according to the present invention for use with the application described with reference to Figs. 27A to 27D; Figures 28A to 28B show a capacitive switch apparatus for use with the integrated circuit of the present invention wherein the circuit of Figure 28C can respond to the capacitance between two electrodes that changes due to an effective change in the surface area of a electrode; Fig. 28C shows a circuit according to the present invention for use with the application described with reference to Figs. 28A to 28B; Figures 29A to 29G show a capacitive switch apparatus which can function as a marking device for use with the integrated circuit of the present invention (Figures 29A to 29D show the electrode configuration of the apparatus in various input stages; at 29F show the pulse output of two types of device rotation and Figure 29G shows a possible integrated circuit configuration for use with the device shown in Figures 29A to 29D); Figures 30A to 30E show another type of capacitive switch marking device for use with the integrated circuit of the present invention wherein an electrode is connected to ground by the user; Figures 30F to 30G show the pulse output of two types of device rotation; Figure 30H shows a diagram of the input connections between the input device of Figures 30A to 30E and an integrated circuit for use with said device; Figures 31A to 31F show the separate construction layers of a touch switch with an integrated matrix control circuit two by two assembled on a substrate; Figure 32 shows an embodiment of the integrated circuit of the present invention using an AC input and low current; Figure 33A shows the input and other portions of an embodiment of the integrated circuit of the present invention for use with electrical field detection applications having an analog output; Figures 33B to 33C show timing diagrams for the integrated circuit shown in Figure 33A; Figure 34 shows a matrix of analog output sensors; Figure 35A is a side elevational view of a push button switch emulation embodiment, in accordance with the present invention; Figure 35B is a bottom plan view of a push button switch emulation embodiment according to the present invention; Figure 35C is a side elevational view of an alternative embodiment of push button switch emulation, in accordance with the present invention; Figure 35D is a side elevational view of another alternative embodiment of a push button switch emulation, in accordance with the present invention; Figure 35E is a bottom plan view of a further alternative embodiment of a push button switch emulation, in accordance with the present invention; Fig. 36A is a side elevation view of an emulation mode of a rocker switch according to the present invention; Fig. 36B is a side elevation view of an emulation mode of a rocker switch according to the present invention; Figure 36C is a bottom plan view of an emulation mode of a rocker switch according to the present invention; Fig. 36D is a side elevation view of an alternative embodiment of a tilter switch emulation according to the present invention; Figure 37A is a side elevational view of a rotary switch emulation according to the present invention; Fig. 37B is a bottom plan view of an embodiment of a rotary switch emulation according to the present invention; Figure 37C is a bottom plan view of another portion of an embodiment of a rotary switch emulation according to the present invention; Fig. 37D is a bottom plan view of another portion of an embodiment of a rotary switch emulation according to the present invention; Fig. 37E is a timing diagram for one embodiment of a rotary switch emulation, in accordance with the present invention; Fig. 37F is a side elevational view of an alternative embodiment of a rotary switch emulation according to the present invention; Figure 37G is a side elevational view of another alternative embodiment of a rotary switch emulation according to the present invention; Fig. 37H is a side elevational view of a further alternative embodiment of a rotary switch emulation according to the present invention; Fig. 371 is a top plan view of a portion of another alternative embodiment of a rotary switch emulation according to the present invention; Fig. 38A is a side elevation view of a further alternative embodiment of a rotary switch emulation according to the present invention; Fig. 38B is a top plan view of a further alternative embodiment of a rotary switch emulation according to the present invention; Figure 38C is a bottom plan view of a portion of a further alternative embodiment of a rotary switch emulation according to the present invention; Fig. 38D is a partial cross-sectional view of a portion of a further alternative embodiment of a rotary switch emulation according to the present invention; Fig. 39A is a side elevational view of an embodiment of a rotary switch emulation or an angular position sensor according to the present invention and a schematic representation of the electrode structure for use in connection therewith; Figure 39B is a schematic representation of an alternative electrode structure for use in relation to the embodiment illustrated in Figure 39A; Fig. 40 is a side elevational view of a further alternative embodiment of a rotary switch emulation according to the present invention; Figure 41A is a side elevation view of an oscillating switch emulation embodiment according to the present invention; Figure 41B is a side elevational view of an alternative embodiment of an oscillating switch emulation in accordance with the present invention; Figure 41C is a side elevational view of another alternative embodiment of an oscillating switch emulation in accordance with the present invention; Figure 42A is a side elevational view of a slider switch emulation embodiment according to the present invention; Figure 42B is a side elevational view of an alternative embodiment of a slide switch emulation according to the present invention; Figure 42C is a side elevational view of another alternative embodiment of a slide switch emulation according to the present invention; Figure 42D is a perspective view of an alternative embodiment of a rotary switch emulation according to the present invention; Figure 42E is a top plan view of a sensor in the x-y position according to the present invention; Figure 43 is a side elevational view of an alternative embodiment of a ball switch emulation according to the present invention; Figure 44 is an illustration of a regulatory control and related sensors that constitute the principles of the present invention; Figure 45A is a perspective view of a tire pressure sensing apparatus according to the present invention; Figure 45B is a side elevational view of a tire pressure sensing apparatus according to the present invention; Figure 46 is a side elevational view of a car seat that includes weight and position sensors, in accordance with the present invention.

DETAILED DESCRIPTION OF THE DRAWINGS The descriptions of the patents of E.U.A. Nos. 5,594,222, 5,856,646, 6,310,611, 6,320,282, 6,713,897 and 6,897,390 and the patent application of E.U.A. serial number 10 / 271,933 titled Intelligent Shelving System, 10 / 272,047 entitled Touch Sensor with Integrated Decoration and 10 / 850,272, entitled Integrated Touch Sensor and Light Apparatus, all filed on October 15, 2002 and all assigned to the beneficiary of the present invention. they are incorporated herein by reference in their entirety. The invention relates to a touch switch apparatus comprising a touchpad having one or more electrodes and a control circuit. Many of the drawings illustrating the control circuit present the large circuit in relation to the touch pad for clarity purposes. However, in typical applications the control circuit will be small compared to the touch pad and preferably in the form of one or more integrated circuit chips. Figure 1 is a perspective representation of a preferred embodiment of a touch switch apparatus 20 of the present invention. The touch switch apparatus 20 comprises a touch pad 22, a control circuit 24 comprising an integrated circuit chip (IC) having eight output terminals, PIN 1 - PIN 8 and first and second resistors R1 and R2. In the embodiment shown, the touch pad 22 comprises a first electrode El and a second electrode E2, although the touch pad may also be constituted by more or less than two electrodes. Although the control circuit 24 can be manufactured using separate electronic components, it is preferable to constitute the control circuit 24 on a single integrated chip, such as the IC chip 26. The control circuit 24, by means of the PATILLA terminals 1 - . 1 - PIN 8 of the chip 26 IC is electronically coupled and communicates with a first and second resistors Rl and R2, a first and second electrodes El and E2 and an input line 30 which is configured to supply a control or a signal from energy from a remote device (not shown). The control circuit 24 also communicates with a remote device (not shown) using a first output line 32. In some embodiments, a second output line 34 is also used for communication with the remote device (not shown). Figure 2 is a partial cross-sectional view of a typical touch switch 20 of the present invention in which the components comprising the touch switch apparatus 20 are mounted on a dielectric substrate 35 having a front surface 36 and a rear surface 37 opposite. In the embodiment shown, the first and second electrodes El and E2 are mounted on rear surfaces 37 of the substrate 35. The IC chip 26 is also mounted on the rear surface 37 of the substrate 35., next to the first and second electrodes El and E2. As can be seen from both figures 1 and 2, in the preferred embodiment it is contemplated that the IC chip 26 comprises a control circuit 24 that is mounted in close proximity to the touch pad 22. The substrate 35 typically consists of a relatively rigid dielectric material such as glass, plastic, ceramic material or any other suitable dielectric material. However, the substrate 35 may also comprise other suitable dielectric material including flexible materials. An example of a suitable flexible substrate is a polyether material from Consolidated Graphics No. HS-500, type 561, level 2 with a thickness of 127 μm (0.0005 inches). In embodiments wherein the components of the touch switch apparatus are driven on a flexible substrate, the flexible carrier is often applied to another generally more rigid substrate. In a preferred embodiment, the substrate 35 is made of glass having a uniform thickness of about 3 mm. In other embodiments, the thickness of the substrate 35 may vary, depending on the type of material used, its mechanical and electrical properties and the physical strength as well as the electrical sensitivity needed for a particular application. The maximum functional thickness for glass and plastic substrates is in the order of several inches. However, in most practical applications the glass substrates vary in thickness from about 1.1 mm to about 5 mm while the plastic substrates can be even thinner. In a preferred embodiment, as shown in Figures 1 and 2, the second electrode E2 substantially surrounds the first electrode El. A space 28 is located between the first electrode El and the second electrode E2. The first electrode El can be dimensioned so that it can be "covered" by the tip of a user's finger or other human appendage when the user touches the corresponding portion of the front surface 36 of the substrate 35. In a preferred embodiment, the first The electrode is square and the second electrode E2 is distributed in a square pattern around and adapting to the shape of the first electrode El. Although the touch pad geometry illustrated in FIGS. 1 and 2 represents a preferred distribution of the first and second electrodes El and E2, it should be recognized that the electrode distribution can vary widely to suit a wide variety of applications. For example, the size of the electrode, shape and placement may vary to suit the size of the appendage or other stimulus with which the tactile switch 20 is intended to be operated. For example, a particular application may require that a hand, rather than a finger, be the one that provides the stimulus to operate the touch switch 20. In said application, the first and second electrodes El and E2 will be much larger and will be further apart. The first electrode The can acquire any of numerous different geometric shapes that include but are not limited to rectangles, trapezoids, circles, ellipses, triangles, hexagons and octagons. Regardless of the shape of the first electrode El, the second electrode E2 can be configured to surround at least partially the first electrode El in a separate relationship. However, it is not necessary for the second electrode El to surround the first electrode even partially in order to obtain the benefits of the invention. For example, the first and second electrodes El and E2 may be adjacent to each other, as shown in Figure 3. In alternative embodiments, the second electrode E2 may be omitted. In addition, the configuration of the electrode does not need to be coplanar but can be three-dimensional to fit a sphere, cube or other geometric shape. This design flexibility allows the invention to be used in a wide variety of applications with substrates of varied shapes and composition. In some applications it may not be necessary to actually touch the substrate 35 on or within which is the touchpad 22 and the control circuit 24. For example, Figure 8 illustrates a touch switch apparatus 20 wherein the first and second electrodes El and E2 are mounted on an outer surface 113 of a first plate 111 of a thermoplate window 110 and which can be operated by a user that generates a suitable stimulus 115 proximate an outer surface 114 of an opposite plate 112 of the window. As indicated in the above, the first and second electrodes El and E2 do not need to be coplanar; they can be mounted on different sides or surfaces of a substrate or on completely different substrates. For example, Figure 9 illustrates a touch switch apparatus 20 wherein the first electrode El is mounted on a first surface 36 of a substrate 35 and a second electrode E2 and the IC chip 26 are mounted on a second opposing surface 37 of the substrate 35. In applications where the first and second electrodes El and E2 are on the same side of a substrate, the IC chip 26 can be mounted on the same side of the substrate as the electrodes or on another side of the substrate. If the first and second electrodes are mounted on different surfaces of a substrate or on different substrates completely, the IC chip 26 can be mounted on the same surface as either of the electrodes or on a different surface or a completely different substrate. However, it is preferred that the IC chip 26 be mounted in close proximity to the electrodes. Preferably, the first electrode El is a solid conductor. However, the first electrode El may also have a plurality of openings or may have a mesh or grid pattern. In some embodiments, the second electrode E2 will take the form of a narrow strip partially surrounding the first electrode E2. In other embodiments, for example where the first and second electrodes El and E2 are simply adjacent to each other, the second electrode E2 may also be a solid conductor or may have a mesh or grid pattern. The control circuit 24 can be designed in many different ways and can be used with a variety of energy sources such as AC, DC that varies periodically (such as a square wave), continuous DC or others. Figures 4 to 7 illustrate a preferred control circuit design which can be easily adapted for use to a wide variety of power supplies in a variety of modes of operation. The modality of figure 4 uses square wave DC energy in a differential input, in operation mode with reference mark; the mode of Figure 5 uses continuous DC power in a differential input, a continuous DC mode; the embodiment of Figure 6 utilizes square wave DC energy in a single-ended, single-ended entry mode; and the embodiment of Figure 7 uses continuous DC power in a single-ended continuous DC input mode. It is evident from FIGS. 4 to 7 that the control circuit 24 can be easily adapted for various different modes of operation. The four preceding modes of operation will be described in detail to demonstrate the design flexibility allowed by the invention. However, it must be recognized that the invention is in no way limited to these four modes of operation. The particular mode of operation as well as the source of energy used in a specific application depends mainly on the requirements and specifications of the controlled device. The areas and rectangles Bl and B2 in Figures 4 to 7 indicate the demarcation between contemplated components that are to be located on the IC chip 26 and components that are located outside the IC chip 26, such as electrodes El and E2, resistors R1 and R2, the controlled device (not shown) and the input and output lines 30 and 32, respectively. The portions of Figures 4 to 7 which are outside the areas in rectangles Bl and B2 are contemplated to be located on chip IC 26 and are identical for all of the four figures and modes of operation presented in this document. . The area in rectangle B6 contains the input portion of the control circuit. Various configurations of the input portion contained in the area in rectangle B6 are set forth with reference to figures Ha A 18E subsequently. Figures 4 to 7 illustrate a control circuit 24 comprising a start and polarization section 40, a pulse generating and logic circuit section 50, a decision circuit section 60 and a self-feeding inverter section 70, the functions of which will be described later. Each of the preceding circuit sections 40, 50, 60 and 70 can be designed in numerous different ways, as will be known to those skilled in the art in the design of electronic integrated circuits. The control circuit 24 also comprises a first, second and third transistors Pl, P2 and P3. In the embodiments described herein, transistors Pl through P3 are P-MOS devices, although N-MOS devices, bipolar devices or other types of transistors may also be used. The control circuit 24 further comprises an inverter II, a first, second and third diodes DI to D3, first and second capacitors Cl and C2, first, second, third and fourth transistor switches SW1 to SW4 and third and fourth resistors R3 and R4 It should be recognized that the third and the resistors R3 and R4 can be replaced with sources. of current or active loads. In each of the embodiments illustrated in Figures 4 to 7, the source terminal 37 of the third transistor P3 and the energy input terminals 41, 51, 61 and 71 of the start and polarization section 40, pulse a section 50 generator and logic circuit, a decision circuit 60 and a self-feeding inverter section 70, respectively, and are coupled directly to the PATILL terminal 8 of the IC chip 26. The PATILLA terminal 8 in turn is electrically coupled to the control circuit 24 of the energy input line 30 which in turn is electrically coupled to a power source 25. Typically, the power source 25 is located in the controlled device (not shown). A bias output terminal 43 from the starting and biasing section 40 is electrically coupled to the gate terminals G2 and G4 of a second and fourth transistor switches SW2 and SW4, respectively. In the preferred embodiment and as described herein with respect to Figures 4 to 7, the first to fourth transistor switches SW1 to SW4 are N-MOS devices. Although other types and combinations of transistors can also be used, as shown in Figures HA to 18E. An ignition reset output 44 from the starting and biasing section 40 is electrically coupled to an energy at the reset input 54 in the pulse generator and logic circuit section 50. The ignition reset output 44 of the starting and biasing section 40 is also electrically coupled to the gate terminals Gl and G3 of the first and third transistor switches SW1 and SW3. The reference output 42 connected to the internal ground of the starting and polarizing section 40 is electrically coupled to low potential plates 102 and 103 of the first and second capacitors Cl and C2, source terminals SI, S2, S3 and S4 from the first to the second. fourth transistor switches SW1 to SW4, respectively, the reference output 52 connected to internal ground of the pulse generator and logic circuit section 50, the reference output 62 connected to internal ground of the decision circuit 60, the anode 98 of the third diode D3, the low potential ends 96 and 97 of the third and fourth resistors R3 and R4 and the PIN terminal 6 of the IC chip 26. The node described in this manner in the following will sometimes be referred to as the reference CHIP VSS connected to internal ground. A pulse output 53 from the output 50 of the pulse generator and logic circuit section is electrically coupled to a terminal 80 and 81 source of the first and second transistors Pl and P2, respectively and to the PIN 2 terminal of IC 26. The terminal 82 of the first transistor Pl is electrically coupled to the PATILLE 1 terminal of IC 26. The gate terminal 83 of the second transistor P2 is electrically coupled to the PATILLE 3 terminal of IC 26. The drain terminal 84 of the first transistor Pl is coupled electrically to the anode 90 of the first diode DI and to a high potential end 94 of the third resistor R3. The drain terminal 85 of the second transistor P2 is electrically coupled to the anode 91 of the second diode D2 and the high potential end 95 of the fourth resistor R4. The cathode 92 of the first diode D I is electrically coupled to the input terminal 64 MAS (plus) of the decision circuit 60, terminals 86 and 87 drain the first and second transistor switches SW1 and SW2 and a high potential plate 100 of the first capacitor Cl. The cathode 93 of the second diode D2 is electrically coupled to the input terminal 66 LESS (minus) the decision circuit 60, terminals 88 and 89 drain the third and fourth transistor switches SW3 and SW4 and a high potential plate 101 of the second capacitor C2. The output 63 of the logic circuit of the decision circuit 60 is electrically coupled to the input 75 of the inverter II and to the input 73 of the inverter trigger of the self-feeding inverter section 70. The output 72 of the self-feeding inverter section 70 is electrically coupled to the PATILLE 4 terminal of IC 26. In the illustrated modes, the section 60 of the decision circuit is designed so that its output 63 is at a low potential when its inputs 64 and 66 MAS and LESS respectively are in substantially equal potentials or when the 66 LESS input is at a substantially greater potential than the 64 MAS input. The output 63 of the section 60 of the decision circuit is at a high potential only when the input 64 MAS is at a substantially higher potential compared to the input 66 LESS. The self-feeding inverter section 70 is designed so that no current flows through the inverting section 70 from the control circuit 24 of the power supply 25 to the reference CHIP VSS connected to the internal ground and through the third diode D3 when the output 63 of the logic circuit of section 60 of the decision circuit is at a low potential. However, when the output 63 of the logic circuit of the decision circuit section 60 is at a high potential, the inverter trigger input 73 is at a high potential and therefore the inverter trigger circuit 70 and enables the current to flow through the inverter section 70 from the power supply 25 of the control circuit 24 to the reference CHIP VSS connected to internal ground and through the third diode D3, by means of the energy input and output terminals 71 and 72 of the inverter 70, respectively. Once the inverter 70 has tripped, it remains tripped, or sealed and activated, until the power is removed from the control circuit 24. The design and construction of the inverting section which operates in a manner is known to those skilled in the art and need not be described in detail here. The output terminal 76 of the inverter II is electrically coupled to the gate terminal 78 of the third transistor P3. The drain terminal 79 of the third transistor P3 is electrically coupled to the PATILLE 7 terminal of IC 26. The third diode D3 is provided to prevent back-biasing of the control circuit 24 when the touch switch apparatus 20 is applied in multiplexed applications. It can be omitted in applications where only one touch pad 22 is used or where multiple touch pads 22 are used, but not multiplexed. The preceding description of the basic design of the control circuit 24 is identical for each of the four modes of operation presented in FIGS. 4 to 7. The distinctions in the general configuration of the apparatus between the four modes of operation are mainly based on the external terminal connections of IC 26, as will be described in detail in the following. Figure 4 illustrates a touch switch apparatus 20 configured for operation in the differential input reference mark mode, as described in the following. The control circuit 24 for operation in this mode is configured as described in the foregoing for figures 4 to 7 in a general manner. The PATILLA 2 terminal of IC 26 is electrically coupled to the high potential ends 104 and 105 of the first and second resistors Rl and R2 respectively. The PATILLA 1 terminal of IC 26 is electrically coupled to both the low potential end 106 of the first resistor Rl and the first electrode El. The PATILLA 3 terminal of IC 26 is electrically coupled to both the low potential end 107 of the second resistor R2 and the second electrode E2. The circuit elements represented as C3 and C4 in Figures 4 to 7 are not separate electrical components. Instead of this, the reference characters C3 and C4 represent the ground capacitance of the first and second electrodes El and E2, respectively. The PATILLE terminal 8 of IC 26 is electrically coupled to the input line 30 which in turn is electrically coupled to a power signal source 25, for example, in the controlled device (not shown). The PATILLA terminal 4 of IC 26 is electrically coupled to the PATILLE terminal 6 of IC 26 and thus electrically coupled to the output terminal 72 of the inverter 70 to the reference CHIP VSS connected to the internal ground and to the anode 98 of the third diode. D3. The PATILLA 7 terminal of the IC chip 26 does not externally terminate in this mode, the PATILLA 5 terminal of IC 26 is electrically coupled to the output line 32 which in turn is electrically coupled to the high potential end 108 of the fifth resistor R5 and the output line 120 which is connected to the controlled device (not shown) either directly or by means of a processor or other intermediate device (not shown). The low potential end 109 of the resistor R5 is electrically coupled to the system ground. In a typical application, the resistor R5 will be at a substantial distance from the other components comprising the touch switch apparatus 20. That is, in the preferred embodiment, the resistor R5 is contemplated to be not close to the touchpad 22 and the control circuit 24. Figure 5 illustrates a typical tactile switch control circuit 24 configured for operation in continuous differential input DC mode as described in the following. The circuit and control apparatus in general is identical to that described in Figure 4 above, with three exceptions. First, in the embodiment of FIG. 5, the PATILLE 7 terminal of IC 26 is electrically coupled to the high potential end 108 of the resistor R5 and to the output line 120, which is connected to the controlled device (not shown) and either directly or by means of a processor or other intermediate device (not shown) while the PIN-7 terminal does not terminate externally in the mode of FIG. 4. Second, in the embodiment of FIG. 5, the PIN-4 terminals and PIN 6 of IC 26 are not electrically coupled together or terminated externally in some other way, unlike as they are in the embodiment of Fig. 4. Third, in the embodiment of Fig. 5, the PIN-5 terminal of IC 26 is electrically coupled to the low potential end 109 of the resistor R5, while in the embodiment of FIG. 4, the PIN 5 terminal of IC 26 is electrically coupled to the high potential end 108 of the fifth resistor and controlled device (not shown). As in the embodiment of Figure 4, the fifth resistor R5 will typically be at a substantial distance from the other components comprising the touch switch apparatus 20. Figure 6 illustrates a typical tactile switch control circuit configured for operation in a mode with single-ended input reference mark, as described in the following. The control circuit 24 is configured as described in the foregoing for figures 4 to 7 in a general manner. The PATILLA 2 terminal of IC 26 is electrically coupled to the high potential ends 104 and 105 of the first and second resistors Rl and R2, respectively. The PATILLA 1 terminal of IC 26 is electrically coupled to both the low potential end 106 of the first resistor R1 and the first electrode El. The PATILLA 3 terminal of IC 26 is electrically coupled to both the low potential end 107 of the second resistor R2 and the high potential end 110 of the sixth resistor electrode R6, so that the second resistor R2 and the sixth resistor R6 of a splitter of voltage. The low potential end 111 of the sixth resistor R6 is electrically coupled to the reference CHIP VSS connected to the internal ground, typically at a point near the PATILLE 5 terminal of IC 26. In Figure 6 the electrical connection of the sixth resistor R6 to the reference CHIP VSS connected to internal ground is represented by a dashed line "A-A" for clarity. The PATILLE terminal 8 of IC 26 is electrically coupled to the input line 30, which in turn is electrically coupled to the power signal source 25. The PATILLA terminal 5 of IC26 is electrically coupled to the output line 32, which in turn is electrically coupled to the high potential end 108 of the fifth resistor R5 and to the output line 120. The output line 120 is electrically coupled to the controlled device (not shown), either directly or by means of a processor or other intermediate device. The PATILLA terminal 4 of IC 26 is electrically coupled to the PATILLE 6 terminal of IC 26. The PATILLA 7 terminal of IC 26 does not terminate externally in this mode. In a typical application, the fifth resistor R5 will be at a substantial distance from the other components comprising the touch switch apparatus 20. Figure 7 illustrates a typical tactile switch control circuit configured for operation in continuous single-ended input DC mode, as described in the following. The control circuit 24 is configured as described in the foregoing for figures 4 to 7 in a general manner. The circuit and control apparatus in general is identical to that described in Figure 6 above, with three exceptions. First, in the embodiment of FIG. 7, the PATILLE 7 terminal of IC 26 is electrically coupled to the high potential end 108 of the fifth resistor R5 and to the output line 120 which in turn is connected to the controlled device (not shown). ), typically by means of a microprocessor or other controller (not shown). The PATILLA 7 terminal of IC 26 does not terminate externally in the embodiment of Figure 6. Secondly, in the embodiment of Figure 7, the PIN 4 and PATILLE 6 terminals of IC 26 do not electrically couple or terminate in some other way. externally, as they do in the embodiment of figure 6. Third, in the embodiment of figure 7, the PATILLE terminal 5 of IC 26 is electrically coupled to the low potential end 109 of the fifth resistor R5 while in the embodiment of Figure 6, the PATILLE 5 terminal of IC 26 is electrically coupled to the high potential end 108 of the fifth resistor and the output line 120. In a typical application, the fifth resistor R5 will be at a substantial distance from the other components comprising the touch switch apparatus 20. In Figure 7, the electrical connection of the sixth resistor R6 to the reference CHIP VSS connected to internal ground is represented by a dashed line "A-A" for clarity. A touch switch apparatus 20 configured for the mode with differential input reference mark operates as follows. With reference to Figure 4, an energy / control signal 25 is provided to the PATILLE terminal 8 of IC 26 and, in turn, to the power input terminals 41, 51, 61 and 71 of the starting section 40. and polarization, the pulse generator and logic circuit section 50, the decision circuit section 60 and the self-feeding inverter section 70, respectively. As it is equipped with energy and after a suitable delay interval to allow stabilization (approximately 25 microseconds are sufficient but may be shorter or longer, depending on the application), the starting and polarizing section 40 transmits a re-establishment signal short-term ignition from the output terminal 44 to the gate terminals Gl and G3 of the first transistor switch SW1 and the third transistor switch SW3, respectively, which causes the first and third transistor switches SW1 and SW3 to be turn on and therefore provide a current path from the high potential plates 100 and 101 of the first and second capacitors Cl and C2, respectively, to the reference CHIP VSS connected to internal ground. The duration of the ignition reset signal is sufficient to allow any charge present in the first and second capacitors Cl and C2 to be substantially completely discharged to the reference CHIP VSS connected to the internal ground. In this way, inputs 64 and 66 MAS and LESS to section 60 of the decision circuit reach a state of initial low potential. Substantially at the same time, the starting and biasing circuit 40 sends an ignition reset signal from the output 44 to the input 54 of the pulse generator and logic circuit section 50, and thus initializes it. After a suitable delay to allow the pulse generator and logic circuit section 50 to stabilize, the pulse generator and logic circuit section 50 generates a pulse and transmits it from the pulse output terminal 53 to the first and second electrodes The and E2 by means of the first and second resistors R1 and R2 and to the source terminals 80 and 81 of the first and second transistors Pl and P2, respectively. The pulse can be of any suitable waveform, such as a square wave pulse. The starting and biasing circuit 40 also transmits a bias voltage from the bias output 43 to the gate terminals G2 and G4 of the second and fourth transistor switches SW2 and SW4, respectively. The bias voltage is out of phase with the pulse output for the first and second electrodes El and E2. That is, when the pulse output is in a high state, the bias voltage output is in a low state, and when the pulse output is in a low state, the bias voltage output is in a high state .

When a pulse is applied to the first and second electrodes El and E2 through first and second resistors R1 and R2, respectively, the voltage at gate terminals 82 and 83 of the first and second transistors Pl and P2 is initially at a potential less than that of the source terminals 80 and 81 of the first and second transistors Pl and P2 respectively, and therefore polarize the first and second transistors Pl and P2 causing them to turn on. With the first and second transistors Pl and P2 turned on, current will flow through the third and fourth transistors R3 and R4, and therefore a peak potential is generated at the anode terminals 90 and 91 of the first and second diodes DI and D2 , respectively. If the peak potential and the anodes 90 and 91 of the first and second diodes DI and D2 are greater than the potential through the first and second capacitors Cl and C2, a peak current is established through the first and second diodes DI and D2. which causes the first and second capacitors Cl and C2 to charge and stabilize a peak potential of each of inputs 64 and 66 MAS and LESS to section 60 of the decision circuit. This situation will occur, for example, after pressing for the first time after the control circuit 24 has been initialized because the first and second capacitors Cl and C2 open discharged from the start, as described above. As is evident to a person skilled in the art, the polarization of the first and second transistors Pl and P2, the current through the third and fourth resistors R3 and R4, the peak potential generated in the anodes 90 and 91 of the first and second diodes DI and D2 as well as the peak potential generated in each of the inputs 64 and 66 MAS and LESS to the decision circuit 60 are proportional to the condition of the electric field in the first and second electrodes El and E2. The condition of the electric field near the electrodes El and E2 will vary in response to the stimuli present near the electrodes. With the control circuit 24 activated, as described above and without present stimuli next to either the first and second electrodes El and E2, the potentials in each of the inputs 64 and 66 MAS and LESS to the decision circuit 60 are in what is called a neutral state. In the neutral state, the potentials of each of the inputs 64 and 66 MAS and LESS can be substantially equal. However, in order to avoid unwanted drives, it may be desirable to adjust the control circuit 24 so that the neutral state of the LESS input 66 is at a slightly higher potential than the neutral state of the MAS 64 input. This adjustment can be carried out by making several configurations of the first and second electrodes El and E2 and the values of the first and second resistors R1 and R2 to obtain the desired neutral state potentials. Regardless of the neutral state potentials, it is contemplated that the output 63 of the decision circuit 60 be at a low potential unless the 64 MAS input is at a substantially higher potential than ... With the 63 output of the 60 circuit of decision at a low potential, the inverter II causes the potential at the gate terminal 78 of the third transistor P3 to be at a high level, substantially equal to the potential at the source terminal 77. In this state, the third transistor P3 will not be polarized and will remain off. However, in this mode, the PATILLA 7 terminal of IC 26 is not finished. The drain terminal 79 of the third transistor P3 is therefore in an open circuit condition and the state of the third transistor P3 has no consequence for the function of the apparatus. Furthermore, with the output 63 of the decision circuit 60 and consequently with the input 73 of the inverter trigger, in a low state, the self-feeding inverter circuit 70 will not trip and no current will flow through the inverter 70 from the supply 25 of energy to the reference CHIP VSS connected to internal ground and through the third diode D3. During a period of time which is determined by the pulse voltage, the values of the first and second resistors Rl and R2, and the ground capacitance of the first and second electrodes El and E2 (represented in the figures as virtual capacitors C3 and C4), the potential in the first and second electrodes El and E2 finally they rise to be substantially equal to the pulse voltage and therefore to the voltage at terminals 80 and 81 sources of the first and second transistors Pl and P2, therefore depolarize the first and second transistors Pl and P2. When this state is reached, the first and second transitions Pl and P2 are inactivated and the potentials at the anodes 90 and 91 of the first and second diodes DI and D2 start to decrease at a substantially equal rate towards the level of the connected reference CHIP VSS to internal earth. Finally, the potential of the anode in each of the first and second diodes DI and D2 is likely to fall below the respective cathode potential. At this point, diodes DI and D2 reverse polarize and prevent the first and second capacitors Cl and C2 from discharging. When the pulse at output 53 advances to a low state, the bias voltage output advances to a high state relative to the reference CHIP VSS connected to internal ground and applies the high bias voltage to gate terminals G2 and G4 of the second and fourth transistor switches SW2 and SW4. In this state, the second and fourth transistor switches SW2 and SW4 are slightly polarized and light enough to carry out a slow and controlled discharge of the first and second capacitors Cl and C2 to the reference CHIP VSS connected to the internal ground. When the pulse then advances to a high state, the bias voltage will return to a low state, the second and fourth transistor switches SW2 and SW4 will turn off and the circuit will respond as initially described. If a stimulus is present at or near the second electrode E2 when the pulse of the pulse generating and logic circuit section 50 advances to a high potential, first the transistor Pl will operate as described above. That is, the first transistor Pl will initially be polarized and will allow some current to flow through the third resistor R3, which generates a peak potential at the anode 90 of the first diode DI and which allows the peak current to flow through the first diode DI, and therefore the first capacitor Cl is charged and a peak potential is established at the 64 MAS input for the decision circuit 60. Once the voltage on the first electrode El has stabilized in response to the incoming pulse, the first transistor Pl will be unpolarized and shut off. The second transistor P2 operates in a very similar manner, except that the presence of the second electrode E2 close to the stimulus will alter the RC time constant for that circuit segment and therefore lengthens the time it takes for the potential to stabilize. the second electrode E2. As a consequence, the second transistor P2 will remain polarized for a longer period of time compared to the first transistor Pl, which allows a greater peak current to flow through the fourth resistor R4 than does the one flowing through the third resistor R3 and therefore generates a potential peak at the anode 91 of the second diode D2 which is greater than the peak potential present at the anode 90 of the first diode DI. Consequently, a peak current will flow through the second diode D2, which causes the second capacitor C2 to be charged, and which ultimately results in a peak potential at the input 66 LESS to the decision circuit 60 which is larger than the peak potential at the 64 MAS input to the decision circuit. Since the decision circuit 60 is configured so that its output will be at a low potential when the potential at the LESS input 66 is greater than or substantially equal to the potential at the 64 MAS input of the decision circuit 60, the exit terminal 63 will be at a low potential. With the decision circuit 60 of the output terminal 63 and consequently the inverter trigger input terminal 73, at a low potential, the self-feeding inverter 70 will not be activated. The inverter II and the third transistor P3 will operate as previously described although, again, the state of the third transistor P3 has no relevance in this configuration. In the event that a contaminant or foreign object or other stimulus substantially covers or applies to both the first and the second electrodes El and E2, the system will respond in a very similar way in the case where a stimulus is not present either in the first electrode or in the second electrode. However, with contaminants or a foreign object present near both electrodes El and E2, the RC time constant for these segments of the circuit will be altered in such a way that they will require more time for the voltage at the first and second electrodes El and E2, respectively, to be substantially equal to the pulse voltage. Accordingly, the first and second transistors Pl and P2 will turn on and allow more current to flow through the third and fourth resistors R3 and R4 than would be expected in a condition where none of the first or second electrodes El or E2 are affected by a stimulus. However, the first and second transistors Pl and P2 will be substantially polarized equally. Therefore, a substantially equal peak potential will develop at anodes 90 and 91 of both the first and the second diodes DI and D2, which causes a substantially equal peak current to flow through the first and second diodes DI and D2, charging the first and second capacitors Cl and C2 and establishing a substantially equal peak potential in both inputs 64 and 66 MAS and LESS to the decision circuit 60. In this state, the section 60 of the decision circuit of the output terminal 63 will be at a low potential, the inverter activation input terminal 73 of the self-feeding swingarm 70 will be at a low potential and the inverter 70 will remain without activation. As previously described, the state of the inverter II and the third transistor P3 is not relevant in this mode. In the situation where a stimulus is applied close to the first electrode El, but not to the second electrode, the second transistor P2 will initially be biased and turned on, establishing a current through the fourth resistor R4 and generating a peak potential at the terminal 90 of anode of the second diode D2. A peak current will flow through the second diode D2, charging the second capacitor C2 and setting a peak potential at the input 66 LESS in section 60 of the decision circuit. As the voltage at the gate terminal 81 of the second transistor P2 rises to the level of the pulse voltage, the second transistor P2 will become non-polarized and turn off. The second diode D2 will then reverse polarize and prevent the second capacitor C2 from discharging. As is evident to a person skilled in the art, the presence of a stimulus close to the first electrode will lengthen the time necessary for the potential to stabilize at the first electrode El. As a consequence, the first transistor Pl will remain polarized for a longer period of time compared to the second transistor P2, which allows a larger peak current to flow through the third resistor R3 than through the fourth resistor R4, and therefore a potential peak is generated at the anode 90 of the first diode DI, which is greater than the potential present at the anode 91 of the second diode D2. Consequently, a peak current of greater magnitude or duration will flow through the first DI diode as compared to that flowing through the second diode D2, which causes the first capacitor Cl to be charged and which ultimately results in a peak potential at the 64 MAS input to the decision circuit 60 which is substantially greater than the peak potential at the 66 input LESS to the decision circuit 60. Since the decision circuit 60 is configured so that the output terminal 63 will be in a high state when the potential at the input 64 MAS is greater than the potential at the input 66 LESS, the output 63 of the decision circuit 60 will be at a high potential. With the output 63 of the decision circuit 60 at a high potential, the inverter II will cause the potential at the gate terminal 78 of the third transistor P3 to be low relative to the potential at the source terminal 77, and therefore polarize to the third P3 transistor which causes it to turn on. However, since the PATILLA 7 terminal of IC 26 is not terminated in this mode, the state of the third transistor P3 is not relevant.

With the output terminal 63 of the decision circuit 60 at a high potential, the self-powered inverter circuit 70 activates the input terminal 73 and will also be at a high potential, and therefore activates the inverter 70. When the inverter is activated 70 of self-feeding, a current path is established from the power supply 25 to the reference CHIP VSS connected to the internal ground and through the third diode D3, effectively placing in short circuit the rest of the control circuit 24, which includes the section 40 of start and polarization, section 50 of pulse generator and logic circuit and section 60 of decision circuit. In this state, those sections of the control circuit 24 are substantially de-energized and stop functioning. Once activated, the self-feeding inverter 70 will remain activated, regardless of the subsequent state of the stimuli proximate to either or both of the El and E2 electrodes. The inverter 70 will be reset when the power from the power supply 25 advances to an almost zero state, for example when the square wave reference mark signal of the power supply 25 of this example drops to zero. Although the self-powered inverter 70 is in an activated state, a steady state signal will be supplied through the fifth resistor R5 and back to the controlled device (not shown). In this way, the touch switch apparatus 20 emulates the state change associated with a mechanical contact switch maintained.

Referring now to FIG. 5, the operation of a touch switch apparatus 20 configured for a differential DC continuous input mode as follows. The control circuit 24, up to and including the decision circuit 60, operates in substantially the same manner as when configured for the operation mode with differential input reference mark as described above with reference to FIG. 4. That is, without present stimulus next to either the first or second electrodes El and E2, with a stimulus present next to both the first and second electrodes El and E2 or with a stimulus present close to the second electrode E2, but not to the first electrode. , the output 63 of the decision circuit 60 will be at a low potential. With a stimulus present close to the first electrode El but not to the second electrode E2, the output 63 of the decision circuit 60 will be at a high level. As can easily be seen in Figure 5, the output 72 of the self-feeding inverter circuit 70 does not end in this mode, and the self-feeding inverter 70 is therefore not operative in the differential input DC mode. However, the drain terminal 79 of the third transistor P3 is electrically coupled to the reference CHIP VSS connected to the internal ground and to an output line 32 in this mode and therefore becomes an operative part of the control circuit 24. When the output 63 of the decision circuit 60 is at a low potential, the inverter II causes the potential at the gate terminal 78 of the third transistor P3 to be at a high potential, substantially equal to the terminal 77 of the potential source. In this state, the third transistor P3 is not polarized and does not turn on. When the output 63 of the decision circuit 60 is at a high potential, the inverter II causes the potential at the gate terminal 78 of the third transistor P3 to be at a low potential as compared to the potential at the source terminal 77. In this state, the third transistor P3 is biased and turned on, which allows current to be established through the third transistor P3 and the fifth resistor R5. The output line resistor R5 limits the current through the third transistor P3 in such a way that the balance of the control circuit 24 is not shorted and remains operative. In the DC mode shown in figure 5, the control circuit 24 also responds to the withdrawal of the stimulus from the proximity of the first electrode El. To the extent that the stimulus remains present close to the first electrode El but not to the second electrode E2, each time the pulse advances to a high state, a peak potential will be generated at the anode 90 of the first DI diode which is greater than the peak potential at the anode 91 of the second diode D2. Accordingly, the peak potential at the 64 MAS input to the decision circuit 60 will be at a level greater than the peak potential at the 66 LESS input and the control circuit 24 will behave as described above. However, when the stimulus is removed and when there is no stimulus present next to either the first electrode El or the second electrode E2, the load on the first capacitor Cl will eventually be discharged to a neutral state by means of the polarization function of the second transistor switch SW2. At this point, the potential at the 64 MAS input of the decision circuit 60 will no longer be high or substantially higher than the potential at the LESS input 66 and the output 63 of the decision circuit 60 will return to a low state. In this way, the touch switch apparatus 20 operating in a differential input DC mode emulates a momentary contact of a mechanical push switch to close and release to open. It should be recognized that, with minor revisions, the control circuit can be configured to emulate a push-type mechanical switch to open and release to close. Referring now to Figure 6, the touch switch apparatus 20 configured for the operation mode with single-ended input reference mark will operate as follows. When a pulse is applied to the first electrode El and the first and second resistors El and E2, the current flows through the second resistor R2 and the sixth resistor R6. The second and sixth resistors R2 and R6 are configured as a voltage divider; that is, when the pulse output is in a high state, the gate terminal 83 of the second resistor P2 will be at a smaller potential than the source terminal 81 of the second transistor P2. Therefore, when the pulse output 53 is in a high state, the second transistor P2 will be continuously biased and will allow a constant current to flow through the fourth resistor R4 and therefore a reference potential at the anode 91 is generated. of the second diode D2. The reference potential at the anode 91 of the second diode D2 will establish a current through the second diode D2, which causes the second capacitor C2 to be charged and therefore a reference potential is generated at the input 66 LESS to the circuit 60 of decision When the reference potential at the LESS input 66 becomes substantially equal to the reference potential at the anode 91 of the second diode D2, the current through the second diode D2 will cease. Concurrently, without stimulus present in the first electrode El, the pulse applied to the terminal 80 source of the first transistor Pl and the first electrode will initially cause the first transistor Pl to become polarized and turn on. In this way a current will be established through the third resistor R3 and a peak potential will be generated at the anode 90 of the first DI diode. The peak potential will establish a peak current through the first DI diode, charging the first capacitor Cl and creating a peak potential at the 64 MAS input of the decision circuit. Resistors R1, R2, R3, R4 and R6 are selected such that when a proximal stimulus is not present, the first electrode El, the reference potential at the LESS input 66 of the decision circuit 60 will be greater than or equal to the peak potential. in terminal 64 MAS of decision circuit 60. In this state, the output 63 of the decision circuit 60 will be at a low potential and the self-feeding inverter 70 will not be activated. In addition, the inverter II will cause the potential at the gate terminal 78 of the third transistor P3 to be in a high state, substantially equal to the potential of the source terminal 77, so that the third transistor P3 is not biased and remains off. However, this is not relevant since the drain terminal 79 of the third transistor P3 is in an open circuit condition in this mode. This mode does not require a second electrode, although a two electrode touch pad can be adapted for use in this mode. In the case in which a two-electrode touchpad is adapted for use in this mode of operation, the presence or absence of a stimulus proximate to the second electrode has no effect on the operation of the circuit. In the event that a stimulus is present close to the first electrode El, the operation of the second transistor P2 is the same as that described above for this modality. However, the presence of the stimulus near the first electrode El will cause a longer time for the voltage at the gate terminal 82 of the first transistor Pl to be equalized to match the potential of the source terminal 80 in the first transistor. Accordingly, the first transistor Pl will turn on and allow a relatively greater current to flow through the third resistor R3, as compared to the current that allows the second transistor P2 to flow through the fourth resistor R4. As a result, the peak potential at the anode 90 of the first diode DI will be greater than the reference potential at the anode 91 of the second diode D2. As a result, the peak potential at the 64 MAS input of the decision circuit 60 will be larger than the reference potential at the LESS input 66 of the decision circuit 60 and the output 63 of the decision circuit 60 will therefore be in a state high. With the output 63 of the decision circuit 60 in a high state, the inverter II causes the potential in the gate terminal 78 of the third transistor P3 to be in a low state, and therefore turn on transistor P3. However, since the drain terminal 79 of the third transistor P3 has not ended effectively, this is not relevant. With the output 63 of the decision circuit 60 in a high state, the inverter trigger input 73 is in a high state and the self-feeding inverter 79 is activated and therefore a current path is established through the section 70 of inverter, from the power supply 25 to the reference CHIP VSS connected to internal ground and through the third diode D3, by which it effectively short-circuits the balance of the control circuit 24. The self-powered inverter 70 will remain in this state until energy is removed in the inverter input terminal 71. Until the inverter 70 is reset in this manner, a continuous digital control signal is transmitted to the controlled device (not shown). In this way, the touch switch apparatus 20 emulates a state change associated with a mechanical switch. Referring now to FIG. 7, a touch switch apparatus 20 configured for operation in single-ended continuous DC input mode operates as follows. The operation and functionality of the control circuit 24 is substantially the same as that described for single-ended input, in the mode with reference mark as described above with reference to FIG. 6. However, in the mode Single-ended input DC, output 72 of the self-feeding inverter is of an open circuit and the self-feeding inverter 70 is therefore not operative. If stimulus applied to the first electrode El, the output 63 of the decision circuit 60 is at a low potential. Accordingly, the output 76 of the inverter II to the gate terminal 78 of the third transistor P3 is at a high potential. With the gate terminal 78 of the third transistor P3 at a high potential, similar to the potential at the source terminal 77, the third transistor P3 is unpolarized and does not turn on, and therefore no current flows through the third transistor P3 or through the fifth resistor R5. With a stimulus close to the first electrode El, the output 63 of the decision circuit 60 and consequently the input 75 to the inverter II is in a high state. The inverter II changes the input of the high level to a low level output and provides an output 76 to the potential of the gate terminal 78 of the third transistor P3. With the gate terminal 78 at a low potential as compared to the source terminal 77, the third transistor P3 is biased, turned on and the current flows through the third transistor P3 and the fifth resistor R5. This generates a high potential at the anode 108 of the fifth resistor R5 which can be used as an input to the controlled device (not shown) via the output line 120. In the continuous DC mode of FIG. 7, the control circuit responds to the withdrawal of the stimulus from the proximity of the first electrode El. Insofar as the stimulus remains present close to the first electrode El, each time the pulse advances to the high state, the peak potential will be created at the anode 90 of the first DI diode which is greater than the reference potential at the anode 91 of the second diode D2. Accordingly, the peak potential at the 64 MAS input to the decision circuit 60 will be at a level greater than the reference potential at the LESS input 66 and the control circuit 24 will behave as described above. When the stimulus is removed from the first electrode El, the load on the first capacitor Cl will finally be discharged to a neutral state by means of the biasing function of the second transistor switch SW2. At this point, the peak potential at the 64 MAS input of the decision circuit 60 will no longer be greater or substantially greater than the reference potential at the 66 LESS input or the 63 output of the decision circuit 60 will return to a low state. In this way, the touch switch apparatus 20 operating in single-ended DC input mode emulates a momentary metal contact switch. With minor revisions, the control circuit can be configured to emulate a mechanical push switch to open and release to close. So far, this specification has described the physical construction and operation of a unique touch switch. Typical applications of a touch switch often involve a plurality of touch switches which are used to carry out a control on a device. Figure 10 shows a panel of switches comprising nine touch switches 20, wherein the nine touch switches 20 are distributed in a three-by-three array. Rectangle B4 represents components of the touch panel, while rectangle B5 represents components in the controlled device. Although theoretically any number of tactile switches can be placed in any way, matrix distributions such as this are easily multiplexable, reducing the number of input and output lines required of the controlled device, and are the preferred ones. Rectangle B6 in Figure 4 presents an input portion of a touch switch control circuit which includes active devices Pl and P2, diodes DI and D2, resistors R3 and R4 and capacitors Cl to C2. Figures HA to 18E present other configurations for the input portion of a touch switch control circuit that involves active devices and peak detector circuits that satisfy some of the objectives described in the foregoing of the present invention that include the supply of damping low impedance, reduction in the size and cost of the integrated circuit, linearization of the input signals, dissipation of the parasitic capacitance and blocking of the damage of the current trajectories. The configurations presented in FIGS. HA to 18E correspond basically to the configuration in the area in rectangle B6 of FIG. 4 as will be understood by those skilled in the art of circuit design. Specifically, the active devices Ml and M2 in figure HA, for example, correspond to the active devices Pl and P2 in figure 4; the active devices Ql and Q2 in the figures HA to 18E correspond to the diodes DI and D2 in figure 4; the resistors R7 and R8 in FIG. HA, for example, correspond to the resistors R3 and R4 in FIG. 4; and capacitors C9 and CIO in FIGS. HA to 18E correspond to capacitors Cl and C2 in FIG. 4. In addition, electrodes El and E2 and resistors Rl and R2 are the same in FIG. 4 as in those of FIGS. HA to 18E where they appear. The OSCB, I RNG and O_RNG pins in which they are shown in Figures HA to 18E, when presented, correspond to the pins PATILE 2, PIN 1 and PIN 3 of Figure 4. The switches SW2 and SW4 in Figure 4 correspond to active devices M3 and M4 in figure HA, for example. The discharge signal DSCHGB in FIGS. HA to 18E corresponds to the current polarization in trace 43 from the start and polarization circuits 40 of FIG. 4. The POS and NEG lines of FIGS. HA to 18E correspond to lines 64 and 66 of figure 4, respectively. Finally, the trace OSCB in the figures HA to 18E corresponds to the trace 53 of the pulse generator and the logic circuits 50 of figure 4. In this way, the input portions of the figures HA to 18E can be understood to be compatible with the circuit configurations described with reference to figures 4 to 7. Figure HA illustrates the inner electrode El and the outer electrode E2 electrically coupled to an oscillating signal generator OSCB through the OSCB pin and the resistors Rl and R2 , respectively. Figure HA further shows capacitance C6 between electrodes. Capacities C7 and C8 which represent a binding pad and inherent wiring bond capacitances when coupling electrical components to an integrated control circuit are also shown. The capacitances C7 and C8 can also represent other capacitances due to metallization under the stroke, traces of redistribution and the like in micropad and other applications that do not involve bond pad wires as are known to those skilled in the art. In FIG. HA, the electrodes El and E2 are electrically coupled to the input portion of the touch switch control circuit in the gates of the active devices M1 and M2, respectively, through the pins I RNG and O RNG respectively. In Figure HA, the active devices Ml and M2 are shown as MOSFET devices of type N. The drains of the active devices Ml and M2 are electrically coupled to the voltage source VDD through resistors R7 and R8, respectively and their sources to OSCB oscillating signal. The drains of the active devices Ml and M2 are also electrically coupled to the respective peak detection circuits consisting of active devices M3, M4, Q1 and Q2 and capacitors C9 and CIO which, as discussed above, correspond to the peak detection circuits shown in Figure 4, which have switches of components SW2 and SW4, diodes DI and D2 and capacitors Cl and C2, except that, since the active input devices Ml and M2 are active devices N-MOS, wherein the active devices Pl and P2 in figure 4 are P-MOS devices, the capacitances C9 and CIO and the sources of the active devices Ml and M2, through the resistors R7 and R8 are coupled to the VDD signal, instead of the VSS voltage signal. The peak detection circuit in FIG. HA associated with the active device Ml includes the active device Ql, whose base is electrically coupled to the source of the active device Ml via the SNEG trace and also, through the resistor R7, to the signal of voltage VDD, the emitter of which is electrically coupled to the drain of the active device M3 and to the capacitor C9, and the pick-up of which is coupled to the voltage signal VSS; the capacitance C9, one terminal of which is electrically coupled to the voltage source VSS and the other terminal of which is electrically coupled to the emitter of the active device Q1 and the drain of the active device M3; the active device M3, the drain of which is electrically coupled to the emitter of the active device Q1, the source of which is coupled to the voltage source VDD and the base of which is electrically coupled to the discharge signal DCHGB. The configuration of the peak detection circuit associated with the active device M2 is analogous and involves the active devices Q2 and M4 and the CIO capacitance. In FIG. HA, the active devices Ql and Q2 are bipolar P-type transistors and the active devices M3 and M4 are M-type P-devices. The emitters of the active devices Ql and Q2 are electrically coupled as inputs to the circuit component of the device. decision (not shown) of the control circuit through NEG and POS traces, respectively. The operation of the decision circuit component is as described above with respect to Figures 4 to 7. In Figure HA, the resistors R7 and R8 serve to convert the drain currents to voltages in the drains of the active devices Ml and M2, respectively. These voltages are related to changes in the electric fields of the El and E2 electrodes caused by tactile or other stimuli. The voltage potential at the respective nodes of the drains of the active devices Ml and M2 are communicated to the peak detectors via the SNEG and SPOS traces, respectively. The peak detectors can convert the peak negative value of very fast transient pulses in the SPOS and SNEG traces to DC signals of the POS and NEG traces, respectively, which makes it easier for the decision circuit to process it. Therefore, Figure HA illustrates a dual input system having negative pulse peak detection circuits. A similar positive pulse peak detection system is described in the US patent. number 5,594,222 for a single channel. The detection circuit that generates these negative pulses can include a N-type MOSFET device that can be capable of low and high speed pulling and a high current source that is extracted in a smoother manner. The active devices Ml and M2 in figure HA will turn on and off, by oscillating the OSCB signal communicated through both electrodes El and E2 and the I_RNG and O_RNG pins, to provide transient negative advance pulses in the SNEG and SPOS traces , respectively. The negative peak levels of these pulses will be proportional to the strength of the electric fields at the input electrodes El and E2, which can change when the electrodes El and E2 are stimulated by touch or in some other way. The signals in the SNEG and SPOS traces are then communicated to the respective bases of the active devices Q1 and Q2 of the peak detection circuits corresponding to the active devices M1 and M2. A low signal communicated to the bases of the active devices Ql and Q2 will deviate them by activating them and will present the maximum negative voltage in the drains of the active devices Ml and M2 to NEG and POS traces, respectively. The C9 and CIO capacitors initially loaded in VDD, decrease the speed of this voltage change in the POS and NEG traces and therefore convert the transient pulses of the SPOS and SNEG traces to DC pulses in the POS and NEG traces, as show in the synchronization diagram of figure HA. The active devices Q1 and Q2 then isolate the capacitors C9 and CIO from the load once the transient signal has elapsed. The active devices M3 and M4, controlled by the DCHGB download signal, can then restore the initial VDD load of the C9 and CIO capacitors, respectively. Using pulses of short duration advantageously allows the touch sensor to maintain a low impedance. In addition, the control circuit consumes low power on average. For example, the peak current through the capacitance of the input electrode can be as high as several milliamperes. This may correspond to a very low impedance during the period of time when the peak current persists. If each pulse was active even for 20 nanoseconds and was sampled once every 50 microseconds, then the average continuous current would be 0.8 microamps for each channel and 1.6 microamps for both channels. In addition, the input portion provides statistical filtering and periodic sampling of the detected signals when the DCHGB download signal is not active. These low impedance and low power consumption characteristics can improve the performance of stimulus interpretation of the touch sensor, as described in the U.S. patent. No. 5,594,222 and it may be advantageous when replacing mechanical switches, membrane switches and the like with tactile sensing devices. Mechanical and other true switches do not allow the current to pass when they are open. Solid switches with low impedance and low power that mimic the characteristics of true switches in this way can allow the direct replacement of mechanical switches without the risk of unacceptable amounts of leakage current through an "open" solid-state switch . In addition, the peak detector circuits of the low impedance and low power touch switches are compatible with the use of relatively low gain bandwidth product amplifiers and op amplifiers in the decision and other circuits and DC and a relatively low gain and low bandwidth devices for signal generating circuits. Figure 11B shows an input portion of an integrated control circuit where the active devices M and M2 are P-type MOSFET devices, active devices M3 and M4 are N-type MOSFET devices and the active devices Q1 and Q2 are N-type bipolar devices Figure 11B otherwise has the same configuration as Figure HA, except that the resistors R7 and R8 and the sources of active devices M3 and M4 are coupled to the voltage signal VSS and the collectors of the active devices Q1 and Q2. they are coupled to the voltage source VDD. Figure HB in this manner illustrates a modality utilizing positive forward and DC transient pulses, as shown in the synchronization diagram of the HB figures. Figures HC and HD show the input portions where the active devices Ml and M2 of figure HA have been replaced by active devices Q3 and Q4 which are of type N in figure HC and of type P in figure HD. Figure HC shows the peak detection circuit of figure HA, which involves active devices type P, Ql, Q2, M3 and M4, and figure HD shows the peak detection circuit of figure HB, whose active devices are all Type-N devices. The operation of these input portion configurations is parallel to the operation described above with respect to the HA figures and will be understood by those skilled in circuit design art. Figures HA to HD show the use of resistors R7 and R8 which provide for the conversion of the drain and collector currents (of any of the active devices M1 and M2 or Q3 and Q4, respectively) to voltages proportional to the current in the drain or collector. Thus, in the HA to HD figures, this drain or collector voltage will be equal to V- (Ir) (R). Other ways to provide this voltage conversion are shown in Figures 12A to 15D. In these drawings, resistors R7 and R8 have been replaced with active devices. The use of active devices as current-to-voltage converters, as shown in Figures 12A to 12D, for example, allows high gain outputs with replacement of resistive components and conserves the space of the integrated circuit. Figures 12A to 12D generally correspond to figures HA to HD, respectively. In Figures 12A to 12B, the resistors R7 and R8 of the Figures HA to HB have been replaced by MOSFET devices M5 and M6, where, in Figures 12C to 12D, the resistors R7 and R8 of the Figures HC to HD are have been replaced by bipolar devices Q5 and Q6. Figures 13A to 13D generally correspond to Figures 12A to 12D except that the current sources of the P-type active device of Figures 12A to 12D have been replaced with N-type active device current sources in Figures 13A to 13D (and similarly, the N-type active device current sources of Figures 12A to 12D have been replaced with P-type active device current sources in Figures 13A to 13D). Since the active loads are of the same type as the input devices in Figures 13A to 13D, these active devices can be incorporated into the integrated circuit during the same manufacturing step. This provides a better match. The output gain is determined by the size of the device and the reference voltage, Vref used. Vref can be established by a polarization circuit that allows the currents to be reflected by expanding the sizes of the gate widths, when using MOSFET devices, or emitting areas, when using bipolar devices. In the embodiments shown in Figures 12e to 12H and 13E to 13H, the resistors R7 and R8 of the HA to HD figures have been replaced with the active devices M5 and M6 (Figures 12E to 12F and 13E to 13F) or Q5 and Q6 (Figures 12G to 12H and 13G to 13H) as well as cascading active devices M7 and M8 (Figure 12E to 12F and 13E to 13F) of Q7 and Q8 (Figures 12G to 12H and 13G to 13H). In this way, cascade polarization helps to immunize the control circuit against energy supply and process variations. Figures 14A to 14D show modalities using complementary device types. For example, in Figure 14A, the active square root extraction M9 and MIO devices are P-type MOSFET devices and the active input devices M1 and M2 are N-type MOSFET devices. Figures 14B to 14D show modalities that use types of complementary devices which correspond to figures HB to HD. In Figures 14C to 14D the active square root extraction devices Q9 and Q10 are bipolar devices. The embodiments shown in Figures 14A to 14D provide better stability despite changes in temperature, power supply, common mode noise and process variations during the manufacture of the integrated circuit. Figures 15A to 15D show modalities using active square root extraction devices and active input devices of the same type. Thus, in Figure 15A, the square root extraction devices M9 and MIO are N-type MOSFET devices, as well as the input devices M1 and M2. Similar configurations are shown in Figures 15B (using N-type MOSFET devices), 15C (using N-type bipolar devices) and 15D (using P-type bipolar devices). The output linearity can be maximized when coupled MOSFET devices, i.e. MOSFET devices of the same type, are used both for input and active square root extraction devices, as shown in Figures 15A to 15B. Figures HA to 15D show input capacitances C7 and C8 at the I-RNG and O_RNG integrated circuit pin input conditions. These input capacitances may vary from one part to another due to manufacturing tolerances and procedures as well as variations may impair the operation of the circuit. These variations tend to add the capacitance of the electric field of the electrodes and can cause variations and deviations in the operation of the control circuit. Since typical applications often require an input detection circuit to separate very small changes in the electric field at the electrodes where the input capacitance at the junction pad input nodes can be relatively large compared to the level of input field effect capacitance signal, the minimization of parasitic capacitance C7 and C8 may be advantageous. One method to minimize the effects of parasitic capacitance variation is to add "dissipative" capacitors to the input circuit. Although this may present a tendency to lose sensitivity of the control circuit, it can stabilize the input against variations due to the input capacitance associated with the attached wires, metallization under blow, traces of redistribution in micro-chip configurations and the like. The use of dissipative capacitance is shown in Figure 16, which generally corresponds to Figure 15A. In Figure 16, the dissipation capacitors Cl 1 and C 12 exist in parallel equivalent to the parasitic capacitance C 7 and C 8, respectively, and are electrically coupled to the voltage signal VSS. It will be understood that the Cl and C12 dissipative capacitors are compatible with all embodiments of the present invention described herein and are not limited to use with the embodiment presented in FIG. 16. Although Cl and C12 dissipative capacitors may improve the operation of the control circuit, will tend to require additional physical space. The space is preserved in the embodiment shown in Figure 17A, which shows the addition of the dissipative capacitance resulting from the suppression capacitance of the diodes D4-D7 at the input of the control circuit, here, the gates of the devices Ml and M2 assets. In Figure 17A, diodes D4 and D6 replace the heatsink capacitor C12 of Figure 16 and diodes D5 and D7 replace the capacitor Cl l heatsink of Figure 16. The amount of capacitance per unit surface area is much larger for configurations of diodes of the kind presented in Figure 17A in comparison with the capacitance per unit area of the poly or metallic type capacitors. In addition, diodes D4 to D7 can be used to protect both positive and negative high voltage potential discharges. This protection is essentially advantageous for touch input applications. Human input devices, such as keyboards, single-entry switches and others, are disposed to electrostatic transients and may include components such as MOSFET and other devices to protect their sensitive input circuits. This problem is exacerbated when, as shown in Figure 17B, the electrodes El and E2 detectors are located very close to the ICC input circuits. Figures 18A to 18E show other possible configurations of input circuits for touch switches with integrated control circuits. Figures 18A to 18C show various alternatives to the common mode stimulation of the active devices Ml and M2. Figure 18A shows generally the configuration of Figure 17A and also includes the active devices Mi l a M14. In Fig. 18A, the active devices M1 to M14 are electrically coupled to the sources of the input active devices M1 and M2. Those composed of the active devices M13 and M14 are coupled to the oscillating signal OSCB and their drains are coupled to the gate of the active device M12. The gate of the active device Mi l is coupled to a CSBS current source bias signal and its drain is coupled to the source of the active device M12. The configuration shown in Fig. 18A can provide negative feedback in the input stage of the active devices Ml and M2. Figure 18B shows the input circuit portion including active devices M15 and M16, here shown as N-type devices, whose sources are electrically coupled to the input pins I_RNG and O_RNG, respectively, and the gates of which are electrically coupled to the OSCB oscillating signal. The drains of the active devices M15 and M16 are coupled to the source of the active square root extraction devices M9 and MIO, respectively, and to the bases of the active detection circuit devices Q1 and Q2, respectively. In Figure 18B, the active devices M15 and M16, which are stimulated by an OSCB oscillating signal through their gates and accept input signals through their sources, are carried out of the active devices Ml and M2 which have been previously placed as stimulated through their sources and accept entries through their gates. Figure 18C shows generally the configuration of Figure 18B and also includes dissipating diodes D4 to D7, as also shown in Figure 17A. The configuration of Figure 18C can also be used in a single input mode with an electrode and can offer all the benefits of using input diodes that provide dissipative capacitance in suppression mode. Fig. 18D shows the configuration of Fig. 16 which includes the heatsink capacitors Cl and C12, which balance the inputs of the active devices Ml and M2, but in the single electrode mode without an outside electrode E2 or an input pin O RNG Fig. 18E shows the configuration of Fig. 18D, except that the dissipative capacitance is provided by diodes D4 to D7, also shown in Fig. 17A, minimizing the space needed to provide the benefits of the dissipative capacitance, as discussed in FIG. previous. Figure 19 is a schematic electrical representation of a possible configuration of an output circuit portion of the integrated circuits of the present invention showing various output characteristics and their possible configurations, including an inverting output LCH_O and its components, which they can function as a self-powered inverter 70 in Figures 4 to 7. These output characteristics allow the touch cell to duplicate conventional membrane responses or mechanical switches. The output pins NDB O, NE_O and ND_O are outputs of the touch cell and the integrated circuit assembly that will activate the electrically low output through the devices. The integrated control circuit can be configured to pull the electrically low output through the active devices when there is an applied stimulus (for example, a human tactile stimulus) or it can be configured to pull the electrically low output through the active devices when there is a lack of stimuli (for example, non-human tactile stimulation). As shown in Figure 19, the output pin NDB_O is electrically coupled to the drain of the active device M18, whose source is coupled to the voltage signal VSS and whose gate is coupled to the input of the inverter U2, the output of the inverter U2 , the gate of the active device M17 and the voltage signal TP_O. The output pin NE_O is electrically coupled to the light emitters active devices Q13 and Q14, whose bases are coupled to the drain of the active device M9 and whose collectors are coupled to the voltage signal VSS. In turn, the active device M20 coupled in its gate to the output of the inverter U2 and its source to the voltage signal VSS. The output pin ND_O is electrically coupled to the bases of the active devices Q13 and Q14 and not to the drain of the active device M20. The active device M20 can act as a negative down-pull device for the output NE_O and can be polarized in the gate of the active devices Q13 and Q14 for the output ND_O. The PDS O, PD_O and PE_O output pins are tactile cell outputs and integrated circuit assemblies that will pull the electrically high output through the active devices. The integrated control circuit can be configured to pull the electrically high output through the active devices when there is applied stimulus (for example, a human tactile stimulus) or they can be configured to electrically activate the output through the active devices when there is a lack of stimulation (for example without human tactile stimulation). In FIG. 19, the output pin PDS_O is electrically coupled to the Schotky diode SD1 which in turn is coupled to the output pin PD_O. The output pin PD_O is electrically coupled to the base of the active device Q12 and to the drain of the active device M17, whose source is coupled to the voltage signal VDD and whose gate is coupled to the output of the inverter Ul and the input of the inverter U2. The collector of the active device Q12 is coupled to the emitter of the active device Q1, whose collector and base are both coupled to the voltage signal VDD. Figure 19 also shows an emitter of the active device Q12 that is coupled to the output pin PE_O. The integrated control circuit can be applied to the conventional DC mode, a DC matrix, a DC matrix mode by pulses or an inverter matrix mode. Figure 20A illustrates applications where the integrated control circuit is applied to touch cell configurations for DC mode. In all applications using the DC mode, each integrated control circuit is continuously connected to the system voltage signals VDD and VSS. In some cases, the outputs of several touch cells are connected in an electrical OR (OR) logic circuit (for example, TCA-TC3 touch cells using PE_O and TC7-TC9 outputs using NE_O outputs). The rest of the touch cells (TC4 to TC6 and TC10 to TC13) show the use of various outputs, specifically PD S, PD O, PD E, NDB O, NE O and ND_O. For the touch cells TC4-TC6, which can electrically activate the outputs, the output pins are coupled through a grounded resistor where, for the touch cells TC10 to TC13, which can electrically inactivate the outputs, the pins output are coupled via a resistor to the voltage signal VDD. Figure 20B illustrates the application of the touch sensors in the DC array mode pulsed in a negative manner. Each integrated control circuit of the touch cells has its VDD voltage signal connected to the system voltage supply Vsu minister- The VSS connections of each integrated touch cell control circuit are also shown in the row selection signal, ROW SELECTION 1 a ROW SELECTION 2. In Figure 20B, the output pins NE_O of each integrated touch-cell control circuit connected to a column return, either COLUMN RETURN 1 (TS and TS2 touch sensors) or COLUMN RETURN 2 (touch sensors TS3 and TS4). As can be seen from Figure 20B, ROW SELECTIONS and COLUMN RETURNS can activate a unique tactile sensor, a row of tactile sensors or a column of tactile sensors. This is also illustrated in the timing diagram of Figure 20B. The active devices type P, Q13 and Q14 that are shown in figure 19, will inactivate NE_O, when there is an active stimulus applied to the associated input. The input can also be configured so that these active P-type devices in the output will inactivate NE_O when there is no stimulus applied to the associated input. The base-emitting junction of the active devices Q13 and Q14 will block the return current through VSS to other devices in the array when either device is inactivated. Whenever a particular touch-cell integrated control circuit is inactivated, there will be a reduced output (measured from Vsupply for NE_O) to the direct polarized voltage drop of the base-emitter junction of the active output devices Q13 and Q14 . Once the device can be used in place or instead of the two active devices Q13 and Q14, depending on the application. When it is desirable to avoid the fall Vbe of the device or the P devices, then the outputs NDB_O or ND_O, which use MOSFET devices as shown in Figure 19, can be used. A negative pulse DC array mode configuration of the touch sensors with ND_O outputs is shown in FIG. 20C and is substantially similar to that shown in FIG. 10B. The voltage drop across MOSFET devices type N M18 or M20 will be relatively low at low current levels and depends on the RDS resistance multiplied by the current through the channel of the MOSFET device. This current will therefore be established predominantly by the external load resistance. At higher current levels, the voltage drop will be lower, in relation to the corresponding voltage drop for the bipolar P-type transistors. On the other hand, at higher current levels, the bipolar transistors will tend to fall in the direct polarization of the base emitter junction (0.6 to 0.7 volts), while N-type MOSFET devices will tend to have an increased voltage drop due to the approximate linear relationship of RDS on with the drain current: Vca, -da = (RDS on) (Idrained). Therefore, in typical logic circuits where lower current levels are present, a N-type MOSFET output will tend to decrease the voltage less compared to a bipolar device. This makes the MOSFET devices generically more appropriate for other logic circuits. Fig. 20D shows a positive pulsed DC array configuration with tactile sensors having PD O outputs using the P-type MOSFET device M17, as shown in Fig. 19, to which these observations also apply. However, MOSFET devices do not have any inherent blocking characteristics like bipolar devices. Figure 21A illustrates a cross-sectional view of a typical P-type substrate with type N and P materials as impurities used in the construction of typical CMOS circuits. Figure 21B are schematic representations of a MOSFET device type N, NI, which can be used as an output inactivation device for the output pin NBD_O (active device M18 in Figure 19) for an output pin ND_O (device active M20 in figure 19). Figure 21C is a schematic representation of a blocking device N2, connected in series with the NI output device to prevent the development of leakage currents from parasitic devices associated with NI, which may be presented as an undesired consequence of the construction of the MOSFET device due to the suppression regions surrounding the device. Figures 21A to 21C illustrate the manner of construction of a N-type MOSFET device resulting in the creation of a parasitic drain to the bipolar source PD1 diode as blocking the VSS leakage current to the substrate. Typical CMOS integrated circuits make use of P or N type substrates. These substrates are typically electrically connected to the VSS or VDD integrated circuits. In the case of P-type substrates, the substrate is bound to VSS and in the case of N-type substrates, the substrate is bound to VDD. Note that, in Fig. 21B, the N-type MOSFET device source, NI, is tied to the voltage signal VSS and that the anode of the parasitic diode PD1 is also tied to the source node of the device NI. The cathode of the parasitic diode PD1 is tied to the drain of the NI device. As a result of this, when the integrated control circuit is implemented in the negative pulsed DS matrix mode with the active electrical inactivation, using the N-type MOSFET devices (as shown in Figure 20C with ND_O outputs), there is an inherent trajectory for reverse current through the parasitic diode PD1 through the substrate P. When the pulses for the reference mark rows are applied to the matrix and are at a potential that is greater than the output potential of ND O, a current will flow through the diode PD1 parasite from VSS to ND_O. This current path will affect the operation of the array and the power supply; and this low current path will provide a low impedance path that connects VSS to VDD through the reference mark activators. A bipolar diode connected in series with a N-type MOSFET will inactivate the device by preventing reverse current flow but will also negate the advantage of an N-type inactivated MOSFET device, specifically, a low voltage drop at the output. A bipolar diode will also tend to decrease Vbe at the base emitter junction. To block this unwanted current path, one way to implement a blocking device is that it is preferably compatible with the development of a conventional integrated circuit and that it has a minimum voltage drop. By making the appropriate connections between the MOSFET type N, NI and N2 devices, the current and leak path can be blocked so that the substrate P and the voltage signal VSS are isolated from the current leakage paths. of the device ND_O, NI; at the same time, the voltage drop of the control circuit output is minimized. The device N2 in FIG. 21A is the blocking device and is shown schematically in FIG. 21C. The draining and blocking source N2 device is connected to VSS and VSS 1 respectively, as shown in Figures 21A and 21C. The gate of the blocking device N2 is coupled to the voltage signal VDD which can, but need not be, of 3-5 volts, so that it is compatible with most of the microprocessors. When the source of the N2 device is of low potential, such as grounding, the channel resistance can be very low to the extent that the gate voltage is slightly greater than the threshold voltage of the device. Since the gate of device N2 is in VDD, which can be of the order of 3-5 volts (VsuminiStro), its source is at zero volts during the active pulse period and its threshold voltage is less than one volt, the resistance channel will be very low and therefore the channel drop of the device will also be very low (ie, less than a standard bipolar diode). When the source of device N2 is at a voltage equal to (or greater than) VDD, the gate of the source voltage (VG S) will be less than the threshold voltage of the device. This will cause the channel resistance to increase significantly, thereby blocking a substantial current through the channel. In addition, the voltage across the suppressive junction of the source of the device N2 to the parasitic diodes PD of the substrate PS will be less than the barrier potential (approximately 0.6 to 0.7 volts) of the source-drain parasitic diode PD1. The parasitic diode PD1 will therefore block the substantial current through the substrate. In addition, the N2 blocking device can be used for reverse voltage protection in standard integrated circuit applications and will provide all the benefits established in the above. When used in this manner, the N2 blocking device can be connected to the VSS of the integrated circuit in the same manner as described and can protect the circuit from reverse current or voltage damage. Figures 21D to 21F show a BDP2 device for electrical activation devices with high performances PDS_O, PD_O and PE_O, which are shown in Figure 19. The device shown in Figure 21D to 21F is complementary to the device shown in Figures 21A to 21C and will be understood by those skilled in the art, based on the discussion with reference to Figures 21A to 21C. In all the DC mode configurations described, there are three connections to each of the integrated control circuits of the touch cells. VDD and VSS for each integrated touch-cell control circuit need to be connected to a power source for a certain amount of time, in order to process the input stimuli. The integrated control circuit output is in PDS_O, PD_O, PE_O, NDB_O, ND O and NE_O, depending on the desired configurations. These outputs form the third necessary connection for the integrated control circuit. However, in some cases, it may be advantageous to have an integrated circuit that requires only two connections. For example, since two connections per switch are typically used in applications involving membrane switches, having a touch detector switch and an integrated control circuit requiring only two connections can facilitate direct replacement of membrane switches with switches tactile A schematic representation of the two-terminal membrane switch matrix MS1-MS4 is shown in Fig. 22. Figure 22 shows a way to solve and read switches within a matrix. Of course, the matrix in Figure 22 can also be modified to include more rows, more columns, more switches and alternative connections. In all cases, the interconnection to each switch will typically include two types of signal lines: ROW SELECTION AND COLUMN RETURN. Each ROW SELECTION line is a source of potential to allow current to flow through each MS1-MS4 switch as it closes (in the case of membrane switches, by pressure of the finger that causes closure) through the lines COLUMN RETURN. The COLUMN RETURN lines, the COLRl and COLR2 terminating resistors of the COLUMN RETURN lines 1 and 2, respectively, are used to develop the voltage to be processed by the return logic circuits and to limit the current through the the switching devices. Reference mark lines can be sequenced such that only one row of the switches (MS I and MS3 or MS2 and MS4) is active at any given time. When a particular row is selected, the voltage generated through each resistor termination COLR will indicate which switches in the selected row are electrically closed. The COLUMN RETURN lines are usually processed simultaneously. The matrix schemes are efficient in terms of the number of interconnections used to process the number of switch inputs. For example, sixty-four switches can be read with an eight-by-eight array using eight ROW SELECTION lines and eight COLUMN RETURN lines. Typically, a certain kind of logic device is connected to the reference mark and the return lines to determine the state of all the switches over a short period of time. This is a typical matrix scheme that a person skilled in the art can know how to implement. It can be used in controllers, computer keyboards, telephones and other devices that are widely available in the market. A solid-state type sensing device that can detect stimuli acting as a two-terminal switch can be advantageous insofar as it can allow circuits with conventional reference mark and matrix reading to be built without software, logic circuits or additional microprocessors which are susceptible to restoration and other failures. Figure 23 illustrates the implementation of said devices, distributed in a matrix and having only two integrated circuit connections. In this way, the touch sensors TSl to TS4 of figure 23 have been replaced by membrane switches MS I to MS4 of figure 22. In figure 23, each touch sensor TS l to TS4 detects differences in electric field potential. According to the presence or absence of appropriate stimuli, the device (depending on the specific application) will move from a state of high impedance (equivalent to an open switch) to a low impedance state (equivalent to a closed switch) and for it will imitate a conventional membrane switch or other mechanical switch. The main advantage of this device is its ability to mimic the attributes of two-terminal switches. Figures 24A and 24B show the possible circuits for the touch sensors TSl to TS4 of Figure 23. The circuits shown in Figures 24A and 24B are based on an inverter circuit portion of the circuit shown in Figure 19. In the figure 19, the inverter circuit shown includes active devices M19 and Q15-Q19 as well as a resistor R9. The inverter circuit output of the LCH_O pin is shown coupled to the emitter of the active device Q19. The active device Q19 in turn is coupled in its base to the output of the inverter U2, to the drain of the active device Q15 and to the gate of the active device M20; and in its collector to the emitter of the active device Q18, whose base is coupled to the voltage signal VDD and whose collector is coupled to the resistor R9 which in turn is coupled to the voltage signal VDD. The collector of the active device Q18 is also shown coupled to the bases of the active device Q15 and Q16, whose emitters are coupled to the voltage signal VDD and the base of the active device Q17, whose collector is coupled to the voltage signal VSS and the emitter of which is coupled to the collector of the active device Q15. The collector of the active device Q18 is also coupled to the drain of the active device M19, whose gate is coupled to the output pin INITB of the control circuit and the source of which is coupled to the voltage signal VDD. Figures 24A and 24B show various embodiments of the inverter circuit of Figure 19. Both modalities omit the optional active devices Q16 to Q18. Figure 24A shows the implementation of bipolar components Q15 and Q19 in the inverter circuit, as shown in Figure 19, and Figure 24B shows the implementation of MOSFET components in the inverter circuit. Other configurations can be implemented considering the spirit and functionality of a two-terminal device. Figure 24A shows a bipolar inverter circuit operation together with a control circuit, which provides the necessary functions to detect an input stimulus, decision making and activation of a bipolar inverter circuit. The control circuit can provide the energy in the functions of resetting, initializing and sequencing various blocks and internal characteristics. The inputs in the control circuit include those associated with the input detection connections, specifically OSCB, + (MAS) (PLUS)) and - (NEGATIVE) (NEGATIVE)); those associated with the power supply of the control circuit, specifically the voltage signals VDD and VSS; and those associated with the inverter circuit, specifically INIT and TRIGGER. The inverter output is through the output pin LCH_O. When there is a path for current from a system Vsuministro to GND (ground) through the MOSFET device type P of active pull on the line SELECT ROW, the reference mark line SELECT ROW in Figure 24A is active. With power supplied, the control circuit can be operational. When the reference mark pulse is applied for the first time, the control circuit will apply a gate signal, via the INIT line to return to the active M19 device. This will ensure that the base emitter voltage of the active Q15 device is essentially 0 volts, keeping it free of conduction (except for leakage current). With Q15 inactivated, there is no current available for the base of Q19 and therefore Q19 will also be off. With Q19 off, the base voltage of Q15 can be essentially VDD, even after the INIT signal has been removed and M19 is off. With the inverter essentially switched off (ie without current flow), the control circuit will be allowed to operate. When operational, the integrated control circuit is in the high impedance mode and simulates an open switch. The output voltage developed through the Rcoiumna resistor is equal to Vsumist R (integrated control circuit) / ([R (integrated control circuit) + RCOiumna.] The higher the effective resistance of the integrated control circuit, the lower the It will be the percentage of Vsuministro that will descend through RCoiumna and the higher the percentage that will decrease through the integrated control circuit.A perfect switch must have an infinite resistance and a zero current when it is open and therefore Vsumipistro will descend through the switch during a reference mark pulse and zero voltage will descend to RCOiumna due to zero current flow, since an integrated circuit is not a switchIt is important to design the integrated control circuit so that it has as little current as possible when it is applied to the reference brand pulse so that it duplicates the characteristics of an open switch more precisely. An input electrode can be configured to cause the integrated control circuit to remain in this high impedance mode with an applied stimulus or without an applied stimulus. When the integrated control circuit is in the high impedance mode, most of the supply will be applied through an integrated control circuit. This will allow the circuit to operate in a floating mode since internal VDD and VSS will be sufficient to operate the integrated circuit as a whole and also the internal control circuit. The electrode configuration can also be such as to cause the control circuit to generate an activation pulse to the inverter circuit when a stimulus is applied, or alternatively, when a stimulus is not applied. When the control circuit generates an activator pulse, the inverter will activate. The activation pulse in Figure 24A can be a positive pulse moving towards VDD from VSS. An activation pulse can be allowed after the INIT signal is restored, which causes M19 to become inactive. This positive pulse can send bias polarization of the base emitter of the bipolar device type N Q19, causing it to turn on. With the base current flow and the gain transfer of the active device Q19, the current will flow in the collector of the active device Q19 and therefore through the resistor R19. The current flowing through the resistor R9 will generate a voltage potential which will cause the base of the active device Q15 to descend to VSS - sufficient to send the polarization of the base junction of the emitter of the active device Q15 to cause it to turn on. The current gain of the active device Q15 will cause substantial current to flow in the collector of the active device Q15 and also cause the voltage on the base of the active device Q19 to increase enough to send the polarization of the base junction of the device emitter Q19 active, even after removing the activation pulse. The activation pulse will be eliminated, due to the voltage drop across the control circuit, enough to disable the operation of the control circuit. The inverting current will remain activated after the activation pulse is eliminated due to the positive current feedback loop between Q15 and Q19. The voltage drop of the inverter will be determined by the saturation voltage, the bonding resistors, the gains of the active Q15 and Q19 devices and the resistance of Rcoiumna-The inverter circuit within the integrated control circuit needs to remain active once it is The actuator has been eliminated since the control circuit is inoperable and it is important that the inverter falls with as little voltage as possible through a range of currents. In this low impedance mode, it is desirable to obtain these attributes that duplicate as much as possible a closed switch. A perfect closed switch will allow infinite passage of current and a drop of zero volts at all current levels. To better duplicate a perfect switch, for example, one with a low voltage drop, the inverter circuit can preferably be use of bipolar transistors with increased emission areas and low Vbe drops and MOSFETS with high W / channel L ratios, low thresholds and devices with high profits. Figure 24B shows the inverter circuit of Figure 24A where bipolar active devices Q15 and Q19 have been replaced by MOSFET devices M21 and M22. The operation of the integrated control circuit of Fig. 24B is identical to the operation of the integrated control circuit of Fig. 24A. The operation of the inverting portion shown in Fig. 14B is described below. When the INIT pulse is applied, the active device M19 is turned on. This will allow VDD to be applied to the gateway of the active device M21. In this condition, the gate source voltage of the active device M21 will be less than the threshold voltage of the device M21 MOSFET type P, essentially zero volts and therefore the active device M21 will be turned off. With the drain current of the M21 device active at essentially zero amperes (or other leakage current), there will be no voltage developed through the RIO resistor. With the gate of the M22 device active essentially at zero volts, its gate source voltage will be substantially less than the threshold voltage of the device. The drain current of the active device M22 will be essentially zero with this gate source voltage well below the threshold voltage. The zero current through the resistor R9 will cause the gate voltage of the active M21 device to be very close to VDD and therefore the gate source voltage of the active M21 device will be essentially zero as well, even after that the INIT signal has been deleted. This condition will place the inverter circuit in the high impedance state. When an activation pulse that approaches VDD is applied to the gate of the active M22 device, after removing the INIT pulse, its gate source voltage will exceed the threshold voltage of the active M22 device, which causes M22 to turn on. The drain current of the active device M22 will increase, developing a voltage drop across the resistor R9. With the voltage drop across the resistor R9, the gate source voltage of the active device M21 will exceed its threshold voltage, which will cause the active device M21 to turn on. The drain current of the active device M21 will also increase causing the voltage drop across the resistor RIO to increase above the threshold voltage of the active device M22, even after the trigger pulse is removed. Therefore, the inverter will move to a low impedance state and the voltage drop across it will depend on the characteristics of the active devices M21 and M22, the values of the R9 and RIO resistors and the Rcoiumna-El resistance. The rest of the operation of the integrated control circuit in Fig. 24B is similar to that of the integrated control circuit in Fig. 24A. Also shown in both figures are the blocking diodes of Figures 21A to 21C, marked D8 and D9 in Figures 24A and 24B, respectively. Figure 25A illustrates the inverter circuit portion of Figure 19 comprising active devices Q15 to Q19 in a possible configuration constructed within the PS substrate. Figure 25B schematically shows the portion of the inverter circuit. In Fig. 25A, the active devices Q15 and Q16 share a 15-EMITTER EMITTER well 16 with impurities P as an emitter and the collector of the active Q15 device and the emitter of the active Q17 device are the same well COLECTOR Q15 -EMISOR Q17 with impurities P, which is coupled to the gate of the active Q15 device. The active devices Q15, Q16 and Q17 also share the same well with N impurities in their bases BASE Q15, BASE Q16 and BASE Q17 respectively. The PS substrate forms the collectors of the active devices Q16 and Q17, COLECTOR Q16 and COLECTOR Q17, respectively. The active device Q19 is shown in a well with separate impurity N in the substrate PS and is coupled to its well manifold with impurity N, COLLECTOR Q19 to the resistance R9, in its well base with impurity P BASE Q19 to the well with impurity P COLLECTOR Q15 / ISSUER Q17 and in its well emitter with impurities N ISSUER Q19 to the voltage signal VSS at the anode of the diode DIO. In Fig. 25A, the active device M19 is coupled in parallel with the resistor R9. The operation of the configuration shown in Figures 25A and 25B will be understood by those skilled in the art of designing active devices and circuits and from the discussion of the inverter circuit with reference to Figure 24A. The active devices Q16 to Q18 improve the signal supplied to the LCH_O output. The configuration shown in Figure 25A will benefit from a reduced inverted ON (ON) voltage drop, compared to the voltage drop associated with a standard inverter, due to the dynamic impedance of the active Q17 device and the sealing of the VSS current through the PS substrate. The diode DIO, coupled at its cathode to the output LCH O and at its anode to the emitter of the active device Q19 and to the voltage signal VSS, can avoid feedback on the inverting portion of the integrated circuit shown in FIG. 25B. Figure 25C shows the diode DIO coupled at the anode to the voltage signal VSS and the collectors of the active devices Q17 and Q18 and at its cathode to the emitter of the active device Q19 and the output LCH_O. Therefore, the configuration in Figure 25C changes the voltage signal in the emitter of the active device Q19, which can be biased by the TRIG output, from VSS in Figure 25B, to VSS 1. Advantageously, the configuration of the inverter circuit can reduce the voltage drop since, in this case, the voltage drop across the diode DIO is not in series with the base emitting voltage of the active device Q19. The optional active device Q18 in FIGS. 25B and 25C is useful for increasing the inverter trip voltage of the inverter circuit. The integrated circuits of the present invention can respond to capacitive inputs that change in a variety of ways. For example, Figures 26A to 26C show a capacitive input detector apparatus compatible with the integrated circuit of the present invention, wherein the capacitive input changes as a result of a change in distance d between the electrodes GE and SE that form the capacitance Csentdodo shown schematically in Figure 26D. The capacitance Csent is a function of the capacitive constant of the electrodes E0, the relative dielectric constant Er, the surface area of the electrodes s and the distance between them, d. The apparatus shown in Figure 26A, having sensor SE electrodes and an integrated ICC control circuit on the 143 side of the substrate 144 and a grounded GE electrode configured in the buttons 122 generates cavities 121 in the other side 145. Figures 26B and 26B show the separate layers of the apparatus shown in Figure 26A. The cavities 121 in Figure 26A allow the buttons 122 to be depressed, for example, by a human finger or other probe in a manner that alters the distance d between the electrodes GE and SE. The control circuit shown in Fig. 26D can respond to the changed capacitance resulting from the changed distance d. The control circuit of Fig. 26D corresponds to the control circuit shown in Fig. 18D, except that the capacitance C3 in Fig. 18D has been renamed, given in Fig. 26D. Heretofore, this specification has generally described various preferred embodiments of touch sensors (or field effect sensors), in accordance with the present invention. The following are descriptions of various preferred embodiments of practical applications for such sensors. Although it is generally preferred that these applications are implemented using tactile sensors described in the foregoing, these applications may also generally be implemented using other types of tactile sensors, for example, the sensors described in US Patents. Nos. 5,594,222 and 6,310,611, conventional capacitive sensors and other types of sensors, as is known to a person skilled in the art. Figures 27A to 27D show a capacitive input liquid level sensing apparatus compatible with the integrated circuit of the present invention, wherein the capacitive input changes as a result of a change in the dielectric constant Er between two electrodes. This change can occur, for example, when a liquid replaces the air between two electrodes GE and SE1 forming the capacitance Csentido. So, in figure 27A, the ground electrode GE on substrate 123 is separated from the sensor electrode SEl through an air gap that can be filled with liquid 125. Figure 27B shows a substrate 124 forming a reservoir for liquid 125 and the substrate 123 adapted to allow the liquid 125 to fill the air gap between the grounded GE electrode and the electrode of the sensor when the liquid 125 reaches a certain level. FIGS. 27C and 27D illustrate a possible advantageous configuration of the grounded GE electrode and the sensor electrode, coupled to the integrated ICC control circuit. In both Fig. 27C and 27D, the electrodes GE and SEl are long and horizontally positioned, i.e. with their axes longitudinally parallel with the surfaces of the liquid 125 so that a small increase in the level of the liquid 125 will change significantly the capacitance Csent. which is shown schematically in Figure 27D. The control circuit shown in Fig. 27E is the same as shown in Fig. 26D and this is also compatible with the apparatus shown in Fig. 27A through 27D. Figures 28A to 28B show a capacitive input detector apparatus compatible with the integrated circuit of the present invention, wherein the capacitive input changes as a result of a change in the surface area sS3 of the sensor electrode SE3. In Figure 28A, the substrate 126 has a GE electrode connected to the ground and a movable substrate 127 which supports the two sensor electrodes SE2 and SE3 coupled to the integrated ICC control circuit. The sensor electrode SE3 has a surface area s-2 that varies along the direction in which the substrate 127 is adapted to move. Thus, Figure 28B shows a substrate 127 moving upward relative to its position in Figure 28A. The surface area sS3 of the sensing electrode SE3 which is observed by the GE electrode connected to earth therefore decreases. This change in surface area corresponds to a change in capacitance Csentt3. which is shown schematically in Figure 28C. The control circuit shown in Fig. 28C is similar to the circuit shown in Fig. 18E, but has the double electrode structure shown in Fig. HA, where electrodes El and E2 have now been referred to as electrodes SE2 and SE3 and capacitance C6 has been renamed capacitance C23. The operation of the circuit will be understood by those skilled in the art and from the foregoing discussion of Figures HA and 18E. Figures 29A to 29D show a capacitive input sensitive marking apparatus compatible with the integrated circuit of the present invention, wherein the pulse widths of input and the sequence can determine the response of the integrated control circuit. Figures 29A to 29D show an electrode SE4 sensor coupled to the integrated control circuit ICC on the substrate 128 and the electrodes GEl and GE2 connected to ground on the rotating disk 129. In Figures 29A to 29D, the grounded electrodes GEl and GE2 (which include the space between them) together occupy only about half the area of the rotary disk 129 and are spaced apart. These and other similar configurations may allow a control circuit to distinguish between clockwise rotation and counter rotation of the marking device. Figures 29B to 29C show the movement of the rotary disk 129 in relation to the stationary substrate 128. Figs. 29E and 29F show the output pulses of the marking apparatus shown in Figs. 29A to 29D, which may generate a response in an input portion of an integrated control circuit, as shown in Fig. 29G. Fig. 29E shows the relatively wide and separate input pulses resulting from the counterclockwise rotation of the rotating disk 129 at a speed, and Fig. 29F shows the relatively close and close input pulses resulting from the clockwise rotation of the rotating disk 129 at a faster speed. The changes in the capacitance that are formed between the electrodes SE4 and either GEl and GE2 and shown schematically in FIGS. 29G (which is similar to the configuration shown in FIG. 27E) can be detected by embodiments of the integrated control circuitry of the present invention. Figures 30A to 30E show another capacitive detection marking apparatus compatible with the integrated circuit of the present invention, wherein a ground connection is provided by the user. Figure 30A shows a rotary disk 130 having transfer electrodes TE1 to TE8 of various sizes which may correspond to the input pulse widths of various sizes when coupled to ground. Figure 30B shows the transfer electrodes TE1 to TE8 of the rotating disc 130 coupled to the coupling electrode CE located in the cylinder 131. Figure 30C shows the cylinder 132, adapted to be coupled within the cylinder 131 of Figure 30B, which It has SE5 and SE6 sensor electrodes coupled to the integrated ICC control circuit. Figure 30D shows the components shown in Figures 30A to 30C assembled together as a rotary capacitive input device. Figure 30E shows a hand holding cylinder 131. The hand 133 is coupled to the coupling electrode CE and to the transfer electrodes TE1 to TE8 with a virtual ground. Each sensor electrode SE5 and SE6, as shown in FIG. 30C, is adapted to receive a capacitive input from one transfer electrode at a time. As shown in FIGS. 30F to 30H, two input pulses can be fed to the integrated ICC control circuit at the same time. Both the direction and the arc length of the rotation of a marker user comprising a rotary disk 130 and a cylinder 131, can be determined from the inputs shown in FIGS. 30F and 30G. Figure 30F shows in train of pulses that results from two full turns of the marking device, in a counterclockwise direction, wherein Figure 30G shows the train of pulses resulting from two turns in a direction in the clockwise direction. Figure 30H shows a schematic representation of the marking device of Figure 30E, including a hand 133 connected to ground, a coupling electrode CE connected to transfer TE electrodes, which form a capacitance with the sensing electrodes SE5 and SE6, coupled to resistors RIN1 and RIN2, respectively. The integrated ICC control circuit provides an oscillating signal OSC to the sensing electrodes SE5 and SE6 through the resistor RIN1 and RIN2, respectively, and provides the outputs OUT1 and OUT2 to a decision circuit (not shown). The various components of the marking device, including the rotary disk 130 and the cylinders 131 and 132 can be formed according to the invention described in the US patent. No. 6,897,390 entitled "Molded / Integrated Touch Switch / Control Panel Assembly and Method for Making Same" or in other ways. Figures 31A to 31F show separate layers and the construction of a touch switch assembly having an integrated control circuit according to the present invention. Figures 31A to 31E show the individual layers of the assembled touch switch shown in Figure 31F. Figure 31A shows the back side of the substrate 133 including an opaque area 135 and a window or transparent area 136. The opaque area 135 may be a decorative frit, a decorative epoxy resin, an ultraviolet-cured ink or any other decorative layer material. Figure 31B shows the tactile switch electrodes 134 positioned on the rear side of the substrate 133 in the window area 136. The electrodes 134 are shown superimposed on the opaque area 135 and may be comprised of a transparent conductive material including indium tin oxide or other suitable material. Figure 31C shows the lower conductive layer of the touch switch assembly, as seen from the rear side, which includes the circuit traces 138, which may be constituted by silver-filled frits, epoxy silver resins, epoxy-copper resins, electro-coated conductors and the like as well as combinations of the foregoing. Figure 31D shows the dielectric layer of the tactile switch having the dielectric layer areas 140, which can be isolated ceramic frits, ultraviolet inks, epoxy resins and the like. Figure 31E shows the transition layer of the touch switch assembly, as seen from the rear side, which includes transition conductors 137 which may be constituted of materials described with reference to Figure 31C. Figure 31F shows the separate layers shown in Figures 31A to 31E assembled together as a finished touch switch assembly. Figure 31F also provides a view of the rear side of the assembly. Although the embodiments presented in the above have been described constituted in the DC mode, the integrated control circuits of the present invention are also compatible with AC inputs and therefore can also operate in the AC mode. The situation AC is presented in figure 32. Figure 32 shows a tactile switch with an integrated control circuit adapted to receive an AC input. In Figure 32, the AC, AC signal is coupled to the rectifier bridge RB, which includes the diodes D1 to D14 through the RIO and RLOAD resistors. The diodes D1 to D14 of the rectifier bridge RB are coupled in parallel with zener diode Z1 and capacitance C15. The AC, AC signal can stimulate the touch switch with the integrated control circuit, which includes the inverting portion shown in Fig. 24A with the separate diode D8. This configuration can be advantageous in that the integrated circuit can be designed to extract a relatively small current and that the circuit is characterized by a low detection impedance, which provides a floating circuit that does not depend on its ground connection. Although the embodiments of the present invention described above have been described as providing a digital output, many of the benefits of the touch switch with the integrated control circuit configurations described above can also be obtained when the integrated control circuit provides an output. analog In the digital output situation, the output reflects information provided by the input to the electrodes for only two stages, for example stimulated or unstimulated. In some applications it is desirable to provide an output that corresponds to more than two states. For example, in liquid detection applications, similar to the situation described with reference to FIGS. 27A to 27D, it may be desirable to provide an output that reflects not two states but many states that may correspond to many liquid levels. An analog output can correspond to many input states. Figure 33A shows the possible circuits for an analog electric field sensor with an integrated control circuit. The circuit configuration of Fig. 33A corresponds to the circuit presented in Fig. 4 and includes start and bias circuit 40 which provide a current bias to the gates of switches SW2 and SW4 and a pulse generator and logic circuits that provide a Ignition re-set signal BY to the gates of switches SW1 and SW3. The configuration of Fig. 33A also includes an input portion, which includes the active devices Ml, M2, M5 and M6, similar to the introduced portion described with reference to Fig. 12A. The drains of the active devices Ml and M2 are coupled to the traces INPUT1 and INPUT2 and, through the diodes DI and D2, to the traces PKOUT1 and PKOUT2, which provide input to the differential amplification circuit 160. The operation of this circuit can be understood from the description provided with reference to figures 4 to 7. The configuration shown in figure 33A can provide the benefits of the configurations shown in figures 4 to 7, which include a sensor electrode and a damping of the signal with reference mark, the rejection in common mode of the electl interference in the electrodes and circuits, temperature stability and the like. Figure 33B and 33C show timing diagrams for the circuits shown in Figures 33A. Figures 33B and 33C show the oscillating signal OSC and the signals provided in the traces INI, IN2, INPUT1 and INPUT2. Figure 33B shows the signals as a function of time in microseconds and Figure 33C shows the signals as a function of time in nanoseconds. Figure 34 shows a two-by-two matrix in the field sensors of Figure 33A that accept an analog input and provide an analog output. The multiplexed system of Figure 34 is similar to that shown in Figure 10. The trace ROWSELECT1 (ROW SELECTION 1), having a signal provided by the control circuit 141, will be activated for a period of time in which the switches analog ATS and ATS3 have power applied to them. The analog outputs AOUT of the analog switches ATSl and ATS3 will provide an output, provided to the trace COLUMNRETURN1 (RETURN OF COLUMN 1) and fed to the analog interconnection circuit 142, which is proportional to the stimulus provided at the electrodes of the analog switches ATS and ATS3 . These outputs will be stable at temperature, will exhibit good signal-to-noise performance characteristics due to the low impedance of the circuits and will also exhibit common mode rejection properties. The analog signals can be processed in a manner similar to that described in the U.S.A. 5,594,222 or using other analog processing techniques, as will be understood by those skilled in the art of electl circuit design. Figures 35A to 35B illustrate one embodiment 1100 of the present invention wherein a field effect sensor is used in relation to another structure to emulate a mechanical button switch to press. The mode 1100 includes a dielectsubstrate 1102, which may be constituted in any suitable manner. Preferably, the substrate 1102 is substantially rigid. For example, the substrate 1102 may be a conventional printed wiring board or a panel or portion of a larger assembly or component, for example the panel of a door or the dashboard of an automobile or the interior panel of a refrigerator. Alternatively, the substrate 1102 may be a flexible circuit carrier. In such an embodiment, the flexible circuit carrier is preferably applied to a substantially rigid secondary substrate (not shown). The substrate 1102 may acquire any other suitable form, as will be recognized by those skilled in the art. The substrate 1102 defines the aperture 1104. The field effect sensor 1106A is placed on the substrate 1102, in proximity to the aperture 1104. The field effect sensor 1106A is shown in Figure 35A as being placed on one side of the substrate 1102 Alternatively, the field effect sensor 1106A can be placed on the other side of the substrate 1102. Further, in embodiments wherein the field effect sensor 1106A includes two or more electrodes, one or more of said electrodes can be placed on one side of the substrate 1102 and the other of the electrodes can be placed on the other side of the substrate 1102. In other embodiments, the field effect sensor 1106A can be encapsulated within the substrate 1102, as shown with respect to the sensor 1106A of field effect in the figure ID, further explained in the following.

The arrow 1108 is inserted in slidable engagement through the opening 1104. A sleeve, bushing or the like (not shown) can be provided in relation to the opening 1104 to allow the arrow 1108 to slide better through the opening 1104 without let it oscillate. The arrow 1108 preferably includes a knob 1110. In the embodiment illustrated, the arrow 1108 is a threaded plastic bolt, the head of which forms a knob 1110. In other embodiments, the arrow 1108 can take on any suitable shape and can be made from any suitable material, as will be recognized by those skilled in the art. Preferably, the arrow 1108 is made of a non-conductive material such as plastic or resin. The electric field stimulator 1112 joins the arrow 1108 at a predetermined location. The electric field stimulator 1112 is made of a material that easily stimulates or alters an electric field, as discussed above. Preferably, the electric field stimulator 1112 is made of a metal or other conductive material, but other materials are also suitable, as is known to those skilled in the art. In the embodiment of FIG. 35A, the electric field stimulator 1112 is a metal washer secured to the arrow 1108 with threaded plastic washers 1116 on each side of the electric field stimulator 1112. In other embodiments, the electric field stimulator 1112 may acquire other shapes, be made from other materials, and be joined to the arrow 1108 by any suitable means, as will be known to those skilled in the art. The plastic washer 1116 is installed between the substrate 1 102 and the electric field stimulator 1112, preferably it is sufficiently thick to prevent the electric field stimulator 1112 from contacting one or more of the electrodes of the field effect sensor 11106A. Alternatively, other structures (not shown) can be provided to prevent the electric field stimulator 1112 from contacting one or more of the electrodes of the field effect sensor 1106A as known to those skilled in the art. Figure 35A shows the electric field stimulator 1112 which is located on the same side of the substrate 1102 as the field effect sensor 1106A and on the opposite side of the substrate 1102 as the head 1110. Alternatively, the field stimulator 1112 electrical and the field effect sensor 1106A can be located on opposite sides of the substrate 1102, and the electric field stimulator 1112 and the head 1110 can be located on the same side of the substrate 1102. The arrow 1108 is skewed longitudinally so that the electric field stimulator 1112 is normally in a predetermined position relative to the field effect sensor 1106A. The arrow 1108 and therefore the electric field stimulator 1112 can be displaced from their normal positions by applying an appropriate force to the head 1110. In the embodiment of Figure 35A, the deflection is provided by a helical spring 1114 installed around the head. the arrow 1108 between the knob 1110 and a corresponding surface of the substrate 1102, so that the electric field stimulator 1112 is usually close to the field effect sensor 1106A. The electric field stimulator 1112 moves away from the field effect sensor 1106A when a longitudinal force is applied to the arrow 1108. In alternate embodiments, the arrow 1108 can be biased so that the electric field stimulator 1112 is usually away from the field effect sensor 1106A and moves closer to the field effect sensor 1106A when a suitable force is applied to the arrow 1108, as will be recognized by those skilled in the art. In further alternative embodiments, the helical spring 1114 can be replaced with any suitable structure for deflecting the arrow 1108. For example, a layer of flexible or resilient material (not shown) may be placed on the substrate 1102 around the opening 1104, or the substrate 1102 itself may be formed of a flexible or resilient material that deforms when the knob 1110 is pressed against the same and return to its original position when released and therefore returning knob 1110, arrow 1108 and electric field stimulator 1112 to their original positions. Any number of other structures can be used to deflect the arrow 1108, as will be known to those skilled in the art. In operation, an electric field is generated around the field effect sensor 1106A, as discussed above. With the arrow 1108 in the normal position, as shown in FIG. 35A, the electric field stimulator 1112 is coupled to this electric field. The detection circuits (not shown) associated with the field effect sensor H06A detect this coupling, as discussed above. When the arrow 1108 moves longitudinally in response, for example, to a user pressing down on the knob 1110, the electric field stimulator 1112 moves away from the field effect sensor 1106A and disconnects from the electric field around the sensor 1106A of field effect. The corresponding detection circuits (not shown) detect this decoupling and provide a signal to a control circuit which, in turn, can provide a control signal to a controlled device, as set forth in the foregoing. In this manner, the 1100 mode emulates a mechanical push button switch. Figure 35C illustrates an alternative embodiment 1140 of the present invention wherein a mechanical pull switch is emulated. The mode 1140 is structurally similar to that of the mode 1100, except that the arrow 1108 is offset so that the electric field stimulator 1112 is normally placed at a predetermined distance from the field effect sensor 1106A. As such, the electric field stimulator 1112 is normally decoupled from the electric field around the field effect sensor 1106A. In order to operate the field effect sensor 1106A, a user must pull the knob 110 and thereby pull the electric field stimulator 1112 near the field effect sensor 1106 and cause the electric field stimulator 1112 to engage. with the electric field around the field effect sensor 1106A. Preferably, a mechanical stop, for example a mechanical stop 1119, attached to the arrow 1108 at a predetermined location is provided to limit the displacement of the arrow 1108 by the helical spring 1114 or other deflection means. Figure 35D illustrates another alternative embodiment 1160 of the present invention that emulates a mechanical push button switch. The mode 1160 includes a post 1118 positioned on the substrate 1102. The field effect sensor 1106A is encapsulated within the substrate 1102 in proximity to the post 1118. In other embodiments, the field effect sensor 1106E can be placed on any surface of the substrate 1102 in the manner of the field effect sensor 1106A, as illustrated and discussed in relation to Figure 35A. The push button 1120 having a bearing surface 1122 slidably engages with the post 1118. The electric field stimulator 1112 is associated with a lower portion of the push button 1120 closest to the substrate 1102. The push button 1120 and the electric field stimulator 1112 may, although not necessarily need to be so, have separate structures. Actually, the push button 1120 and the electric field stimulator 1112 can be constituted as a single monolithic structure. The helical spring 1114 deflects the push button 1120 so that the field effect stimulator 1112 is normally located at a predetermined distance from the field effect sensor 1106E. The application of an appropriate force to the support surface 1112 displaces the field effect stimulator 1112 towards the field effect sensor 1106E. The field effect sensor 1106E and the associated detection circuits respond as set forth in the foregoing. In a significative way, this mode 1160 does not include an opening in the substrate 1102. As such, the mode 1160 may be particularly preferable for use in applications where it is desirable to prevent the entry of fluids or contaminants through the substrate 1102. The 1160 mode can easily be modify to function as a pull switch, as will be recognized by those skilled in the art. More than one field effect sensor can be used in relation to any of the above modalities. Figure 35E illustrates a mode using four field effect sensors 1106A through 1106D distributed around the aperture 1104 of the modes 1100 and 1140. The mode 1160 can be modified in a similar manner. Other modes can use more or less than four field effect sensors. In embodiments utilizing a plurality of field effect sensors 1106A through 1106n, the sensors and corresponding detection and control circuits can be configured so that the electric field stimulator 1112 is coupled or decoupled from the electric field or electric fields on which each individual field effect sensor 1106i substantially simultaneously as the electric field stimulator 1112 moves toward or away from the field effect sensors 1106A-1106n. Alternatively, such modalities can be configured (via, for example, sensory and / or stimulatory geometry) so that the electric field stimulator 1112 is coupled or decoupled from the electric field or fields around each sensor 1106i of field effect as the electric field stimulator 1112 reaches different points in its travel towards or away from the field effect sensors H06A-1106n. Other modifications of the above modalities are possible.

For example, the biasing means of any of the above embodiments may be omitted so that the arrow 1108 or push button 1120 remains in the last position in which it was placed by the user. In addition, although arrow 1108 and post 1118 are shown substantially perpendicular to substrate 1102, arrow 1108 and post 1118 can be configured at other angles to substrate 1102, as will be known to those skilled in the art. Figures 36A to 36B illustrate an embodiment 1200 of the present invention that emulates a mechanical rocker switch. The mode 1200 includes a substrate 1202 defining an opening 1204. The field effect sensor 1206A is placed on the substrate 1202 in proximity to the opening 1204. The arrow 1208 extends through and is pivotally connected to the substrate 1202 in the opening 1204. The arrow 1208 may, but need not necessarily be, include a knob 1210. A bearing (not shown) may be provided, for example, a cylindrical or spherical bearing or other means (not shown) in the opening 1204 to provide support or limit the degree and direction of movement of the arrow 1208. For example, in a mode designed for use as a simple on-off switch, it may be desirable to limit the arrow 1208 so that it can be moved only in a single plane , for example, to the left and to the right in the modality of Figure 36A. The electric field stimulator 1212 is attached to the arrow 1208 at a predetermined location, as discussed above in relation to the mode 1100. In the embodiment of FIGS. 36A to 36B, a helical spring 1214 is inserted between the head 1210 and the substrate 1202, by deflecting the arrow 1208 to a centered position where the arrow 1208 is substantially perpendicular to the substrate 1202. In other embodiments other means may be used to deflect the arrow 1208 to a centered position or other desired position, as will be recognized those skilled in the art. Alternatively, such deflection means may be omitted so that arrow 1208 normally remains in the last position to which it will move. In operation, an electric field is generated around the sensor 1206A of field effect, as stated in the above. With the arrow 1208 in the centered position, the electric field stimulator 1202 is sufficiently separated from this electric field so that the electric field stimulator 1212 does not alter this electric field. When the arrow 1208 is moved, for example, by a user applying a force perpendicular to the arrow 1208, the electric field stimulator 1212 is shifted so that at least a portion of the electric field stimulator 1212 moves closer to the sensor 1206A field effect, and therefore alters the electric field around the field effect sensor 1206A. The detection circuits associated with the field effect sensor 1206A detect this alteration and, in turn, send an output signal to the corresponding control circuits, as discussed above. The mode 1200 can be easily modified to provide a combination mode of a tilting / thrusting motor (not shown) by adapting the connection between the arrow 1208 and the opening 1204 so that the arrow 1208 can vascularize around and slide to through opening 1204, as will be understood by those skilled in the art. Figure 36C illustrates an alternative embodiment that includes four field effect sensors 1206A-1206D that are located in proximity to the aperture 1204 and that are spaced apart from each other around the aperture 1204 at 90 ° intervals. Each field effect sensor 1206A-1206D includes corresponding field generation and detection circuits. A particular field effect sensor 1206i is activated when, in response to the tilting action of the arrow 1208, the electric field stimulator 1212 approaches sufficiently so that the field effect sensor 1206i alters the electric field around the sensor 1206i field effect. Typically, only a field effect sensor 1206i is activated at any time. However, the field effect sensors 1206A-1206D (and their corresponding field generation and detection circuits) can be adapted so that two (or more) adjacent field effect sensors 1206i are driven simultaneously when the stimulator 1212 of Electric field is placed near them. For example, in the embodiment of Figure 36C, the electric field stimulator 1212 can be coupled to both field effect sensors 1206A and 1206B when the arrow 1208 is pivoted such that at least a portion of the stimulator is placed. 1212 electric field between field effect sensors 1206A and 1206B. In alternative embodiments, more or less than four field effect sensors may be distributed over the substrate 1202 around the aperture 1204 in any desired distribution, as will be recognized by those skilled in the art. Figure 36D illustrates another embodiment 1240 of the present invention that emulates a mechanical rocker switch. Mode 1240 includes an arrow 1208 connected to substrate 1202 at pivot point 1224. In this embodiment, the arrow 208 does not penetrate the substrate 1202. The electric field stimulator 1212 joins the arrow 1208 at a predetermined distance from the pivot point 1224. A deflection means (not shown) can be provided to deflect the arrow 1208 to any desired position. Figures 37A to 37D illustrate a mode 1300 of the present invention emulating a mechanical rotary switch. The substrate 1302 defines an opening 1304. An internal field effect sensor 1306A and an external field effect sensor 1306B are placed on a surface of the substrate 1302 at a first and second predetermined distances, respectively, from the aperture 1304. The arrow 1308 is inserted through and is free to rotate within the opening 1304. A bushing, bearing or other means (not shown) can be provided to allow the arrow 1308 to rotate better within the opening 1304 and to prevent the arrow 1308 from rotating. slides through the opening 1304. Preferably, the arrow 1308 includes a knob 1310 to facilitate gripping and rotation of the arrow 1308 by a user. The electric field stimulator mounting plate 1330 is attached to the date 1308 at a predetermined distance from the substrate 1302 by any suitable means, as is known to those skilled in the art. The indoor electric field stimulators 1332 are mounted on the electric field stimulator mounting plate 1330 in an annular distribution at a predetermined distance from the center of the electric field stimulator mounting plate 1330. This predetermined distance corresponds and preferably equals a predetermined distance from the center of the opening 1304 to the inner field effect sensor 1306A. Similarly, the outdoor electric field stimulators 1334 are mounted on the electric field stimulator mounting plate 1330 in an annular distribution at a predetermined distance from the center of the plate 1330 of the corresponding electric field stimulator assembly and which preferably is equal to the predetermined distance from the center of the opening 1304 of the outdoor field effect sensor 1306B. Preferably, the angular separation between adjacent indoor electric field stimulators 1332 is the same. Similarly, the angular separation between adjacent external electrical field stimulators 1334 are also preferably the same. In operation, a user rotates knob 1310, which in turn rotates arrow 1308 and electric field stimulator mounting plate 1330. As the electric field stimulator mounting plate 1330 rotates, each internal electric field stimulator 1332 is alternately coupled and uncoupled from the electric field around the indoor field effect sensor 1306A. In a similar wayeach external electric field stimulator 1334 is coupled and uncoupled from the electric field around the outdoor field effect sensor 1306BA. The detection circuits associated with the field effect sensors 1306A, 1306B detect this coupling and decoupling and provide output signals corresponding to a control circuit (not shown). The control circuit can be adapted to recognize the degree and rotation of the knob 1310 based on these signals.

Preferably, the inner electric field stimulators 1332 are not radially aligned or angularly centered between adjacent external electric field stimulators 1334. In this way, the indoor electric field stimulators 1332 will couple and disengage from the electric field around the indoor field effect sensor 1306A at certain angular shifts of the knob 1310 and the outdoor electric field stimulators 1334 will be coupled and decoupled from the electric field around the outdoor field effect sensor 1306B at different angular displacements of the knob 1310. Figure 37E illustrates typical signals of output signals of the detection circuits associated with field effect sensors 1306A, 1306B as the knob 1310 is rotated in a particular direction. Based on these signals, a microprocessor can determine if the knob 1310 is being turned clockwise or in a counterclockwise direction, as will be recognized by those skilled in the art. In alternative embodiments, one of the indoor field effect sensor 1306A and the outdoor field effect sensor 1306B may be omitted. In such embodiments, the corresponding inner electric field stimulators 1332 or the external electric field stimulators 1334 will preferably also be omitted. In other alternative embodiments, the arrow 1308 may be adapted to slide longitudinally through, as well as rotate within the opening 1304 and a means may be provided to deflect the arrow 1308 longitudinally as set forth in the foregoing in connection with the emulation modes of pushbutton switch for mechanical pressing and thus a combination of a rotary / push or pull switch emulation mode is provided. Such embodiments may include one or more additional field effect sensors or electric field stimulators to facilitate said push or pull switch functionality, as will be understood by a person skilled in the art. Figure 37F illustrates an alternate rotary switch emulation mode 1350 of the present invention. The mode 1350 includes a second substrate 1340 in predetermined spatial relationship with the substrate 1302. The second indoor and outdoor field effect sensors 1306C, 1306D are placed on the second substrate 1340. The second indoor and outdoor electric field stimulators 1342 and 1344 they are placed on a second surface of the electric field stimulator mounting plate 1330 opposite the surface on which the inner and outer electric field stimulators 1332 and 1334 are placed. The arrow 1308 is free to slide, as well as to rotate within the aperture 1304. FIG. 37F illustrates the electric field stimulator mounting plate 1330 in a first position wherein the inner and outer electric field stimulators 1332 and 1334 are in relatively close proximity to the substrate 1302 (and therefore the rings in which the indoor and outdoor field effect sensors 1306A and 1306B are located) and the seconds indoor and outdoor electric field stimulators 1342, 1344 are relatively remote from the second substrate 1340. In this position, rotation of the knob 1310 causes the inner and outer electric field stimulators 1332 and 1334 to alternately engage and disengage from the electric fields around the indoor and outdoor field effect sensors 1306A and 1306B, respectively. In this position, the second inner and outer electric field stimulators 1332 and 1334 remain sufficiently far from the second inner and outer field effect sensor 1306C and 1306D so that the second inner and outer electrical field stimulators 1342 and 1344 do not couple. and decoupling from the electric fields around the respective field effect sensors 1306C and 1306D. By pressing the 1310 knob, a user can move the electric field stimulator mounting plate 1330 to a second position where the inner and outer electric field stimulators 1332 and 1334 are relatively far from the substrate 1302 and the second inner and outer electric field stimulators 1332 and 1334 are relatively in close proximity to the second substrate 1340 (and therefore, the rings in which the second indoor and outdoor field effect sensors 1306C and 1306D are located). In this position, rotation of the knob 1310 causes the second inner and outer electric field stimulators 1342 and 1344 to alternately couple and uncouple the electric fields around the second inner and outer field effect sensors 1306C and 1306D , respectively. In this position, the indoor and outdoor electric field stimulators 1332 and 1334 remain sufficiently far from the indoor and outdoor electric field sensors 1306A and 1306B so that the indoor and outdoor effect sensors 1306A and 1306B do not couple and uncouple from the electric fields around the respective field effect sensors 1306A and 1306B. Helical spring 1314 can be provided to polarize the mounting plate 1330 of the electric field stimulator to a "normal" position, as illustrated in Figure 37F. In other embodiments, the electric field stimulator mounting plate 1330 may be biased to a different "normal" position. In further embodiments, the helical spring 1314 can be omitted, so that the electric field stimulator mounting plate 1330 remains in any desired position between the substrate 1302 and the second substrate 1340. In addition, the 1350 mode can be adapted accordingly. that both sets of indoor and outdoor electric field stimulators 1332 and 1334, and 1342 and 1344 can be coupled to the electric fields around respective field effect sensors 1306A, 1306B, 1306C and 1306D when the stimulator mounting plate 1330 The electric field is placed substantially mid-way between the substrate 1302 and the second substrate 1340. Alternatively, the mode 1350 can be adapted so that an electric field stimulator can not be coupled to its respective field effect sensor when The mounting plate of the electric field stimulator is placed in this way. All of the foregoing modalities are suitable for use in connection with detection and analog or digital control circuits, as will be understood by a person skilled in the art. Figure 37G illustrates an alternative embodiment 1360 of the present invention that emulates a mechanical rotary switch that is particularly suitable for use in connection with detection and analog control circuits. The mode 1360 includes a substrate 1302 which defines an opening 1304. The field effect sensor 1306 is placed on the substrate 1302 in proximity to the opening 1304. The arrow 1308 is inserted through and is free to rotate within the opening 1304 In the embodiment illustrated, the arrow 1308 is fixed longitudinally. In other embodiments, the arrow 1308 may be adapted to slide through the opening 1304. The arrow 1308 preferably includes a knob 1310 at one end. The electric field stimulator 1328 is attached to the arrow 1308 at a predetermined distance from the substrate 1302. The electric field stimulator 1328 is preferably tapered similar to a driving blade so that the distance between the field effect sensor 1306 and the electric field stimulator 1328 varies with rotation of knob 1310 and arrow 1308. Alternatively, electric field stimulator 1328 can be substantially flat and parallel to substrate 1302 and have a width or thickness that varies with the distance from the arrow 1308, as illustrated in figure 371. As such, the degree of coupling of the electric field stimulator 1328 with the electric field around the field effect sensor 1306 varies with the rotation of the knob 1310 as a function of the distance between the electric field stimulator 1328 and the effect sensor 1306 field or the effective area of the electric field stimulator 1328 in proximity to the field effect sensor 1306. By using appropriate analog detection and control circuits, the 1360 mode can emulate, for example, a potentiometer. Figure 37H illustrates another alternative embodiment 1380 of the present invention that is particularly suitable for use in connection with analog detection and control circuits. The mode 1380 includes the substrate 1302 which defines the opening 1304 having internal threads 1305. The field effect sensor 1306 is placed on the substrate 1302 in proximity to the opening 1304. The threaded shaft 1308 has a knob 1310 at one end and is screwed into the opening 1304. The electric field stimulator 1312 is attached to the arrow 1308 in a predetermined place. As the knob 1310 rotates clockwise, the electric field stimulator 1312 moves away from the field effect sensor 1306. Conversely, as the knob 1310 rotates counterclockwise, the electric field stimulator 1312 moves closer to the field effect sensor 1306. As such, the rotation of the knob 1310 affects the coupling changes between the electric field stimulator 1312 and the field effect sensor 1306.

These coupling changes can be detected and easily processed by analog control detection circuits, as is known to those skilled in the art. Figures 38A to 38D illustrate another additional embodiment 1400 of the present invention that emulates a rotary switch. The mode 1400 includes a substrate 1402. The inner and outer knobs 1450 and 1452 are attached to the substrate 1402 by any suitable means such that each knob 1450, 1452 can rotate about an axis substantially perpendicular to the substrate 1402, as will be recognized by the experts in the art. One or more electric field stimulators 1412 are placed in the base of each of the inner and outer knobs 1450 and 1452. The inner and outer field effect sensors 1406A and 1406B are placed on the substrate 1402 substantially in alignment with stimulators 1412 electric field placed in respective inner and outer knobs 1450 and 1452 so that each electric field stimulator 1412 is alternately coupled and decoupled from the electric field around the respective field effect sensor 1406A, 1406B before the rotation of the 1450 knob , 1452 inside and outside respectively. The electric field stimulator 1412 can be constituted in various ways. For example, each electric field stimulator 1412 may be a conductive mass 1413, for example a ball bearing, which is placed at the bottom of the respective knob 1450, 1452. Alternatively, each electric field stimulator 1412 may be a projection 1417 on a ring 1415 inserted in the lower part of the respective knob 1450, 1452, as shown in Figure 38D. In a preferred embodiment, ring 1415 is made of beryllium and copper having patterned projections 1517. Figures 39A to 39B illustrate an alternative embodiment 1500 of the present invention emulating a rotary switch. This mode is particularly suitable for angular position detection applications. These modalities use a single field effect sensor with multiple detection electrodes. This embodiment includes the substrate 1502 on which they are placed in a detection 1503 of generally circular distribution and a sequence of electrodes in detectors 1505A to 1505H interposed with the resistors R1 to R7. In alternative embodiments, the detection circuit 1503 can be located remotely and with more or fewer detection electrodes and resistors to those shown are those that can be used. The knob 1510 is connected to the substrate 1502 so that the knob 1510 can rotate about an axis substantially perpendicular to the substrate 1502. In the embodiment of Figure 39A, the arrow 1508 is inserted inside and is free to rotate within the opening 1504 defined by the substrate 1502 and the knob 1510 is fixed to the arrow 1508. In other embodiments, the arrow 1508 can be fixed to the substrate 1502 and the knob 1510 can rotate about the arrow 1508. The field effect stimulator 1512 is embedded inside or is associated in some other way with the knob 1510 so that the field effect stimulator 1512 rotates with the knob 1510 through an arc substantially corresponding to the circular distribution in which the electrodes 1505A to 1505H and the resistors R1 to R7 are placed on the substrate 1502. In operation, as a user rotates knob 1510, the electric field stimulator 1512 alternately couples and disconnects from the field s electrical around the corresponding electrodes 1505A to 1505H. Analogue detection circuits can be adapted to determine the degree, rate and direction of rotation of the knob 1510, as will be understood by a person ordinarily skilled in the art. In a preferred embodiment, the sensing circuit 1503 may take the form shown in Fig. 33A, with electrodes 1505A and 1505H of the embodiment of Fig. 39A taking the place of electrodes E2 and El., respectively, which are shown in Figure 33A. The forces of the signals at the inputs (+) and (-) and, therefore, the sum of the adder 160 will have unique predetermined values for each position of the electric field stimulator 1512 with respect to the electrodes 1505A to 1505H, as will recognize an expert in the art. (A detection circuit of the shape shown in Figure 33A can also be used to detect variations in the distance between two conductive sheets, as will be recognized by a person skilled in the art.) As such, the detection circuit of the figure 33A that can be used in relation to a pair of distributed conductor sheets such as a vibration sensor, a sound pressure sensor, an air pressure sensor, a similar position sensor.In some embodiments, a conductive foam layer is you can place between the conductive sheets). The lead 1507 having variable impedance on its length, as shown in Fig. 39B, can be used in place of the electrode-resistor chain shown in Fig. 39A. The continuously varying impedance of the conductor 1507 provides a continuously varying output for the detection circuit 1503, as the field effect stimulator 1512 changes position in response to the rotation of the knob 1510. As such, the use of the conductor 1507 it may be preferred when a fine resolution is required, for example, in the angular position. The embodiment of Figures 39A to 39B can be easily adapted for use as an angular position sensor, as can be recognized by those skilled in the art. The principles of the embodiments of Figs. 39A to 39B can be easily adapted to provide a slidable switch or slidable potentiometer simply by distributing field effect sensors 1505A to 1505H and resistors Rl to R7 linearly and when replacing knob 1510 with a slide, as shown, for example, in Figure 42A. These principles can be extended further to detect the position of a stimulus in an X-Y distribution by creating a detection circuit distribution and electrode-resistor chains, as shown in Figure 42E. Figure 40 illustrates another additional embodiment 1600 of the invention emulating a rotary switch. The mode 1600 includes a substrate 1602 and an arrow 1608 having an outer knob 1610 and an inner knob 1611. The substrate 1602 is formed, for example, by molding, to retain the inner knob 1611 and encapsulate the field effect sensor 1606. The electric field coupling element 1612 is encapsulated or embedded in some other way within the outer knob 1610. Alternatively, the electric field coupling element 1612 may be encapsulated or otherwise embedded within the inner knob 1611. A light emitting device 1621 can be encapsulated within the substrate 1602. By selecting transparent or translucent materials for at least the portions of the substrate 1602, the inner and outer knobs 1610 and 1611 and the arrow 1608, the light emitting device 1621 is it can be used to selectively illuminate at least a portion of the outer knob 1610. Figure 41A illustrates a mode 1700 of the present invention that emulates a mechanical rocker switch. The modality 1700 includes a substrate 1702, two field effect sensors 1706A and 1706B positioned on a surface of the substrate 1702, and electric field stimulators 1713A and 1713B in the form of an oscillator 1713 attached to a substrate 1702. In the illustrated embodiment, the oscillator 1713 is a piece of curved spring steel fixed to the substrate 1702 and the electric field stimulators 1713A and 1713B are monolithic portions of the oscillator 1713. In alternative embodiments, the oscillator 1713 can be made from other materials and acquire other forms and stimulators 1713A and 173B electric field can be separate elements, for example, ball bearings, embedded within oscillator 1713, as will be recognized by a person skilled in the art. In operation, a user depresses either any stimulators 1713A electric field corresponding to the left side of the oscillator 1713 or stimulator 1713B electric field corresponding to the right side of the oscillator 1713, to the substrate 1702. As the stimulator 1713A or 1713B of electric field approaches or makes contact with the substrate 1702, the electric field stimulator 1713A or 1713B is coupled to the corresponding field effect sensor 1706A, 1706B. In the illustrated embodiment, both electric field stimulators 1713A and 1713B can be moved to the substrate 1702 at the same time. Preferably, the oscillator 1713 is configured such that only one of the electric field stimulators 1713A and 1713B can move toward the substrate 1702 at any time. Figure 41B illustrates another embodiment 1750 of the present invention that emulates a mechanical rocker switch. The 1750 mode is similar to the 1700 mode, except that the 1750 mode uses a rigid 1713 oscillator. In some embodiments, the oscillator 1713 can be made from a material that provides sufficient coupling to the field effect sensors 1706A, 1706B when depressed. In such embodiments, the conductive masses 1715 can be embedded in appropriate places, in oscillator 1713 to improve such coupling, as will be recognized by those skilled in the art. In other embodiments, the oscillator 1713 can be formed and may be sized so that the user's finger on the oscillator 1713 provides the coupling to the electric field around the sensor 1706A or 1706B field effect when the user presses the corresponding portion of oscillator 1713 to substrate 1702. A biasing means may be provided to bias oscillator 1713 to a predetermined "normal" position. In Figure 41B, the biasing means is included as a pair of tabs 1725 attached to the substrate plastic 1702. The tabs 1725 they are plastic flexible enough to flex when the oscillator 1713 is depressed, and sufficiently resilient to return to the oscillator 1713 to position "normal" when the oscillator 1713 is released. Any other suitable polarization means can be used, as will be recognized by those skilled in the art. Alternatively, as illustrated in Figure 41C, oscillator 1713 and the tabs may be adapted to secure oscillator 1713 in a particular position until it is repositioned by a user. In such embodiment, tabs 1725 preferably include buttons 1725 projecting toward the ends of oscillator 1713 and oscillator 1713 preferably includes concavities 1727 at their ends to receive buttons 1725.

Figure 42A illustrates an embodiment 1800 of the present invention that emulates a sliding mechanical switch. The mode 1800 includes a substrate 1802. One or more field effect sensors 1806 are placed on the substrate 1802. The electric field stimulator 1812, for example, a conductive cylinder or a ball bearing, is attached to the slide 1811. The slider 1811 is coupled with the rails 1803 attached to the substrate 1802. In operation, a user slides the slider 1811 back and forth along the substrate 1802. As the electric field stimulator 1812 is brought into proximity with a sensor 1806 of particular field effect, the electric field stimulator 1812 is coupled to the electric field around said field effect sensor 1806. In the same way, when the electric field stimulator 1812 moves away from a particular field effect sensor 1806, the electric field stimulator 1812 is decoupled from the electric field around said electric field sensor 1806. In an alternative embodiment, the slider 1811 can be replaced with the slider 1817 having a cutout 1819 designed to accommodate a user's finger. In this mode, the user's finger functions as the electric field stimulator 1812. In a further alternative embodiment, the slide 1811 can be completely eliminated. The same principles can be applied to a rotary switch emulation by distributing the field effect sensors 1806 around the periphery of a cylinder (not shown) or a truncated cone 1807, as illustrated in FIG. 42D. In some embodiments, the portions of the slide 1811 can be illuminated. Such embodiments preferably include a light tube 1821 and a light source (not shown) for illuminating the light tube 1821 relative to the substrate 1802, such as a light channel 1823 positioned on the slide 1811 that can receive light from the 1821 light tube. In other embodiments, another means may be used to illuminate the slide 1811 or portions thereof. Figure 42B illustrates another embodiment 1850 of the present invention emulating a slide switch. The 1850 mode is similar to the 1800 mode, except that the mode eliminates the 1811 slider altogether. The flexible sheet 1827 below the rails 1803 so as to be superimposed on the substrate 1802. Preferably, the sheet 1827 is easily replaceable and may include graphics indicating, for example, the location of the field effect sensors (not shown) placed on the substrate 1802 under the sheet 1827. Usually there is an air gap between the sheet 1827 and a field effect sensor (not shown) placed on the substrate 1802 below the sheet 1827. When a user touches the sheet 1827 to actuate said field effect sensor, the air moves from the air gap, which allows and improves the coupling of the user's fingers to the electric field around the field effect sensor. Figure 42C illustrates an alternative embodiment 1860 of the present invention that emulates a slide switch. The field effect sensor 1806 is placed on the substrate 1802. The substrate 1802 includes rails 1803. The slider 1811 is slidably coupled to the substrate 1802 via rails 1803. The electric field stimulator 1812 is preferably a conductive mass placed on the slide 1811. In the embodiment of Figure 41C, the cross-sectional area of the electric field stimulator 1812 varies from one end of the slide 1811 to the other. With the slide 1811 in the position shown in FIG. 42C, the electric field stimulator 1812 is remote from the field effect sensor 1806 and does not mesh with the electric field around the field effect sensor 1806. As the slider 1811 moves to the right, for example, by a user's finger, the electric field stimulator finally moves close enough to the field effect sensor 1806 to couple to the electric field around the sensor 1806 of effect. countryside. Initially, said coupling is small due to the small area of the electric field stimulator 1812 which is in proximity with the field effect sensor 1806 and the corresponding electric field. As the slide 1811 moves farther to the right, a larger portion of the electric field stimulator 1812 is brought into proximity with the field effect sensor 1806 and the corresponding electric field and the coupling of the electric field stimulator 1812 to the electric field it increases. The analog detection circuits can discern the variable coupling state and provide an analog output corresponding to a corresponding control circuit. A biasing means, for example, a helical spring 1814, may be provided to keep the slider 1811 in a "normal" position in the absence of a force-shifting slider 1811 from said "normal" position. Figure 43 illustrates a mode 1900 of the present invention that emulates a mechanical spherical switch or a control ball. The mode 1900 includes a substrate 1902 that forms a housing 1962 for the ball 1960. One or more field effect sensors 1906 are distributed on the surface or embedded within the substrate 1902. The perimeter of the 1960 ball includes field stimulators 1912 distributed in a unique non-repetitive pattern. In operation, as the ball 1960 rotates within the housing 1962, the electric field stimulators are coupled and decoupled from the electric fields around the field effect sensors 1906. The detection and control circuits associated with the field effect sensors 1906 can be adapted to determine the degree and direction of rotation of the ball 1960, as will be recognized by those skilled in the art. In an alternative embodiment, the ball 1960 can be fixed and the substrate 1902 and the housing 1962 can be allowed to rotate or move in some other way around the housing 1962. This mode can be used, for example, to detect inclination or vibration. Figure 44 illustrates a specific embodiment 2000 of a mechanical switch emulation according to the present invention, in particular, a regulator for a snow vehicle or a personal watercraft. A 2006 field effect sensor is placed inside or encapsulated within a handle 2002. The electric field stimulator 2012 in the form of a conductive mass is placed on the regulating lever 2016. As the user presses and releases the regulating lever 2016, the electric field stimulator 2012 moves closer to or away from the field effect sensor 2006, respectively. Analog detection and control can be used to determine the regulatory pressure based on signals received from a field effect sensor 2006. In the preferred embodiments, the additional field-effect sensors 2031, 2033 and 2035 can be placed on the handle 2002. These additional sensors can include, for example, a redundant sensor 2031 for controller control, a hand-position sensor 2033. that disables the regulator unless it detects the driver's hand on the 2002 handle and a 2035 water sensor that disables the regulator when submerged in water due, for example, to watercraft reversal. Figures 45A to 45B illustrate a specific embodiment of a tire pressure sensor 2100 according to the present invention. In preferred embodiments, a compressible and preferably conductive foam substrate 2104 is placed on a substrate surface 2102. A plurality of field effect sensors 2106 are distributed in a matrix array on the other surface of the substrate 2102. In operation, the tire 2108, for example of an automobile (not shown) is placed on the foam substrate 2104 and thus the portion of the foam substrate 2104 is compressed in contact with the tire 2108. The compressed portion of the substrate 2104 of foam is coupled with the electric fields around the corresponding field effect sensors 2106, and in this way these sensors operate, as will be understood by those skilled in the art. A microprocessor (not shown) programmed with the weight of the load in the tire 2108 can determine the air pressure in the tire 2108 based on the signals it receives from the field effect sensors 2106 corresponding to the compressed foam area by the tire 2108. In other embodiments, the foam substrate 2104 may be omitted, such that the tire 2108 carry out the coupling of the field effect sensors 2106. Figure 46 illustrates the passenger seat 2202 of an automobile having a seat portion 2202A and a backrest portion 2202B. The seat 2202 is preferably placed or padded using compressible foam 2204 into which a plurality of field effect sensors 2206A are embedded to detect the weight placed on the seat 2202 and a plurality of field effect sensors 2206B to detect the dimensions of a person or article placed on the seat 2202. The field effect stimulators 2212, constituted as seat support posts in the embodiment illustrated, is located in a predetermined spatial relationship with the field effect sensors 2206A. With the seat 2202 empty, the field effect sensors 2206A are at a predetermined distance from the effect stimulators 2212 in such a way that the field effect sensors 2206A are not actuated. When a load, for example a person or a package, is placed on the seat 2202, the foam 2204 in the seat portion 2202A is compressed, moving the field effect sensors 2206A closer to the field effect stimulators 2212, which causes the field effect stimulators 2212 to alter the electric field around the field effect sensors 2206A. The heavier the load placed in the seat 2202, the greater the compression of the foam 2204 in the seat portion 2202B and the corresponding displacement of the field effect sensors 2206A. An analog detection and control circuit (not shown) that receives the signals emitted from the field effect sensors 2206A can determine from these signals the displacement of the field effect sensors 2206A in response to the load placed on the seat 2202. The control circuit can determine the weight of the affected person or of the article placed on the seat 2202 based on these displacement data and the compressibility characteristics of the foam 2204. Furthermore, with the seat 2202 empty, no stimulus is given. It couples to the electric fields around the field effect sensors 2206B. When a person sits or places a package in the seat 2202, portions of the person or package in proximity to any of the field effect sensors 2206B are coupled to the electric fields around these sensors. An analog or digital detection and control circuit that receives the output signals from the field effect sensors 2206B can determine the physical contour of the load (person or package) in the seat 2202. The control circuit can use this data in relation to the weight data derived from the signals received from the field effect sensors 2206A, as discussed above, to determine whether the load on the seat 2202 is a person or a packet. If the control circuit determines that the load is a package and not a person, it can deactivate the passenger's airbag. If the control circuit determines that the load is a person and not a package, it can adapt the deployment speed of the airbag and adapt to the size and weight of the person occupying the seat 2202. Although several modalities of the present invention, it will be evident to a person skilled in the art that numerous modifications can be made without departing from the spirit of the appended claims therein.

Claims

1. An apparatus that emulates a mechanical switch, characterized in that it comprises: a substrate; a first field effect sensor placed on the substrate; and a first electric field stimulator, wherein the first electric field stimulator can be moved along a predetermined path, the predetermined path has at least one first point relatively close to the first field effect sensor and at least a second point relatively remote from the first field effect sensor.

2. The apparatus according to claim 1, characterized in that it further comprises a second field effect sensor placed on the substrate, wherein the first electric field stimulator can be moved along a predetermined path, the predetermined path has at minus a third point relatively close to the second field effect sensor and at least a fourth point relatively remote from the second field effect sensor.

3. The apparatus according to claim 1, characterized in that it further comprises: a second field effect sensor placed on the substrate; a second electric field stimulator, wherein the second electric field stimulator can be moved along a predetermined path, the predetermined path has at least a third point relatively close to the second field effect sensor and at least one fourth point relatively far from the second field effect sensor.